General Description
The MAX16812 is a peak-current-mode LED driver with
an integrated 0.2Ωpower MOSFET designed to control
the current in a single string of high-brightness LEDs
(HB LEDs). The MAX16812 can be used in multiple
converter topologies such as buck, boost, or buck-
boost. The MAX16812 operates over a 5.5V to 76V wide
supply voltage range.
The MAX16812 features a low-frequency, wide-range
brightness adjustment (100:1), analog and PWM dim-
ming control input, as well as a resistor-programmable
EMI suppression circuitry to control the rise and fall
times of the internal switching MOSFET. A high-side
LED current-sense amplifier and a dimming MOSFET
driver are also included, simplifying the design and
reducing the total component count.
The MAX16812 uses peak-current-mode control,
adjustable slope compensation that allows for addition-
al design flexibility. The device has two current regula-
tion loops. The first loop controls the internal switching
MOSFET peak current, while the second current regula-
tion loop controls the LED current. Switching frequency
can be adjusted from 125kHz to 500kHz.
Additional features include adjustable UVLO, soft-start,
external enable/disable input, thermal shutdown, a
1.238V 1% accurate buffered reference, and an on-
chip oscillator. An internal 5.2V linear regulator supplies
up to 20mA to power external devices.
The MAX16812 is available in a thermally enhanced
5mm x 5mm, 28-pin TQFN-EP package and is specified
over the automotive -40°C to +125°C temperature range.
Applications
Automotive Lighting:
DRL, Fog Lights
Rear Combination Lights
Front and Rear Signal Lights
Interior Lighting
Warning and Emergency Lighting
Architectural and Industrial Lighting
Features
oIntegrated 76V, 0.2Ω(typ) Power MOSFET
o5.5V to 76V Wide Input Range
oAdjustable LED Current with 5% Accuracy
oFloating Differential LED Current-Sense Amplifier
oFloating Dimming N-Channel MOSFET Driver
oPWM LED Dimming with:
PWM Control Signal
Analog Control Signal
Chopped VIN Input
oPeak-Current-Mode Control
o125kHz to 500kHz Adjustable Switching Frequency
oAdjustable UVLO and Soft-Start
oOutput Overvoltage Protection
o5µs LED Current Rise/Fall Times During Dimming
Minimize EMI
oOvertemperature and Short-Circuit Protection
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
________________________________________________________________
Maxim Integrated Products
1
VIN
CIN
RTGRM
CH_REG
ROV1
ROV2
CTGRM
RT
DIM
TGRM
VOUT
L_REG
RT
EN
IN
LV
RCS
DOUT
VOUT
SRC
GT
DRV
SLP
COMP
RCOMP1
BUCK-BOOST CONFIGURATION
CCOMP1
CSLP
RSRC
COUT
RCOMP2
CS-
CS+
DGT
DD
H_REG
HV
LX
OV
SGND
AGND
REFI
REF
CS_OUT
FB
MAX16812
Simplified Diagram
19-0880; Rev 0; 7/07
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
EVALUATION KIT
AVAILABLE
Typical Application Circuit and Pin Configuration appear at
end of data sheet.
Ordering Information
+
Denotes a lead-free package.
*
EP = Exposed pad.
PART TEMP RANGE PIN-
PACKAGE
PKG
CODE
MAX16812ATI+ -40°C to +125°C 28 TQFN-EP* T2855-8
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, TA= TJ= -40°C to +125°C, unless oth-
erwise noted. Typical values are at TA= +25°C.)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
(All voltages are referenced to AGND, unless otherwise noted.)
SGND ....................................................................-0.3V to +0.3V
IN, EN, LX, DIM ......................................................-0.3V to +80V
L_REG, GT, DRV ......................................................-0.3V to +6V
RT, REF, REFI, CS_OUT, FB, COMP, SRC,
SLP, TGRM, OV ....................................................-0.3V to +6V
LV, HV, CS-, CS+, DGT, DD, H_REG ....................-0.3V to +80V
CS+, DGT, H_REG to LV ........................................-0.3V to +12V
CS- to LV ...............................................................-0.3V to +0.3V
CS+ to CS- .............................................................-0.3V to +12V
DD to LV ....................................................................-1V to +80V
Maximum Current into Any Pin (except LX, SRC) ............±20mA
Maximum Current into LX and SRC.......................................+2A
Continuous Power Dissipation (TA= +70°C)
28-Pin TQFN 5mm x 5mm
(derate 34.65mW/°C* above +70°C) .........................2759mW
Operating Temperature Range .........................-40°C to +125°C
Junction Temperature......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Input Voltage Range VIN 5.5 76.0 V
Quiescent Supply IQVTGRM = 1V, VDIM = 0V 0.3 2.5 mA
Shutdown Supply Current ISHDN VEN ≤ 300mV 20 45 µA
Internal MOSFET On-Resistance RDSON ILX = 1A, VIN > 10V, VGT = VDRV = 5V 0.2 0.4 Ω
Output Current Accuracy ILED ILED = 350mA, RCS = 1Ω-5 +5 %
Peak Switch Current Limit ILXLIM 2.6 3.1 3.6 A
Hiccup Switch Current 6A
Switch Leakage Current ILXLEAK VEN = 0V, VLX = 76V, VGT = 0V 1 10 µA
UNDERVOLTAGE LOCKOUT
IN Undervoltage Lockout UVLO VIN rising 4.6 4.9 5.3 V
UVLO Hysteresis 100 mV
EN Threshold Voltage VEN_THUP VEN rising 1.2 1.38 1.6 V
EN Hysteresis 100 mV
REFERENCE (REF) AND LOW-SIDE LINEAR REGULATOR (L_REG)
Startup Response Time tPOR VIN or VEN rising 50 µs
Reference Voltage VREF IREF = 10µA 1.190 1.238 1.288 V
Reference Soft-Start Charging
Current IREF_SLEW VREF = 0V 25 40 60 µA
L_REG Supply Voltage VIN = 7.5V, IL_REG = 1mA 4.9 5.2 5.5 V
L_REG Load Regulation IL_REG = 20mA 20 Ω
L_REG Dropout Voltage IL_REG = 25mA 400 mV
*As per JEDEC51 standard (multilayer board).
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, TA= TJ= -40°C to +125°C, unless oth-
erwise noted. Typical values are at TA= +25°C.)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
PWM COMPARATOR
COMP Input Leakage Current ILKCOMP VCOMP = 1V, VSRC = 0.5V, VTGRM = 1V,
VDIM = 0.5V -0.10 +0.10 µA
SRC Input Leakage Current ILKSRC VCOMP = 0V, VSRC = 0.5V, VTGRM = 0V,
VDIM = 0.5V -5 +5 µA
Comparator Offset Voltage VOS
(
EA
)
(VCOMP - VSRC) = VOS 860 mV
Input Voltage Range VSRC VCOMP = VSRC + 860mV 0 1.23 V
Propagation Delay tPD 50mV overdrive 100 ns
ERROR AMPLIFIER
FB Input Current VFB = 1V, VREFI = 1.2V -100 +100 nA
REFI Input Current VFB = 1V, VREFI = 1V -100 +100 nA
Error-Amplifier Offset Voltage VOS VFB = VCOMP = 1.2V -23 +23 mV
Input Common-Mode Range VFB = (VCOMP - 0.9V) 0 1.5 V
Source Current ICOMP (VREFI - VFB) ≥ 0.5V 300 µA
Sink Current (VFB - VREFI) ≥ 0.5V 80 µA
COMP Clamp Voltage VCOMP VREF = 1.2V, VFB = 0V 1.20 2.56 V
DC Gain 72 dB
Unity-Gain Bandwidth 0.8 MHz
ELECTRICAL CHARACTERISTICS
(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, RCS = 1Ω, TA= TJ= -40°C to +125°C,
unless otherwise noted. Typical values are at TA= +25°C.)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
HIGH-SIDE UNDERVOLTAGE LOCKOUT AND LINEAR REGULATOR (H_REG) ((VHV - VLV) = 21V)
H_REG Input-Voltage Threshold VH_REG is rising 3.60 3.887 4.20 V
H_REG Supply Voltage IH_REG = 0 4.75 5 5.40 V
H_REG Load Regulation IH_REG = 0 to 3mA 80 Ω
Dropout Voltage IH_REG = 5mA 820 mV
HIGH-SIDE CURRENT-SENSE AMPLIFIERS (VHV - VLV) = 21V
CS- Input Bias Current ICS- VCS- = VLV, (VCS+ - VCS-) = -0.1V 500 µA
CS+ Input Bias Current ICS+ VCS- = VLV, (VCS+ - VCS-) = 0.1V -1 +1 µA
Input Voltage Range VCS- = VLV 0 0.25 V
Sinking 25
Minimum Output Current ICS_OUT Sourcing 400 µA
Output Voltage Range VCS_OUT 0 1.5 V
DC Voltage Gain 4 V/V
Unity-Gain Bandwidth 0.8 MHz
Maximum REFI Input Voltage VREFI 1.0 V
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
4 _______________________________________________________________________________________
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
HIGH-SIDE DIMMING LINEAR REGULATOR ((VHV - VLV) = 21V)
VLV = VCS-, (VCS+ - VCS-) = 0.3V,
(VDD - VLV) = 1V, VDIM = 1V, VTGRM = 0V,
VDGT = 1V, VREFI = 1.0V, sinking
1.2
Minimum Output Current IDGT
VLV = VCS-, (VCS+ - VCS-) = 0.2V,
(VDD - VLV) = 1V, VTGRM = 0V, VDGT = 3V,
VREFI = 1.0V, VDIM = 1V, sourcing
1.2
mA
Output Voltage Range 0.2 5.0 V
DC Gain CDGT = 1nF to LV 60 dB
DD Input Bias Current IDD (VDD - VCS-) = 0.5V -3 +3 µA
DD Input Low Threshold VTGRM = 0V, VDIM = 1V, VREFI = 1.2V,
(VDGT - VLV) > 1.5V, VDD falling 0.25 0.50 0.75 V
DIMMING ((VHV - VLV) = 21V)
DIM Input Bias Current IDIM VDIM = 1.1V -1 +1 µA
TGRM Input High Threshold 1.18 1.23 1.27 V
TGRM Reset High-to-TGRM Low
Pulse Width 1µs
TGRM Reset Switch RDS(ON) VTGRM = 1.3V 20 Ω
Dimming Rise and Fall LED
Current Times 5µs
OVERVOLTAGE PROTECTION (OV)
OV Input High Threshold VOV rising 1.180 1.230 1.292 V
OV Input Threshold Hysteresis 14 mV
OV Input Bias Current IOV VOV = 1.1V -1 +1 µA
INTERNAL OSCILLATOR CLOCK
RT = 2MΩ to AGND 470 525 570
Internal Clock Frequency fOSC RT = 50kΩ to AGND 105 125 155 kHz
SLOPE COMPENSATION INPUT (SLP)
SLP Input Current ISLP VSLP = 0V 150 µA
LOW-SIDE GATE DRIVE (DRV)
DRV Output Low Impedance RDRV_LO DRV sinking 20mA 3 30 Ω
DRV Output High Impedance RDRV_HI DRV sourcing 20mA 10 45 Ω
INTERNAL POWER MOSFET
GT Input Leakage Current VGT = 0 to 5V -1 +1 µA
Internal MOSFET Gate-to-Source
Threshold Voltage VTH 2.5 V
Internal MOSFET Gate Charge QgVLX = 50V 8 nC
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, CREF = 47nF, VTGRM = 0V, RSRC = 0.2Ω, RCS = 1Ω, TA= TJ= -40°C to +125°C,
unless otherwise noted. Typical values are at TA= +25°C.)
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
_______________________________________________________________________________________
5
SHUTDOWN CURRENT
vs. TEMPERATURE
MAX16812 toc04
TEMPERATURE (°C)
SHUTDOWN CURRENT (μA)
1109580655035205-10-25
5
10
15
20
25
30
0
-40 125
VREF vs. TEMPERATURE
MAX16812 toc05
TEMPERATURE (°C)
VREF (V)
1109580655035205-10-25
1.22
1.23
1.24
1.25
1.21
-40 125
IREF = 10μA
IN UVLO THRESHOLD
vs. TEMPERATURE
MAX16812 toc06
TEMPERATURE (°C)
IN UVLO THRESHOLD (V)
1109580655035205-10-25
5.05
5.10
5.15
5.20
5.00
-40 125
VIN RISING
IN UVLO THRESHOLD
vs. TEMPERATURE
MAX16812 toc07
TEMPERATURE (°C)
IN UVLO (V)
1109580655035205-10-25
5.01
5.04
5.07
5.10
5.09
5.08
5.06
5.05
5.03
5.02
5.00
-40 125
VIN FALLING
EN UVLO THRESHOLD
vs. TEMPERATURE
MAX16812 toc08
TEMPERATURE (°C)
EN UVLO (V)
1109580655035205-10-25
1.05
1.20
1.35
1.50
1.45
1.40
1.30
1.25
1.15
1.10
1.00
-40 125
VEN RISING
RDS(ON) vs. ILX
MAX16812 toc01
ILX (A)
RDS(ON) (Ω)
2.52.01.5
0.05
0.10
0.15
0.20
0.25
0.30
0.35
0.40
0.45
0
1.0 3.0
TA = +125°C
TA = +25°C
TA = -40°C
RDS(ON) vs. VGT
MAX16812 toc02
VGT (V)
RDS(ON) (Ω)
6.45.84.6 5.23.4 4.02.8
0.2
0.4
0.6
0.8
1.0
1.2
1.4
1.6
1.8
2.0
0
2.2 7.0
TA = +25°C
SWITCH CURRENT LIMIT
vs. TEMPERATURE
MAX16812 toc03
TEMPERATURE (°C)
SWITCH CURRENT LIMIT (A)
11095-25 -10 5 35 50 6520 80
2.950
3.000
3.050
3.100
3.150
3.200
3.250
3.300
2.900
-40 125
Typical Operating Characteristics
(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, VTGRM = 0V, TA= +25°C, unless otherwise noted.)
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
6 _______________________________________________________________________________________
Typical Operating Characteristics (continued)
(VIN = VEN = 12V, CL_REG = 3.3µF, CH_REG = 1µF, VTGRM = 0V, TA= +25°C, unless otherwise noted.)
VL_REG vs. IL_REG
MAX16812 toc10
IL_REG (mA)
VL_REG (V)
181612 144 6 8 102
4.6
4.7
4.8
4.9
5.0
5.1
5.2
5.3
5.4
5.5
4.5
020
TA = +125°C
TA = +25°C
TA = -40°C
VIN = 7.5V
OSCILLATOR FREQUENCY
vs. TEMPERATURE
MAX16812 toc11
TEMPERATURE (°C)
OSCILLATOR FREQUENCY (kHz)
1109580655035205-10-25
100
200
300
400
500
600
0
-40 125
RT = 2MΩ
RT = 180kΩ
RT = 50kΩ
EN UVLO THRESHOLD
vs. TEMPERATURE
MAX16812 toc09
TEMPERATURE (°C)
EN UVLO (V)
1109580655035205-10-25
1.05
1.20
1.35
1.50
1.45
1.40
1.30
1.25
1.15
1.10
1.00
-40 125
VEN FALLING
OSCILLATOR FREQUENCY vs. RT
MAX16812 toc12
RT (MΩ)
OSCILLATOR FREQUENCY (kHz)
10.1
100
200
300
400
500
600
0
0.01 10
VH_REG THRESHOLD
vs. TEMPERATURE
MAX16812 toc13
TEMPERATURE (°C)
VH_REG THRESHOLD (V)
1109580655035205-10-25
3.6
3.9
4.2
4.1
4.0
3.8
3.7
3.5
3.4
-40 125
VH_REG vs. IH_REG
MAX16812 toc14
IH_REG (mA)
VH_REG (V)
2.52.01.51.00.5
4.55
4.60
4.65
4.70
4.75
4.80
4.85
4.90
4.95
5.00
4.50
0 3.0
(VHV - VLV) = 6V
VIN = 12V
VH_REG IS MEASURED
WITH RESPECT TO VLV
VH_REG vs. TEMPERATURE
MAX16812 toc15
TEMPERATURE (°C)
VH_REG (V)
1109580655035205-10-25
4.3
4.6
4.9
5.2
5.1
5.0
4.8
4.7
4.5
4.4
4.2
-40 125
(VHV - VLV) = 21V
ILOAD = 3mA
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
_______________________________________________________________________________________ 7
Pin Description
PIN NAME FUNCTION
1 FB Low-Side Error Amplifier’s Inverting Input
2 COMP Low-Side Error Amplifier’s Output. Connect a compensation network from COMP to FB for stable operation.
3 REFI Reference Input. VREFI provides the reference voltage for the high-side current-sense amplifier to set the
LED current.
4 REF +1.23V Reference Output. Connect an appropriate soft-start capacitor from REF to AGND.
5 CS_OUT High-Side Current-Sense Amplifier Output. VCS_OUT is proportional to the current through RCS.
6 AGND Analog Ground
7EN
Enable Input/Undervoltage Lockout. Connect EN to IN through a resistive voltage-divider to program the
UVLO threshold. Connect EN directly to IN to set up the device for 5V internal threshold. Apply a logic-
level input to EN to enable/disable the device.
8 IN Positive Power-Supply Input. Bypass with a 1µF ceramic capacitor to AGND.
9 L_REG 5V Low-Side Regulator Output. Bypass with a 3.3µF ceramic capacitor to AGND.
10 SGND Signal Ground
11 DD MOSFET’s Drain Voltage-Sense Input. Connect DD to the drain of the external dimming MOSFET.
12 DGT External Dimming MOSFET’s Gate Drive
13 CS+ High-Side Current-Sense Amplifier’s Positive Input. Connect RCS between CS+ and CS-. CS+ is
referenced to LV.
14 CS- High-Side Current-Sense Amplifier’s Negative Input. Connect RCS between CS- and CS+. CS- is
referenced to LV.
15 LV High-Side Reference Voltage Input. A DC voltage at LV sets the lowest reference point for the high-side
current-sense and dimming MOSFET control circuitry.
16 H_REG High-Side Regulator Output. H_REG provides a regulated supply for high-side circuitry. Bypass with a 1µF
ceramic capacitor to LV.
17 HV High-Side Positive Supply Voltage Input. HV provides power for dimming and LED current-sense circuitry.
HV is referenced to LV.
18 DRV Internal MOSFET Gate Driver Output. Connect to a resistor between DRV and GT to set the rise and fall
times at LX.
19 GT Internal MOSFET GATE. Connect a resistor between GT and DRV to set the rise and fall times at LX.
20, 21 LX Internal MOSFET Drain
22, 23 SRC Internal Power MOSFET Source
24 SLP Slope Compensation Setting. Connect an appropriate external capacitor from SLP to AGND to generate a
ramp signal for stable operation.
25 TGRM Dimming Comparator’s Reference/Ramp Generator
26 DIM Dimming Control Input
27 RT Resistor-Programmable Internal Oscillator Setting. Connect a resistor from RT to AGND to set the internal
oscillator frequency.
28 OV
Overvoltage Protection Input. Connect OV to HI through a resistive voltage-divider to AGND to set the
overvoltage limit for the load. When the voltage at OV exceeds the 1.238V (typ) threshold, the gate drive
(DRV) for the switching MOSFET is disabled. Once VOV goes below 1.238V by 14mV, the switching
MOSFET turns on again.
—EP
Exposed Pad. Connect EP to a large-area ground plane for effective power dissipation. Do not use as the
IC ground connection.
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
8 _______________________________________________________________________________________
LDOH
POR
3.88V 1.2X
1.1X
1X
PREG BG
VREF
LDOL
OSC
tD = 200ns
UVLO/
POR
DIM
RAMP
REF
CMP
CMP
IHI CSA
ADIM
DS
DRMP
0.5V
G1
LATCH
QS
R
1X
LOGIC
CONTROL
EN
ERROR
AMPLIFIER
AND
DIMMING
S/H
VREFI = 1.2V
VRAMP = 0.3V
CMP
OVP
CMP
1.238V
1.238V
2μs PULSE
LOW TO DISCHARGE
DIM
SIGNAL
2.5V
VDD
1.2V
SGND
HICCUP
0.6V
ILIM
PWM
VBE
X0.2
X1
MAX16812
HV
H_REG
LV
IN
L_REG
EN
REF
RT
DIM
TGRM
OV
SGND AGND
DD
CS+
CS-
LX
LX
SRC
SRC
GT
DRV
SLP
COMP
FB
CS_OUT
REFI
DGT
Figure 1. Functional Diagram
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
_______________________________________________________________________________________ 9
Detailed Description
The MAX16812 is a current-mode PWM LED driver
with an integrated 0.2Ωpower MOSFET for use in dri-
ving HB LEDs. By using two current regulation loops,
5% LED current accuracy is achieved. One current reg-
ulation loop controls the internal MOSFET peak current
through a sense resistor (RSRC) from SRC to ground,
while the other current regulation loop controls the
average LED current in a single LED string through
another sense resistor (RCS) in series with the LEDs.
The MAX16812 includes a cycle-by-cycle current limit
that turns off the gate drive to the internal MOSFET dur-
ing an overcurrent condition. The MAX16812 features a
programmable oscillator that simplifies and optimizes
the design of magnetics. The MAX16812 is well suited
for inputs from 5.5V to 76V. An external resistor in
series with the internal MOSFET gate can control the
rise and fall times on the drain of the internal switching
MOSFET, therefore minimizing EMI problems.
The MAX16812 high-frequency, current-mode PWM
HB LED driver integrates all the necessary building
blocks for driving a series LED string in an adjustable
constant current mode with PWM dimming. Current-
mode control with leading-edge blanking simplifies
control-loop design, and an external adjustable slope-
compensation control stabilizes the inner current-mode
loop when operating at duty cycles above 50%.
An input undervoltage lockout (UVLO) programs the
input supply startup voltage. An external voltage-
divider on EN programs the supply startup voltage. If
EN is directly connected to the input, the UVLO is set at
5V. A single external resistor from RT to AGND pro-
grams the switching frequency from 125kHz to 500kHz.
Wide contrast (100:1) PWM dimming can be achieved
with the MAX16812. A DC input on DIM controls the
dimming duty cycle. The dimming frequency is set by
the sawtooth ramp frequency on TGRM (see the
PWM
Dimming
section). In addition, PWM dimming can be
achieved by applying a PWM signal to DIM with TGRM
set to a DC voltage less than 1.238V. A floating high-
voltage driver drives an external n-channel MOSFET in
series with the LED string. REFI allows analog dimming
of the LED current, further increasing the effective dim-
ming range over PWM alone. The MAX16812 has a 5µs
preprogrammed LED current rise and fall time.
A nonlatching overvoltage protection limits the voltage
on the internal switching MOSFET under open-circuit
conditions in the LED string. The internal thermal shut-
down circuit protects the device if the junction tempera-
ture should exceed +165°C.
Current-Mode Control
The MAX16812 offers a current-mode control operation
feature with leading-edge blanking that blanks the
sensed current signal applied to the input of the PWM
current-mode comparator. In addition, a current-limit
comparator monitors the same signal at all times and
provides cycle-by-cycle current limit. An additional hic-
cup comparator limits the absolute peak current to two
times the cycle-by-cycle current limit. The leading-edge
blanking of the current-sense signal prevents noise at
the PWM comparator input from prematurely terminat-
ing the on-cycle. The switch current-sense signal con-
tains a leading-edge spike that results from the
MOSFET gate-charge current, and the capacitive and
diode reverse-recovery current of the power circuit. The
MAX16812’s capacitor-adjustable slope-compensation
feature allows for easy stabilization of the inner switch-
ing MOSFET current-mode loop. Upon triggering the
hiccup current limit, the soft-start capacitor on REF is
discharged and the gate drive to DRV is disabled.
Once the inductor current falls below the hiccup cur-
rent limit, the soft-start capacitor is released and it
begins to charge after 10µs.
Slope Compensation
The MAX16812 uses an internal ramp generator for
slope compensation. The internal ramp signal resets at
the beginning of each cycle and slews at the rate pro-
grammed by the external capacitor connected at SLP
and an internal ISLP current source of 150µA. An inter-
nal attenuator attenuates the actual slope compensa-
tion signal by a factor of 0.2. Adjust the MAX16812
slew-rate capacitor by using the following equation:
where ISLP is the charging current in mA and CSLOPE is
the slope compensation capacitance on the SLP in µF,
and SR is the designed slope in mV/µs.
When using the MAX16812 for internal switching MOS-
FET duty cycles greater than 50%, the following condi-
tions must be met to avoid current-loop subharmonic
oscillations.
where RSRC is in mΩ, VIND_OFF is in volts, and L is in
µH. L is the inductor connected to the LX pin of the
internal switching MOSFET and VIND_OFF is the voltage
across the inductor during the off-time of the internal
MOSFET.
SR RV
LmV s
SRC IND OFF
. /
_
≥××05 µ
CI
SR
SLOPE SLP
. =×02
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
10 ______________________________________________________________________________________
Undervoltage Lockout
The MAX16812 features an adjustable UVLO through
the enable input (EN). Connect EN directly to IN to use
the 5V default UVLO. Connect EN to IN through a resis-
tive divider to ground to set the UVLO threshold. The
MAX16812 is enabled when VEN exceeds the 1.38V
(typ) threshold.
Calculate the EN UVLO resistor-divider values as fol-
lows (see Figure 2):
where RUV1 is in the 20kΩrange, VEN is the 1.38V (typ)
EN threshold voltage, and VUVLO is the desired input-
voltage UVLO threshold in volts. Due to the 100mV hys-
teresis of the UVLO threshold, capacitor CEN is
required to prevent chattering at the UVLO threshold
due to line impedance drops at power-up and during
dimming. If the undervoltage setting is very close to the
required minimum operating voltage, there can be
jumps in the voltage at IN while dimming. CEN should
be large enough to limit the ripple on EN to less than
100mV (EN hysteresis) under these conditions so that it
does not turn on and off due to the ripple on IN.
Soft-Start
The soft-start feature of the MAX16812 allows the LED
string current to ramp up in a controlled manner, thus
minimizing output-voltage overshoot. While the part is in
UVLO, CREF is discharged (Figure 3). Upon coming out
of UVLO, an internal current source starts charging CREF
during the soft-start cycle. Use the following equation to
calculate total soft-start time:
where IREF is 40µA, CREF is in µF, and tST is in sec-
onds. Operation begins when REF ramps above 0.6V.
Once the soft-start is complete, REF is regulated to
1.238V, the internal voltage reference.
Low-Side Internal
Switching MOSFET Driver Supply (L_REG)
L_REG is the regulated (5.2V) internal supply voltage
capable of delivering 20mA. L_REG provides power to
the gate drive of the internal switching power MOSFET.
VL_REG is referenced to AGND. Connect a 3.3µF
ceramic capacitor from L_REG to AGND.
High-Side Regulator (H_REG)
H_REG is a low-dropout linear regulator referenced to
LV. H_REG provides the gate drive for the external
n-channel dimming MOSFET and also powers up the
MAX16812’s LED current-sense circuitry. Bypass
H_REG to LV with a 1µF ceramic capacitor.
tC I
ST REF REF
.
=×
1 238
R R x V
V- V
UV1 UV2 EN
UVLO EN
=⎛
⎝
⎜
⎞
⎠
⎟
MAX16812
VIN
IN
AGND
EN
RUV2
RUV1
CEN
Figure 2. UVLO Threshold Setting
MAX16812
VIN
IN
AGND
REF
CREF
Figure 3. Soft-Start Setting
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
______________________________________________________________________________________ 11
High-Side Current-Sense Output (CS_OUT)
A high-side transconductance amplifier converts the
voltage across the LED current-sense resistor (RCS)
into an internal current output. This current flows
through an internal resistor connected to AGND. The
voltage gain for the LED current-sense signal is 4. The
amplified signal is then buffered and connected
through an internal switch to CS_OUT.
Internal Error Amplifier
The MAX16812 includes a built-in voltage-error amplifi-
er, which can be used to close the feedback loop. The
internal LED current-sense output signal is buffered
internally and then connected to CS_OUT through an
internal switch. CS_OUT is connected to the inverting
input (FB) pin of the error amplifier through a resistor.
See Figures 4 and 5. The reference voltage for the out-
put current is connected to REFI, the noninverting input
of the error amplifier. When the internal dimming signal
is low, COMP is disconnected from the output of the
error amplifier and CS_OUT is simultaneously discon-
nected from the buffered LED current-sense output sig-
nal (Figure 5). When the internal dimming signal is high,
the output of the op amp is connected to COMP and
CS_OUT is connected to the buffered LED current-
sense signal at the same time (Figure 4). This enables
the compensation capacitor to hold the charge when
the DIM signal has turned off the internal switching
MOSFET gate drive. To maintain the charge on the
compensation capacitors CCOMP1 and CCOMP2, the
capacitors should be of the low-leakage ceramic type.
When the internal dimming signal is enabled, the voltage
on the compensation capacitor forces the converter into
steady state almost instantaneously. The voltage on
COMP is subtracted from the internal slope compensa-
tion signal and is then connected to one of the inputs of
the PWM comparator. The PWM comparator input is of
the CMOS type with very low bias currents.
REFI
COMP
OUT
RCOMP2
RCOMP1
CCOMP1
CCOMP2
STATE A
X1
EA
Figure 4. Internal Error Amplifier Connection (Dimming Signal High)
REFI
COMP
OUT
STATE B
X1
EA
RCOMP2
RCOMP1
CCOMP1
CCOMP2
Figure 5. Internal Error Amplifier Connections (Dimming Signal Low)
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
12 ______________________________________________________________________________________
Analog Dimming
The MAX16812 offers analog dimming of the LED cur-
rent by allowing the application of an external voltage
at REFI. The output current is proportional to the volt-
age at REFI. Use a potentiometer from REF or directly
apply an external voltage source at REFI.
PWM Comparator
The PWM comparator uses the instantaneous switch
current, the error-amplifier output, and the slope com-
pensation to determine when the gate drive DRV to the
internal n-channel switching MOSFET turns off. In nor-
mal operation, gate drive DRV to the n-channel MOS-
FET turns off when:
ISW x RSRC ≥VCOMP - VOFFSET - VSCOMP
where ISW is the current through the internal n-channel
switching MOSFET, RSRC is the switch current-sense
resistor, VCOMP is the output voltage of the internal
amplifier, VOFFSET is the internal DC offset, which is a
VBE drop, and VSCOMP is the ramp function that starts
at zero and slews at the programmed slew rate (SR).
Internal Switching MOSFET Current Limit
The current-sense resistor (RSRC), connected between
the source of the internal MOSFET and ground, sets the
current limit. The SRC input has a voltage trip level
(VSRC) of 600mV for the cycle-by-cycle current limit. Use
the following equation to calculate the value of RSRC:
where ILXLIM is the peak current that flows through the
switching MOSFET at full load and low line. When the
voltage produced by this current (through the current-
sense resistor) exceeds the current-limit (ILIM) com-
parator threshold, the MOSFET driver (DRV) quickly
terminates the current on-cycle. The 200ns leading-
edge blanking circuit suppresses the leading-edge
spike on the current-sense waveform from appearing at
the current-limit comparator. There is also a hiccup
comparator (HICCUP) that limits the peak current in the
internal switch set at twice the peak limit setting.
Internal n-Channel
Switching MOSFET Driver (DRV)
L_REG provides power for the DRV output. Connect a
resistor from DRV to gate GT of the internal switching
MOSFET to control the switching MOSFET rise and fall
times, if necessary.
External Dimming
MOSFET Gate Drive (DGT)
DGT is the gate drive to the external dimming MOSFET
referenced to LV. H_REG provides the power to the
gate drive.
Overvoltage Protection
The overvoltage protection (OVP) comparator com-
pares the voltage at OV with a 1.238V (typ) internal ref-
erence. When the voltage at OV exceeds the internal
reference, the OVP comparator terminates PWM switch-
ing and no further energy is transferred to the load.
Connect OV to HV through a resistive voltage-divider to
ground to set the overvoltage threshold at the output.
Setting the Overvoltage Threshold
Connect OV to HV or to the high-side of the LEDs
through a resistive voltage-divider to set the overvolt-
age threshold at the output (Figure 6).
RV
I
SRC SRC
LXLIM
=
MAX16812
VLED+
AGND
OV
HV
ROV1
ROV2
MAX16812
VLED+
AGND
OV
ROV1
ROV2
Figure 6. OVP Setting
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
______________________________________________________________________________________ 13
The overvoltage protection (OVP) comparator com-
pares the voltage at OV with a 1.238V (typ) internal ref-
erence. Use the following equation to calculate resistor
values:
where VOV is the 1.238V OV threshold. Choose ROV1
and ROV2 to be reasonably high-value resistors to pre-
vent the discharge of filter capacitors. This prevents
degraded performance during dimming.
Internal Oscillator Switching Frequency
The oscillator switching frequency is programmed by a
resistor connected from RT to AGND. To program the
oscillator frequency above 125kHz, choose the appro-
priate resistor RT from the curves shown in the
Oscillator Frequency vs. RTgraph in the
Typical
Operating Characteristics
section.
PWM Dimming
PWM dimming can be achieved by driving DIM with an
analog voltage less than VREF. See Figure 7. An exter-
nal resistor on TGRM from L_REG in conjunction with
the ramp capacitor, CTGRM, from TGRM to AGND cre-
ates a sawtooth ramp that is compared with the DC
voltage on DIM. The output of the comparator is a pul-
sating dimming signal. The frequency fRAMP of the
sawtooth signal on TGRM is given by:
Use the following formula to calculate the voltage VDIM,
necessary for a given output duty cycle, D:
VDIM = D x 1.238V
where VDIM is the DC voltage applied to DIM in volts.
The DC voltage for DIM can also be created by con-
necting DIM to REF through a resistive voltage-divider.
Using the required dimming input voltage, VDIM, calcu-
late the resistor values for the divider string using the
following equation:
RDIM2 = [VDIM / (VREF - VDIM)] x RDIM1
where VREF is the voltage on REF.
PWM dimming can also be achieved by connecting
TGRM to a DC voltage less than VREF and applying the
PWM signal at DIM. The moment the internal dimming
signal goes low, gate drive DRV to the internal switching
MOSFET is turned off. The error amplifier goes to state B
(see the
Internal Error Amplifier
section and Figures 4
and 5). The peak current in the inductor prior to dis-
abling DRV is ILX. Gate drive DGT to the external dim-
ming MOSFET is held high. Then after a switchover
period, gate voltage VDGT on the external dimming
MOSFET is linearly controlled to reduce the LED current
to 0. The fall time of the LED current is controlled by an
internal timing circuit to 5µs for the MAX16812. During
this period, the gate (DRV) to the internal switching
MOSFET is enabled. After the fall time, the gate drive to
the external dimming MOSFET is turned off and the gate
drive to the internal switching MOSFET is still held high
after the switchover period. The peak current in the
inductor is controlled at ILX. Then after a time period of
20µs, the gate drive is disabled. The scope shots in
Figures 8–11 show the dimming waveforms.
fCR
RAMP TGRM TGRM
.
≅×
367
R R x
VV
V
OV1 OV2 OV_LIM OV
OV
=−
⎛
⎝
⎜
⎞
⎠
⎟
MAX16812
AGND
DIM
REF
TGRM
L_REG
RDIM1 RTGRM
CTGRM
RDIM2
Figure 7. PWM Dimming from REF
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
14 ______________________________________________________________________________________
When the DIM signal goes high, the LED current is
gradually increased to the programmed value. The rise
time of the LED current is controlled to 5µs for the
MAX16812 by controlling the voltage on DGT. After the
rise time, an internal sensing circuit monitors the volt-
age across the drain to the source of the external dim-
ming MOSFET. The LED current is now controlled at the
programmed value by a linear current regulating cir-
cuit. Once the voltage across the drain to source of the
dimming MOSFET drops below 0.5V, the reference for
the linear current regulating circuit is increased to 1.1
times the programmed value. The gate drive (DRV) to
the internal switching MOSFET is enabled and the error
amplifier is returned to state A (see the
Internal Error
Amplifier
section and Figures 4 and 5).
Fault Protection
The MAX16812 features built-in overvoltage protection
and thermal shutdown. Connect a resistive voltage-
divider between HV, OV, and AGND to program the over-
voltage protection. In the case of a short circuit across
the LED string, the temperature of the external dimming
MOSFET could exceed the maximum allowable junction
temperature. This is due to excess power dissipation in
the MOSFET. Use the fault protection circuit shown in
Figure 12 to protect the external dimming MOSFET.
Internal thermal shutdown in the MAX16812 safely turns
off the IC when the junction temperature exceeds
+165°C.
MAX16812 fig08
10μs/div
ILED
VOUT
VDRV 0V
2V/div
0A, 0V
100mA/div
10V/div
Figure 8. LED Current, Output Voltage, and DRV Waveforms
when DIM Signal Goes Low
MAX16812 fig09
10μs/div
VDIM
5V/div
0V
VDRV
2V/div
0A, 0V
ILED
100mA/div
Figure 9. LED Current, DIM Signal, and DRV Waveforms when
DIM Signal Goes Low
MAX16812 fig10
10μs/div
ILED
VOUT
VDRV 0V
2V/div
0A, 0V
100mA/div
10V/div
Figure 10. LED Current, Output Voltage, and DRV Waveforms
when DIM Signal Goes High
MAX16812 fig11
10μs/div
0V
VDRV
2V/div
0A, 0V
100mA/div
VDIM
5V/div
ILED
Figure 11. LED Current, DIM Signal, and DRV Waveforms when
DIM Signal Goes High
Inductor Selection
The minimum required inductance is a function of the
operating frequency, the input-to-output voltage differ-
ential and the peak-to-peak inductor current (ΔIL).
Higher ΔILallows for a lower inductor value while a
lower ΔILrequires a higher inductor value. A lower
inductor value minimizes size and cost, improves large-
signal transient response, but reduces efficiency due to
higher peak currents and higher peak-to-peak output
ripple voltage for the same output capacitor. On the
other hand, higher inductance increases efficiency by
reducing the ripple current, ΔIL. However, resistive
losses due to the extra turns can exceed the benefit
gained from lower ripple current levels, especially when
the inductance is increased without allowing for larger
inductor dimensions. A good compromise is to choose
ΔILequal to 30% of the full load current. The inductor
saturating current specification is also important to
avoid runaway current during output overload and con-
tinuous short-circuit conditions.
Buck Configuration: In a buck configuration (Figure
13), the average inductor current does not vary with the
input. The worst-case peak current occurs at the high-
est input voltage. In this case, the inductance, L, for
continuous conduction mode is given by:
where VINMAX is the maximum input voltage, fSW is the
switching frequency, and VOUT is the output voltage.
Boost Configuration: In the boost converter, the aver-
age inductor current varies with the input voltage and
the maximum average current occurs at the lowest
input voltage. For the boost converter, the average
inductor current is equal to the input current. In this
case, the inductance, L, is calculated as:
where VINMIN is the minimum input voltage, VOUT is the
output voltage, and fSW is the switching frequency. See
Figure 14.
Buck-Boost Configuration: In a buck-boost converter
(see the
Typical Application Circuit
), the average
inductor current is equal to the sum of the input current
and the LED current. In this case, the inductance, L, is:
where VINMIN is the minimum input voltage, VOUT is the
output voltage, and fSW is the switching frequency.
L Vx V
V V x f x I
OUT INMIN
OUT INMIN SW L
=+
()
Δ
L
V x V V
V x f x I
INMIN OUT INMIN
OSWL
=−
()
UT Δ
L
Vx V V
V x f x I
OUT INMAX OUT
INMAX SW L
=−
()
Δ
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
______________________________________________________________________________________ 15
MAX6501
GND
4.7μF
100kΩ
5.1V
ZENER
GND
VCC
TO L_REG PIN
OF MAX16812
TO EN PIN OF
MAX16812
VIN
TOVER
Figure 12. Dimming MOSFET Protection
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
16 ______________________________________________________________________________________
MAX16812
H_REG
EN
RT
TGRM
L_REG
HV LX LV DD DGT CS- CS+
RRT
RTGRM
CL_REG
CH_REG
CREF
ROV1
RCOMP1
RREF1
RREF2
RCOMP2
CCOMP1
CCOMP2
VOUT
ROV2
GT
DRV
SLP
COMP
OV SGND AGND REFI
REF FBCS_OUT
VIN
DIM
CTGRM
CIN
CSLP
RG
SRC
RSRC
DOUT RCS
COUT
IN
Figure 13. Buck Configuration
MAX16812
LV SRC
IN
RT
EN
TGRM
L_REG
CS- CS+ DGT DD H_REG HV LX
RCS
RRT
RTGRM
CH_REG
CL_REG
CREF
ROV1
RCOMP1
RREF1
RREF2
RCOMP2
CCOMP1
CCOMP2
VOUT
VIN
ROV2
GT
DRV
SLP
COMP
OV SGND AGND REFI
REF FBCS_OUT
VIN
CIN1
COUT
DIM
CTGRM
CSLP
DOUT
RSRC
RG
VOUT
Figure 14. Boost Configuration
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
______________________________________________________________________________________ 17
Output Capacitor
The function of the output capacitor is to reduce the
output ripple to acceptable levels. The ESR, ESL, and
the bulk capacitance of the output capacitor contribute
to the output ripple. In most of the applications, the out-
put ESR and ESL effects can be dramatically reduced
by using low-ESR ceramic capacitors. To reduce the
ESL effects, connect multiple ceramic capacitors in
parallel to achieve the required capacitance.
In a buck configuration, the output capacitance, COUT,
is calculated using the following equation:
where ΔVRis the maximum allowable output ripple.
In a boost configuration, the output capacitance, COUT,
is calculated as:
where COUT is the output capacitor.
In a buck-boost configuration, the output capacitance,
COUT is:
where VOUT is the voltage across the load and IOUT is
the output current.
Input Capacitor
An input capacitor connected between IN and ground
must be used when configuring the MAX16812 as a
buck converter. Use a low-ESR input capacitor that can
handle the maximum input RMS ripple current.
Calculate the maximum RMS ripple using the following
equation:
When using the MAX16812 in a boost or buck-boost
configuration, the input capacitor’s RMS current is low
and the input capacitance can be small. However, an
additional electrolytic capacitor may be required to pre-
vent oscillations due to line impedances.
I
IVV - V
V
IN(RMS)
OUT OUT INMIN OUT
INMIN
=×× ( )
C 2 V I
VV V f
OUT
OUT OUT
R OUT INMIN SW
≥××
×+ ×
( ) Δ
C
VV I
VV f
OUT
OUT INMIN OUT
R OUT SW
≥−××
××
( )
2
Δ
C VVV
VL Vf
OUT
INMAX OUT OUT
R INMAX SW
≥−×
××× ×
( )
Δ22
MAX16812
CS-
CS+
DGT
DD
H_REG
HV
LX
OV
SGND
AGND
REFI
FB
REF
CS_OUT
RREF2
CH_REG
L2
L1
RREF1
CSLP
COUT
CS
RG
RCOMP1
RCOMP2
ROV1
ROV2
RSRC
VOUT
RT
CL_REG
RCS
RTGRM
COMP
SLP
SRC
GT
DRV
CCOMP1
CCOMP2
DOUT
VOUT
VIN
CIN1
CTGRM
LV
VIN
IN
L_REG
TGRM
DIM
EN
RT
Figure 15. SEPIC Configuration
MAX16812
Layout Recommendations
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain cur-
rent often form high di/dt loops. Similarly, the drain of
the internal MOSFET connected to the LX pin presents
a dv/dt source. Keep all PCB traces carrying switching
currents as short as possible to minimize current loops.
Use ground planes for best results.
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer
board whenever possible for better noise immunity and
power dissipation. Follow these guidelines for good
PCB layout:
•Use a large copper plane under the MAX16812
package. Ensure that all heat-dissipating compo-
nents have adequate cooling. Connect the exposed
pad of the device to the ground plane.
•Isolate the power components and high-current
paths from sensitive analog circuitry.
•Keep the high-current paths short, especially at the
ground terminals. This practice is essential for stable,
jitter-free operation. Keep switching loops short.
•Connect AGND and SGND to a ground plane.
Ensure a low-impedance connection between all
ground points.
•Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs to enhance full-load efficiency.
•Ensure that the feedback connection to FB is short
and direct.
•Route high-speed switching nodes away from the
sensitive analog areas.
•To prevent discharge of the compensation capaci-
tors, CCOMP1 and CCOMP2, during the off-time of
the dimming cycle, ensure that the PCB area close
to these components has extremely low leakage.
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
18 ______________________________________________________________________________________
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
______________________________________________________________________________________ 19
MAX16812
LV
BUCK-BOOST CONFIGURATION
SRC
IN
RT
EN
TGRM
L_REG
CS- CS+ DGT DD H_REG HV LX
RCS
RT
RTGRM
CH_REG
CL_REG
ROV1
RCOMP1
RREF1
RREF2
CREF
RCOMP2
CCOMP1
CCOMP2
VOUT
ROV2
GT
DRV
SLP
COMP
OV SGND AGND REFI
REF FBCS_OUT
VIN
CIN1
COUT
DIM
CTGRM
CSLP
DOUT
RSRC
RG
VOUT
Typical Application Circuit
Chip Information
PROCESS: BiCMOS
TRANSISTOR COUNT: 8699
MAX16812
TQFN
+
TOP VIEW
26
27
25
24
10
9
11
COMP
REF
CS_OUT
AGND
EN
12
FB
LX
DRV
HV
LX
H_REG
LV
12
TGRM
4567
2021 19 17 16 15
DIM
RT
DGT
DD
SGND
L_REG
REFI GT
3
18
28 8
OV IN
*EP
*EP = EXPOSED PAD
SLP
23 13 CS+
SRC
22 14 CS-
SRC
Pin Configuration
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
20 ______________________________________________________________________________________
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
QFN THIN.EPS
PACKAGE OUTLINE,
21-0140 2
1
K
16, 20, 28, 32, 40L THIN QFN, 5x5x0.8mm
MAX16812
Integrated High-Voltage LED Driver
with Analog and PWM Dimming Control
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________
21
© 2007 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.
Heaney
Package Information (continued)
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
PACKAGE OUTLINE,
21-0140 2
2
K
16, 20, 28, 32, 40L THIN QFN, 5x5x0.8mm
