MAX16818ATI+T - Maxim

General Description

The MAX16818 pulse-width modulation (PWM) LED dri-

ver controller provides high-output-current capability in

a compact package with a minimum number of external

components. The MAX16818 is suitable for use in syn-

chronous and nonsynchronous step-down (buck)

topologies, as well as in boost, buck-boost, SEPIC, and

Cuk LED drivers. The MAX16818 is the first LED driver

controller that enables Maxim’s technology for fast LED

current transients of up to 20A/µs and 30kHz dimming

frequency.

This device utilizes average-current-mode control that

enables optimal use of MOSFETs with optimal charge

and on-resistance characteristics. This results in the

minimized need for external heatsinking even when

delivering up to 30A of LED current. True differential

sensing enables accurate control of the LED current. A

wide dimming range is easily implemented to accom-

modate an external PWM signal. An internal regulator

enables operation over a wide input voltage range:

4.75V to 5.5V or 7V to 28V and above with a simple

external biasing device. The wide switching frequency

range, up to 1.5MHz, allows for the use of small induc-

tors and capacitors.

The MAX16818 features a clock output with 180° phase

delay to control a second out-of-phase LED driver to

reduce input and output filter capacitors size or to mini-

mize ripple currents. The MAX16818 offers programma-

ble hiccup, overvoltage, and overtemperature protection.

The MAX16818ETI+ is rated for the extended tempera-

ture range (-40°C to +85°C) and the MAX16818ATI+ is

rated for the automotive temperature range (-40°C to

+125°C). This LED driver controller is available in a

lead-free, 0.8mm high, 5mm x 5mm 28-pin TQFN pack-

age with exposed paddle.

Applications

Front Projectors/Rear Projection TVs

Portable and Pocket Projectors

Automotive, Bus/Truck Exterior Lighting

LCD TVs and Display Backlight

Automotive Emergency Lighting and Signage

Features

♦High-Current LED Driver Controller IC, Up to 30A

Output Current

♦Average-Current-Mode Control

♦True-Differential Remote-Sense Input

♦4.75V to 5.5V or 7V to 28V Input Voltage Range

♦Programmable Switching Frequency or External

Synchronization from 125kHz to 1.5MHz

♦Clock Output for 180° Out-of-Phase Operation

♦Integrated 4A Gate Drivers

♦Output Overvoltage and Hiccup Mode

Overcurrent Protection

♦Thermal Shutdown

♦Thermally Enhanced 28-Pin Thin QFN Package

♦-40°C to +125°C Operating Temperature Range

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

________________________________________________________________

Maxim Integrated Products

19-0666; Rev 2; 3/09

For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,

or visit Maxim’s website at www.maxim-ic.com.

EVALUATION KIT

AVAILABLE

Pin Configuration appears at end of data sheet.

Ordering Information

Denotes a lead(Pb)-free/RoHS-compliant package.

EP = Exposed pad.

PART TEMP RANGE PIN-PACKAGE

MAX16818ATI+ -40°C to +125°C 28 TQFN-EP*

MAX16818ETI+ -40°C to +85°C 28 TQFN-EP*

HIGH-FREQUENCY

PULSE TRAIN

LED

7V TO 28V

CSP

NOTE: MAXIM TOPOLOGY

PGND

ILIM

OVI

CLP

MAX16818

Simplified Diagram

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

2 _______________________________________________________________________________________

ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS

(VCC = 5V, VDD = VCC, TA= TJ= TMIN to TMAX, unless otherwise noted. Typical specifications are at TA= +25°C.) (Note 1)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional

operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to

absolute maximum rating conditions for extended periods may affect device reliability.

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

SYSTEM SPECIFICATIONS

728

Input Voltage Range VIN Short IN and VCC together for 5V input

operation 4.75 5.50 V

Quiescent Supply Current IQEN = VCC or SGND, not switching 2.7 5.5 mA

LED CURRENT REGULATOR

No load, VIN = 4.75V to 5.5V, fSW = 500kHz 0.594 0.6 0.606

SENSE+ to SENSE- Accuracy

(Note 2) No load, VIN = 7V to 28V, fSW = 500kHz 0.594 0.6 0.606 V

Soft-Start Time tSS 1024 Clock

Cycles

STARTUP/INTERNAL REGULATOR

VCC Undervoltage Lockout UVLO VCC rising 4.1 4.3 4.5 V

VCC Undervoltage Hysteresis 200 mV

VCC Output Voltage VIN = 7V to 28V, ISOURCE = 0 to 60mA 4.85 5.1 5.30 V

MOSFET DRIVERS

Output Driver Impedance RON Low or high output, ISOURCE/SINK = 20mA 1.1 3.0 Ω

Output Driver Source/Sink Current IDH,IDL 4A

Nonoverlap Time tNO CDH/DL = 5nF 35 ns

OSCILLATOR

Switching Frequency Range 125 1500 kHz

Switching Frequency RT = 500kΩ121 125 129

Switching Frequency RT = 120kΩ495 521 547

Switching Frequency

fSW

RT = 39.9kΩ1515 1620 1725

kHz

120kΩ ≤ RT ≤ 500kΩ-5 +5

Switching Frequency Accuracy 40kΩ ≤ RT ≤ 120kΩ-8 +8 %

IN to SGND.............................................................-0.3V to +30V

BST to SGND..........................................................-0.3V to +35V

BST to LX..................................................................-0.3V to +6V

DH to LX .......................................-0.3V to [(VBST - VLX_) + 0.3V]

DL to PGND................................................-0.3V to (VDD + 0.3V)

VCC to SGND............................................................-0.3V to +6V

VCC, VDD to PGND ...................................................-0.3V to +6V

SGND to PGND .....................................................-0.3V to +0.3V

All Other Pins to SGND...............................-0.3V to (VCC + 0.3V)

Continuous Power Dissipation (TA= +70°C)

28-Pin TQFN (derate 34.5mW/°C above +70°C) .......2758mW

Operating Temperature Range

MAX16818ATI+..............................................-40°C to +125°C

MAX16818ETI+................................................-40°C to +85°C

Maximum Junction Temperature .....................................+150°C

Storage Temperature Range .............................-60°C to +150°C

Lead Temperature (soldering, 10s) .................................+300°C

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

_______________________________________________________________________________________ 3

ELECTRICAL CHARACTERISTICS (continued)

(VCC = 5V, VDD = VCC, TA= TJ= TMIN to TMAX, unless otherwise noted. Typical specifications are at TA= +25°C.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

CLKOUT Phase Shift φ_CLKOUT With respect to DH, fSW = 125kHz 180 D eg r ees

CLKOUT Output Low Level VCLKOUTL ISINK = 2mA 0.4 V

CLKOUT Output High Level VCLKOUTH ISOURCE = 2mA 4.5 V

SYNC Input-High Pulse Width tSYNC 200 ns

SYNC Input Clock High Threshold VSYNCH 2.0 V

SYNC Input Clock Low Threshold VSYNCL 0.4 V

SYNC Pullup Current ISYNC_OUT VRT/SYNC = 0V 250 750 µA

SYNC Power-Off Level VSYNC_OFF 0.4 V

INDUCTOR CURRENT LIMIT

Average Current-Limit Threshold VCL CSP to CSN 24.0 26.9 28.2 mV

Reverse Current-Limit Threshold VCLR CSP to CSN -3.2 -2.3 -0.1 mV

Cycle-by-Cycle Current Limit CSP to CSN 60 mV

Cycle-by-Cycle Overload VCSP to VCSN = 75mV 260 ns

Hiccup Divider Ratio LIM to VCM, no switching 0.547 0.558 0.565 V/V

Hiccup Reset Delay 200 ms

LIM Input Impedance LIM to SGND 55.9 kΩ

CURRENT-SENSE AMPLIFIER

CSP or CSN Input Resistance RCS 4kΩ

Common-Mode Range VCMR

(

)

VIN = 7V to 28V 0 5.5 V

Input Offset Voltage VOS

(

)

0.1 mV

Amplifier Gain AV(CS) 34.5 V/V

3dB Bandwidth f3dB 4 MHz

CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)

Transconductance gm550 µS

Open-Loop Gain AVOL

(

)

No load 50 dB

DIFFERENTIAL VOLTAGE AMPLIFIER FOR LED CURRENT (DIFF)

Common-Mode Voltage Range VCMR

(

DIFF

)

0 +1.0 V

DIFF Output Voltage VCM VSENSE+ = VSENSE- = 0V 0.6 V

Input Offset Voltage VOS

(

DIFF

)

-1 +1 mV

Amplifier Gain AV

(

DIFF

)

0.994 1 1.006 V/V

3dB Bandwidth f3dB CDIFF = 20pF 3 MHz

Minimum Output-Current Drive IOUT

(

DIFF

)

4mA

SENSE+ to SENSE- Input RVS VSENSE- = 0V 50 100 kΩ

V_IOUT AMPLIFIER

Gain-Bandwidth Product VV_IOUT = 2.0V 4 MHz

3dB Bandwidth VV_IOUT = 2.0V 1 MHz

Output Sink Current 30 µA

Output Source Current 90 µA

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

4 _______________________________________________________________________________________

Note 1: Specifications at TA= +25°C are 100% tested. Specifications over the temperature range are guaranteed by design.

Note 2: Does not include an error due to finite error amplifier gain. See the

Voltage-Error Amplifier (EAOUT)

section.

ELECTRICAL CHARACTERISTICS (continued)

(VCC = 5V, VDD = VCC, TA= TJ= TMIN to TMAX, unless otherwise noted. Typical specifications are at TA= +25°C.) (Note 1)

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

Maximum Load Capacitance 50 pF

V_IOUT Output to IOUT Transfer

Function RSENSE = 1mΩ, 100mV ≤ V_IOUT ≤ 5.5V 132.3 135 137.7 mV/A

Offset Voltage 1mV

VOLTAGE-ERROR AMPLIFIER (EAOUT)

Open-Loop Gain AVOLEA 70 dB

Unity-Gain Bandwidth fGBW 3 MHz

EAN Input Bias Current IB(EA) VEAN = 2.0V -0.2 +0.03 +0.2 µA

Error Amplifier Output Clamping

Voltage VCLAMP

(

)

With respect to VCM 883 930 976 mV

POWER-GOOD AND OVERVOLTAGE PROTECTION

PGOOD Trip Level VUV PGOOD goes low when VOUT is below this

threshold 87.5 90 92.5 %VOUT

PGOOD Output Low Level VPGLO ISINK = 4mA 0.4 V

PGOOD Output Leakage Current IPG PGOOD = VCC 1µA

OVI Trip Threshold OVPTH With respect to SGND 1.244 1.276 1.308 V

OVI Input Bias Current IOVI 0.2 µA

ENABLE INPUT

EN Input High Voltage VEN EN rising 2.437 2.5 2.562 V

EN Input Hysteresis 0.28 V

EN Pullup Current IEN 13.5 15 16.5 µA

THERMAL SHUTDOWN

Thermal Shutdown Temperature rising 150 °C

Thermal Shutdown Hysteresis 30 °C

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

_______________________________________________________________________________________ 5

SUPPLY CURRENT (IQ) vs. FREQUENCY

MAX16818 toc01

FREQUENCY (kHz)

SUPPLY CURRENT (mA)

13001100900700500300

100 1500

EXTERNAL CLOCK

NO DRIVER LOAD

VIN = 12V

VIN = 24V

VIN = 5V

SUPPLY CURRENT vs. TEMPERATURE

MAX16818 toc02

TEMPERATURE (°C)

SUPPLY CURRENT (mA)

603510-15

-40 85

VIN = 12V

fSW = 250kHz

CDL/CDH = 22nF

CURRENT-SENSE THRESHOLD

vs. OUTPUT VOLTAGE

MAX16818 toc03

VOUT (V)

(VCSP - VCSN) (mV)

4321

26.5

27.0

27.5

28.0

28.5

29.0

26.0

VIN = 12V

fSW = 250kHz

HICCUP CURRENT LIMIT vs. REXT

MAX16818 toc04

REXT (MΩ)

CURRENT LIMIT (A)

161284

23.5

24.0

24.5

25.0

25.5

26.0

23.0

020

VIN = 12V

fSW = 250kHz

R1 = 1mΩ

VOUT = 1.5V

VCC LOAD REGULATION

vs. INPUT VOLTAGE

MAX16818 toc05

VCC LOAD CURRENT (mA)

VCC (V)

125100755025

4.85

4.95

5.05

5.15

5.25

4.75

0 150

VIN = 12V

VIN = 5V

VIN = 24V

DRIVER RISE TIME

vs. DRIVER LOAD CAPACITANCE

MAX16818 toc06

CAPACITANCE (nF)

tR (ns)

2116116

100

VIN = 12V

fSW = 250kHz

DRIVER FALL TIME

vs. DRIVER LOAD CAPACITANCE

MAX16818 toc07

CAPACITANCE (nF)

tF (ns)

2116116

100

VIN = 12V

fSW = 250kHz

HIGH-SIDE DRIVER (DH) SINK

AND SOURCE CURRENT

MAX16818 toc08

2A/div

100ns/div

CLOAD = 22nF

VIN = 12V

LOW-SIDE DRIVER (DL) SINK

AND SOURCE CURRENT

MAX16818 toc09

3A/div

100ns/div

CLOAD = 22nF

VIN = 12V

Typical Operating Characteristics

(TA= +25°C, using Figure 5

unless otherwise noted.)

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

6 _______________________________________________________________________________________

HIGH-SIDE DRIVER (DH) RISE TIME

MAX16818 toc10

2V/div

40ns/div

CLOAD = 22nF

VIN = 12V

HIGH-SIDE DRIVER (DH) FALL TIME

MAX16818 toc11

2V/div

40ns/div

CLOAD = 22nF

VIN = 12V

LOW-SIDE DRIVER (DL) RISE TIME

MAX16818 toc12

2V/div

40ns/div

CLOAD = 22nF

VIN = 12V

LOW-SIDE DRIVER (DL) FALL TIME

MAX16818 toc13

2V/div

40ns/div

CLOAD = 22nF

VIN = 12V

FREQUENCY vs. RT

MAX16818 toc14

(

kΩ

)

fSW (kHz)

470

430

390

350

310

270

230

190

150

110

1000

10,000

100

30 510

VIN = 12V

FREQUENCY vs. TEMPERATURE

MAX16818 toc15

TEMPERATURE (°C)

fSW (kHz)

603510-15

242

244

246

248

250

252

254

256

258

260

240

-40 85

VIN = 12V

SYNC, CLKOUT, AND LX WAVEFORM

MAX16818 toc16

1µs/div

CLKOUT

5V/div

VIN = 12V

fSW = 250kHz

10V/div

SYNC

5V/div

Typical Operating Characteristics (continued)

(TA= +25°C, using Figure 5

unless otherwise noted.)

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

_______________________________________________________________________________________ 7

Pin Description

PIN NAME FUNCTION

1 PGND Power-Supply Ground

2, 7 N.C. No Connection. Not internally connected.

3 DL Low-Side Gate Driver Output

4 BST Boost Flying Capacitor Connection. Reservoir capacitor connection for the high-side MOSFET driver

supply. Connect a ceramic capacitor between BST and LX.

5 LX Source connection for the high-side MOSFET.

6 DH High-Side Gate Driver Output. Drives the gate of the high-side MOSFET.

8, 22, 25 SGND Signal Ground. Ground connection for the internal control circuitry. Connect SGND and PGND

together at one point near the IC.

9 CLKOUT Oscillator Output. Rising edge of CLKOUT is phase-shifted from the rising edge of DH by 180°.

10 PGOOD Power-Good Output

11 EN

Output Enable. Drive high or leave unconnected for normal operation. Drive low to shut down the

power drivers. EN has an internal 15µA pullup current. Connect a capacitor from EN to SGND to

program the hiccup-mode duty cycle.

12 RT/SYNC

Switching Frequency Programming and Chip-Enable Input. Connect a resistor from RT/SYNC to

SGND to set the internal oscillator frequency. Drive RT/SYNC to synchronize the switching frequency

with external clock.

13 V_IOUT Voltage Source Output Proportional to the Inductor Current. The voltage at V_IOUT = 135 x ILED x RS.

14 LIM Current-Limit Setting Input. Connect a resistor from LIM to SGND to set the hiccup current-limit

threshold. Connect a capacitor from LIM to SGND to ignore short output overcurrent pulses.

15 OVI

Overvoltage Protection. Connect OVI to DIFF. When OVI exceeds 12.7% above the programmed

output voltage, DH is latched low and DL is latched high. Toggle EN or recycle the input power to

reset the latch.

16 CLP Current-Error Amplifier Output. Compensate the current loop by connecting an RC network to ground.

17 EAOUT Voltage-Error Amplifier Output. Connect to the external compensation network.

18 EAN Voltage-Error Amplifier Inverting Input

19 DIFF Differential Remote-Sense Amplifier Output. DIFF is the output of a precision unity-gain amplifier

whose inputs are SENSE+ and SENSE-.

20 CSN Current-Sense Differential Amplifier Negative Input. The differential voltage between CSN and CSP is

amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

8 _______________________________________________________________________________________

Pin Description (continued)

PIN NAME FUNCTION

21 CSP Current-Sense Differential Amplifier Positive Input. The differential voltage between CSN and CSP is

amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.

23 SENSE- Differential LED Current-Sensing Negative Input. SENSE- is used to sense the LED current. Connect

SENSE- to the negative side of the LED current-sense resistor.

24 SENSE+ Differential LED Current-Sensing Positive Input. SENSE+ is used to sense the LED current. Connect

SENSE+ to the positive side of the LED current-sense resistor.

26 IN Supply Voltage Connection. Connect IN to VCC for a +5V system.

27 VCC Internal +5V Regulator Output. VCC is derived from the IN voltage. Bypass VCC to SGND with 4.7µF

and 0.1µF ceramic capacitors.

28 VDD

Supply Voltage for Low-Side and High-Side Drivers. Connect a parallel combination of 0.1µF and 1µF

ceramic capacitors to PGND and a 1Ω resistor to VCC to filter out the high peak currents of the driver

from internal circuitry.

—EP

Exposed Paddle. Connect the exposed paddle to a copper pad (SGND) to improve power

dissipation.

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

_______________________________________________________________________________________ 9

1314

22 23 24 25 26 27 28

VLED

C1 LED

STRING

VIN

7V TO 28V

C10

R12

R4 R5

VCC

ON/OFF

R11

R10

PGND

N.C.

BST

LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND

SGND SENSE- SENSE+ SGND IN V

OVI

CLP

EAOUT

EAN

DIFF

CSN

CSP

VCC

MAX16818

VLED

C6 C5 C4

Typical Application Circuits

Figure 1. Typical Application Circuit for a Boost LED Driver (Nonsynchronous)

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

10 ______________________________________________________________________________________

1314

22 23 24 25 26 27 28

VLED

LED

STRING

1 TO 6

LEDS

VIN

7V TO 28V

C10

R12

R4 R5

VCC

ON/OFF

R11

R10

PGND

N.C.

BST

SGND SENSE- SENSE+ SGND IN V

OVI

CLP

EAOUT

EAN

DIFF

CSN

CSP

VCC

MAX16818

VLED

C6 C5 C4

VCC

RS+

RS- OUT

MAX4073T

LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND

Typical Application Circuits (continued)

Figure 2. Typical Application Circuit for an Input-Referred Buck-Boost LED Driver (Input: 7V to 28V, Output: 1 to 6 LEDs in Series)

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

______________________________________________________________________________________ 11

1314

22 23 24 25 26 27 28

VLED

C2 LED

STRING

VIN

7V TO 28V

C11

C10

R12

R4 R5

VCC

ON/OFF

R11

R10

PGND

N.C.

BST

SGND SENSE- SENSE+ SGND IN V

OVI

CLP

EAOUT

EAN

DIFF

CSN

CSP

VCC

MAX16818

VLED

C7 C6 C5

LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND

Typical Application Circuits (continued)

Figure 3. Typical Application Circuit for a SEPIC LED Driver

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

12 ______________________________________________________________________________________

1314

22 23 24 25 26 27 28

VLED

LED

STRING

VIN

7V TO 18V

C11

C10

R12

R4 R5

VCC

ON/OFF

R11

R10

PGND

N.C.

BST

SGND SENSE- SENSE+ SGND IN V

OVI

CLP

EAOUT

EAN

DIFF

CSN

CSP

VCC

MAX16818

VLED

C7 C6 C5

LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND

Typical Application Circuits (continued)

Figure 4. Application Circuit for a Ground-Referred Buck-Boost LED Driver

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

______________________________________________________________________________________ 13

1314

22 23 24 25 26 27 28

LED

STRING

VIN

7V TO 28V

C11

C10

R10

VCC

ON/OFF

PGND

N.C.

BST

LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND

SGND SENSE- SENSE+ SGND IN V

OVI

CLP

EAOUT

EAN

DIFF

CSN

CSP

VCC

MAX16818

C7 C6 C5

Typical Application Circuits (continued)

Figure 5. Application Circuit for a Buck LED Driver

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

14 ______________________________________________________________________________________

Functional Diagram

2 x fS (V/s)

RAMP

RT/SYNC

CSP

CSN

SGND

SENSE-

SENSE+

CLP

LIM

VDD

BST

PGND

PGOOD

AV = 34.5

AV = 4

100kΩ

126.7kΩ

PWM

COMPARATOR

0.5V x VCC

TO INTERNAL

CIRCUITS HICCUP MODE

CURRENT LIMIT

V_IOUT

gm = 500µS

DIFF

CLKOUT

CLK

CPWM

CEA

VCLAMP

HIGH

VCLAMP

LOW

VCC

0.1 x VREF

+0.6V

VREF = 0.6V

VCM (0.6V)

OVP COMP

0.12 x VREF

LATCH

RAMP

GENERATOR

SOFT-

START

OSCILLATOR

CLEAR ON UVLO RESET OR

ENABLE LOW

OVP LATCH

DIFF

AMP

EAN

EAOUT

OVI

VEA

ERROR AMP

0.5 x VCLAMP

VCM

VCC

UVLO

POR

TEMP SENSOR

LDO

REGULATOR

MAX16818

Figure 6. MAX16818 Functional Diagram

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

______________________________________________________________________________________ 15

Detailed Description

The MAX16818 is a high-performance average-current-

mode PWM controller for high-power, high-brightness

LEDs (HBLEDs). Average current-mode control is the

ideal method for driving HBLEDs. This technique offers

inherently stable operation, reduces component derat-

ing and size by accurately controlling the inductor cur-

rent. The device achieves high efficiency at high

current (up to 30A) with a minimum number of external

components. The high- and low-side drivers source

and sink up to 4A for lower switching losses while dri-

ving high-gate-charge MOSFETs. The MAX16818’s

CLKOUT output is 180° out-of-phase with respect to the

high-side driver. CLKOUT drives a second MAX16818

LED driver out of phase, reducing the input-capacitor

ripple current.

The MAX16818 consists of an inner average current loop

representing inductor current and an outer voltage loop

voltage-error amplifier (VEA) that directly controls LED

current. The combined action of the two loops results in

a tightly regulated LED current. The inductor current is

sensed across a current-sense resistor. The differential

amplifier senses LED current through a sense resistor in

series with the LEDs and the resulting sensed voltage is

compared against an internal 0.6V reference at the error-

amplifier input. The MAX16818 will adjust the LED cur-

rent to within 1% accuracy to maintain emitted spectrum

of the light in HBLEDs.

IN, VCC, and VDD

The MAX16818 accepts either a 4.75V to 5.5V or 7V to

28V input voltage range. All internal control circuitry

operates from an internally regulated nominal voltage of

5V (VCC). For input voltages of 7V or greater, the inter-

nal VCC regulator steps the voltage down to 5V. The

VCC output voltage is a regulated 5V output capable of

sourcing up to 60mA. Bypass the VCC to SGND with

4.7µF and 0.1µF low-ESR ceramic capacitors for high-

frequency noise rejection and stable operation.

The MAX16818 uses VDD to power the low-side and

high-side drivers. Isolate VDD from VCC with a 1Ωresis-

tor and put a 1µF capacitor in parallel with a 0.1µF

capacitor to ground to prevent high-current noise spikes

created by the driver from disrupting internal circuitry.

The TQFN is a thermally enhanced package and can

dissipate up to 2.7W. The high-power packages allow

the high-frequency, high-current converter to operate

from a 12V or 24V bus. Calculate power dissipation in

the MAX16818 as a product of the input voltage and the

total VCC regulator output current (ICC). ICC includes qui-

escent current (IQ) and gate-drive current (IDD):

PD= VIN x ICC

ICC = IQ+ [fSW x (QG1 + QG2)]

where QG1 and QG2 are the total gate charge of the

low-side and high-side external MOSFETs at VGATE =

5V, IQis 3.5mA (typ), and fSW is the switching frequen-

cy of the converter.

Undervoltage Lockout (UVLO)

The MAX16818 includes an undervoltage lockout with

hysteresis and a power-on-reset circuit for converter

turn-on. The UVLO rising threshold is internally set at

4.35V with a 200mV hysteresis. Hysteresis at UVLO

eliminates chattering during startup.

Most of the internal circuitry, including the oscillator,

turns on when the input voltage reaches 4V. The

MAX16818 draws up to 3.5mA of current before the

input voltage reaches the UVLO threshold.

Soft-Start

The MAX16818 has an internal digital soft-start for a

monotonic, glitch-free rise of the output current. Soft-

start is achieved by the controlled rise of the error

amplifier dominant input in steps using a 5-bit counter

and a 5-bit DAC. The soft-start DAC generates a linear

ramp from 0 to 0.7V. This voltage is applied to the error

amplifier at a third (noninverting) input. As long as the

soft-start voltage is lower than the reference voltage,

the system converges to that lower reference value.

Once the soft-start DAC output reaches 0.6V, the refer-

ence takes over and the DAC output continues to climb

to 0.7V, assuring that it does not interfere with the refer-

ence voltage.

Internal Oscillator

The internal oscillator generates a clock with the fre-

quency proportional to the inverse of RT. The oscillator

frequency is adjustable from 125kHz to 1.5MHz with

better than 8% accuracy using a single resistor con-

nected from RT/SYNC to SGND. The frequency accura-

cy avoids the over-design, size, and cost of passive

filter components like inductors and capacitors. Use

the following equation to calculate the oscillator fre-

quency:

For 120kΩ≤RT≤500kΩ:

For 40kΩ≤RT≤120kΩ:

TSW

=640 10

TSW

=625 10

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

16 ______________________________________________________________________________________

The oscillator also generates a 2VP-P voltage-ramp sig-

nal for the PWM comparator and a 180° out-of-phase

clock signal for CLKOUT to drive a second LED regula-

tor out-of-phase.

Synchronization

The MAX16818 can be easily synchronized by connect-

ing an external clock to RT/SYNC. If an external clock is

present, then the internal oscillator is disabled and the

external clock is used to run the device. If the external

clock is removed, the absence of clock for 32µs is

detected and the circuit starts switching from the inter-

nal oscillator. Pulling RT/SYNC to ground for at least

50µs disables the converter. Use an open-collector

transistor to synchronize the MAX16818 with the exter-

nal system clock.

Control Loop

The MAX16818 uses an average-current-mode control

scheme to regulate the output current (Figure 7). The

main control loop consists of an inner current loop for

controlling the inductor current and an outer current

loop for regulating the LED current. The inner current

loop absorbs the inductor pole reducing the order of the

outer current loop to that of a single-pole system. The

current loop consists of a current-sense resistor (RS), a

current-sense amplifier (CA), a current-error amplifier

(CEA), an oscillator providing the carrier ramp, and a

PWM comparator (CPWM) (Figure 7). The precision CA

amplifies the sense voltage across RSby a factor of

34.5. The inverting input to the CEA senses the CA out-

put. The CEA output is the difference between the volt-

age-error amplifier output (EAOUT) and the amplified

voltage from the CA. The RC compensation network

connected to CLP provides external frequency compen-

sation for the CEA. The start of every clock cycle

enables the high-side drivers and initiates a PWM on-

cycle. Comparator CPWM compares the output voltage

from the CEA with a 0V to 2V ramp from the oscillator.

The PWM on-cycle terminates when the ramp voltage

exceeds the error voltage. Compensation for the outer

LED current loop varies based upon the topology.

The MAX16818 outer LED current control loop consists

of the differential amplifier (DIFF AMP), reference volt-

age, and VEA. The unity-gain differential amplifier pro-

vides true differential remote sensing of the voltage

across the LED current set resistor, RLS. The differential

amplifier output connects to the inverting input (EAN) of

the VEA. The DIFF AMP is bypassed and the inverting

input is available to the pin for direct feedback. The

noninverting input of the VEA is internally connected to

an internal precision reference voltage, set to 0.6V. The

VEA controls the inner current loop (Figure 6). A feed-

back network compensates the outer loop using the

EAOUT and EAIN pins.

DRIVE

Z COMP

CPWM

CEA

CLP

VREF + VCM = 1.2V

CSPCSN

CCFF

VIN

COUT

RLS

LED

STRING

CCF RCF

VEA

600mV

SENSE+

EAOUT

SENSE-

EAN

DIFF

MAX16818

DIFF

AMP

Figure 7. MAX16818 Control Loop

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

______________________________________________________________________________________ 17

Inductor Current-Sense Amplifier

The differential current-sense amplifier (CA) provides a

DC gain of 34.5. The maximum input offset voltage of

the current-sense amplifier is 1mV and the common-

mode voltage range is 0 to 5.5V (IN = 7V to 28V). The

current-sense amplifier senses the voltage across a

current-sense resistor. The maximum common-mode

voltage is 3.6V when VIN = 5V.

Inductor Peak-Current Comparator

The peak-current comparator provides a path for fast

cycle-by-cycle current limit during extreme fault condi-

tions, such as an inductor malfunction (Figure 8). Note

the average current-limit threshold of 26.9mV still limits

the output current during short-circuit conditions. To

prevent inductor saturation, select an inductor with a

saturation current specification greater than the average

current limit. Proper inductor selection ensures that only

the extreme conditions trip the peak-current comparator,

such as an inductor with a shorted turn. The 60mV

threshold for triggering the peak-current limit is twice the

full-scale average current-limit voltage threshold. The

peak-current comparator has only a 260ns delay.

Current-Error Amplifier

(For Inductor Currents)

The MAX16818 has a transconductance current-error

amplifier (CEA) with a typical gmof 550µS and 320µA

output sink- and source-current capability. The current-

error amplifier output CLP serves as the inverting input

to the PWM comparator. CLP is externally accessible to

provide frequency compensation for the inner current

loops (Figure 7). Compensate (CEA) so the inductor

current negative slope, which becomes the positive

slope to the inverting input of the PWM comparator, is

less than the slope of the internally generated voltage

ramp (see the

Compensation

section).

PWM Comparator and R-S Flip-Flop

The PWM comparator (CPWM) sets the duty cycle for

each cycle by comparing the output of the current-error

amplifier to a 2VP-P ramp. At the start of each clock

cycle, an R-S flip-flop resets and the high-side driver

(DH) goes high. The comparator sets the flip-flop as

soon as the ramp voltage exceeds the CLP voltage,

thus terminating the on-cycle (Figure 8).

Figure 8. MAX16818 Phase Circuit

2 x fS (V/s)

RAMP

CLK

CSP

CSN

EAN

EAOUT

SHDN

CLP

VDD

BST

PGND

AV = 34.5

PEAK-CURRENT

COMPARATOR

60mV

gm = 550µS

CPWM

SET

CLR

CEA

MAX16818

VEA

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

18 ______________________________________________________________________________________

Differential Amplifier

The DIFF AMP facilitates remote sensing at the load

(Figure 7). It provides true differential LED current

(through the RLS sense resistor) sensing while rejecting

the common-mode voltage errors due to high-current

ground paths. The VEA provides the difference

between the differential amplifier output (DIFF) and the

desired LED current-sense voltage. The differential

amplifier has a bandwidth of 3MHz. The difference

between SENSE+ and SENSE- is regulated to 0.6V.

Connect SENSE+ to the positive side of the LED current-

sense resistor and SENSE- to the negative side of the

LED current-sense resistor (which is often PGND).

MOSFET Gate Drivers (DH, DL)

The high-side (DH) and low-side (DL) drivers drive the

gates of external n-channel MOSFETs (Figures 1–5).

The drivers’ 4A peak sink- and source-current capabili-

ty provides ample drive for the fast rise and fall times of

the switching MOSFETs. Faster rise and fall times result

in reduced cross-conduction losses. Due to physical

realities, extremely low gate charges and RDS(ON)

resistance of MOSFETs are typically exclusive of each

other. MOSFETs with very low RDS(ON) will have a high-

er gate charge and vice versa. Choosing the high-side

MOSFET (Q1) becomes a trade-off between these two

attributes. Applications where the input voltage is much

higher than the output voltage result in a low duty cycle

where conduction losses are less important than

switching losses. In this case, choose a MOSFET with

very low gate charge and a moderate RDS(ON).

Conversely, for applications where the output voltage is

near the input voltage resulting in duty cycles much

greater than 50%, the RDS(ON) losses become at least

equal, or even more important than the switching losses.

In this case, choose a MOSFET with very low RDS(ON)

and moderate gate charge. Finally, for the applications

where the duty cycle is near 50%, the two loss compo-

nents are nearly equal, and a balanced MOSFET with

moderate gate charge and RDS(ON) work best.

In a buck topology, the low-side MOSFET (Q2) typically

operates in a zero voltage switching mode, thus it does

not have switching losses. Choose a MOSFET with very

low RDS(ON) and moderate gate charge.

Size both the high-side and low-side MOSFETs to han-

dle the peak and RMS currents during overload condi-

tions. The driver block also includes a logic circuit that

provides an adaptive nonoverlap time to prevent shoot-

through currents during transition. The typical nonover-

lap time between the high-side and low-side MOSFETs

is 35ns.

BST

The MAX16818 uses VDD to power the low- and high-

side MOSFET drivers. The high-side driver derives its

power through a bootstrap capacitor and VDD supplies

power internally to the low-side driver. Connect a

0.47µF low-ESR ceramic capacitor between BST and

LX. Connect a Schottky rectifier from BST to VDD. Keep

the loop formed by the boost capacitor, rectifier, and IC

small on the PCB.

Protection

The MAX16818 includes output overvoltage protection

(OVP). During fault conditions when the load goes to

high impedance (opens), the controller attempts to

maintain LED current. The OVP protection disables the

MAX16818 whenever the voltage exceeds the thresh-

old, protecting the external circuits from undesirable

voltages.

Current Limit

The VEA output is clamped to 930mV with respect to

the common-mode voltage (VCM). Average-current-

mode control has the ability to limit the average current

sourced by the converter during a fault condition. When

a fault condition occurs, the VEA output clamps to

930mV with respect to the common-mode voltage

(0.6V) to limit the maximum current sourced by the con-

verter to ILIMIT = 26.9mV / RS. The hiccup current limit

overrides the average current limit. The MAX16818

includes hiccup current-limit protection to reduce the

power dissipation during a fault condition. The hiccup

current-limit circuit derives inductor current information

from the output of the current amplifier. This signal is

compared against one half of VCLAMP(EA). With no

resistor connected from the LIM pin to ground, the hic-

cup current limit is set at 90% of the full-load average

current limit. Use REXT to increase the hiccup current

limit from 90% to 100% of the full load average limit.

The hiccup current limit can be disabled by connecting

LIM to SGND. In this case, the circuit follows the aver-

age current-limit action during overload conditions.

Overvoltage Protection

The OVP comparator compares the OVI input to the

overvoltage threshold. A detected overvoltage event

latches the comparator output forcing the power stage

into the OVP state. In the OVP state, the high-side

MOSFET turns off and the low-side MOSFET latches on.

Connect OVI to the center tap of a resistor-divider from

VLED to SGND. In this case, the center tap is compared

against 1.276V. Add an RC delay to reduce the sensitivity

of the overvoltage circuit and avoid nuisance tripping of

the converter. Disable the overvoltage function by con-

necting OVI to SGND.

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

______________________________________________________________________________________ 19

Applications Information

Application Circuit Descriptions

This section provides some detail regarding the appli-

cation circuits in the

Simplified Diagram

and Figures

1–5. The discussion includes some description of the

topology as well as basic attributes.

High-Frequency LED Current Pulser

The

Simplified Diagram

shows the MAX16818 providing

high-frequency, high-current pulses to the LEDs. The

basic topology must be a buck, since the inductor

always connects to the load in that configuration (in all

other topologies, the inductor disconnects from the

load at one time or another). The design minimizes the

current ripple by oversizing the inductor, which allows

for a very small (0.01µF) output capacitor. When MOS-

FET Q3 turns on, it diverts the current around the LEDs

at a very fast rate. Q3 also discharges the output

capacitor, but since the capacitor is so small, it does

not stress the MOSFET. Resistor R1 senses the LED/Q3

current and there is no reaction to the short that Q3

places across the LEDs. This design is superior in that

it does not attempt to actually change the inductor cur-

rent at high frequencies and yet the current in the LEDs

varies from zero to full in very small periods of time. The

efficiency of this technique is very high. Q3 must be

able to dissipate the LED current applied to its RDS(ON)

at some maximum duty cycle. If the circuit needs to

control extremely high currents, use paralleled

MOSFETs. PGOOD is low during LED pulsed-current

operation.

Boost LED Driver

In Figure 1, the external components configure the

MAX16818 as a boost converter. The circuit applies the

input voltage to the inductor during the on-time, and

then during the off-time the inductor, which is in series

with the input capacitor, charges the output capacitor.

Because of the series connection between the input

voltage and the inductor, the output voltage can never

go lower than the input voltage. The design is nonsyn-

chronous, and since the current-sense resistor con-

nects to ground, the power supply can go to any output

voltage (above the input) as long as the components are

rated appropriately. R2 again provides the sense voltage

the MAX16818 uses to regulate the LED current.

Input-Referenced LED Driver

The circuit in Figure 2 shows a step-up/step-down reg-

ulator. It is similar to the boost converter in Figure 1 in

that the inductor is connected to the input and the

MOSFET is essentially connected to ground. However,

rather than going from the output to ground, the LEDs

span from the output to the input. This effectively

removes the boost-only restriction of the regulator in

Figure 1, allowing the voltage across the LEDs to be

greater than or less than the input voltage. LED current

sensing is not ground-referenced, so a high-side cur-

rent-sense amplifier is used to measure current.

SEPIC LED Driver

Figure 3 shows the MAX16818 configured as a SEPIC

LED driver. While buck topologies require the output to

be lesser than the input, and boost topologies require

the output to be greater than the input, a SEPIC topolo-

gy allows the output voltage to be greater than, equal

to, or less than the input. In a SEPIC topology, the volt-

age across C1 is the same as the input voltage, and L1

and L2 are the same inductance. Therefore, when Q1

conducts (on-time), both inductors ramp up current at

the same rate. The output capacitor supports the output

voltage during this time. During the off-time, L1 current

recharges C1 and combines with L2 to provide current

to recharge C2 and supply the load current. Since the

voltage waveform across L1 and L2 are exactly the

same, it is possible to wind both inductors on the same

core (a coupled inductor). Although voltages on L1 and

L2 are the same, RMS currents can be quite different

so the windings may have a different gauge wire.

Because of the dual inductors and segmented energy

transfer, the efficiency of a SEPIC converter is some-

what lower than standard bucks or boosts. As in the

boost driver, the current-sense resistor connects to

ground, allowing the output voltage of the LED driver to

exceed the rated maximum voltage of the MAX16818.

Ground-Referenced Buck/Boost LED Driver

Figure 4 depicts a buck/boost topology. During the on-

time with this circuit, the current flows from the input

capacitor, through Q1, L1, and Q3 and back to the

input capacitor. During the off-time, current flows up

through Q2, L1, D1, and to the output capacitor C1.

This topology resembles a boost in that the inductor sits

between the input and ground during the on-time.

However, during the off-time the inductor resides

between ground and the output capacitor (instead of

between the input and output capacitors in boost

topologies), so the output voltage can be any voltage

less than, equal to, or greater than the input voltage. As

compared to the SEPIC topology, the buck/boost does

not require two inductors or a series capacitor, but it

does require two additional MOSFETs.

Buck Driver with Synchronous Rectification

In Figure 5, the input voltage can go from 7V to 28V and,

because of the ground-based current-sense resistor, the

output voltage can be as high as the input. The synchro-

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

20 ______________________________________________________________________________________

nous MOSFET keeps the power dissipation to a minimum,

especially when the input voltage is large when com-

pared to the voltage on the LED string. It is important to

keep the current-sense resistor, R1, inside the LC loop,

so that ripple current is available. To regulate the LED

current, R2 creates a voltage that the differential amplifier

compares to 0.6V. If power dissipation is a problem in R2,

add a noninverting amplifier and reduce the value of the

sense resistor accordingly.

Inductor Selection

The switching frequencies, peak inductor current, and

allowable ripple at the output determine the value and

size of the inductor. Selecting higher switching frequen-

cies reduces the inductance requirement, but at the

cost of lower efficiency. The charge/discharge cycle of

the gate and drain capacitances in the switching

MOSFETs create switching losses. The situation wors-

ens at higher input voltages, since switching losses are

proportional to the square of the input voltage. The

MAX16818 can operate up to 1.5MHz, however for

VIN > +12V, use lower switching frequencies to limit the

switching losses.

The following discussion is for buck or continuous

boost-mode topologies. Discontinuous boost, buck-

boost, and SEPIC topologies are quite different in

regards to component selection.

Use the following equations to determine the minimum

inductance value:

Buck regulators:

Boost regulators:

where VLED is the total voltage across the LED string.

As a first approximation choose the ripple current, ∆IL,

equal to approximately 40% of the output current.

Higher ripple current allows for smaller inductors, but it

also increases the output capacitance for a given volt-

age ripple requirement. Conversely, lower ripple cur-

rent increases the inductance value, but allows the

output capacitor to reduce in size. This trade-off can be

altered once standard inductance and capacitance val-

ues are chosen. Choose inductors from the standard

surface-mount inductor series available from various

manufacturers.

For example, for a buck regulator and 2 LEDs in series,

calculate the minimum inductance at VIN(MAX) = 13.2V,

VLED = 7.8V, ∆IL= 400mA, and fSW = 330kHz:

Buck regulators:

For a boost regulator with four LEDs in series, calculate

the minimum inductance at VIN(MAX) = 13.2V, VLED =

15.6V, ∆IL=400mA, and fSW = 330kHz:

Boost regulators:

The average-current-mode control feature of the

MAX16818 limits the maximum peak inductor current

and prevents the inductor from saturating. Choose an

inductor with a saturating current greater than the

worst-case peak inductor current. Use the following

equation to determine the worst-case inductor current:

where RSis the inductor sense resistor and VCL =

0.0282V.

Switching MOSFETs

When choosing a MOSFET for voltage regulators, con-

sider the total gate charge, RDS(ON), power dissipation,

and package thermal impedance. The product of the

MOSFET gate charge and on-resistance is a figure of

merit, with a lower number signifying better perfor-

mance. Choose MOSFETs optimized for high-frequency

switching applications.

The average current from the MAX16818 gate-drive

output is proportional to the total capacitance it drives

at DH and DL. The power dissipated in the MAX16818

is proportional to the input voltage and the average

drive current. See the

IN, V

, and V

section to

determine the maximum total gate charge allowed from

the combined driver outputs. The gate-charge and

drain-capacitance (CV2) loss, the cross-conduction loss

in the upper MOSFET due to finite rise/fall times, and

the I2R loss due to RMS current in the MOSFET

RDS(ON) account for the total losses in the MOSFET.

ILPEAK CL

∆

xkx H

MIN

(. .) .

. . .=−=µ

15 6 13 2 13 2

15 6 330 0 4 15 3

xkx H

MIN

(. .) .

. . .=−=µ

13 2 7 8 7 8

13 2 330 0 4 24 2

LVV xV

VxfxI

MIN LED INMAX INMAX

LED SW L

()

=−

∆

LVVxV

VxfxI

MIN INMAX LED LED

INMAX SW L

( )

=−

∆

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

______________________________________________________________________________________ 21

Buck Regulator

Estimate the power loss (PDMOS_) caused by the high-side

and low-side MOSFETs using the following equations:

where QG, RDS(ON), tR, and tFare the upper-switching

MOSFET’s total gate charge, on-resistance at maximum

operating temperature, rise time, and fall time, respectively.

For the buck regulator, D = VLEDs / VIN, IVALLEY =

(IOUT - ∆IL/ 2) and IPK = (IOUT + ∆IL/ 2).

For example, from the typical specifications in the

Applications Information

section with VOUT = 7.8V, the

high-side and low-side MOSFET RMS currents are

0.77A and 0.63A, respectively, for a 1A buck regulator.

Ensure that the thermal impedance of the MOSFET

package keeps the junction temperature at least +25°C

below the absolute maximum rating. Use the following

equation to calculate the maximum junction tempera-

ture: TJ= (PDMOS x θJA) + TA, where θJA and TAare

the junction-to-ambient thermal impedance and ambi-

ent temperature, respectively.

To guarantee that there is no shoot-through from VIN to

PGND, the MAX16818 produces a nonoverlap time of

35ns. During this time, neither high- nor low-side MOS-

FET is conducting, and since the output inductor must

maintain current flow, the intrinsic body diode of the

low-side MOSFET becomes the conduction path. Since

this diode has a fairly large forward voltage, a Schottky

diode (in parallel to the low-side MOSFET) diverts current

flow from the MOSFET body diode because of its lower

forward voltage, which, in turn, increases efficiency.

Boost Regulator

Estimate the power loss (PDMOS_) caused by the MOS-

FET using the following equations:

For a boost regulator in continuous mode, D = VLEDs /

(VIN + VLEDs), IVALLEY = (IOUT - ∆IL/ 2) and IPK = (IOUT

+ ∆IL/ 2).

The voltage across the MOSFET:

VMOSFET = VLED + VF

where VFis the maximum forward voltage of the diode.

The output diode on a boost regulator must be rated to

handle the LED series voltage, VLED. It should also

have fast reverse-recovery characteristics and should

handle the average forward current that is equal to the

LED current.

Input Capacitors

For buck regulator designs, the discontinuous input

current waveform of the buck converter causes large

ripple currents in the input capacitor. The switching fre-

quency, peak inductor current, and the allowable peak-

to-peak voltage ripple reflected back to the source

dictate the capacitance requirement. Increasing switch-

ing frequency or paralleling out-of-phase converters

lowers the peak-to-average current ratio, yielding a

lower input capacitance requirement for the same LED

current. The input ripple is comprised of ∆VQ(caused

by the capacitor discharge) and ∆VESR (caused by the

ESR of the capacitor). Use low-ESR ceramic capacitors

with high-ripple-current capability at the input. Assume

the contributions from the ESR and capacitor discharge

are equal to 30% and 70%, respectively. Calculate the

input capacitance and ESR required for a specified ripple

using the following equation:

ESR V

IN ESR

OUT L

⎛

⎝

⎜⎞

⎠

⎟

∆

PD Q x V x f

VxI xt txf RxI

IIIIxIx

FET G DD SW

IN OUT R F SW DS ON RMS HI

RMS HI VALLEY PK VALLEY PK

( )

( ) ( )

( )

()

⎛

⎝

⎜⎞

⎠

⎟+

=++

−

( )

()

RxI

IIIIxIx

DS ON RMS LO

RMS LO VALLEY PK VALLEY PK

−

−=++ −

22 1

PD Q x V x f

MOS LO G DD SW−=+ ( )

IIIIxIx

RMS HI VALLEY PK VALLEY PK−=++ ( )

VxI xt txf

RxI

IN OUT R F SW

DS ON RMS HI

( )

()

⎛

⎝

⎜⎞

⎠

⎟

+−

PD Q x V x f

MOS HI G DD SW−=+ ( )

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

22 ______________________________________________________________________________________

Buck:

where IOUT is the output current of the converter. For

example, at VIN = 13.2V, VLED = 7.8V, IOUT = 1A, ∆IL=

0.4A, and fSW = 330kHz, the ESR and input capaci-

tance are calculated for the input peak-to-peak ripple of

100mV or less yielding an ESR and capacitance value

of 25mΩand 10µF.

For boost regulator designs, the input-capacitor current

waveform is dominated by the inductor, a triangle wave

a magnitude of ∆IL. For simplicity’s sake, the current

waveform can be approximated by a square wave with

a magnitude that is half that of the triangle wave.

Calculate the input capacitance and ESR required for a

specified ripple using the following equation:

Boost:

Duty cycle, D, for a boost regulator is equal to (VOUT -

VIN) / VOUT. As an example, at VIN = 13.2V, VLED =

15.6V, IOUT = 1A, ∆IL= 0.4A, and fSW = 330kHz, the

ESR and input capacitance are calculated for the input

peak-to-peak ripple of 100mV or less yielding an ESR

and capacitance value of 250mΩand 1µF, respectively.

Output Capacitor

For buck converters, the inductor always connects to

the load, so the inductance controls the ripple current.

The output capacitance shunts a fraction of this ripple

current and the LED string absorbs the rest. The

capacitor reactance (which includes the capacitance

and ESR) and the dynamic impedance of the LED

diode string form a conductance divider that splits the

ripple current between the LEDs and the capacitor. In

many cases, the capacitor is very large as compared to

the ESR, and this divider reduces to the ESR and the

LED resistance.

Boost converters place a harsher requirement on the

output capacitors as they must sustain the full load dur-

ing the on-time of the MOSFET and are replenished

during the off-time. The ripple current in this case is the

full load current, and the holdup time is equal to the

duty cycle times the switching period.

Current Limit

In addition to the average current limit, the MAX16818

also has hiccup current limit. The hiccup current limit is

set to 10% below the average current limit to ensure that

the circuit goes in hiccup mode during continuous out-

put short circuit. Connecting a resistor from LIM to

ground increases the hiccup current limit, while shorting

LIM to ground disables the hiccup current-limit circuit.

Average Current Limit

The average-current-mode control technique of the

MAX16818 accurately limits the maximum output current.

The MAX16818 senses the voltage across the sense

resistor and limit the peak inductor current (IL-PK)

accordingly. The on-cycle terminates when the current-

sense voltage reaches 25.5mV (min). Use the following

equation to calculate the maximum current-sense resis-

tor value:

where PDRis the dissipation in the series resistors.

Select a 5% lower value of RSto compensate for any

parasitics associated with the PCB. Also, select a non-

inductive resistor with the appropriate power rating.

Hiccup Current Limit

The hiccup current-limit value is always 10% lower than

the average current-limit threshold, when LIM is left

unconnected. Connect a resistor from LIM to SGND to

increase the hiccup current-limit value from 90% to

100% of the average current-limit value. The average

current-limit architecture accurately limits the average

output current to its current-limit threshold. If the hiccup

current limit is programmed to be equal or above the

average current-limit value, the output current does not

reach the point where the hiccup current limit can trig-

ger. Program the hiccup current limit at least 5% below

the average current limit to ensure that the hiccup cur-

rent-limit circuit triggers during overload. See the

Hiccup Current Limit vs. REXT graph in the

Typical

Operating Characteristics

PD x

SOUT

=−

0 0255

075 10 3

IxD

Vxf

QSW

∆

ESR V

IN ESR

=∆

∆

CIxDD

Vxf

IN OUT

QSW

()

=−1

∆

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

______________________________________________________________________________________ 23

Compensation

The main control loop consists of an inner current loop

(inductor current) and an outer LED current loop. The

MAX16818 uses an average current-mode control

scheme to regulate the LED current (Figure 7). The VEA

output provides the controlling voltage for the current

source. The inner current loop absorbs the inductor pole

reducing the order of the LED current loop to that of a

single-pole system. The major consideration when

designing the current control loop is making certain that

the inductor downslope (which becomes an upslope at

the output of the CEA) does not exceed the internal

ramp slope. This is a necessary condition to avoid sub-

harmonic oscillations similar to those in peak current

mode with insufficient slope compensation. This requires

that the resistance, RCF, at the output of the CEA be lim-

ited, based on the following equation (Figure 6):

Buck:

where VRAMP = 2V, gm= 550µS, and AV= 34.5.

Boost:

The crossover frequency of the inner current loop is

expressed as:

Buck:

When AV = 34.5, gm = 550µS, and VRAMP = 2V, this

becomes:

Boost:

which becomes:

For adequate phase margin, place the zero formed by

RCF and CCZ not more than 1/3 to 1/5 of the crossover

frequency. The pole formed by RCF and CCP may not

be required in most applications but can be added to

minimize noise at a frequency at or above the switching

frequency.

Power Dissipation

The TQFN is a thermally enhanced package and can dis-

sipate about 2.7W. The high-power package makes the

high-frequency, high-current LED driver possible to oper-

ate from a 12V or 24V bus. Calculate power dissipation in

the MAX16818 as a product of the input voltage and the

total VCC regulator output current (ICC). ICC includes qui-

escent current (IQ) and gate drive current (IDD):

where QG1 and QG2 are the total gate charge of the low-

side and high-side external MOSFETs at VGATE = 5V, IQ

is estimated from the Supply Current (IQ) vs. Frequency

graph in the

Typical Operating Characteristics

, and fSW

is the switching frequency of the LED driver. For boost

drivers, only consider one gate charge, QG1.

Use the following equation to calculate the maximum

power dissipation (PDMAX) in the chip at a given ambi-

ent temperature (TA):

PDMAX = 34.5 x (150 - TA) mW.

PVxI

IIfxQQ

DINCC

CC Q SW G G

( )

=+ +

[]

fmS V R V R

C boost SLED CF

()

×× ×

9 488

2π

fAgRV R

C boost VmSLEDCF

RAMP

_=××× ×

××2π

fmS V R V R

C buck SIN CF

()

×× ×

9 488

2π

fAgRVR

C buck VmSINCF

RAMP

_=××××

××2π

RVfL

AgR V V

CF RAMP SW

VmS LEDIN

≤××

××× −

()

≤1005 ××

×−

()

RV V

SLEDIN

RfL

CF SW

SLED

≤× ×

105

RVfL

AgRV

CF RAMP SW

VmSLED

≤××

×××

PCB Layout Guidelines

Use the following guidelines to layout the switching volt-

age regulator:

1) Place the IN, VCC, and VDD bypass capacitors

close to the MAX16818.

2) Minimize the area and length of the high current

loops from the input capacitor, upper switching

MOSFET, inductor, and output capacitor back to

the input capacitor negative terminal.

3) Keep short the current loop formed by the lower

switching MOSFET, inductor, and output capacitor.

4) Place the Schottky diodes close to the lower

MOSFETs and on the same side of the PCB.

5) Keep the SGND and PGND isolated and connect

them at one single point.

6) Run the current-sense lines CSP and CSN very

close to each other to minimize the loop area.

Similarly, run the remote voltage-sense lines

SENSE+ and SENSE- close to each other. Do not

cross these critical signal lines through power cir-

cuitry. Sense the current right at the pads of the

current-sense resistors.

7) Avoid long traces between the VDD bypass capaci-

tors, the driver output of the MAX16818, the MOS-

FET gates, and PGND. Minimize the loop formed by

the VCC bypass capacitors, bootstrap diode, boot-

strap capacitor, the MAX16818, and the upper

MOSFET gate.

8) Distribute the power components evenly across the

board for proper heat dissipation.

9) Provide enough copper area at and around the

switching MOSFETs, inductor, and sense resistors

to aid in thermal dissipation.

10) Use wide copper traces (2oz) to keep trace induc-

tance and resistance low to maximize efficiency.

Wide traces also cool heat-generating components.

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

24 ______________________________________________________________________________________

Chip Information

TRANSISTOR COUNT: 5654

PROCESS: BiCMOS

432

MAX16818

TQFN

TOP VIEW

N.C.

PGND

BST

N.C.

VDD

VCC

SGND

SENSE+

SENSE-

SGND

CSP

CSN

DIFF

EAN

EAOUT

CLP

* EXPOSED PAD

OVI

LIM

V_IOUT

RT/SYNC

PGOOD

CLKOUT

SGND

Pin Configuration

Package Information

For the latest package outline information and land patterns, go

to www.maxim-ic.com/packages.

PACKAGE TYPE PACKAGE CODE DOCUMENT NO.

28 TQFN T2855-3 21-0140

MAX16818

1.5MHz, 30A High-Efficiency, LED Driver

with Rapid LED Current Pulsing

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are

implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________

Revision History

REVISION

NUMBER

REVISION

DATE DESCRIPTION PAGES

CHANGED

0 10/06 Initial release —

1 6/08 Replaced Compensation section and corrected Figure 4. 12, 23

2 3/09 Updated formula in Inductor Selection section. 20