LM3406HVMHX/NOPB - NATIONAL SEMICONDUCTOR

GND

DIM

BOOT SW

VOUT

RON

LM3406/06HV

VIN,VINS

RSNS

RON

CIN

VIN

VCC

COMP

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

LM3406 1.5-A, Constant Current, Buck Regulator for Driving High Power LEDs

1 FEATURES 3 DESCRIPTION

The LM3406 family are monolithic switching

1• LM3406HV-Q1 regulators designed to deliver constant currents to

– Automotive Grade Device high power LEDs. Ideal for automotive, industrial, and

– AEC-Q100 Grade 1 Qualified general lighting applications, they contain a high-side

N-channel MOSFET switch with a current limit of

– Operating Ambient Temperature: –40°C to 2.0A (typical) for step-down (Buck) regulators.

125°C Controlled on-time with true average current and an

• Integrated 2.0A MOSFET external current sense resistor allow the converter

• VIN Range 6V to 42V (LM3406) output voltage to adjust as needed to deliver a

constant current to series and series-parallel

• VIN Range 6V to 75V (LM3406HV) connected LED arrays of varying number and type.

• VIN Range 6V to 75V (LM3406HV-Q1) LED dimming via pulse width modulation (PWM) is

• True Average Output Current Control achieved using a dedicated logic pin or by PWM of

the power input voltage. The product feature set is

• 1.7A Minimum Output Current Limit Over rounded out with low-power shutdown and thermal

Temperature shutdown protection.

• Cycle-by-Cycle Current Limit The LM3406HV-Q1 is AEC-Q100 grade 1 qualified.

• PWM Dimming with Dedicated Logic Input

• PWM Dimming with Power Input Voltage

• Simple Control Loop Compensation

• Low Power Shutdown

• Supports All-Ceramic Output Capacitors and

Capacitor-less Outputs

• Thermal Shutdown Protection

• TSSOP-14 Package

2 APPLICATIONS

• LED Driver

• Constant Current Source

• Automotive Lighting

• General Illumination

• Industrial Lighting

3.1 TYPICAL APPLICATION

An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

DAP

BOOT

VOUT

GND

VIN

VCC

VINS

RON

COMP

DIM

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

www.ti.com

3.1 Electrostatic Discharge Caution

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

3.1 CONNECTION DIAGRAM

LM3406 family

14-Lead Exposed Pad Plastic TSSOP Package

See Package Number PWP0014A

PIN DESCRIPTIONS

Pin(s) Name Description Application Information

1,2 SW Switch pin Connect these pins to the output inductor and Schottky diode.

3 BOOT MOSFET drive bootstrap pin Connect a 22 nF ceramic capacitor from this pin to the SW pins.

4 NC No Connect No internal connection. Leave this pin unconnected.

5 VOUT Output voltage sense pin Connect this pin to the output node where the inductor and the first LED's

anode connect.

6 CS Current sense feedback pin Set the current through the LED array by connecting a resistor from this pin

to ground.

7 GND Ground pin Connect this pin to system ground.

8 DIM Input for PWM dimming Connect a logic-level PWM signal to this pin to enable/disable the power

MOSFET and reduce the average light output of the LED array. Logic high

= output on, logic low - output off.

9 COMP Error amplifier output Connect a 0.1 µF ceramic capacitor with X5R or X7R dielectric from this pin

to ground.

10 RON On-time control pin A resistor connected from this pin to VIN sets the regulator controlled on-

time.

11 VCC Output of the internal 7V linear Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor with

regulator X5R or X7R dielectric.

12 VINS Input voltage PWM dimming Connect this pin to the anode of the input diode to allow dimming by PWM

comparator input of the input voltage

13,14 VIN Input voltage pin Nominal operating input range for this pin is 6V to 42V (LM3406) or 6V to

75V (LM3406HV, LM3406HV-Q1).

DAP DAP Thermal Pad Connect to ground. Place 4-6 vias from DAP to bottom layer ground plane.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406, LM3406HV, LM3406HV-Q1

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

4 ABSOLUTE MAXIMUM RATINGS (1)

If Military/Aerospace specified devices are required, contact the Texas Instruments Semiconductor Sales Office/

Distributors for availability and specifications. LM3406 –-0.3V to 45V

VIN to GND LM3406HV, LM3406HV-Q1 –-0.3V to 76V

LM3406 –-0.3V to 45V

VINS to GND LM3406HV, LM3406HV-Q1 –-0.3V to 76V

LM3406 –-0.3V to 45V

VOUT to GND LM3406HV, LM3406HV-Q1 –-0.3V to 76V

LM3406 –-0.3V to 59V

BOOT to GND LM3406HV, LM3406HV-Q1 –-0.3V to 76V

LM3406 –1.5V to 45V

SW to GND LM3406HV, LM3406HV-Q1 –1.5V to 76V

LM3406 –-0.3V to 45V

BOOT to VCC LM3406HV, LM3406HV-Q1 –-0.3V to 76V

BOOT to SW –-0.3V to 14V

VCC to GND –-0.3V to 14V

DIM to GND –-0.3V to 7V

COMP to GND –-0.3V to 7V

CS to GND –-0.3V to 7V

RON to GND –-0.3V to 7V

Junction Temperature 150°C

Storage Temp. Range -65°C to 125°C

ESD Rating (2) 2kV

Soldering Information

Lead Temperature (Soldering, 10sec) 260°C

Infrared/Convection Reflow (15sec) 235°C

(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of

device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or

other conditions beyond those indicated in the Operating Ratings is not implied. The recommended Operating Ratings indicate

conditions at which the device is functional and the device should not be operated beyond such conditions.

(2) The human body model is a 100 pF capacitor discharged through a 1.5-kΩresistor into each pin.

5 RECOMMENDED OPERATING CONDITIONS(1)

LM3406 6V to 42V

VIN LM3406HV, LM3406HV-Q1 6V to 75V

LM3406, LM3406HV −40°C to +125°C

Junction Temperature Range LM3406HV-Q1 −40°C to +150°C

Ambient Temperature Range LM3406HV-Q1(2) −40°C to +125°C

Thermal Resistance θJA (TSSOP-14 Package)(3) 50°C/W

(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of

device reliability and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or

other conditions beyond those indicated in the Operating Ratings is not implied. The recommended Operating Ratings indicate

conditions at which the device is functional and the device should not be operated beyond such conditions.

(2) The LM3406HV-Q1 can operate at an ambient temperature of up to +125°C as long as the junction temperature maximum of +150°C is

not exceeded.

(3) θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1-oz. copper on the top or bottom PCB layer.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

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6 ELECTRICAL CHARACTERISTICS LM3406/LM3406HV/LM3406HV-Q1

VIN = 24V unless otherwise indicated. Unless otherwise specified, datasheet typicals and limits apply to LM3406, LM3406HV

and LM3406HV-Q1. Typicals and limits appearing in plain type apply for TA= TJ= +25°C (1). Limits appearing in boldface

type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design, test, or

statistical analysis. Parameter Test Conditions Min Typ Max Units

REGULATION COMPARATOR AND ERROR AMPLIFIER

187.5 200 210

VREF CS Regulation Threshold CS Decreasing, SW turns on mV

191.0(2) 210.0(2)

V0V CS Over-voltage Threshold CS Increasing, SW turns off 300 mV

ICS CS Bias Current CS = 0V 0.9 µA

IVOUT VOUT Bias Current VOUT = 24V 83 µA

ICOMP COMP Pin Current CS = 0V 25 µA

Gm-CS Error Amplifier Transconductance 150 mV < CS < 250 mV 145 µS

SHUTDOWN

Shutdown Threshold RON Increasing 0.3 0.7 1.05

VSD-TH V

Shutdown Threshold (LM3406HV-Q1) RON Increasing 0.3 0.7 1.066

VSD-HYS Shutdown Hysteresis RON Decreasing 40 mV

ON AND OFF TIMER

tOFF-MIN Minimum Off-time CS = 0V 230

Programmed On-time VIN = 24V, VO= 12V, RON = 200kΩ800 1300 1800

tON ns

Programmed On-time (LM3406HV-Q1) VIN = 24V, VO= 12V, RON = 200kΩ800 1300 1850

tON-MIN Minimum On-time 280

VINS COMPARATOR

VINS-TH VINS Pin Threshold VINS decreasing 70 %VIN

IIN-2WD VINS Pin Input Current VINS = 24V * 0.7 25 µA

INTERNAL REGULATOR

VCC Regulated Output 0 mA < ICC < 5 mA 6.4 77.4

VCC-REG V

VCC Regulated Output (LM3406HV-Q1) 0 mA < ICC < 5 mA 6.4 77.5

ICC = 5 mA, 6.0V < VIN < 8.0V,

VIN-DO VIN - VCC 300 mV

Non-switching

VCC-BP-TH VCC Bypass Threshold VIN Increasing 8.8 V

VCC-LIM VCC Current Limit VIN = 24V, VCC = 0V 420 mA

VCC-UV-TH VCC Under-voltage Lock-out Threshold VCC Increasing 5.3 V

VCC-UV-HYS VCC Under-voltage Lock-out Hysteresis VCC Decreasing 150 mV

IIN-OP IIN Operating Current Non-switching, CS = 0.5V 1.2 mA

IIN-SD IIN Shutdown Current RON = 0V 240 350 µA

CURRENT LIMIT

Current Limit Threshold 1.7 2.1 2.7

ILIM A

Current Limit Threshold (LM3406HV-Q1) 1.65 2.1 2.60

DIM COMPARATOR

VIH Logic High DIM Increasing 2.2 V

VIL Logic Low DIM Decreasing 0.8 V

IDIM-PU DIM Pull-up Current DIM = 1.5V 80 µA

MOSFET AND DRIVER

RDS-ON Buck Switch On Resistance ISW = 200 mA, BOOT = 6.3V 0.37 0.75 Ω

VDR-UVLO BOOT Under-voltage Lock-out Threshold BOOT–SW Increasing 1.7 2.9 4.3 V

VDR-HYS BOOT Under-voltage Lock-out Hysteresis BOOT–SW Decreasing 370 mV

THERMAL SHUTDOWN

TSD Thermal Shutdown Threshold 165 °C

TSD-HYS Thermal Shutdown Hysteresis 25 °C

(1) Typical values represent most likely parametric norms at the conditions specified.

(2) Specified with junction temperature from 0°C - 125°C.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406, LM3406HV, LM3406HV-Q1

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

ELECTRICAL CHARACTERISTICS LM3406/LM3406HV/LM3406HV-Q1 (continued)

VIN = 24V unless otherwise indicated. Unless otherwise specified, datasheet typicals and limits apply to LM3406, LM3406HV

and LM3406HV-Q1. Typicals and limits appearing in plain type apply for TA= TJ= +25°C (1). Limits appearing in boldface

type apply over full Operating Temperature Range. Datasheet min/max specification limits are specified by design, test, or

statistical analysis. Parameter Test Conditions Min Typ Max Units

THERMAL RESISTANCE

θJA Junction to Ambient TSSOP-14 Package (3) 50 °C/W

(3) θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1-oz. copper on the top or bottom PCB layer.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

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7 TYPICAL PERFORMANCE CHARACTERISTICS

Figure 1. Efficiency vs. Number of InGaN LEDs in Series Figure 2. Efficiency Vs. Output Current

(1) (1)

Figure 3. VREF vs Temperature Figure 4. VREF vs VIN, LM3406

Figure 5. VREF vs VIN, LM3406HV/LM3406HV-Q1 Figure 6. Current Limit vs Temperature

(1) VIN = 24V, IF= 1A, TA= 25°C, and the load consists of three InGaN LEDs in series unless otherwise noted. See the Bill of Materials

table at the end of the datasheet.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406, LM3406HV, LM3406HV-Q1

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

TYPICAL PERFORMANCE CHARACTERISTICS (continued)

Figure 7. Current Limit vs VIN, LM3406 Figure 8. Current Limit vs VIN, LM3406HV/LM3406HV-Q1

Figure 9. VCC vs VIN Figure 10. VO-MAX vs VIN, LM3406

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

BOOT

VCC

VIN

DIM

RON

GND

VIN

SENSE

7V BIAS

REGULATOR

BYPASS

SWITCH

VCC

UVLO THERMAL

SHUTDOWN

ON TIMER

Complete

Start

300 ns MIN

OFF TIMER

Complete

Start

LOGIC

CURRENT

LIMIT OFF

TIMER

BUCK

SWITCH

CURRENT

SENSE

LEVEL

SHIFT

GATE DRIVE

UVLO VIN

-2.0A

0.7V

0.2V

1.5V

75 éA

COMP

On-Time

Current

Generator

VOUT

VIN

VINS +

x 0.7

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

www.ti.com

7.1 BLOCK DIAGRAM

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406/06HV

CS VSNS

RSNS

One-shot

LED 1

LED n

0.2V

COMP

tON = 1 x 10-11 x RON x VO

VIN

LM3406, LM3406HV, LM3406HV-Q1

www.ti.com

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

8 APPLICATION INFORMATION

8.1 THEORY OF OPERATION

The LM3406, LM3406HV and LM3406HV-Q1 are buck regulators with a wide input voltage range, low voltage

reference, and a fast output enable/disable function. These features combine to make them ideal for use as a

constant current source for LEDs with forward currents as high as 1.5A. The controlled on-time (COT)

architecture uses a comparator and a one-shot on-timer that varies inversely with input and output voltage

instead of a fixed clock. The LM3406 family also employs an integrator circuit that averages the output current.

When the converter runs in continuous conduction mode (CCM) the controlled on-time maintains a constant

switching frequency over changes in both input and output voltage. These features combine to give the LM3406

family an accurate output current, fast transient response, and constant switching frequency over a wide range of

conditions.

8.2 CONTROLLED ON-TIME OVERVIEW

shows a simplified version of the feedback system used to control the current through an array of LEDs. A

differential voltage signal, VSNS, is created as the LED current flows through the current setting resistor, RSNS.

VSNS is fed back by the CS pin, where it is integrated and compared against an error amplifier-generated

reference. The error amplifier is a transconductance (Gm) amplifier which adjusts the voltage on COMP to

maintain a 200 mV average at the CS pin. The on-comparator turns on the power MOSFET when VSNS falls

below the reference created by the Gmamp. The power MOSFET conducts for a controlled on-time, tON, set by

an external resistor, RON, the input voltage, VIN and the output voltage, VO. On-time can be estimated by the

following simplified equation (for the most accurate version of this expression see the Appendix):

(1)

At the conclusion of tON the power MOSFET turns off and must remain off for a minimum of 230 ns. Once this

tOFF-MIN is complete the CS comparator compares the integrated VSNS and reference again, waiting to begin the

next cycle.

Comparator and One-Shot

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

fSW =1

1 x 10-11 x RON

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

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8.3 SWITCHING FREQUENCY

The LM3406 family does not contain a clock, however the on-time is modulated in proportion to both input

voltage and output voltage in order to maintain a relatively constant frequency. On-time tON, duty cycle D and

switching frequency fSW are related by the following expression:

fSW = D / tON (2)

D = (VO+ VD) / (VIN - VSW + VD) (3)

VD= Schottky diode (typically 0.5V) (4)

VSW = IFx RDSON (5)

The LM3406 family regulators should be operated in continuous conduction mode (CCM), where inductor current

stays positive throughout the switching cycle. During steady-state CCM operation, the converter maintains a

constant switching frequency that can be estimated using the following equation (for the most accurate version,

particularly for applications that will have an input or output voltage of less than approximately 12V, see the

Appendix):

(6)

(7)

8.4 SETTING LED CURRENT

LED current is set by the resistor RSNS, which can be determined using the following simple expression due to

the output averaging:

RSNS = 0.2 / IF(8)

8.5 MAXIMUM NUMBER OF SERIES LEDS

LED driver designers often want to determine the highest number of LEDs that can be driven by their circuits.

The limit on the maximum number of series LEDs is set by the highest output voltage, VO-MAX, that the LED driver

can provide. A buck regulator cannot provide an output voltage that is higher than the minimum input voltage,

and in pratice the maximum output voltage of the LM3406 family is limited by the minimum off-time as well. VO-

MAX determines how many LEDs can be driven in series. Referring to the illustration in , output voltage is

calculated as:

VO-MAX = VIN-MIN x (1 - fSW x tOFF-MIN) (9)

tOFF-MIN = 230 ns

Once VO-MAX has been calculated, the maximum number of series LEDs, nMAX, can be calculated by the following

espression and rounding down:

nMAX = VO-MAX / VF(10)

VF= forward voltage of each LED

At low switching frequency VO-MAX is higher, allowing the LM3406 family to regulate output voltages that are

nearly equal to input voltage, and this can allow the system to drive more LEDs in series. Low switching

frequencies are not always desireable, however, because they require larger, more expensive components.

8.6 CALCULATING OUTPUT VOLTAGE

Even though output current is the controlled parameter in LED drivers, output voltage must still be calculated in

order to design the complete circuit. Referring to the illustration in , output voltage is calculated as:

VO= n x VF+ VSNS (11)

VSNS = sense voltage of 200 mV, n = number of LEDs in series

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

8.7 MINIMUM ON-TIME

The minimum on-time for the LM3406 family is 280 ns (typical). One practical example of reaching the minimum

on-time is when dimming the LED light output with a power FET placed in parallel to the LEDs. When the FET is

on, the output voltage drops to 200 mV. This results in a small duty cycle and in most circuits requires an on-time

that would be less than 280 ns. In such a case the LM3406 family keeps the on-time at 280 ns and increases the

off-time as much as needed, which effectively reduces the switching frequency.

8.8 HIGH VOLTAGE BIAS REGULATOR (VCC)

The LM3406 family contains an internal linear regulator with a 7V output, connected between the VIN and the

VCC pins. The VCC pin should be bypassed to the GND pin with a 0.1 µF ceramic capacitor connected as close

as possible to the pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical) and then regulates at 7V as

VIN increases. The LM3406 family comes out of UVLO and begins operating when VCC crosses 5.3V. This is

shown graphically in the Typical Performance curves.

Connecting an external supply to VCC to power the gate drivers is not recommended. However, it may be done if

certain precautions are taken. Be sure that the external supply will not violate any absolute maximum conditions

and will at no point exceed the voltage applied to the VIN pins. Under certain conditions, some ringing may be

present on the SW and BOOT pins when VCC is driven with an external supply. It is important to ensure that the

absolute maximum ratings of these pins are not violated during the ringing or else damage to the device may

occur.

8.9 INTERNAL MOSFET AND DRIVER

The LM3406 family features an internal power MOSFET as well as a floating driver connected from the SW pin

to the BOOT pin. Both rise time and fall time are 20 ns each (typical) and the approximate gate charge is 9 nC.

The high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-voltage diode and

an external 22 nF capacitor, CB. VCC charges CBthrough the internal diode while the power MOSFET is off.

When the MOSFET turns on, the internal diode reverse biases. This creates a floating supply equal to the VCC

voltage minus the diode drop to drive the MOSFET when its source voltage is equal to VIN.

8.10 FAST LOGIC PIN FOR PWM DIMMING

The DIM pin is a TTL compatible input for PWM dimming of the LED. A logic low (below 0.8V) at DIM will disable

the internal MOSFET and shut off the current flow to the LED array. While the DIM pin is in a logic low state the

support circuitry (driver, bandgap, VCC) remains active in order to minimize the time needed to turn the LED

array back on when the DIM pin sees a logic high (above 2.2V). A 75 µA (typical) pull-up current ensures that the

LM3406 family is on when DIM pin is open circuited, eliminating the need for a pull-up resistor. Dimming

frequency, fDIM, and duty cycle, DDIM, are limited by the LED current rise time and fall time and the delay from

activation of the DIM pin to the response of the internal power MOSFET. In general, fDIM should be at least one

order of magnitude lower than the steady state switching frequency in order to prevent aliasing.

8.11 INPUT VOLTAGE COMPARATOR FOR PWM DIMMING

Adding an external input diode and using the internal VINS comparator allows the LM3406 family to sense and

respond to dimming that is done by PWM of the input voltage. This method is also referred to as "Two-Wire

Dimming", and a typical application circuit is shown in . If the VINS pin voltage falls 70% below the VIN pin

voltage, the LM3406 family disables the internal power FET and shuts off the current to the LED array. The

support circuitry (driver, bandgap, VCC) remains active in order to minimize the time needed to the turn the LED

back on when the VINS pin voltage rises and exceeds 70% of VIN. This minimizes the response time needed to

turn the LED array back on.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

GND

DIM

BOOT SW

VOUT

RON

LM3406/06HV

VIN,VINS

RSNS

RON

CIN

VIN IF

VCC

COMP

GND

BOOT SW

VOUT

RON

LM3406/06HV

VIN

RSNS

RON

CIN

VIN

VCC

COMP

VINS

DIM

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

www.ti.com

INPUT VOLTAGE COMPARATOR FOR PWM DIMMING (continued)

Typical Application using Two-Wire Dimming

8.12 PARALLEL MOSFET FOR HIGH-SPEED PWM DIMMING

For applications that require dimming at high frequency or with wide dimming duty cycle range neither the VINS

comparator or the DIM pin are capable of slewing the LED current from 0 to the target level fast enough. For

such applications the LED current slew rate can by increased by shorting the LED current with a N-MOSFET

placed in parallel to the LED or LED array, as shown in . While the parallel FET is on the output current flows

through it, effectively reducing the output voltage to equal the CS pin voltage of 0.2V. This dimming method

maintains a continuous current through the inductor, and therefore eliminates the biggest delay in turning the

LED(s) or and off. The trade-off with parallel FET dimming is that more power is wasted while the FET is on,

although in most cases the power wasted is small compared to the power dissipated in the LEDs. Parallel FET

circuits should use no output capacitance or a bare minimum for noise filtering in order to minimize the slew rate

of output voltage. Dimming FET Q1 can be driven from a ground-referenced source because the source stays at

0.2V along with the CS pin.

Dimming with a Parallel FET

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

GND

BOOT SW

VOUT

RON

LM3406/6HV

VIN,VINS

RSNS

RON

CIN

VIN

VCC

COMP

DIM ROUT

LM3406, LM3406HV, LM3406HV-Q1

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

8.13 PEAK CURRENT LIMIT

The current limit comparator of the LM3406 family will engage whenever the power MOSFET current (equal to

the inductor current while the MOSFET is on) exceeds 2.1A (typical). The power MOSFET is disabled for a cool-

down time that of approximately 100 µs. At the conclusion of this cool-down time the system re-starts. If the

current limit condition persists the cycle of cool-down time and restarting will continue, creating a low-power

hiccup mode, minimizing thermal stress on the LM3406 family and the external circuit components.

8.14 OVER-VOLTAGE/OVER-CURRENT COMPARATOR

The CS pin includes an output over-voltage/over-current comparator that will disable the power MOSFET

whenever VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current

overshoot is limited to 300 mV / RSNS by this comparator during transients. The OVP/OCP comparator limits the

maximum ripple voltage at the CS pin to 200 mVP-P.

8.15 OUTPUT OPEN-CIRCUIT

The most common failure mode for power LEDs is a broken bond wire, and the result is an output open-circuit.

When this happens the feedback path is disconnected, and the output voltage will attempt to rise. In buck

converters the output voltage can only rise as high as the input voltage, and the minimum off-time requirement

ensures that VO(MAX) is slightly less than VIN. shows a method using a zener diode, Z1, and zener limiting

resistor, RZ, to limit output voltage to the reverse breakdown voltage of Z1 plus 200 mV. The zener diode reverse

breakdown voltage, VZ, must be greater than the maximum combined VFof all LEDs in the array. The maximum

recommended value for RZis 1 kΩ.

The output stage (SW and VOUT pins) of the LM3406 family is capable of withstanding VO(MAX) indefinitely as

long as the output capacitor is rated to handle the full input voltage. When an LED fails open-circuit and there is

no output capacitor present the surge in output voltage due to the collapsing magnetic field in the output inductor

can exceed VIN and can damage the LM3406 family IC. As an alternative to the zener clamp method described

previously, a diode can be connected from the output to the input of the regulator circuit that will clamp the

inductive surge to one VDabove VIN.

Regardless of which protection method is used a resistance in series with the VOUT pin, ROUT, is recommended

to limit the current in the event the VOUT pin is pulled below ground when the LED circuit is reconnected. This

can occur frequently if the lead lengths to the LEDs are long and the inductance is significant. A resistor between

1 kΩand 10 kΩis recommended.

Two Methods of Output Open Circuit Protection

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

GND

BOOT SW

VOUT

RON

LM3406/06HV

VIN,VINS

RSNS

RON

CIN

VIN

VCC

COMP

2N7000 or

equivalent

ON/OFF

DIM

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

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8.16 LOW POWER SHUTDOWN

The LM3406 family can be placed into a low power state (IIN-SD = 240 µA) by grounding the RON pin with a

signal-level MOSFET as shown in . Low power MOSFETs like the 2N7000, 2N3904, or equivalent are

recommended devices for putting the LM3406 family into low power shutdown. Logic gates can also be used to

shut down the LM3406 family as long as the logic low voltage is below the over temperature minimum threshold

of 0.3V. Noise filter circuitry on the RON pin can cause a few pulses with longer on-times than normal after RON

is grounded or released. In these cases the OVP/OCP comparator will ensure that the peak inductor or LED

current does not exceed 300 mV / RSNS.

Low Power Shutdown

8.17 THERMAL SHUTDOWN

Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction

temperature is exceeded. The threshold for thermal shutdown is 165°C with a 25°C hysteresis (both values

typical). During thermal shutdown the MOSFET and driver are disabled.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

VIN - VO

'iL = 'iF = x tON

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8.1 DESIGN CONSIDERATIONS

8.1.1 SWITCHING FREQUENCY

Switching frequency is selected based on the trade-offs between efficiency (better at low frequency), solution

size/cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower

frequency.) Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the

LM3406 family can be programmed for switching frequencies ranging from the 10’s of kHz to over 1 MHz. This

on-time varies in proportion to both VIN and VOin order to maintain first-order control over switching frequency,

however in practice the switching frequency will shift in response to large swings in VIN or VO. The maximum

switching frequency is limited only by the minimum on-time and minimum off-time requirements.

8.1.2 LED RIPPLE CURRENT

Selection of the ripple current, ΔiF, through the LED array is similar to the selection of output ripple voltage in a

standard voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC

output voltage, LED manufacturers generally recommend values for ΔiFranging from ±5% to ±20% of IF. Higher

LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all.

Lower ripple current requires more output inductance, higher switching frequency, or additional output

capacitance, and may be necessary for applications that are not intended for human eyes, such as machine

vision or industrial inspection.

8.1.3 BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS

The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to

the load during the entire switching cycle. By definition an inductor will control the rate of change of current that

flows through it, and this control over current ripple forms the basis for component selection in both voltage

regulators and current regulators. A current regulator such as the LED driver for which the LM3406 family was

designed focuses on the control of the current through the load, not the voltage across it. A constant current

regulator is free of load current transients, and has no need of output capacitance to supply the load and

maintain output voltage. Referring to the Typical Application circuit on the front page of this datasheet, the

inductor and LED can form a single series chain, sharing the same current. When no output capacitor is used,

the same equations that govern inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For a

controlled on-time converter such as LM3406 family the ripple current is described by the following expression:

(12)

The triangle-wave inductor current ripple flows through RSNS and produces a triangle-wave voltage at the CS pin.

To provide good signal to noise ratio (SNR) the amplitude of CS pin ripple voltage, ΔvCS, should be at least 25

mVP-P.ΔvCS is described by the following:

ΔvCS =ΔiFx RSNS (13)

8.1.4 BUCK CONVERTERS WITH OUTPUT CAPACITORS

A capacitor placed in parallel with the LED(s) can be used to reduce the LED current ripple while keeping the

same average current through both the inductor and the LED array. With an output capacitor the output

inductance can be lowered, making the magnetics smaller and less expensive. Alternatively, the circuit could be

run at lower frequency but keep the same inductor value, improving the power efficiency. Both the peak current

limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ΔiLcan be

even if ΔiFis made very small. Adding a capacitor that reduces ΔiFto well below the target provides headroom

for changes in inductance or VIN that might otherwise push the peak LED ripple current too high.

shows the equivalent impedances presented to the inductor current ripple when an output capacitor, CO, and its

equivalent series resistance (ESR) are placed in parallel with the LED array. Note that ceramic capacitors have

so little ESR that it can be ignored. The entire inductor ripple current still flows through RSNS to provide the

required 25 mV of ripple voltage for proper operation of the CS comparator.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

IIN(rms) = IF x D(1 - D)

CIN (MIN) ='VIN (MAX)

IF x tON

'iF =

ZC = ESR +

'iL

1 + ZC

2Sx fSW x CO

'iL

ESR

RSNS

'iC'iF

'iL

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DESIGN CONSIDERATIONS (continued)

LED and CORipple Current

To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED

dynamic resistance is not always specified on the manufacturer’s datasheet, but it can be calculated as the

inverse slope of the LED’s VFvs. IFcurve. Note that dividing VFby IFwill give an incorrect value that is 5x to 10x

too high. Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rDof one

device multiplied by n. Inductor ripple current is still calculated with the expression from Buck Regulators without

Output Capacitors. The following equations can then be used to estimate ΔiFwhen using a parallel capacitor:

(14)

The calculation for ZCassumes that the shape of the inductor ripple current is approximately sinusoidal.

Small values of COthat do not significantly reduce ΔiFcan also be used to control EMI generated by the

switching action of the LM3406 family. EMI reduction becomes more important as the length of the connections

between the LED and the rest of the circuit increase.

8.1.5 INPUT CAPACITORS

Input capacitors at the VIN pin of the LM3406 family are selected using requirements for minimum capacitance

and rms ripple current. The input capacitors supply pulses of current approximately equal to IFwhile the power

MOSFET is on, and are charged up by the input voltage while the power MOSFET is off. All switching regulators

have a negative input impedance due to the decrease in input current as input voltage increases. This inverse

proportionality of input current to input voltage can cause oscillations (sometimes called ‘power supply

interaction’) if the magnitude of the negative input impedance is greater the the input filter impedance. Minimum

capacitance can be selected by comparing the input impedance to the converter’s negative resistance; however

this requires accurate calculation of the input voltage source inductance and resistance, quantities which can be

difficult to determine. An alternative method to select the minimum input capacitance, CIN(MIN), is to select the

maximum input voltage ripple which can be tolerated. This value, ΔvIN(MAX), is equal to the change in voltage

across CIN during the converter on-time, when CIN supplies the load current. CIN(MIN) can be selected with the

following:

(15)

A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. A minimum input

capacitance of 2x the CIN(MIN) value is recommended for all LM3406 family circuits. To determine the rms current

rating, the following formula can be used:

(16)

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DESIGN CONSIDERATIONS (continued)

Ceramic capacitors are the best choice for the input to the LM3406 family due to their high ripple current rating,

low ESR, low cost, and small size compared to other types. When selecting a ceramic capacitor, special

attention must be paid to the operating conditions of the application. Ceramic capacitors can lose one-half or

more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes in temperature.

A DC voltage rating equal to twice the expected maximum input voltage is recommended. In addition, the

minimum quality dielectric which is suitable for switching power supply inputs is X5R, while X7R or better is

preferred.

8.1.6 RECIRCULATING DIODE

The LM3406 family is a non-synchronous buck regulator that requires a recirculating diode D1 (see the Typical

Application circuit) to carrying the inductor current during the MOSFET off-time. The most efficient choice for D1

is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must be rated to handle the

maximum input voltage plus any switching node ringing when the MOSFET is on. In practice all switching

converters have some ringing at the switching node due to the diode parasitic capacitance and the lead

inductance. D1 must also be rated to handle the average current, ID, calculated as:

ID= (1 – D) x IF(17)

This calculation should be done at the maximum expected input voltage. The overall converter efficiency

becomes more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the

load current for an increasing percentage of the time. This power dissipation can be calculating by checking the

typical diode forward voltage, VD, from the I-V curve on the product datasheet and then multiplying it by ID. Diode

datasheets will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to estimate

the operating die temperature of the device. Multiplying the power dissipation (PD= IDx VD) by θJA gives the

temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below

the operational maximum.

8.2 Transient Protection Considerations

Considerations need to be made when external sources, loads or connections are made to the switching

converter circuit due to the possibility of Electrostatic Discharge (ESD) or Electric Over Stress (EOS) events

occurring and damaging the integrated circuit (IC) device. All IC device pins contain zener based clamping

structures that are meant to clamp ESD. ESD events are very low energy events, typically less than 5µJ

(microjoules). Any event that transfers more energy than this may damage the ESD structure. Damage is

typically represented as a short from the pin to ground as the extreme localized heat of the ESD / EOS event

causes the aluminum metal on the chip to melt, causing the short. This situation is common to all integrated

8.2.1 CS PIN PROTECTION

When hot swapping in a load (e.g. test points, load boards, LED stack), any residual charge on the load will be

immediately transferred through the output capacitor to the CS pin, which is then damaged as shown in below.

The EOS event due to the residual charge from the load is represented as VTRANSIENT.

From measurements, we know that the 8V ESD structure on the CS pin can typically withstand 25mA of direct

current (DC). Adding a 1kΩresistor in series with the CS pin, shown in , results in the majority of the transient

energy to pass through the discrete sense resistor rather than the device. The series resistor limits the peak

current that can flow during a transient event, thus protecting the CS pin. With the 1kΩresistor shown, a 33V,

49A transient on the LED return connector terminal could be absorbed as calculated by:

V = 25mA * 1kΩ+ 8V = 33V (18)

I = 33V / 0.67Ω= 49A (19)

This is an extremely high energy event, so the protection measures previously described should be adequate to

solve this issue.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406

GND

VTRANSIENT

~ 0.675

Module

Connector

Module

Connector

1k5

LM3406

GND

VTRANSIENT

~ 0.675

Module

Connector

Module

Connector

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Transient Protection Considerations (continued)

CS Pin, Transient Path

CS Pin, Transient Path with Protection

Adding a resistor in series with the CS pin causes the observed output LED current to shift very slightly. The

reason for this is twofold: (1) the CS pin has about 20pF of inherent capacitance inside it which causes a slight

delay (20ns for a 1kΩseries resistor), and (2) the comparator that is watching the voltage at the CS pin uses a

pnp bipolar transistor at its input. The base current of this pnp transistor is approximately 100nA which will cause

a 0.1mV change in the 200mV threshold. These are both very minor changes and are well understood. The shift

in current can either be neglected or taken into consideration by changing the current sense resistance slightly.

8.2.2 CS PIN PROTECTION WITH OVP

When designing output overvoltage protection into the switching converter circuit using a zener diode, transient

protection on the CS pin requires additional consideration. As shown in , adding a zener diode from the output to

the CS pin (with the series resistor) for output overvoltage protection will now again allow the transient energy to

be passed through the CS pin’s ESD structure thereby damaging it.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

LM3406

GND

VTRANSIENT

~ 0.675

Module

Connector

Module

Connector

50051 k5

LM3406

GND

VTRANSIENT

~ 0.675

Module

Connector

Module

Connector

1 k5

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Transient Protection Considerations (continued)

Adding an additional series resistor to the CS pin as shown in will result in the majority of the transient energy to

pass through the sense resistor thereby protecting the LM340X device.

CS Pin with OVP, Transient Path

CS Pin with OVP, Transient Path with Protection

8.2.3 VIN PIN PROTECTION

The VIN pin also has an ESD structure from the pin to GND with a breakdown voltage of approximately 80V. Any

transient that exceeds this voltage may damage the device. Although transient absorption is usually present at

the front end of a switching converter circuit, damage to the VIN pin can still occur.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

VIN

LM3406

GND

80V

VIN

Module

Connector

Module

Connector

TVS CIN

Board Trace

Inductance

TVS or

smaller

zener diode

VIN

LM3406

GND

80V

VIN

Module

Connector

Module

Connector

TVS CIN

Board Trace

Inductance

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Transient Protection Considerations (continued)

When VIN is hot swapped in, the current that rushes in to charge CIN up to the VIN value also charges (energizes)

the circuit board trace inductance as shown in . The excited trace inductance then resonates with the input

capacitance (similar to an under-damped LC tank circuit) and causes voltages at the VIN pin to rise well in

excess of both VIN and the voltage at the module input connector as clamped by the input TVS. If the resonating

voltage at the VIN pin exceeds the 80V breakdown voltage of the ESD structure, the ESD structure will activate

and then “snap-back” to a lower voltage due to its inherent design. If this lower snap-back voltage is less than

the applied nominal VIN voltage, then significant current will flow through the ESD structure resulting in the IC

being damaged.

An additional TVS or small zener diode should be placed as close as possible to the VIN pins of each IC on the

board, in parallel with the input capacitor as shown in . A minor amount of series resistance in the input line

would also help, but would lower overall conversion efficiency. For this reason, NTC resistors are often used as

inrush limiters instead.

VIN Pin with Typical Input Protection

VIN Pin with Additional Input Protection

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

tON = 1 x 10-11 x RON x VO

VIN

fSW =1

1 x 10-11 x RON

RON =1

fSW x 1 x 10-11

GND

DIM

BOOT SW

VOUT

RON

LM3406

VIN

RSNS

RON

CIN

VIN = 24V ±10% IF = 1.5A ±5%

VCC

COMP

LED1

LEDn

One to

five

LEDs

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Transient Protection Considerations (continued)

8.2.4 GENERAL COMMENTS REGARDING OTHER PINS

Any pin that goes “off-board” through a connector should have series resistance of at least 1kΩto 10kΩin series

with it to protect it from ESD or other transients. These series resistors limit the peak current that can flow (or

cause a voltage drop) during a transient event, thus protecting the pin and the device. Pins that are not used

should not be left floating. They should instead be tied to GND or to an appropriate voltage through resistance.

8.3 Design Example 1

The first example circuit uses the LM3406 to create a flexible LED driver capable of driving anywhere from one to

five white series-connected LEDs at a current of 1.5A ±5% from a regulated DC voltage input of 24V ±10%. In

addition to the ±5% tolerance specified for the average output current, the LED ripple current must be controlled

to 10%P-P of the DC value, or 150 mAP-P. The typical forward voltage of each individual LED at 1.5A is 3.9V,

hence the output voltage ranges from 4.1V to 19.7V, adding in the 0.2V drop for current sensing. A complete bill

of materials can be found in Table 1 at the end of this datasheet.

Schematic for Design Example 1

8.3.1 RON and tON

A moderate switching frequency of 500 kHz will balance the requirements of inductor size and overall power

efficiency. The LM3406 will allow some shift in switching frequency when VOchanges due to the number of LEDs

in series, so the calculation for RON is done at the mid-point of three LEDs in series, where VO= 11.8V. Note that

the actual RON calculation is done with the high accuracy expression listed in the Appendix.

(20)

RON = 144 kΩ(21)

The closest 1% tolerance resistor is 143 kΩ. The switching frequency and on-time of the circuit should be

checked for one, three and five LEDs using the equations relating RON and tON to fSW. As with the RON

calculation, the actual fSW and tON values have been calculated using the high accuracy expressions listed in the

Appendix.

(22)

fSW(1 LED) = 362 kHz (23)

fSW(3 LEDs) = 504 kHz (24)

fSW(5 LEDs) = 555 kHz (25)

(26)

tON(1 LED) = 528 ns (27)

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

ÂIF

ÂVF

'iFx rD

'iL - 'iF

ZC =

L =VIN - VO

'iLx tON

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Design Example 1 (continued)

tON(3 LEDs) = 1014 ns (28)

tON(5 LEDs) = 1512 ns (29)

8.3.2 OUTPUT INDUCTOR

Since an output capacitor will be used to filter some of the AC ripple current, the inductor ripple current can be

set higher than the LED ripple current. A value of 40%P-P is typical in many buck converters:

ΔiL= 0.4 x 1.5 = 0.6AP-P (30)

With the target ripple current determined the inductance can be chosen:

(31)

LMIN = [(24 – 11.8) x 1.01 x 10-6] / (0.6) = 20.5 µH (32)

The closest standard inductor value is 22 µH. The average current rating should be greater than 1.5A to prevent

overheating in the inductor. Inductor current ripple should be calculated for one, three and five LEDs:

ΔiL(1 LED) = [(24 - 4.1) x 5.28 x 10-7] / 22 x 10-6 = 478 mAP-P (33)

ΔiL(3 LEDs) = [(24 - 11.8) x 1.01 x 10-6] / 22 x 10-6 = 560 mAP-P (34)

ΔiL(5 LEDs) = [(24 - 19.7) x 1.51 x 10-6] / 22 x 10-6 = 295 mAP-P (35)

The peak LED/inductor current is then estimated. This calculation uses the worst-case ripple current which

occurs with three LEDs.

IL(PEAK) = IL+ 0.5 x ΔiL(MAX) (36)

IL(PEAK) = 1.5 + 0.5 x 0.56 = 1.78A (37)

In order to prevent inductor saturation the inductor’s peak current rating must be above 1.8A. A 22 µH off-the

shelf inductor rated to 2.1A (peak) and 1.9A (average) with a DCR of 59 mΩwill be used.

8.3.3 USING AN OUTPUT CAPACITOR

This application does not require high frequency PWM dimming, allowing the use of an output capacitor to

reduce the size and cost of the output inductor while still meeting the 10%P-P target for LED ripple current. To

select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-arranged to

yield the following:

(38)

The dynamic resistance, rD,of one LED can be calculated by taking the tangent line to the VFvs. IFcurve in the

LED datasheet. shows an example rDcalculation.

Calculating rDfrom the VFvs. IFCurve

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Design Example 1 (continued)

Extending the tangent line to the ends of the plot yields values for ΔVFand ΔIFof 0.7V and 2000 mA,

respectively. Dynamic resistance is then:

rD=ΔVF/ΔIF= 0.5V / 2A = 0.25Ω(39)

The most filtering (and therefore the highest output capacitance) is needed when rDis lowest, which is when

there is only one LED. Inductor ripple current with one LED is 478 mAP-P. The required impedance of COis

calculated:

ZC= [0.15 / (0.478 - 0.15] x 0.35 = 0.114Ω(40)

A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 362 kHz:

CO= 1/(2 x πx 0.16 x 3.62 x 105) = 3.9 µF (41)

This calculation assumes that COwill be a ceramic capacitor, and therefore impedance due to the equivalent

series resistance (ESR) and equivalent series inductance (ESL) of of the device is negligible. The closest 10%

tolerance capacitor value is 4.7 µF. The capacitor used should be rated to 25V or more and have an X7R

dielectric. Several manufacturers produce ceramic capacitors with these specifications in the 1206 case size. A

typical value for ESR of 3 mΩcan be read from the curve of impedance vs. frequency in the product datasheet.

8.3.4 RSNS

Using the expression for RSNS:

RSNS = 0.2 / IF(42)

RSNS = 0.2 / 1.5 = 0.133Ω(43)

Sub-1Ωresistors are available in both 1% and 5% tolerance. A 1%, 0.13Ωdevice is the closest value, and a

0.33W, 1210 size device will handle the power dissipation of 290 mW. With the resistance selected, the average

value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. From the

expression for average LED current:

IF= 0.2 / 0.13 = 1.54A, 3% above the target current (44)

8.3.5 INPUT CAPACITOR

Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 24V x 2%P-P = 480 mV. The

minimum required capacitance is calculated for the largest tON, corresponding to five LEDs:

CIN(MIN) = (1.5 x 1.5 x 10-6) / 0.48 = 4.7 µF (45)

As with the output capacitor, this required value is low enough to use a ceramic capacitor, and again the effective

capacitance will be lower than the rated value with 24V across CIN. Reviewing plots of %C vs. DC Bias for

several capacitors reveals that a 4.7 µF, 1812-size capacitor in X7R rated to 50V loses about 40% of its rated

capacitance at 24V, hence two such caps are needed.

Input rms current is high in buck regulators, and the worst-case is when the duty cycle is 50%. Duty cycle in a

buck regulator can be estimated as D = VO/ VIN, and when this converter drives three LEDs the duty cycle will

be nearly 50%.

IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA (46)

Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2A, so two of them in parallel can

tolerate more than enough for this design.

8.3.6 RECIRCULATING DIODE

The input voltage of 24V ±5% requires Schottky diodes with a reverse voltage rating greater than 30V. The next

highest standard voltage rating is 40V. Selecting a 40V rated diode provides a large safety margin for the ringing

of the switch node and also makes cross-referencing of diodes from different vendors easier.

The next parameters to be determined are the forward current rating and case size. The lower the duty cycle the

more thermal stress is placed on the recirculating diode. When driving one LED the duty cycle can be estimated

as: D = 4.1 / 24 = 0.17 (47)

The estimated average diode current is then:

ID= (1 - 0.17) x 1.54 = 1.28A (48)

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Design Example 1 (continued)

A 2A-rated diode will be used. To determine the proper case size, the dissipation and temperature rise in D1 can

be calculated as shown in the Design Considerations section. VDfor a case size such as SMB in a 40V, 2A

Schottky diode at 1.5A is approximately 0.4V and the θJA is 75°C/W. Power dissipation and temperature rise can

be calculated as:

PD= 1.28 x 0.4 = 512 mW TRISE = 0.51 x 75 = 38°C (49)

8.3.7 CB, CCAND CF

The bootstrap capacitor CBshould always be a 22 nF ceramic capacitors with X7R dielectric. A 25V rating is

appropriate for all application circuits. The COMP pin capacitor CCand the linear regulator filter capacitor CF

should always be 100 nF ceramic capacitors, also with X7R dielectric and a 25V ratings.

8.3.8 EFFICIENCY

To estimate the electrical efficiency of this example the power dissipation in each current carrying element can

be calculated and summed. Electrical efficiency, η, should not be confused with the optical efficacy of the circuit,

which depends upon the LEDs themselves. One calculation will be detailed for three LEDs in series, where VO=

11.8V, and these calculations can be repeated for other numbers of LEDs.

Total output power, PO, is calculated as:

PO= IFx VO= 1.54 x 11.8 = 18.2W (50)

Conduction loss, PC, in the internal MOSFET:

PC= (IF2x RDSON) x D = (1.542x 0.75) x 0.5 = 890 mW (51)

Gate charging and VCC loss, PG, in the gate drive and linear regulator:

PG= (IIN-OP + fSW x QG) x VIN PG= (600 x 10-6 + 5 x 105x 9 x 10-9) x 24 = 122 mW (52)

Switching loss, PS, in the internal MOSFET:

PS= 0.5 x VIN x IFx (tR+ tF) x fSW PS= 0.5 x 24 x 1.54 x 40 x 10-9 x 5 x 105= 370 mW (53)

AC rms current loss, PCIN, in the input capacitor:

PCIN = IIN(rms)2x ESR = 0.7520.003 = 2 mW (negligible) (54)

DCR loss, PL, in the inductor

PL= IF2x DCR = 1.542x 0.06 = 142 mW (55)

Recirculating diode loss, PD= (1 - 0.5) x 1.54 x 0.4 = 300 mW

Current Sense Resistor Loss, PSNS = 293 mW

Electrical efficiency, η= PO/ (PO+ Sum of all loss terms) = 18.2 / (18.2 + 2.1) = 89%

Temperature Rise in the LM3406 IC is calculated as:

TLM3406 = (PC+ PG+ PS) x θJA = (0.89 + 0.122 + 0.37) x 50 = 69°C (56)

8.4 Design Example 2

The second example circuit uses the LM3406 to drive a single white LED at 1.5A ±10% with a ripple current of

20%P-P in a typical 12V automotive electrical system. The two-wire dimming function will be employed in order to

take advantage of the legacy 'theater dimming' method which dims and brightens the interior lights of

automobiles by chopping the input voltage with a 200Hz PWM signal. As with the previous example, the typical

VFof a white LED is 3.9V, and with the current sense voltage of 0.2V the total output voltage will be 4.1V. The

LED driver must operate to specifications over an input range of 9V to 16V as well as operating without suffering

damage at 28V for two minutes (the 'double battery jump-start' test) and for 300 ms at 40V (the 'load-dump' test).

The LED driver must also be able to operate without suffering damage at inputs as low as 6V to satisfy the 'cold

crank' tests. A complete bill of materials can be found in Table 2 at the end of this datasheet.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

tON = 1 x 10-11 x RON x VO

VIN

fSW =1

1 x 10-11 x RON

RON =1

fSW x 1 x 10-11

GND

BOOT SW

VOUT

RON

LM3406

VIN

RSNS

RON

CIN

VIN = 6V (cold-crank)

VIN = 9V to 16V (nominal)

VIN = 28V (2 minutes)

VIN = 40V (300 ms) IF = 1.5A

VCC

COMP

VINS

LED1 CO

DIM

LM3406, LM3406HV, LM3406HV-Q1

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

Design Example 2 (continued)

Schematic for Design Example 2

8.4.1 RON and tON

A switching frequency of 450 kHz helps balance the requirements of inductor size and overall power efficiency,

but more importantly keeps the switching frequency below 530 kHz, where the AM radio band begins. This

design will concentrate on meeting the switching frequency and LED current requirements over the nominal input

range of 9V to 16V, and will then check to ensure that the transient conditions do not cause the LM3406 to

overheat. The LM3406 will allow a small shift in switching frequency when VIN changes, so the calculation for

RON is done at the typical expected condition where VIN = 13.8V and VO= 4.1V. The actual RON calculation uses

the high accuracy equation listed in the Appendix.

(57)

RON = 124 kΩ(58)

The closest 1% tolerance resistor is 124 kΩ. The switching frequency and on-time of the circuit should be

checked at VIN-MIN and VIN-MAX which are 9V and 16V, respectively. The actual fSW and tON values have been

calculated with the high accuracy equations in the APPENDIX.

(59)

fSW(VMIN) = 463 kHz (60)

fSW(VMAX) = 440 kHz (61)

(62)

tON(VMIN) = 1090 ns (63)

tON(VMAX) = 650 ns (64)

8.4.2 OUTPUT INDUCTOR

Since an output capacitor will be used to filter some of the LED ripple current, the inductor ripple current can be

set higher than the LED ripple current. A value of 40%P-P is typical in many buck converters:

ΔiL= 0.4 x 1.5 = 0.6AP-P (65)

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

ÂIF

ÂVF

'iFx rD

'iL - 'iF

ZC =

L =VIN - VO

'iLx tON

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

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Design Example 2 (continued)

The minimum inductance required to ensure a ripple current of 600 mAP-P or less is calculated at VIN-MAX:

(66)

LMIN = [(16 – 4.1) x 6.5 x 10-7] / (0.6) = 12.9 µH (67)

The closest standard inductor value is 15 µH. The average current rating should be greater than 1.5A to prevent

overheating in the inductor. Inductor current ripple should be calculated for VIN-MIN and VIN-MAX:

ΔiL(VMIN) = [(9 - 4.1) x 6.5 x 10-7] / 15 x 10-6 = 357 mAP-P (68)

ΔiL(VMAX) = [(16 - 4.1) x 1.09 x 10-6] / 15 x 10-6 = 516 mAP-P (69)

The peak LED/inductor current is then estimated. This calculation uses the worst-case ripple current which

occurs at VIN-MAX.

IL(PEAK) = IL+ 0.5 x ΔiL(MAX) (70)

IL(PEAK) = 1.5 + 0.5 x 0.516 = 1.76A (71)

In order to prevent inductor saturation the inductor’s peak current rating must be above 1.8A. A 15 µH off-the

shelf inductor rated to 2.4A (peak) and 2.2A (average) with a DCR of 47 mΩwill be used.

8.4.3 USING AN OUTPUT CAPACITOR

This application does not require high frequency PWM dimming, allowing the use of an output capacitor to

reduce the size and cost of the output inductor while still meeting the 20%P-P (300 mA) target for LED ripple

current. To select the proper output capacitor the equation from Buck Regulators with Output Capacitors is re-

arranged to yield the following:

(72)

The dynamic resistance, rD,of one LED can be calculated by taking the tangent line to the VFvs. IFcurve in the

LED datasheet. shows an example rDcalculation.

Calculating rDfrom the VFvs. IFCurve

Extending the tangent line to the ends of the plot yields values for ΔVFand ΔIFof 0.7V and 2000 mA,

respectively. Dynamic resistance is then:

rD=ΔVF/ΔIF= 0.5V / 2A = 0.25Ω(73)

The most filtering (and therefore the highest output capacitance) is needed when ΔILis highest, which occurs at

VIN-MAX. Inductor ripple current with one LED is 516 mAP-P. The required impedance of COis calculated:

ZC= [0.3 / (0.516 - 0.3] x 0.35 = 0.35Ω(74)

A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 440 kHz:

CO= 1/(2 x πx 0.49 x 4.4 x 105) = 1.03 µF (75)

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Design Example 2 (continued)

This calculation assumes that COwill be a ceramic capacitor, and therefore impedance due to the equivalent

series resistance (ESR) and equivalent series inductance (ESL) of of the device is negligible. The closest 10%

tolerance capacitor value is 1.5 µF. The capacitor used should have an X7R dielectric and should be rated to

50V. The high voltage rating ensures that COwill not be damaged if the LED fails open circuit and a load dump

occurs. Several manufacturers produce ceramic capacitors with these specifications in the 1206 case size. With

only 4V of DC bias a 50V rated ceramic capacitor will have better than 90% of it's rated capacitance, which is

more than enough for this design.

8.4.4 RSNS

Using the expression for RSNS:

RSNS = 0.2 / IF(76)

RSNS = 0.2 / 1.5 = 0.133Ω(77)

Sub-1Ωresistors are available in both 1% and 5% tolerance. A 1%, 0.13Ωdevice is the closest value, and a

0.33W, 1210 size device will handle the power dissipation of 290 mW. With the resistance selected, the average

value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. From the

expression for average LED current:

IF= 0.2 / 0.13 = 1.54A, 3% above the target current (78)

8.4.5 INPUT CAPACITOR

Controlling input ripple current and voltage is critical in automotive applications where stringent conducted

electromagnetic interference tests are required. ΔvIN(MAX) will be limited to 300 mVP-P or less. The minimum

required capacitance is calculated for the largest tON, 1090 ns, which occurs at the minimum input voltage. Using

the equations from the Input Capacitors section:

CIN(MIN) = (1.5 x 1.09 x 10-6) / 0.3 = 5.5 µF (79)

As with the output capacitor, this required value is low enough to use a ceramic capacitor, and again the effective

capacitance will be lower than the rated value with 16V across CIN. Reviewing plots of %C vs. DC Bias for

several capacitors reveals that a 3.3 µF, 1210-size capacitor in X7R rated to 50V loses about 22% of its rated

capacitance at 16V, hence two such caps are needed.

Input rms current is high in buck regulators, and the worst-case is when the duty cycle is 50%. Duty cycle in a

buck regulator can be estimated as D = VO/ VIN, and when VIN drops to 9V the duty cycle will be nearly 50%.

IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA (80)

Ripple current ratings for 1210 size ceramic capacitors are typically higher than 2A, so two of them in parallel can

tolerate more than enough for this design.

8.4.6 RECIRCULATING DIODE

To survive an input voltage transient of 40V the Schottky diode must be rated to a higher voltage. The next

highest standard voltage rating is 60V. Selecting a 60V rated diode provides a large safety margin for the ringing

of the switch node and also makes cross-referencing of diodes from different vendors easier.

The next parameters to be determined are the forward current rating and case size. The lower the duty cycle the

more thermal stress is placed on the recirculating diode. When driving one LED the duty cycle can be estimated

as: D = 4.1 / 13.8 = 0.3 (81)

The estimated average diode current is then:

ID= (1 - 0.3) x 1.54 = 1.1A (82)

A 2A-rated diode will be used. To determine the proper case size, the dissipation and temperature rise in D1 can

be calculated as shown in the Design Considerations section. VDfor a case size such as SMB in a 60V, 2A

Schottky diode at 1.5A is approximately 0.4V and the θJA is 75°C/W. Power dissipation and temperature rise can

be calculated as:

PD= 1.1 x 0.4 = 440 mW TRISE = 0.44 x 75 = 33°C (83)

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

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Design Example 2 (continued)

8.4.7 CB, CCAND CF

The bootstrap capacitor CBshould always be a 22 nF ceramic capacitors with X7R dielectric. A 25V rating is

appropriate for all application circuits. The COMP pin capacitor CCand the linear regulator filter capacitor CF

should always be 100 nF ceramic capacitors, also with X7R dielectric and a 25V ratings.

8.4.8 EFFICIENCY

To estimate the electrical efficiency of this example the power dissipation in each current carrying element can

be calculated and summed. One calculation will be detailed for the nominal input voltage of 13.8V, and these

calculations can be repeated for other numbers of LEDs.

Total output power, PO, is calculated as:

PO= IFx VO= 1.54 x 4.1 = 6.3W (84)

Conduction loss, PC, in the internal MOSFET:

PC= (IF2x RDSON) x D = (1.542x 0.75) x 0.3 = 530 mW (85)

Gate charging and VCC loss, PG, in the gate drive and linear regulator:

PG= (IIN-OP + fSW x QG) x VIN PG= (600 x 10-6 + 4.5 x 105x 9 x 10-9) x 13.8 = 64 mW (86)

Switching loss, PS, in the internal MOSFET:

PS= 0.5 x VIN x IFx (tR+ tF) x fSW PS= 0.5 x 13.8 x 1.54 x 40 x 10-9 x 4.5 x 105= 190 mW (87)

AC rms current loss, PCIN, in the input capacitor:

PCIN = IIN(rms)2x ESR = 0.7520.003 = 2 mW (negligible) (88)

DCR loss, PL, in the inductor

PL= IF2x DCR = 1.542x 0.05 = 120 mW (89)

Recirculating diode loss, PD= (1 - 0.3) x 1.54 x 0.4 = 430 mW

Current Sense Resistor Loss, PSNS = 293 mW

Electrical efficiency, η= PO/ (PO+ Sum of all loss terms) = 6.3 / (6.3 + 1.6) = 80%

Temperature Rise in the LM3406 IC is calculated as:

TLM3406 = (PC+ PG+ PS) x θJA = (0.53 + 0.06 + 0.19) x 50 = 39°C (90)

8.5 Thermal Considerations During Input Transients

The error amplifier of the LM3406 ensures that average LED current is controlled even at the transient load-

dump voltage of 40V, leaving thermal considerations as a primary design consideration during high voltage

transients. A review of the operating conditions at an input of 40V is still useful to make sure that the LM3406 die

temperature is not exceeded. Switching frequency drops to 325 kHz, the on-time drops to 350 ns, and the duty

cycle drops to 0.12. Repeating the calculations for conduction, gate charging and switching loss leads to a total

internal loss of 731 mW, and hence a die temperature rise of 37°C. The LM3406 should operate properly even if

the ambient temperature is as high a 85°C.

8.6 Layout Considerations

The performance of any switching converter depends as much upon the layout of the PCB as the component

selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and

minimum generation of unwanted EMI.

8.6.1 COMPACT LAYOUT

Parasitic inductance can be reduced by keeping the power path components close together and keeping the

area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In

particular, the switch node (where L1, D1, and the SW pin connect) should be just large enough to connect all

three components without excessive heating from the current it carries. The LM3406 family operates in two

distinct cycles whose high current paths are shown in :

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Layout Considerations (continued)

Buck Converter Current Loops

The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop

represents the high current path during the off-time.

8.6.2 GROUND PLANE AND SHAPE ROUTING

The diagram of is also useful for analyzing the flow of continuous current vs. the flow of pulsating currents. The

circuit paths with current flow during both the on-time and off-time are considered to be continuous current, while

those that carry current during the on-time or off-time only are pulsating currents. Preference in routing should be

given to the pulsating current paths, as these are the portions of the circuit most likely to emit EMI. The ground

plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as any other circuit

path. The continuous current paths on the ground net can be routed on the system ground plane with less risk of

injecting noise into other circuits. The path between the input source and the input capacitor and the path

between the recirculating diode and the LEDs/current sense resistor are examples of continuous current paths.

In contrast, the path between the recirculating diode and the input capacitor carries a large pulsating current.

This path should be routed with a short, thick shape, preferably on the component side of the PCB. Do not place

any vias near the anode of Schottky diode. Instead, multiple vias in parallel should be used right at the pad of the

input capacitor to connect the component side shapes to the ground plane. A second pulsating current loop that

is often ignored is the gate drive loop formed by the SW and BOOT pins and capacitor CB. To minimize this loop

and the EMI it generates, keep CBclose to the SW and BOOT pins.

8.6.3 CURRENT SENSING

The CS pin is a high-impedance input, and the loop created by RSNS, RZ(if used), the CS pin and ground should

be made as small as possible to maximize noise rejection. RSNS should therefore be placed as close as possible

to the CS and GND pins of the IC.

8.6.4 REMOTE LED ARRAYS

In some applications the LED or LED array can be far away (several inches or more) from the LM3406 family, or

on a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large

or separated from the rest of the converter, the output capacitor should be placed close to the LEDs to reduce

the effects of parasitic inductance on the AC impedance of the capacitor. The current sense resistor should

remain on the same PCB, close to the LM3406 family.

Remote LED arrays and high speed dimming with a parallel FET must be treated with special care. The parallel

dimming FET should be placed on the same board and/or heatsink as the LEDs to minimize the loop area

between them, as the switching of output current by the parallel FET produces a pulsating current just like the

switching action of the LM3406's internal power FET and the Schottky diode. shows the path that the inductor

current takes through the LED or through the dimming FET. To minimize the EMI from parallel FET dimming the

parasitic inductance of the loop formed by the LED and the dimming FET (where only the dark grey arrows

appear) should be reduced as much as possible. Parasitic inductance of a loop is mostly controlled by the loop

area, hence making this loop as physically small (short) as possible will reduce the inductance.

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

Buck Inductor is

Continuous

Current Source

LM3406, LM3406HV, LM3406HV-Q1

SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

www.ti.com

Layout Considerations (continued)

Parallel FET Dimming Current Loops

Table 1. BOM for Design Example 1

ID Part Number Type Size Parameters Qty Vendor

U1 LM3406 LED Driver eTSSOP-14 42V, 2A 1 NSC

L1 SLF10145T-220M1R-PF Inductor 10 x 10 x 4.5mm 22 µH, 1.9A, 59 mΩ1 TDK

D1 CMSH2-40 Schottky Diode SMB 40V, 2A 1 Central Semi

Cc, Cf VJ0603Y104KXXAT Capacitor 0603 100 nF 10% 2 Vishay

Cb VJ0603Y223KXXAT Capacitor 0603 22 nF 10% 1 Vishay

Cin1 C4532X7R1H475M Capacitor 1812 4.7 µF, 50V 2 TDK

Cin2

Co C2012X7R1E105M Capacitor 0805 1.0 µF, 25V 1 TDK

Rsns ERJ14RQFR13V Resistor 1210 0.13Ω1% 1 Panasonic

Ron CRCW08051433F Resistor 0805 143 kΩ1% 1 Vishay

Table 2. BOM for Design Example 2

ID Part Number Type Size Parameters Qty Vendor

U1 LM3406 LED Driver eTSSOP-14 42V, 2A 1 NSC

L1 SLF10145T-150M2R2-P Inductor 10 x 10 x 4.5mm 15 µH, 2.2A, 47 mΩ1 TDK

D1 CMSH2-60 Schottky Diode SMB 60V, 2A 1 Central Semi

Cc, Cf VJ0603Y104KXXAT Capacitor 0603 100 nF 10% 2 Vishay

Cb VJ0603Y223KXXAT Capacitor 0603 22 nF 10% 1 Vishay

Cin1 C3225X7R1H335M Capacitor 1210 3.3 µF, 50V 2 TDK

Cin2

Co C3216X7R1H105M Capacitor 1206 0.15 µF, 50V 1 TDK

Rsns ERJ14RQFR13V Resistor 1210 0.13Ω1% 1 Panasonic

Ron CRCW08051243F Resistor 0805 124 kΩ1% 1 Vishay

Rpd CRCW08051002F Resistor 0805 10 kΩ1% 1 Vishay

Table 3. Bill of Materials for Efficiency Curves

ID Part Number Type Size Parameters Qty Vendor

U1 LM3406 Buck LED Driver eTSSOP-14 42V, 1.5A 1 NSC

Q1 Si3458DV N-MOSFET SOT23-6 60V, 2.8A 1 Vishay

D1 CMSH2-60M Schottky Diode SMA 60V, 2A 1 Central Semi

L1 VLF10045T-330M2R3 Inductor 10 x 10 x 4.5mm 33 µH, 2.3A, 70 mΩ1 TDK

Cin1 C4532X7R1H685M Capacitor 1812 6.8 µF, 50V 2 TDK

Cin2

Co C3216X7R1H474M Capacitor 1206 470 nF, 50V 1 TDK

Cf ,Cc VJ0603Y104KXXAT Capacitor 0603 100 nF 10% 2 Vishay

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

D x (VIN ± 1.5)

fSW = + 1.75 x 10-7 x (VIN ± 1.5)

9.92 x 10-12 x (VO + 0.65) x RON

(D ± fSW x 1.75 x 10-7) x (VIN ± 1.5)

RON = 9.92 x 10-12 x fSW x (VO + 0.65)

9.92 x 10-12 x (VO + 0.65) x RON

tON = VIN ± 1.5 + 1.75 x 10-7

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SNVS512E –SEPTEMBER 2008–REVISED MARCH 2014

Table 3. Bill of Materials for Efficiency Curves (continued)

ID Part Number Type Size Parameters Qty Vendor

Cb VJ0603Y223KXXAT Capacitor 0603 22 nF 10% 1 Vishay

R3.5 ERJ6RQFR56V Resistor 0805 0.56Ω1% 1 Panasonic

R.7 ERJ6RQFR62V Resistor 0805 0.62Ω1% 1 Panasonic

R1 ERJ6RQFR30V Resistor 0805 0.3Ω1% 1 Panasonic

R1.5 ERJ6RQFR16V Resistor 0805 0.16Ω1% 1 Panasonic

Ron CRCW08051433F Resistor 0805 143kΩ1% 1 Vishay

Rpd Rout CRCW06031002F Resistor 0603 10 kΩ1% 2 Vishay

OFF* 160-1512 Terminal 0.062" 3 Cambion

DIM1

DIM2

VIN GND 160-1026 Terminal 0.094" 2 Cambion

CS/LED-

Vo/LED+

8.7 APPENDIX

The following expressions provide the best accuracy for users who wish to create computer-based simulations or

circuit calculators:

(91)

(92)

(93)

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9 Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Revision D (April 2013) to Revision E Page

• Added availability of LM3406HV-Q1, the automotive grade device throughout the data sheet. ........................................... 1

Changes from Revision C (May 2013) to Revision D Page

• Changed layout of National Data Sheet to TI format .......................................................................................................... 31

Product Folder Links: LM3406 LM3406HV LM3406HV-Q1

PACKAGE OPTION ADDENDUM

www.ti.com 10-Dec-2020

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead finish/

Ball material

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM3406HVMH/NOPB ACTIVE HTSSOP PWP 14 94 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3406

HVMH

LM3406HVMHX/NOPB ACTIVE HTSSOP PWP 14 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3406

HVMH

LM3406HVQMHQ1 ACTIVE HTSSOP PWP 14 94 RoHS & Green SN Level-1-260C-UNLIM -40 to 150 LM3406Q

HVMH

LM3406HVQMHXQ1 ACTIVE HTSSOP PWP 14 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 150 LM3406Q

HVMH

LM3406MH/NOPB ACTIVE HTSSOP PWP 14 94 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3406

LM3406MHX/NOPB ACTIVE HTSSOP PWP 14 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3406

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance

do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may

reference these types of products as "Pb-Free".

RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.

Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based

flame retardants must also meet the <=1000ppm threshold requirement.

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

PACKAGE OPTION ADDENDUM

www.ti.com 10-Dec-2020

Addendum-Page 2

(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two

lines if the finish value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

OTHER QUALIFIED VERSIONS OF LM3406HV, LM3406HV-Q1 :

•Catalog: LM3406HV

•Automotive: LM3406HV-Q1

NOTE: Qualified Version Definitions:

•Catalog - TI's standard catalog product

•Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

LM3406HVMHX/NOPB HTSSOP PWP 14 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1

LM3406HVQMHXQ1 HTSSOP PWP 14 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1

LM3406MHX/NOPB HTSSOP PWP 14 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 6-Mar-2019

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

LM3406HVMHX/NOPB HTSSOP PWP 14 2500 367.0 367.0 35.0

LM3406HVQMHXQ1 HTSSOP PWP 14 2500 367.0 367.0 35.0

LM3406MHX/NOPB HTSSOP PWP 14 2500 367.0 367.0 35.0

PACKAGE MATERIALS INFORMATION

www.ti.com 6-Mar-2019

Pack Materials-Page 2

MECHANICAL DATA

PWP0014A

www.ti.com

MXA14A (Rev A)

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