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LM26001
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LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
LM26001/-Q1 1.5-A Switching Regulator With High-Efficiency Sleep Mode
1 Features 3 Description
The LM26001 is a switching regulator designed for
1• LM26001-Q1 is an Automotive-Grade Product that the high-efficiency requirements of applications with
is AEC-Q100 Grade 1 Qualified (–40°C to +125°C standby modes. The device features a low-current
Operating Junction Temperature) sleep mode to maintain efficiency under light-load
• High-Efficiency Sleep Mode conditions and current-mode control for accurate
regulation over a wide input voltage range. Quiescent
• 40-µA Typical Iq in Sleep Mode current is reduced to 10 µA typically in shutdown
• 10-µA Typical Iq in Shutdown Mode mode and less than 40 µA in sleep mode. Forced
• 3.0-V Minimum Input Voltage PWM mode is also available to disable sleep mode.
• 4.0-V to 38-V Continuous Input Range The LM26001 can deliver up to 1.5 A of continuous
• 1.5% Reference Accuracy load current with a fixed current limit, through the
• Cycle-by-Cycle Current Limit internal N-channel switch. The part has a wide input
voltage range of 4.0 V to 38 V and can operate with
• Adjustable Frequency (150 kHz to 500 kHz) input voltages as low as 3 V during line transients.
• Synchronizable to an External Clock Operating frequency is adjustable from 150 kHz to
• Power Good Flag 500 kHz with a single resistor and can be
• Forced PWM Function synchronized to an external clock.
• Adjustable Soft-Start Other features include Power Good, adjustable soft-
• HTSSOP-16 Exposed Pad Package start, enable pin, input undervoltage protection, and
• Thermal Shut Down an internal bootstrap diode for reduced component
count.
2 Applications Device Information(1)
• Automotive Telematics PART NUMBER PACKAGE BODY SIZE (NOM)
• Navigation Systems LM26001 HTSSOP (16) 5.00 mm x 4.40 mm
• In-Dash Instrumentation LM26001-Q1
• Battery-Powered Applications (1) For all available packages, see the orderable addendum at
• Standby Power for Home Gateways/Set-top the end of the datasheet.
Boxes
Typical Application Circuit
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM26001
,
LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
www.ti.com
Table of Contents
7.4 Device Functional Modes........................................ 16
1 Features.................................................................. 18 Applications and Implementation ...................... 17
2 Applications ........................................................... 18.1 Application Information............................................ 17
3 Description............................................................. 18.2 Typical Application.................................................. 17
4 Revision History..................................................... 29 Power Supply Recommendations...................... 23
5 Pin Configuration and Functions......................... 310 Layout................................................................... 23
6 Specifications......................................................... 410.1 Layout Guidelines ................................................. 23
6.1 Absolute Maximum Ratings ...................................... 410.2 Layout Example .................................................... 24
6.2 ESD Ratings - LM26001........................................... 410.3 Thermal Considerations and TSD......................... 24
6.3 ESD Ratings - LM26001-Q1..................................... 411 Device and Documentation Support................. 25
6.4 Recommended Operating Conditions....................... 511.1 Documentation Support ........................................ 25
6.5 Thermal Information.................................................. 511.2 Related Links ........................................................ 25
6.6 Electrical Characteristics........................................... 511.3 Trademarks........................................................... 25
6.7 Typical Characteristics.............................................. 811.4 Electrostatic Discharge Caution............................ 25
7 Detailed Description............................................ 11 11.5 Glossary................................................................ 25
7.1 Overview................................................................. 11 12 Mechanical, Packaging, and Orderable
7.2 Functional Block Diagram....................................... 11 Information ........................................................... 25
7.3 Feature Description................................................. 12
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision H (November 2014) to Revision I Page
• Update made to the Power Dissipation description in Section 6.1 ....................................................................................... 4
• Changed ESD Ratings to ± and moved storage temp to Absolute Maximum Ratings.......................................................... 4
Changes from Revision G (April 2013) to Revision H Page
• Added Pin Configuration and Functions section, Handling Rating table, Feature Description section, Device
Functional Modes,Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information
section ................................................................................................................................................................................... 1
Changes from Revision F (April 2013) to Revision G Page
• Changed layout of National Data Sheet to TI format ........................................................................................................... 24
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Exposed Pad
Connect to GND
GND 8
VIN 1
VIN 2
PGOOD 3
EN 4
SS 5
COMP 6
FB 7 9 FREQ
10 FPWM
11 SYNC
12 VBIAS
13 VDD
14 BOOT
15 SW
16 SW
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,
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SNVS430I –MAY 2006–REVISED MARCH 2015
5 Pin Configuration and Functions
16-Pin
HTSSOP Package
Top View
Pin Functions
PIN I/O DESCRIPTION
NO. NAME
1 VIN A Power supply input
2 VIN A Power supply input
3 PGOOD O Power Good pin. An open-drain output which goes high when the output voltage is greater than
92% of nominal.
4 EN I Enable is an analog level input pin. When pulled below 0.8 V, the device enters shutdown mode.
5 SS A Soft-start pin. Connect a capacitor from this pin to GND to set the soft-start time.
6 COMP A Compensation pin. Connect to a resistor capacitor pair to compensate the control loop.
7 FB A Feedback pin. Connect to a resistor divider between Vout and GND to set output voltage.
8 GND G Ground
9 FREQ A Frequency adjust pin. Connect a resistor from this pin to GND to set the operating frequency.
10 FPWM I FPWM is a logic level input pin. For normal operation, connect to GND. When pulled high, sleep
mode operation is disabled.
11 SYNC I Frequency synchronization pin. Connect to an external clock signal for synchronized operation.
SYNC must be pulled low for non-synchronized operation.
12 VBIAS A Connect to an external 3-V or greater supply to bypass the internal regulator for improved efficiency.
If not used, VBIAS should be tied to GND.
13 VDD A The output of the internal regulator. Bypass with a minimum 1.0-µF capacitor.
14 BOOT A Bootstrap capacitor pin. Connect a 0.1-µF minimum ceramic capacitor from this pin to SW to
generate the gate drive bootstrap voltage.
15 SW A Switch pin. The source of the internal N-channel switch.
16 SW A Switch pin. The source of the internal N-channel switch.
EP EP G Exposed Pad thermal connection. Connect to GND.
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6 Specifications
6.1 Absolute Maximum Ratings(1)(2)
MIN MAX UNIT
VIN –0.3 40 V
SW(3) –0.5 40 V
VDD –0.3 7 V
VBIAS –0.3 10 V
FB –0.3 6 V
Voltages BOOT SW-0.3 SW+7 V
from the
indicated PGOOD –0.3 7 V
pins to FREQ –0.3 7 V
GND SYNC –0.3 7 V
EN –0.3 40 V
FPWM –0.3 y7 V
SS –0.3 7 V
Power Dissipation(4)(5) 2.6 W
Recomme Vapor Phase (70s) 215 °C
nded Lead Infrared (15s) 220 °C
Temperatu
re
Storage Tstg –65 150 °C
temperatur
e
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Recommended Operating Conditions indicate
conditions for which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications
and test conditions, see the Electrical Characteristics.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) The absolute maximum specification applies to DC voltage. An extended negative voltage limit of -2V applies for a pulse of up to 1 µs,
and –1 V for a pulse of up to 20 µs.
(4) The maximum allowable power dissipation is a function of the maximum junction temperature, TJ_MAX, the junction-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD_MAX = (TJ_MAX - TA) /θJA. The maximum power dissipation of 2.6W is determined using TA = 25°C, θJA = 38°C/W, and
TJ_MAX = 125°C. The number stated here reflects the maximum power dissipation for the package and not the device.
(5) For Device Power Dissipation, please refer to section 10.3.
6.2 ESD Ratings - LM26001 VALUE UNIT
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins(1) ± 2 kV
Charged device model (CDM), per JEDEC specification JESD22-C101, ± 1
V(ESD) Electrostatic discharge all pins(2)
Machine model ± 200 V
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 ESD Ratings - LM26001-Q1 VALUE UNIT
Human body model (HBM), per AEC Q100-002(1) ± 2
Corner pins (1, 8, 9, and ± 1 kV
Charged device model (CDM), per AEC 16)
V(ESD) Electrostatic discharge Q100-011 Other pins ± 1
Machine model ± 200 V
(1) AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
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6.4 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted) MIN MAX UNIT
Operating Junction Temp. –40 125 °C
Supply Voltage(1) 3.0 38 V
(1) Below 4.0-V input, power dissipation may increase due to increased RDS(ON). Therefore, a minimum input voltage of 4.0 V is required to
operate continuously within specification. A minimum of 3.9 V (typical) is also required for startup.
6.5 Thermal Information LM26001
THERMAL METRIC(1) PWP UNIT
16 PINS
RθJA Junction-to-ambient thermal resistance 38.8
RθJC(top) Junction-to-case (top) thermal resistance 23.0
RθJB Junction-to-board thermal resistance 16.7 °C/W
ψJT Junction-to-top characterization parameter 0.6
ψJB Junction-to-board characterization parameter 16.4
RθJC(bot) Junction-to-case (bottom) thermal resistance 1.7
(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
6.6 Electrical Characteristics
Unless otherwise stated, Vin=12 V. Minimum and Maximum limits are ensured through test, design, or statistical correlation.
Typical values represent the most likely parametric norm at TJ= 25°C, and are provided for reference purposes only.(1)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
SYSTEM
ISD(2) Shutdown Current EN = 0 V 10.8 µA
EN = 0 V, –40°C ≤TJ≤125°C 20
Iq_Sleep_VB(2) Quiescent Current Sleep mode, VBIAS = 5 V 38 µA
Sleep mode, VBIAS = 5 V, –40°C 70
≤TJ≤125°C
Iq_Sleep_VDD Quiescent Current Sleep mode, VBIAS = GND 75 µA
Sleep mode, VBIAS = GND, 125
–40°C ≤TJ≤125°C
Iq_PWM_VB Quiescent Current PWM mode, VBIAS = 5 V 150 230 µA
Iq_PWM_VDD Quiescent Current PWM mode, VBIAS = GND 0.65 0.85 mA
IBIAS_Sleep(2) Bias Current Sleep mode, VBIAS = 5 V 33 µA
Sleep mode, VBIAS = 5 V, –40°C 85
≤TJ≤125°C
IBIAS_PWM Bias Current PWM mode, VBIAS = 5 V 0.5 0.70 mA
VFB Feedback Voltage 5 V < Vin < 38 V 1.234 V
5 V < Vin < 38 V, –40°C ≤TJ≤1.2155 1.2525
125°C
IFB FB Bias Current ±200 nA
ΔVOUT/ΔVIN Vout line regulation 5 V < Vin < 38 V 0.001 %/V
ΔVOUT/ΔIOUT Vout load regulation 0.8 V < VCOMP < 1.15 V 0.07%
VDD VDD output voltage 7 V < Vin < 35 V, IVDD= 0 mA to 5 5.95 V
mA
7 V < Vin < 35 V, IVDD= 0 mA to 5 5.50 6.50
mA, –40°C ≤TJ≤125°C
(1) All room temperature limits are 100% production tested. All limits at temperature extremes are ensured through correlation using
standard Statistical Quality Control (SQC) methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Iq and ISD specify the current into the VIN pin. IBIAS is the current into the VBIAS pin when the VBIAS voltage is greater than 3 V. All
quiescent current specifications apply to non-switching operation.
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Electrical Characteristics (continued)
Unless otherwise stated, Vin=12 V. Minimum and Maximum limits are ensured through test, design, or statistical correlation.
Typical values represent the most likely parametric norm at TJ= 25°C, and are provided for reference purposes only.(1)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
ISS_Source Soft-start source current 2.2 µA
–40°C ≤TJ≤125°C 1.5 4.6
Vbias_th VBIAS On Voltage Specified at IBIAS = 92.5% of full 2.64 2.9 3.07 V
value
SWITCHING
RDS(ON) Switch on Resistance Isw = 1A 0.2 Ω
Isw = 1A, –40°C ≤TJ≤125°C 0.12 0.42
Isw_off Switch off state leakage current Vin = 38 V, VSW = 0 V 0.002 µA
Vin = 38 V, VSW = 0 V, –40°C ≤5.0
TJ≤125°C
fsw Switching Frequency RFREQ = 62k, 124k, 240k ±10%
VFREQ FREQ voltage 1.0 V
fSW range Switching Frequency range –40°C ≤TJ≤125°C 150 500 kHz
VSYNC Sync pin threshold SYNC rising 1.2 V
SYNC rising, –40°C ≤TJ≤125°C 1.6
SYNC falling 1.1
SYNC falling, –40°C ≤TJ≤125°C 0.8
Sync pin hysteresis 114 mV
ISYNC SYNC leakage current 6 nA
FSYNC_UP Upper frequency synchronization range As compared to nominal fSW, 30%
–40°C ≤TJ≤125°C
FSYNC_DN Lower frequency synchronization range As compared to nominal fSW, –20%
–40°C ≤TJ≤125°C
TOFFMIN Minimum Off-time 365 ns
TONMIN Minimum On-time 155 ns
THSLEEP_HYS Sleep mode threshold hysteresis VFB rising, % of THWAKE 101.2%
THWAKE Wake up threshold Measured at falling FB, COMP = 1.234 V
0.6 V
IBOOT BOOT pin leakage current BOOT = 16 V, SW = 10 V 0.0006 µA
BOOT = 16 V, SW = 10 V, –40°C 5.0
≤TJ≤125°C
PROTECTION
ILIMPK Peak Current Limit 2.5 A
–40°C ≤TJ≤125°C 1.85 3.2
VFB_SC Short circuit frequency foldback Measured at FB falling 0.87 V
threshold
F_min_sc Min Frequency in foldback VFB < 0.3 V 71 kHz
VTH_PGOOD Power Good Threshold Measured at FB, PGOOD rising 92%
Measured at FB, PGOOD rising, 89% 95%
–40°C ≤TJ≤125°C
PGOOD hysteresis 2% 7% 8%
IPGOOD_HI PGOOD leakage current PGOOD = 5 V 0.2 nA
RDS_PGOOD PGOOD on resistance PGOOD sink current = 500 µA 64 Ω
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Electrical Characteristics (continued)
Unless otherwise stated, Vin=12 V. Minimum and Maximum limits are ensured through test, design, or statistical correlation.
Typical values represent the most likely parametric norm at TJ= 25°C, and are provided for reference purposes only.(1)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
VUVLO Under-voltage Lock-Out Threshold Vin falling , shutdown, VDD = VIN 2.9 V
Vin falling , shutdown, VDD = VIN, 2.60 3.20
–40°C ≤TJ≤125°C
Vin rising, soft-start, VDD = VIN 3.9
Vin rising, soft-start, VDD = VIN, 3.60 4.20
–40°C ≤TJ≤125°C
TSD Thermal Shutdown Threshold 160 °C
θJA Thermal resistance Power dissipation = 1W, 0 lfpm air 38 °C/W
flow
LOGIC
VthEN Enable Threshold voltage 1.2 V
–40°C ≤TJ≤125°C 0.8 1.4
Enable hysteresis 120 mV
IEN_Source EN source current EN = 0 V 4.5 µA
VTH_FPWM FPWM threshold 1.2 V
–40°C ≤TJ≤125°C 0.8 1.6
IFPWM FPWM leakage current FPWM = 5 V 35 nA
EA
gm Error amp trans-conductance 670 µmho
–40°C ≤TJ≤125°C 400 1000
ICOMP COMP source current VCOMP = 0.9 V 56 µA
COMP sink current VCOMP = 0.9 V 56 µA
VCOMP COMP pin voltage range 0.64 1.27 V
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VIN (V)
2.5
2.7
2.9
3.1
3.3
3.5
3.7
3.9
4.1
TEMPERATURE (ºC)
-40 -20 0 20 40 60 80 100 120 140
On Threshold
Off Threshold
100
101
102
-40 0 40 80 120
TEMPERATURE (ºC)
SWITCHING FREQUENCY (%)
100
98
99
140
60
20
-20
TEMPERATURE (ºC)
CURRENT (PA)
50
70
90
-40 0 40 80 120
100
20
30
140
6020-20
40
60
80 IQ
(VBIAS=0V)
IQ
(VBIAS=5V)
IVBIAS
(VBIAS=5V)
TEMPERATURE (ºC)
VFB (V)
-40 0 40 80 120
100 140
60
20
-20
1.228
1.230
1.232
1.234
1.236
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6.7 Typical Characteristics
Unless otherwise specified the following conditions apply: VIN = 12 V, TJ= 25°C.
Figure 1. VFB vs Temperature Figure 2. VFB vs Vin (IDC = 300 mA)
Figure 4. IQ and IVBIAS vs Temperature (PWM Mode)
Figure 3. IQ and IVBIAS vs Temperature (Sleep Mode)
Figure 5. Normalized Switching Frequency vs Temperature Figure 6. UVLO Threshold vs Temperature (VDD = VIN)
(300 kHz)
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1 Ps/DIV
Vout
1V/Div
PGOOD
5V/Div
SS
1V/Div
EN
10V/Div
Vout
40 mV/Div
Iout
500 mA/Div
200 Ps/DIV
EFFICIENCY (%)
1
IDC (mA)
10 100 1000 10000
5Vout
3.3Vout
FPWM
mode
50
60
70
80
90
100
EFFICIENCY (%)
1
IDC (mA)
10 100 1000 10000
5Vout
3.3Vout
FPWM
mode
50
60
70
80
90
100
VFB (V)
SWITCHING FREQUENCY (kHz)
0
50
100
150
200
250
300
350
0.2 0.4 0.6 0.8 1.0 1.2
TEMPERATURE (ºC)
ILIM PEAK (A)
2.0
2.2
2.4
2.6
2.8
3.0
-40 -20 0 20 40 60 80 100 120 140
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Typical Characteristics (continued)
Unless otherwise specified the following conditions apply: VIN = 12 V, TJ= 25°C.
Figure 7. Peak Current Limit vs Temperature Figure 8. Short Circuit Foldback Frequency vs VFB (325 kHz
Nominal)
Figure 9. Efficiency vs Load Current (330 kHz) Figure 10. Efficiency vs Load Current (500 kHz)
Figure 11. Startup Waveforms Figure 12. Load Transient Response
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0
50
100
150
200
250
300
350
400
450
500
3 3.5 4 4.5 5 5.5
VIN (V)
VIN-VOUT DROPOUT (mV)
1A 125°C
1A 25°C
1A -40°C
40mA -40°C
40mA
25 to 125°C
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Typical Characteristics (continued)
Unless otherwise specified the following conditions apply: VIN = 12 V, TJ= 25°C.
Figure 13. Low Input Voltage Dropout Nominal VOUT = 5 V
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sleep
wake
0.6V
2 PA
+
-
0.9V
V clamp
+
-
+
-
SS
logic
+
-
TSD
VDD_low
ss end
blanking
qn
qn
fpwm
Sync and
bootstrap
control
fpwm
LG
LG
5 PA
VDD_low
EN
FB
SW
SW
FPWM
COMP
PGOOD
SS
GND
SYNC
BOOT
VIN
VBIAS
VDD
VIN
SD
+
-
+
-
+
-
Clock / Sync
BG
BG
+
I Sense
PWM
Comp
BG
IREF
TSD
FPWM / Sleep
Peak Current
Control
Sleep
Reset
EA
PG
0.92BG
+
-
Corrective
Ramp
PWM Control
Logic
+
-
+
-Sleep
Set
LDO
UVLO
on
Switchover
control
VREG
sleep
EP
frequency
foldback
FREQ
soft start on
SD
ff
ff
1 Ps/DIV
Vout
10 mV/Div
IL
500 mA/Div
ID
1A/Div
VSW
5V/Div
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7 Detailed Description
7.1 Overview
The LM26001 is a current mode PWM buck regulator. At the beginning of each clock cycle, the internal high-side
switch turns on, allowing current to ramp up in the inductor. The inductor current is internally monitored during
each switching cycle. A control signal derived from the inductor current is compared to the voltage control signal
at the COMP pin, derived from the feedback voltage. When the inductor current reaches the threshold, the high-
side switch is turned off and inductor current ramps down. While the switch is off, inductor current is supplied
through the catch diode. This cycle repeats at the next clock cycle. In this way, duty cycle and output voltage are
controlled by regulating inductor current. Current mode control provides superior line and load regulation. Other
benefits include cycle by cycle current limiting and a simplified compensation scheme. Typical PWM waveforms
are shown in Figure 14.
Figure 14. PWM Waveforms 1-A Load, Vin = 12 V
7.2 Functional Block Diagram
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010 20 30 40
VIN (V)
0
50
100
150
200
250
IDC (mA)
22 PH
15 PH
22 PH
15 PH
500 kHz
330 kHz
ISleep = Imin + 0.13 PVin - Vout
L
2
xfsw x L
D x 2 x (Vin ± Vout)
Vout
50 mV/Div
VSW
5V/Div
IL
200 mA/Div
100 Ps/DIV
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7.3 Feature Description
7.3.1 Sleep Mode
In light load conditions, the LM26001 automatically switches into sleep mode for improved efficiency. As loading
decreases, the voltage at FB increases and the COMP voltage decreases. When the COMP voltage reaches the
0.6-V (typical) clamp threshold, and the FB voltage rises 1% above nominal, sleep mode is enabled and
switching stops. The regulator remains in sleep mode until the FB voltage falls to the reset threshold, at which
point switching resumes. This 1% FB window limits the corresponding output ripple to approximately 1% of
nominal output voltage. The sleep cycle will repeat until load current is increased. Figure 15 shows typical
switching and output voltage waveforms in sleep mode.
Figure 15. Sleep Mode Waveforms 25-mA Load, Vin = 12 V
In sleep mode, quiescent current is reduced to less than 40 µA when not switching. The DC sleep mode
threshold can be calculated according to the equation below:
(1)
Where Imin = Ilim/16 (2.5A/16 typically) and D = duty cycle, defined as (Vout + Vdiode)/Vin.
When load current increases above this limit, the LM26001 is forced back into normal PWM operation. The sleep
mode threshold varies with frequency, inductance, and duty cycle as shown in Figure 16.
Figure 16. Sleep Mode Threshold vs Vin Vout = 3.3 V
Below the sleep threshold, decreasing load current results in longer sleep cycles, which can be quantified as
shown below:
Dwake = Iload/Isleep (2)
Where Dwake is the percentage of time awake when the load current is below the sleep threshold.
Sleep mode combined with low IQ operation minimizes the input supply current. Input supply current in sleep
mode can be calculated based on the wake duty cycle, as shown below:
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10 mV/Div
VSW
5V/Div
IL
200 mA/Div
1 Ps/DIV
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Feature Description (continued)
Iin = Iq + (IQG x Dwake) + (Io x D) (3)
Where IQG is the gate drive current, calculated as:
IQG = (4.6 x 10-9)xfSW
And Io is the sum of Iload, Ibias, and current through the feedback resistors.
Because this calculation applies only to sleep mode, use the Iq_Sleep_VB and IBIAS_SLEEP values from the Electrical
Characteristics. If VBIAS is connected to ground, use the same equation with Ibias equal to zero and Iq_Sleep_VDD.
7.3.2 FPWM
Pulling the FPWM pin high disables sleep mode and forces the LM26001 to always operate in PWM mode. Light
load efficiency is reduced in PWM mode, but switching frequency remains stable. The FPWM pin can be
connected to the VDD pin to pull it high. In FPWM mode, under light load conditions, the regulator operates in
discontinuous conduction mode (DCM) . In discontinuous conduction mode, current through the inductor starts at
zero and ramps up to its peak, then ramps down to zero again. Until the next cycle, the inductor current remains
at zero. At nominal load currents, in FPWM mode, the device operates in continuous conduction mode, where
positive current always flows in the inductor. Typical discontinuous operation waveforms are shown in Figure 17.
Figure 17. Discontinuous Mode Waveforms 75-mA Load, Vin = 12 V
At very light load, in FPWM mode, the LM26001 may enter sleep mode. This is to prevent an over-voltage
condition from occurring. However, the FPWM sleep threshold is much lower than in normal operation.
7.3.3 Enable
The LM26001 provides a shutdown function via the EN pin to disable the device when the output voltage does
not need to be maintained. EN is an analog level input with typically 120 mV of hysteresis. The device is active
when the EN pin is above 1.2 V (typical) and in shutdown mode when EN is below this threshold. When EN goes
high, the internal VDD regulator turns on and charges the VDD capacitor. When VDD reaches 3.9 V (typical), the
soft-start pin begins to source current. In shutdown mode, the VDD regulator shuts down and total quiescent
current is reduced to 10 µA (typical). Because the EN pin sources 4.5 µA (typical) of pull-up current, this pin can
be left open for always-on operation. When open, EN will be pulled up to VIN.
If EN is connected to VIN, it must be connected through a 10 kΩresistor to limit noise spikes. EN can also be
driven externally with a maximum voltage of 38V or VIN + 15V, whichever is lower.
7.3.4 Soft-Start
The soft-start feature provides a controlled output voltage ramp up at startup. This reduces inrush current and
eliminates output overshoot at turn-on. The soft-start pin, SS, must be connected to GND through a capacitor. At
power-on, enable, or UVLO recovery, an internal 2.2 µA (typical) current charges the soft-start capacitor. During
soft-start, the error amplifier output voltage is controlled by both the soft-start voltage and the feedback loop. As
the SS pin voltage ramps up, the duty cycle increases proportional to the soft-start ramp, causing the output
voltage to ramp up. The rate at which the duty cycle increases depends on the capacitance of the soft-start
capacitor. The higher the capacitance, the slower the output voltage ramps up. The soft-start capacitor value can
be calculated with the following equation:
Copyright © 2006–2015, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Links: LM26001 LM26001-Q1
(Vin ± Vout) x Vout
fsw x L x Vin
Iripple =
Iripple
2
Iloadmax = 1.85A -
Iss x tss
1.234V
Css =
LM26001
,
LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
www.ti.com
Feature Description (continued)
(4)
Where tss is the desired soft-start time and Iss is the soft-start source current. During soft-start, current limit and
synchronization remain in effect, while sleep mode and frequency foldback are disabled. Soft-start mode ends
when the SS pin voltage reaches 1.23 V typical. At this point, output voltage control is transferred to the FB pin
and the SS pin is discharged.
7.3.5 Current Limit
The peak current limit is set internally by directly measuring peak inductor current through the internal switch. To
ensure accurate current sensing, VIN should be bypassed with a minimum 1-µF ceramic capacitor placed directly
at the pin.
When the inductor current reaches the current limit threshold, the internal FET turns off immediately allowing
inductor current to ramp down until the next cycle. This reduction in duty cycle corresponds to a reduction in
output voltage.
The current limit comparator is disabled for less than 100 ns at the leading edge for increased immunity to
switching noise.
Because the current limit monitors peak inductor current, the DC load current limit threshold varies with
inductance and frequency. Assuming a minimum current limit of 1.85A, maximum load current can be calculated
as follows:
(5)
Where Iripple is the peak-to-peak inductor ripple current, calculated as shown below:
(6)
To find the worst case (lowest) current limit threshold, use the maximum input voltage and minimum current limit
specification.
During high over-current conditions, such as output short circuit, the LM26001 employs frequency foldback as a
second level of protection. If the feedback voltage falls below the short circuit threshold of 0.9 V, operating
frequency is reduced, thereby reducing average switch current. This is especially helpful in short circuit
conditions, when inductor current can rise very high during the minimum on-time. Frequency reduction begins at
20% below the nominal frequency setting. The minimum operating frequency in foldback mode is 71 kHz typical.
If the FB voltage falls below the frequency foldback threshold during frequency synchronized operation, the
SYNC function is disabled. Operating frequency versus FB voltage in short circuit conditions is shown in the
Typical Characteristics section.
Under conditions where the on time is close to minimum (less than 200 nsec typically), such as high input
voltage and high switching frequency, the current limit may not function properly. This is because the current limit
circuit cannot reduce the on-time below minimum which prevents entry into frequency foldback mode. There are
two ways to ensure proper current limit and foldback operation under high input voltage conditions. First, the
operating frequency can be reduced to increase the nominal on time. Second, the inductor value can be
increased to slow the current ramp and reduce the peak over-current.
7.3.6 Frequency Adjustment and Synchronization
The switching frequency of the LM26001 can be adjusted between 150 kHz and 500 kHz using a single external
resistor. This resistor is connected from the FREQ pin to ground as shown in the typical application. The resistor
value can be calculated with the following empirically derived equation:
RFREQ = (6.25 x 1010) x fSW-1.042 (7)
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Product Folder Links: LM26001 LM26001-Q1
Sync_Dmin
t
1 - fnom
fsync
050 100 150 200 250 300
RFREQ (k:)
100
200
300
400
500
600
SWITCHING FREQUENCY (kHz)
LM26001
,
LM26001-Q1
www.ti.com
SNVS430I –MAY 2006–REVISED MARCH 2015
Feature Description (continued)
Figure 18. Switching Frequency vs RFREQ
The switching frequency can also be synchronized to an external clock signal using the SYNC pin. The SYNC
pin allows the operating frequency to be varied above and below the nominal frequency setting. The adjustment
range is from 30% above nominal to 20% below nominal. External synchronization requires a 1.2V (typical) peak
signal level at the SYNC pin. The FREQ resistor must always be connected to initialize the nominal operating
frequency. The operating frequency is synchronized to the falling edge of the SYNC input. When SYNC goes
low, the high-side switch turns on. This allows any duty cycle to be used for the sync signal when synchronizing
to a frequency higher than nominal. When synchronizing to a lower frequency, however, there is a minimum duty
cycle requirement for the SYNC signal, given in the equation below:
(8)
Where fnom is the nominal switching frequency set by the FREQ resistor, and fsync is a square wave. If the
SYNC pin is not used, it must be pulled low for normal operation. A 10 kΩpull-down resistor is recommended to
protect against a missing sync signal. Although the LM26001 is designed to operate at up to 500 kHz, maximum
load current may be limited at higher frequencies due to increased temperature rise. See the Thermal
Considerations and TSD section.
7.3.7 VBIAS
The VBIAS pin is used to bypass the internal regulator which provides the bias voltage to the LM26001. When
the VBIAS pin is connected to a voltage greater than 3 V, the internal regulator automatically switches over to the
VBIAS input. This reduces the current into VIN (Iq) and increases system efficiency. Using the VBIAS pin has the
added benefit of reducing power dissipation within the device.
For most applications where 3 V < Vout < 10V, VBIAS can be connected to Vout. If not used, VBIAS should be
tied to GND.
If VBIAS drops below 2.9 V (typical), the device automatically switches over to supply the internal bias voltage
from Vin.
7.3.8 Low VIN Operation and UVLO
The LM26001 is designed to remain operational during short line transients when input voltage may drop as low
as 3.0 V. Minimum nominal operating input voltage is 4.0 V. Below this voltage, switch RDS(ON) increases, due to
the lower gate drive voltage from VDD. The minimum voltage required at VDD is approximately 3.5 V for normal
operation within specification.
VDD can also be used as a pull-up voltage for functions such as PGOOD and FPWM. Note that if VDD is used
externally, the pin is not recommended for loads greater than 1 mA.
Copyright © 2006–2015, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LM26001 LM26001-Q1
Vout
20 mV/Div
VSW
2V/Div
IL
100 mA/Div
4 Ps/DIV
LM26001
,
LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
www.ti.com
Feature Description (continued)
If the input voltage approaches the nominal output voltage, the duty cycle is maximized to hold up the output
voltage. In this mode of operation, once the duty cycle reaches its maximum, the LM26001 can skip a maximum
of seven off pulses, effectively increasing the duty cycle and thus minimizing the dropout from input to output.
Typical off-pulse skipping waveforms are shown in Figure 19.
Figure 19. Off-pulse Skipping Waveforms Vin = 3.5 V, Vnom = 3.3 V, fnom = 305 kHz
UVLO is sensed at both VIN and VDD, and is activated when either voltage falls below 2.9 V (typical). Although
VDD is typically less than 200 mV below VIN, it will not discharge through VIN. Therefore when the VIN voltage
drops rapidly, VDD may remain high, especially in sleep mode. For fast line voltage transients, using a larger
capacitor at the VDD pin can help to hold off a UVLO shutdown by extending the VDD discharge time. By
holding up VDD, a larger cap can also reduce the RDS(ON) (and dropout voltage) in low VIN conditions.
Alternately, under heavy loading the VDD voltage can fall several hundred mV below VIN. In this case, UVLO
may be triggered by VDD even though the VIN voltage is above the UVLO threshold.
When UVLO is activated the LM26001 enters a standby state in which VDD remains charged. As input voltage
and VDD voltage rise above 3.9 V (typical) the device will restart from softstart mode.
7.3.9 PGOOD
A power good pin, PGOOD, is available to monitor the output voltage status. The pin is internally connected to
an open-drain MOSFET, which remains open while the output voltage is within operating range. PGOOD goes
low (low impedance to ground) when the output falls below 85% of nominal or EN is pulled low. When the output
voltage returns to within 92% of nominal, as measured at the FB pin, PGOOD returns to a high state. For
improved noise immunity, there is a 5 µs delay between the PGOOD threshold and the PGOOD pin going low.
7.4 Device Functional Modes
The LM26001 has three basic operation mode: Shutdown, Sleep or light load operation and full operation.
The part enters shutdown mode when the EN pin is pulled low. In this mode the converter is disabled and the
quiescent current minimized See Enable for more details.
The part enters sleep mode when the converter is active (EN high) and the output current is low. Sleep mode is
activated as the COMP voltage naturally falls below a typical 0.6V threshold in light load operation. When
operating in sleep mode, the switching events of the converter are reduced in order to lower the current
consumption of the system. Forcing the FPWM pin high will prevent sleep mode operation. For details of
operation in sleep mode as well as entering and exiting sleep mode. See Sleep Mode.
When the part in enabled and the output load is higher, the part will be in full PWM operation.
In addition to these normal functioning modes, the LM26001 has a frequency foldback operating mode which
reduces the operating frequency to protect from short circuits. See Current Limit.
16 Submit Documentation Feedback Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001
SW
SW
BOOT
FB
COMP
VDD
GND
FPWM
FREQ
SS
SYNC
EN
PGOOD
VIN
2
3
13
9
144
10
6
11
5
8
7
15
16
VIN : 4V - 38V
VOUT : 3.3V
VIN VBIAS
1 12
C1
3.3 PF
50V
C5
10 nF R3
120k
1%
C8
4.7 nF
R5
15k
L
22 PH
3.5A
C4
0.1 PF
R2
33k
1%
R1
56k
1%
C6
100 PF
12 m:
C3
10 PF
R4
D1
3A
60V
VDD
17
EP
R6
10k
EN
SYNC
+C2
47 PF
50V
PGOOD +
C9
47 pF
200k
Ref # Manufacturer Part Number
C1
C2
C6
D1
L1
TDK
Panasonic
Panasonic
NIEC
TDK
3225JB1H335K
EEVFK1H470P
EEFVEOK101R
NSQ03A06
SLF12565T-
220M3R5
LM26001
,
LM26001-Q1
www.ti.com
SNVS430I –MAY 2006–REVISED MARCH 2015
8 Applications and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM26001 offers efficient step-down function meeting the requirements of automotive applications. Current
mode control ensures smooth and safe operation thanks to its cycle by cycle current limiting function. Its low
minimum ON time allows it to operate over a wide range of output voltages. The following sections detail the
steps required for designing a successful application with the LM26001 from component selection to layout.
8.2 Typical Application
Figure 20 shows a complete typical application schematic. The components have been selected based on the
design criteria given in the following sections.
Figure 20. Example Circuit 1.5A Max, 305 kHz
8.2.1 Design Requirements
The following parameters are needed to properly design the application and size the components:
PARAMETERS VALUES
Vout Output voltage
Vin min Maximum input voltage
Vin max Minimum input voltage
Iout max Maximum output current
Fsw Switching Frequency
Fbw Bandwidth of the converter
8.2.2 Detailed Design Procedure
8.2.2.1 Setting Output Voltage
The output voltage is set by the ratio of a voltage divider at the FB pin as shown in the typical application. The
resistor values can be determined by the following equation:
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Product Folder Links: LM26001 LM26001-Q1
'Vt
'It
ReMAX =
(Vin ± Vout) x Vout x Re
Vin x fsw x Vrip
LMIN =
Iripple
2
Ipeak = Iload +
(Vin ± Vout) x Vout
fsw x L x Vin
Iripple =
R2 = Vout
Vfb
©
§¹
·
R1
-1
LM26001
,
LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
www.ti.com
(9)
Where Vfb = 1.234V typically.
A maximum value of 150kΩis recommended for the sum of R1 and R2.
As input voltage decreases towards the nominal output voltage, the LM26001 can skip up to seven off-pulses as
described in the Low VIN Operation and UVLO section. In low output voltage applications, if the on-time reaches
TonMIN, the device will skip on-pulses to maintain regulation. There is no limit to the number of pulses that are
skipped. In this mode of operation, however, output ripple voltage may increase slightly.
8.2.2.2 Inductor
The output inductor should be selected based on inductor ripple current. The amount of inductor ripple current
compared to load current, or ripple content, is defined as Iripple/Iload. Ripple content should be less than 40%.
Inductor ripple current, Iripple, can be calculated as shown below:
(10)
Larger ripple content increases losses in the inductor and reduces the effective current limit.
Larger inductance values result in lower output ripple voltage and higher efficiency, but a slightly degraded
transient response. Lower inductance values allow for smaller case size, but the increased ripple lowers the
effective current limit threshold.
Remember that inductor value also affects the sleep mode threshold as shown in Figure 16.
When choosing the inductor, the saturation current rating must be higher than the maximum peak inductor
current and the RMS current rating should be higher than the maximum load current. Peak inductor current,
Ipeak, is calculated as:
(11)
For example, at a maximum load of 1.5A and a ripple content of 40%, peak inductor current is equal to 1.8A
which is safely below the minimum current limit of 1.85A. By increasing the inductor size, ripple content and peak
inductor current are lowered, which increases the current limit margin.
The size of the output inductor can also be determined using the desired output ripple voltage, Vrip. The
equation to determine the minimum inductance value based on Vrip is as follows:
(12)
Where Re is the ESR of the output capacitors, and Vrip is a peak-to-peak value. This equation assumes that the
output capacitors have some amount of ESR. It does not apply to ceramic output capacitors.
If this method is used, ripple content should still be verified to be less than 40%.
8.2.2.3 Output Capacitor
The primary criterion for selecting an output capacitor is equivalent series resistance, or ESR.
ESR (Re) can be selected based on the requirements for output ripple voltage and transient response. Once an
inductor value has been selected, ripple voltage can be calculated for a given Re using the equation above for
Lmin. Lower ESR values result in lower output ripple.
Re can also be calculated from the following equation:
(13)
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Irms = Vin
Vout x (Vin ± Vout)
Iload x
CMIN = ('Vt)2 - ('It x Re)2
'Vt -
L x ©
§
©
§
Vout x Re
2
LM26001
,
LM26001-Q1
www.ti.com
SNVS430I –MAY 2006–REVISED MARCH 2015
Where ΔVt is the allowed voltage excursion during a load transient, and ΔIt is the maximum expected load
transient. If the total ESR is too high, the load transient requirement cannot be met, no matter how large the
output capacitance. If the ESR criteria for ripple voltage and transient excursion cannot be met, more capacitors
should be used in parallel. For non-ceramic capacitors, the minimum output capacitance is of secondary
importance, and is determined only by the load transient requirement.
If there is not enough capacitance, the output voltage excursion will exceed the maximum allowed value even if
the maximum ESR requirement is met. The minimum capacitance is calculated as follows:
(14)
It is assumed the total ESR, Re, is no greater than ReMAX. Also, it is assumed that L has already been selected.
Generally speaking, the output capacitance requirement decreases with Re, ΔIt, and L. A typical value greater
than 100 µF works well for most applications.
8.2.2.4 Input Capacitor
In a switching converter, very fast switching pulse currents are drawn from the input rail. Therefore, input
capacitors are required to reduce noise, EMI, and ripple at the input to the LM26001. Capacitors must be
selected that can handle both the maximum ripple RMS current at highest ambient temperature as well as the
maximum input voltage. The equation for calculating the RMS input ripple current is shown below:
(15)
For noise suppression, a ceramic capacitor in the range of 1.0 µF to 10 µF should be placed as close as possible
to the VIN pin.
A larger, high ESR input capacitor should also be used. This capacitor is recommended for damping input
voltage spikes during power-on and for holding up the input voltage during transients. In low input voltage
applications, line transients may fall below the UVLO threshold if there is not enough input capacitance. Both
tantalum and electrolytic type capacitors are suitable for the bulk capacitor. However, large tantalums may not be
available for high input voltages and their working voltage must be derated by at least 2X.
8.2.2.5 Bootstrap
The drive voltage for the internal switch is supplied via the BOOT pin. This pin must be connected to a ceramic
capacitor, Cboot, from the switch node, shown as C4 in the typical application. The LM26001 provides the VDD
voltage internally, so no external diode is needed. A maximum value of 0.1 uF is recommended for Cboot.
Values smaller than 0.01 uF may result in insufficient hold up time for the drive voltage and increased power
dissipation.
During low Vin operation, when the on-time is extended, the bootstrap capacitor is at risk of discharging. If the
Cboot capacitor is discharged below approximately 2.5V, the LM26001 enters a high frequency re-charge mode.
The Cboot cap is re-charged via the LG synchronous FET shown in the block diagram. Switching returns to
normal when the Cboot cap has been recharged.
8.2.2.6 Catch Diode
When the internal switch is off, output current flows through the catch diode. Alternately, when the switch is on,
the diode sees a reverse voltage equal to Vin. Therefore, the important parameters for selecting the catch diode
are peak current and peak inverse voltage. The average current through the diode is given by:
IDAVE = Iload x (1-D) (16)
Where D is the duty cycle, defined as Vout/Vin. The catch diode conducts the largest currents during the lowest
duty cycle. Therefore IDAVE should be calculated assuming maximum input voltage. The diode should be rated to
handle this current continuously. For over-current or short circuit conditions, the catch diode should be rated to
handle peak currents equal to the peak current limit.
The peak inverse voltage rating of the diode must be greater than maximum input voltage.
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Product Folder Links: LM26001 LM26001-Q1
B
fzc fpc1
FREQUENCY
GAIN (dB)
fpc
(0Hz)
-20dB/dec
-20dB/dec
0dB/dec
-60
-40
-20
0
20
0.01 0.1 1 10 100
FREQUENCY (kHz)
GAIN (dB)
1000
0
-45
-90
-135
-180
PHASE (º)
fp fz fn
LM26001
,
LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
www.ti.com
A Schottky diode must be used. It's low forward voltage maximizes efficiency and BOOT voltage, while also
protecting the SW pin against large negative voltage spikes.
8.2.2.7 Compensation
The purpose of loop compensation is to ensure stable operation while maximizing dynamic performance. Stability
can be analyzed with loop gain measurements, while dynamic performance is analyzed with both loop gain and
load transient response. Loop gain is equal to the product of control-output transfer function (power stage) and
the feedback transfer function (the compensation network).
For stability purposes, our target is to have a loop gain slope that is -20dB /decade from a very low frequency to
beyond the crossover frequency. Also, the crossover frequency should not exceed one-fifth of the switching
frequency, i.e. 60 kHz in the case of 300 kHz switching frequency.
For dynamic purposes, the higher the bandwidth, the faster the load transient response. A large DC gain means
high DC regulation accuracy (i.e. DC voltage changes little with load or line variations). To achieve this loop gain,
the compensation components should be set according to the shape of the control-output bode plot. A typical
plot is shown in Figure 21.
Figure 21. Control-Output Transfer Function
The control-output transfer function consists of one pole (fp), one zero (fz), and a double pole at fn (half the
switching frequency).
Referring to Figure 21, the following should be done to create a -20dB /decade roll-off of the loop gain:
1. Place a pole at 0 Hz (fpc)
2. Place a zero at fp (fzc)
3. Place a second pole at fz (fpc1)
The resulting feedback (compensation) bode plot is shown in Figure 22. Adding the control-output response to
the feedback response will then result in a nearly continuous –20db/decade slope.
Figure 22. Feedback Transfer Function
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1
2 x S x fz x R5
C9 =
1
2 x S x fPMAX x R5
C8 =
R5 = R1 + R2
R2
©
§¹
·
B
gm x
FB
COMP
SS
6
7
5
C8
R1
R2 C10
C9
R5
To Vout
2
fsw
fn =
1
10 x S x Ro x Co
fp = +0.5
2 x S x L x fsw x Co
1
2S x Re x Co
fz =
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,
LM26001-Q1
www.ti.com
SNVS430I –MAY 2006–REVISED MARCH 2015
The control-output corner frequencies can be determined approximately by the following equations:
(17)
(18)
(19)
Where Co is the output capacitance, Ro is the load resistance, Re is the output capacitor ESR, and fsw is the
switching frequency. The effects of slope compensation and current sense gain are included in this equation.
However, the equation is an approximation intended to simplify loop compensation calculations. To derive the
exact transfer function, use 0.2V/V sense amp gain and 36mVp-p slope compensation.
Since fp is determined by the output network, it shifts with loading. Determine the range of frequencies
(fpmin/max) across the expected load range. Then determine the compensation values as described below and
shown in Figure 23.
Figure 23. Compensation Network
1. The compensation network automatically introduces a low frequency pole (fpc), which is close to 0Hz.
2. Once the fp range is determined, R5 should be calculated using:
(20)
Where B is the desired feedback gain in v/v between fp and fz, and gm is the transconductance of the error
amplifier. A gain value around 10dB (3.3v/v) is generally a good starting point. Bandwidth increases with
increasing values of R5.
3. Next, place a zero (fzc) near fp using C8. C8 can be determined with the following equation:
(21)
The selected value of C8 should place fzc within a decade above or below fpmax, and not less than fpmin. A
higher C8 value (closer to fpmin) generally provides a more stable loop, but too high a value will slow the
transient response time. Conversely, a smaller C8 value will result in a faster transient response, but lower phase
margin.
4. A second pole (fpc1) can also be placed at fz. This pole can be created with a single capacitor, C9. The
minimum value for this capacitor can be calculated by:
(22)
C9 may not be necessary in all applications. However if the operating frequency is being synchronized below the
nominal frequency, C9 is recommended. Although it is not required for stability, C9 is very helpful in suppressing
noise.
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Product Folder Links: LM26001 LM26001-Q1
1 Ps/DIV
Vout
1V/Div
PGOOD
5V/Div
SS
1V/Div
EN
10V/Div
Vout
40 mV/Div
Iout
500 mA/Div
200 Ps/DIV
Vfb
fpff = fzff x Vout
1
2 x S x R1 x C10
fzff =
LM26001
,
LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
www.ti.com
A phase lead capacitor can also be added to increase the phase and gain margins. The phase lead capacitor is
most helpful for high input voltage applications or when synchronizing to a frequency greater than nominal. This
capacitor, shown as C10 in Figure 23, should be placed in parallel with the top feedback resistor, R1. C10
introduces an additional zero and pole to the compensation network. These frequencies can be calculated as
shown below:
(23)
(24)
A phase lead capacitor will boost loop phase around the region of the zero frequency, fzff. fzff should be placed
somewhat below the fpz1 frequency set by C9. However, if C10 is too large, it will have no effect.
8.2.3 Application Curves
Figure 24. Startup Waveforms Figure 25. Load Transient Response
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9 Power Supply Recommendations
The LM26001 is designed to operate from various DC power supply including a car battery. If so, VIN input
should be protected from reversal voltage and voltage dump over 48 Volts. The impedance of the input supply
rail should be low enough that the input current transient does not cause drop below VIN UVLO level. If the input
supply is connected by using long wires, additional bulk capacitance may be required in addition to normal input
capacitor.
10 Layout
10.1 Layout Guidelines
Good board layout is critical for switching regulators such as the LM26001. First, the ground plane area must be
sufficient for thermal dissipation purposes, and second, appropriate guidelines must be followed to reduce the
effects of switching noise.
Switch mode converters are very fast switching devices. In such devices, the rapid increase of input current
combined with parasitic trace inductance generates unwanted Ldi/dt noise spikes at the SW node and also at the
VIN node. The magnitude of this noise tends to increase as the output current increases. This parasitic spike
noise may turn into electromagnetic interference (EMI), and can also cause problems in device performance.
Therefore, care must be taken in layout to minimize the effect of this switching noise.
The current sensing circuit in current mode devices can be easily affected by switching noise. This noise can
cause duty cycle jitter which leads to increased spectral noise. Although the LM26001 has 100ns blanking time at
the beginning of every cycle to ignore this noise, some noise may remain after the blanking time. Following the
important guidelines below will help minimize switching noise and its effect on current sensing.
The switch node area should be as small as possible. The catch diode, input capacitors, and output capacitors
should be grounded to a large ground plane, with the bulk input capacitor grounded as close as possible to the
catch diode anode. Additionally, the ground area between the catch diode and bulk input capacitor is very noisy
and should be somewhat isolated from the rest of the ground plane.
A ceramic input capacitor must be connected as close as possible to the VIN pin and grounded close to the GND
pin. Often this capacitor is most easily located on the bottom side of the pcb. If placement close to the GND pin
is not practical, the ceramic input capacitor can also be grounded close to the catch diode ground. The above
layout recommendations are illustrated below in Figure 26.
It is a good practice to connect the EP, GND pin, and small signal components (COMP, FB, FREQ) to a separate
ground plane, shown in Figure 26 as EP GND, and in the schematics as a signal ground symbol. Both the
exposed pad and the GND pin must be connected to ground. This quieter plane should be connected to the high
current ground plane at a quiet location, preferably near the Vout ground as shown by the dashed line in
Figure 26.
The EP GND plane should be made as large as possible, since it is also used for thermal dissipation. Several
vias can be placed directly below the EP to increase heat flow to other layers when they are available. The
recommended via hole diameter is 0.3mm.
The trace from the FB pin to the resistor divider should be short and the entire feedback trace must be kept away
from the inductor and switch node. See AN-1229 SIMPLE SWITCHER® PCB Layout Guidelines,SNVA054, for
more information regarding PCB layout for switching regulators.
Copyright © 2006–2015, Texas Instruments Incorporated Submit Documentation Feedback 23
Product Folder Links: LM26001 LM26001-Q1
PswAC = Vin x Iload x fsw x Vin x 10-9
1.33
©
§¹
·
Vin Vout
GND
SW
EP
GND
+
LM26001
,
LM26001-Q1
SNVS430I –MAY 2006–REVISED MARCH 2015
www.ti.com
10.2 Layout Example
Figure 26. Example PCB Layout
10.3 Thermal Considerations and TSD
Although the LM26001 has a built in current limit, at ambient temperatures above 80°C, device temperature rise
may limit the actual maximum load current. Therefore, temperature rise must be taken into consideration to
determine the maximum allowable load current.
Temperature rise is a function of the power dissipation within the device. The following equations can be used to
calculate power dissipation (PD) and temperature rise, where total PD is the sum of FET switching losses, FET
DC losses, drive losses, Iq, and VBIAS losses:
PDTOTAL = PswAC + PswDC + PQG + PIq + PVBIAS (25)
(26)
PswDC = D x Iload2x (0.2 + 0.00065 x (Tj- 25)) (27)
PQG = Vin x 4.6 x 10-9 x fsw (28)
PIq = Vin x Iq (29)
PVBIAS = Vbias x IVBIAS (30)
Given this total power dissipation, junction temperature can be calculated as follows:
Tj = Ta + (PDTOTAL xθJA) (31)
Where θJA=38°C/W (typically) when using a multi-layer board with a large copper plane area. θJA varies with
board type and metallization area.
To calculate the maximum allowable power dissipation, assume Tj = 125°C. To ensure that junction temperature
does not exceed the maximum operating rating of 125°C, power dissipation should be verified at the maximum
expected operating frequency, maximum ambient temperature, and minimum and maximum input voltage. The
calculated maximum load current is based on continuous operation and may be exceeded during transient
conditions.
If the power dissipation remains above the maximum allowable level, device temperature will continue to rise.
When the junction temperature exceeds its maximum, the LM26001 engages Thermal Shut Down (TSD). In
TSD, the part remains in a shutdown state until the junction temperature falls to within normal operating limits. At
this point, the device restarts in soft-start mode.
24 Submit Documentation Feedback Copyright © 2006–2015, Texas Instruments Incorporated
Product Folder Links: LM26001 LM26001-Q1
LM26001
,
LM26001-Q1
www.ti.com
SNVS430I –MAY 2006–REVISED MARCH 2015
11 Device and Documentation Support
11.1 Documentation Support
11.1.1 Related Documentation
AN-1229 SIMPLE SWITCHER® PCB Layout Guidelines,SNVA054
IC Package Thermal Metrics application report, SPRA953
11.2 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 1. Related Links
TECHNICAL TOOLS & SUPPORT &
PARTS PRODUCT FOLDER SAMPLE & BUY DOCUMENTS SOFTWARE COMMUNITY
LM26001 Click here Click here Click here Click here Click here
LM26001-Q1 Click here Click here Click here Click here Click here
11.3 Trademarks
All trademarks are the property of their respective owners.
11.4 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.5 Glossary
SLYZ022 —TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
Copyright © 2006–2015, Texas Instruments Incorporated Submit Documentation Feedback 25
Product Folder Links: LM26001 LM26001-Q1
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM26001MXA/NOPB ACTIVE HTSSOP PWP 16 92 Green (RoHS
& no Sb/Br) Call TI | SN Level-1-260C-UNLIM -40 to 125 L26001
MXA
LM26001MXAX/NOPB ACTIVE HTSSOP PWP 16 2500 Green (RoHS
& no Sb/Br) Call TI | SN Level-1-260C-UNLIM -40 to 125 L26001
MXA
LM26001QMXA/NOPB ACTIVE HTSSOP PWP 16 92 Green (RoHS
& no Sb/Br) Call TI | SN Level-1-260C-UNLIM -40 to 125 L26001
QMXA
LM26001QMXAX/NOPB ACTIVE HTSSOP PWP 16 2500 Green (RoHS
& no Sb/Br) Call TI | SN Level-1-260C-UNLIM -40 to 125 L26001
QMXA
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 2
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM26001, LM26001-Q1 :
•Catalog: LM26001
•Automotive: LM26001-Q1
NOTE: Qualified Version Definitions:
•Catalog - TI's standard catalog product
•Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM26001MXAX/NOPB HTSSOP PWP 16 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1
LM26001QMXAX/NOPB HTSSOP PWP 16 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 24-Jun-2019
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM26001MXAX/NOPB HTSSOP PWP 16 2500 367.0 367.0 35.0
LM26001QMXAX/NOPB HTSSOP PWP 16 2500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 24-Jun-2019
Pack Materials-Page 2
www.ti.com
PACKAGE OUTLINE
C
TYP
6.6
6.2
14X 0.65
16X 0.30
0.19
2X
4.55
(0.15) TYP
0 - 8 0.15
0.05
3.3
2.7
3.3
2.7
2X 1.34 MAX
NOTE 5
1.2 MAX
(1)
0.25
GAGE PLANE
0.75
0.50
A
NOTE 3
5.1
4.9
B4.5
4.3
4X 0.166 MAX
NOTE 5
4214868/A 02/2017
PowerPAD HTSSOP - 1.2 mm max heightPWP0016A
PLASTIC SMALL OUTLINE
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm per side.
4. Reference JEDEC registration MO-153.
5. Features may not be present.
PowerPAD is a trademark of Texas Instruments.
TM
116
0.1 C A B
9
8
PIN 1 ID
AREA
SEATING PLANE
0.1 C
SEE DETAIL A
DETAIL A
TYPICAL
SCALE 2.400
THERMAL
PAD
17
www.ti.com
EXAMPLE BOARD LAYOUT
(5.8)
0.05 MAX
ALL AROUND 0.05 MIN
ALL AROUND
16X (1.5)
16X (0.45)
14X (0.65)
(3.4)
NOTE 9
(5)
NOTE 9
(3.3)
(3.3)
( 0.2) TYP
VIA (1.1) TYP
(1.1)
TYP
4214868/A 02/2017
PowerPAD HTSSOP - 1.2 mm max heightPWP0016A
PLASTIC SMALL OUTLINE
SYMM
SYMM
SEE DETAILS
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE:10X
1
89
16
METAL COVERED
BY SOLDER MASK
SOLDER MASK
DEFINED PAD
17
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
9. Size of metal pad may vary due to creepage requirement.
TM
METAL
SOLDER MASK
OPENING
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
PADS 1-16
EXPOSED
METAL
SOLDER MASK
DEFINED
SOLDER MASK
METAL UNDER SOLDER MASK
OPENING
EXPOSED
METAL
www.ti.com
EXAMPLE STENCIL DESIGN
16X (1.5)
16X (0.45)
(3.3)
(3.3)
BASED ON
0.125 THICK
STENCIL
14X (0.65)
(R0.05) TYP
(5.8)
4214868/A 02/2017
PowerPAD HTSSOP - 1.2 mm max heightPWP0016A
PLASTIC SMALL OUTLINE
2.79 X 2.790.175 3.01 X 3.010.15 3.3 X 3.3 (SHOWN)0.125 3.69 X 3.690.1
SOLDER STENCIL
OPENING
STENCIL
THICKNESS
NOTES: (continued)
10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
11. Board assembly site may have different recommendations for stencil design.
TM
SYMM
SYMM
1
89
16
BASED ON
0.125 THICK
STENCIL
BY SOLDER MASK
METAL COVERED SEE TABLE FOR
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
SOLDER PASTE EXAMPLE
EXPOSED PAD
100% PRINTED SOLDER COVERAGE BY AREA
SCALE:10X
17
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