LM3406,LM3406HV
LM3406/06HV 1.5A Constant Current Buck Regulator for Driving High Power
LEDs
Literature Number: SNVS512C
LM3406/06HV
February 8, 2010
1.5A Constant Current Buck Regulator for Driving High
Power LEDs
General Description
The LM3406/06HV are monolithic switching regulators de-
signed to deliver constant currents to high power LEDs. Ideal
for automotive, industrial, and general lighting applications,
they contain a high-side N-channel MOSFET switch with a
current limit of 2.0A (typical) for step-down (Buck) regulators.
Controlled on-time with true average current and an external
current sense resistor allow the converter output voltage to
adjust as needed to deliver a constant current to series and
series-parallel connected LED arrays of varying number and
type. LED dimming via pulse width modulation (PWM) is
achieved using a dedicated logic pin or by PWM of the power
input voltage. The product feature set is rounded out with low-
power shutdown and thermal shutdown protection.
Features
■Integrated 2.0A MOSFET
■VIN Range 6V to 42V (LM3406)
■VIN Range 6V to 75V (LM3406HV)
■True average output current control
■1.7A Minimum Output Current Limit Over Temperature
■Cycle-by-Cycle Current Limit
■PWM Dimming with Dedicated Logic Input
■PWM Dimming with Power Input Voltage
■Simple Control Loop Compensation
■Low Power Shutdown
■Supports All-Ceramic Output Capacitors and Capacitor-
less Outputs
■Thermal Shutdown Protection
■eTSSOP-14 Package
Applications
■LED Driver
■Constant Current Source
■Automotive Lighting
■General Illumination
■Industrial Lighting
Typical Application
30020301
© 2010 National Semiconductor Corporation 300203 www.national.com
LM3406/06HV 1.5A Constant Current Buck Regulator for Driving High Power LEDs
Connection Diagram
LM3406/06HV
30020302
14-Lead Exposed Pad Plastic TSSOP Package
NS Package Number MXA14A
Ordering Information
Order Number Package Type NSC Package Drawing Supplied As
LM3406MH
eTSSOP-14 MXA14A
95 units in anti-static rails
LM3406MHX 2500 units on tape and reel
LM3406HVMH 95 units in anti-static rails
LM3406HVMHX 2500 units on tape and reel
Pin Descriptions
Pin(s) Name Description Application Information
1,2 SW Switch pin Connect these pins to the output inductor and Schottky diode.
3 BOOT MOSFET drive bootstrap pin Connect a 22 nF ceramic capacitor from this pin to the SW pins.
4 NC No Connect No internal connection. Leave this pin unconnected.
5 VOUT Output voltage sense pin Connect this pin to the output node where the inductor and the first
LED's anode connect.
6 CS Current sense feedback pin Set the current through the LED array by connecting a resistor from
this pin to ground.
7 GND Ground pin Connect this pin to system ground.
8 DIM Input for PWM dimming Connect a logic-level PWM signal to this pin to enable/disable the
power MOSFET and reduce the average light output of the LED array.
Logic high = output on, logic low - output off.
9 COMP Error amplifier output Connect a 0.1 µF ceramic capacitor with X5R or X7R dielectric from
this pin to ground.
10 RON On-time control pin A resistor connected from this pin to VIN sets the regulator controlled
on-time.
11 VCC Output of the internal 7V linear
regulator
Bypass this pin to ground with a minimum 0.1 µF ceramic capacitor
with X5R or X7R dielectric.
12 VINS Input voltage PWM dimming
comparator input
Connect this pin to the anode of the input diode to allow dimming by
PWM of the input voltage
13,14 VIN Input voltage pin Nominal operating input range for this pin is 6V to 42V (LM3406) or 6V
to 75V (LM3406HV).
DAP DAP Thermal Pad Connect to ground. Place 4-6 vias from DAP to bottom layer ground
plane.
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LM3406/LM3406HV
Absolute Maximum Ratings
LM3406/LM3406HV (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
VIN to GND -0.3V to 45V
(76V LM3406HV)
VINS to GND -0.3V to 45V
(76V LM3406HV)
VOUT to GND -0.3V to 45V
(76V LM3406HV)
BOOT to GND -0.3V to 59V
(90V LM3406HV)
SW to GND -1.5V to 45V
(76V LM3406HV)
BOOT to VCC -0.3V to 45V
(76V LM3406HV)
BOOT to SW -0.3V to 14V
VCC to GND -0.3V to 14V
DIM to GND -0.3V to 7V
COMP to GND -0.3V to 7V
CS to GND -0.3V to 7V
RON to GND -0.3V to 7V
Junction Temperature 150°C
Storage Temp. Range -65°C to 125°C
ESD Rating (Note 2) 2kV
Soldering Information
Lead Temperature (Soldering,
10sec) 260°C
Infrared/Convection Reflow (15sec) 235°C
Operating Ratings
(Note 1)
VIN 6V to 42V
(75V LM3406HV)
Junction Temperature Range −40°C to +125°C
Thermal Resistance θJA
(eTSSOP-14 Package)
(Note 4) 50°C/W
Electrical Characteristics VIN = 24V unless otherwise indicated. Typicals and limits appearing in plain type apply
for TA = TJ = +25°C (Note 3). Limits appearing in boldface type apply over full Operating Temperature Range. Datasheet min/
max specification limits are guaranteed by design, test, or statistical analysis.
LM3406/LM3406HV
Symbol Parameter Conditions Min Typ Max Units
REGULATION COMPARATOR AND ERROR AMPLIFIER
VREF CS Regulation Threshold CS Decreasing, SW turns on 187.5 200 210 mV
191.0
(Note 5)
210.0
(Note 5)
V0V CS Over-voltage Threshold CS Increasing, SW turns off 300 mV
ICS CS Bias Current CS = 0V 0.9 µA
IVOUT VOUT Bias Current VOUT = 24V 83 µA
ICOMP COMP Pin Current CS = 0V 25 µA
Gm-CS Error Amplifier
Transconductance
150 mV < CS < 250 mV 145 µS
SHUTDOWN
VSD-TH Shutdown Threshold RON Increasing 0.3 0.7 1.05 V
VSD-HYS Shutdown Hysteresis RON Decreasing 40 mV
ON AND OFF TIMER
tOFF-MIN Minimum Off-time CS = 0V 230 ns
tON Programmed On-time VIN = 24V, VO = 12V, RON = 200 kΩ800 1300 1800
tON-MIN Minimum On-time 280
VINS COMPARATOR
VINS-TH VINS Pin Threshold VINS decreasing 70 %VIN
IIN-2WD VINS Pin Input Current VINS = 24V * 0.7 25 µA
INTERNAL REGULATOR
VCC-REG VCC Regulated Output 0 mA < ICC < 5 mA 6.4 77.4 V
VIN-DO VIN - VCC ICC = 5 mA, 6.0V < VIN < 8.0V, Non-
switching
300 mV
VCC-BP-TH VCC Bypass Threshold VIN Increasing 8.8 V
VCC-LIM VCC Current Limit VIN = 24V, VCC = 0V 420 mA
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LM3406/LM3406HV
Symbol Parameter Conditions Min Typ Max Units
VCC-UV-TH VCC Under-voltage Lock-out
Threshold
VCC Increasing 5.3 V
VCC-UV-HYS VCC Under-voltage Lock-out
Hysteresis
VCC Decreasing 150 mV
IIN-OP IIN Operating Current Non-switching, CS = 0.5V 1.2 mA
IIN-SD IIN Shutdown Current RON = 0V 240 350 µA
CURRENT LIMIT
ILIM Current Limit Threshold 1.7 2.1 2.7 A
DIM COMPARATOR
VIH Logic High DIM Increasing 2.2 V
VIL Logic Low DIM Decreasing 0.8 V
IDIM-PU DIM Pull-up Current DIM = 1.5V 80 µA
MOSFET AND DRIVER
RDS-ON Buck Switch On Resistance ISW = 200 mA, BOOT = 6.3V 0.37 0.75 Ω
VDR-UVLO BOOT Under-voltage Lock-out
Threshold
BOOT–SW Increasing 1.7 2.9 4.3 V
VDR-HYS BOOT Under-voltage Lock-out
Hysteresis
BOOT–SW Decreasing 370 mV
THERMAL SHUTDOWN
TSD Thermal Shutdown Threshold 165 °C
TSD-HYS Thermal Shutdown Hysteresis 25 °C
THERMAL RESISTANCE
θJA Junction to Ambient eTSSOP-14 Package (Note 4) 50 °C/W
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur, including inoperability and degradation of device reliability
and/or performance. Functional operation of the device and/or non-degradation at the Absolute Maximum Ratings or other conditions beyond those indicated in
the Operating Ratings is not implied. The recommended Operating Ratings indicate conditions at which the device is functional and the device should not be
operated beyond such conditions.
Note 2: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 3: Typical values represent most likely parametric norms at the conditions specified and are not guaranteed.
Note 4: θJA of 50°C/W with DAP soldered to a minimum of 2 square inches of 1oz. copper on the top or bottom PCB layer.
Note 5: Specified with junction temperature from 0°C - 125°C.
Note 6: VIN = 24V, IF = 1A, TA = 25°C, and the load consists of three InGaN LEDs in series unless otherwise noted. See the Bill of Materials table at the end of
the datasheet.
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LM3406/LM3406HV
Current Limit vs VIN, LM3406
30020339
Current Limit vs VIN, LM3406HV
30020340
VCC vs VIN
30020341
VO-MAX vs VIN, LM3406
30020342
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LM3406/LM3406HV
Block Diagram
30020303
Application Information
THEORY OF OPERATION
The LM3406 and LM3406HV are buck regulators with a wide
input voltage range, low voltage reference, and a fast output
enable/disable function. These features combine to make
them ideal for use as a constant current source for LEDs with
forward currents as high as 1.5A. The controlled on-time
(COT) architecture uses a comparator and a one-shot on-
timer that varies inversely with input and output voltage in-
stead of a fixed clock. The LM3406/06HV also employs an
integrator circuit that averages the output current. When the
converter runs in continuous conduction mode (CCM) the
controlled on-time maintains a constant switching frequency
over changes in both input and output voltage. These features
combine to give the LM3406/06HV an accurate output cur-
rent, fast transient response, and constant switching frequen-
cy over a wide range of conditions.
CONTROLLED ON-TIME OVERVIEW
Figure 1 shows a simplified version of the feedback system
used to control the current through an array of LEDs. A dif-
ferential voltage signal, VSNS, is created as the LED current
flows through the current setting resistor, RSNS. VSNS is fed
back by the CS pin, where it is integrated and compared
against an error amplifier-generated reference. The error am-
plifier is a transconductance (Gm) amplifier which adjusts the
voltage on COMP to maintain a 200 mV average at the CS
pin. The on-comparator turns on the power MOSFET when
VSNS falls below the reference created by the Gm amp. The
power MOSFET conducts for a controlled on-time, tON, set by
an external resistor, RON, the input voltage, VIN and the output
voltage, VO. On-time can be estimated by the following sim-
plified equation (for the most accurate version of this expres-
sion see the Appendix):
At the conclusion of tON the power MOSFET turns off and
must remain off for a minimum of 230 ns. Once this tOFF-MIN
is complete the CS comparator compares the integrated
VSNS and reference again, waiting to begin the next cycle.
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LM3406/LM3406HV
30020306
FIGURE 1. Comparator and One-Shot
SWITCHING FREQUENCY
The LM3406/06HV does not contain a clock, however the on-
time is modulated in proportion to both input voltage and
output voltage in order to maintain a relatively constant fre-
quency. On-time tON, duty cycle D and switching frequency
fSW are related by the following expression:
fSW = D / tON
D = (VO + VD) / (VIN - VSW + VD)
VD = Schottky diode (typically 0.5V)
VSW = IF x RDSON
The LM3406/06HV regulators should be operated in contin-
uous conduction mode (CCM), where inductor current stays
positive throughout the switching cycle. During steady-state
CCM operation, the converter maintains a constant switching
frequency that can be estimated using the following equation
(for the most accurate version, particularly for applications
that will have an input or output voltage of less than approxi-
mately 12V, see the Appendix):
SETTING LED CURRENT
LED current is set by the resistor RSNS, which can be deter-
mined using the following simple expression due to the output
averaging:
RSNS = 0.2 / IF
MAXIMUM NUMBER OF SERIES LEDS
LED driver designers often want to determine the highest
number of LEDs that can be driven by their circuits. The limit
on the maximum number of series LEDs is set by the highest
output voltage, VO-MAX, that the LED driver can provide. A
buck regulator cannot provide an output voltage that is higher
than the minimum input voltage, and in pratice the maximum
output voltage of the LM3406/06HV is limited by the minimum
off-time as well. VO-MAX determines how many LEDs can be
driven in series. Referring to the illustration in Figure 1, output
voltage is calculated as:
VO-MAX = VIN-MIN x (1 - fSW x tOFF-MIN)
tOFF-MIN = 230 ns
Once VO-MAX has been calculated, the maximum number of
series LEDs, nMAX, can be calculated by the following espres-
sion and rounding down:
nMAX = VO-MAX / VF
VF = forward voltage of each LED
At low switching frequency VO-MAX is higher, allowing the
LM3406/06HV to regulate output voltages that are nearly
equal to input voltage, and this can allow the system to drive
more LEDs in series. Low switching frequencies are not al-
ways desireable, however, because they require larger, more
expensive components.
CALCULATING OUTPUT VOLTAGE
Even though output current is the controlled parameter in LED
drivers, output voltage must still be calculated in order to de-
sign the complete circuit. Referring to the illustration in Figure
1, output voltage is calculated as:
VO = n x VF + VSNS
VSNS = sense voltage of 200 mV, n = number of LEDs in series
MINIMUM ON-TIME
The minimum on-time for the LM3406/06HV is 280 ns (typi-
cal). One practical example of reaching the minimum on-time
is when dimming the LED light output with a power FET
placed in parallel to the LEDs. When the FET is on, the output
voltage drops to 200 mV. This results in a small duty cycle
and in most circuits requires an on-time that would be less
than 280 ns. In such a case the LM3406/06HV keeps the on-
time at 280 ns and increases the off-time as much as needed,
which effectively reduces the switching frequency.
HIGH VOLTAGE BIAS REGULATOR (VCC)
The LM3406/06HV contains an internal linear regulator with
a 7V output, connected between the VIN and the VCC pins.
The VCC pin should be bypassed to the GND pin with a 0.1
µF ceramic capacitor connected as close as possible to the
pins of the IC. VCC tracks VIN until VIN reaches 8.8V (typical)
and then regulates at 7V as VIN increases. The
LM3406/06HV comes out of UVLO and begins operating
when VCC crosses 5.3V. This is shown graphically in the
Typical Performance curves.
Connecting an external supply to VCC to power the gate
drivers is not recommended. However, it may be done if cer-
tain precautions are taken. Be sure that the external supply
will not violate any absolute maximum conditions and will at
no point exceed the voltage applied to the VIN pins. Under
certain conditions, some ringing may be present on the SW
and BOOT pins when VCC is driven with an external supply.
It is important to ensure that the absolute maximum ratings of
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LM3406/LM3406HV
these pins are not violated during the ringing or else damage
to the device may occur.
INTERNAL MOSFET AND DRIVER
The LM3406/06HV features an internal power MOSFET as
well as a floating driver connected from the SW pin to the
BOOT pin. Both rise time and fall time are 20 ns each (typical)
and the approximate gate charge is 9 nC. The high-side rail
for the driver circuitry uses a bootstrap circuit consisting of an
internal high-voltage diode and an external 22 nF capacitor,
CB. VCC charges CB through the internal diode while the power
MOSFET is off. When the MOSFET turns on, the internal
diode reverse biases. This creates a floating supply equal to
the VCC voltage minus the diode drop to drive the MOSFET
when its source voltage is equal to VIN.
FAST LOGIC PIN FOR PWM DIMMING
The DIM pin is a TTL compatible input for PWM dimming of
the LED. A logic low (below 0.8V) at DIM will disable the in-
ternal MOSFET and shut off the current flow to the LED array.
While the DIM pin is in a logic low state the support circuitry
(driver, bandgap, VCC) remains active in order to minimize
the time needed to turn the LED array back on when the DIM
pin sees a logic high (above 2.2V). A 75 µA (typical) pull-up
current ensures that the LM3406/06HV is on when DIM pin is
open circuited, eliminating the need for a pull-up resistor.
Dimming frequency, fDIM, and duty cycle, DDIM, are limited by
the LED current rise time and fall time and the delay from
activation of the DIM pin to the response of the internal power
MOSFET. In general, fDIM should be at least one order of
magnitude lower than the steady state switching frequency in
order to prevent aliasing.
INPUT VOLTAGE COMPARATOR FOR PWM DIMMING
Adding an external input diode and using the internal VINS
comparator allows the LM3406/06HV to sense and respond
to dimming that is done by PWM of the input voltage. This
method is also referred to as "Two-Wire Dimming", and a typ-
ical application circuit is shown in Figure 2. If the VINS pin
voltage falls 70% below the VIN pin voltage, the
LM3406/06HV disables the internal power FET and shuts off
the current to the LED array. The support circuitry (driver,
bandgap, VCC) remains active in order to minimize the time
needed to the turn the LED back on when the VINS pin volt-
age rises and exceeds 70% of VIN. This minimizes the re-
sponse time needed to turn the LED array back on.
30020304
FIGURE 2. Typical Application using Two-Wire Dimming
PARALLEL MOSFET FOR HIGH-SPEED PWM DIMMING
For applications that require dimming at high frequency or
with wide dimming duty cycle range neither the VINS com-
parator or the DIM pin are capable of slewing the LED current
from 0 to the target level fast enough. For such applications
the LED current slew rate can by increased by shorting the
LED current with a N-MOSFET placed in parallel to the LED
or LED array, as shown in Figure 3. While the parallel FET is
on the output current flows through it, effectively reducing the
output voltage to equal the CS pin voltage of 0.2V. This dim-
ming method maintains a continuous current through the
inductor, and therefore eliminates the biggest delay in turning
the LED(s) or and off. The trade-off with parallel FET dimming
is that more power is wasted while the FET is on, although in
most cases the power wasted is small compared to the power
dissipated in the LEDs. Parallel FET circuits should use no
output capacitance or a bare minimum for noise filtering in
order to minimize the slew rate of output voltage. Dimming
FET Q1 can be driven from a ground-referenced source be-
cause the source stays at 0.2V along with the CS pin.
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LM3406/LM3406HV
30020327
FIGURE 3. Dimming with a Parallel FET
PEAK CURRENT LIMIT
The current limit comparator of the LM3406/06HV will engage
whenever the power MOSFET current (equal to the inductor
current while the MOSFET is on) exceeds 2.1A (typical). The
power MOSFET is disabled for a cool-down time that of ap-
proximately 100 µs. At the conclusion of this cool-down time
the system re-starts. If the current limit condition persists the
cycle of cool-down time and restarting will continue, creating
a low-power hiccup mode, minimizing thermal stress on the
LM3406/06HV and the external circuit components.
OVER-VOLTAGE/OVER-CURRENT COMPARATOR
The CS pin includes an output over-voltage/over-current
comparator that will disable the power MOSFET whenever
VSNS exceeds 300 mV. This threshold provides a hard limit
for the output current. Output current overshoot is limited to
300 mV / RSNS by this comparator during transients. The OVP/
OCP comparator limits the maximum ripple voltage at the CS
pin to 200 mVP-P.
OUTPUT OPEN-CIRCUIT
The most common failure mode for power LEDs is a broken
bond wire, and the result is an output open-circuit. When this
happens the feedback path is disconnected, and the output
voltage will attempt to rise. In buck converters the output volt-
age can only rise as high as the input voltage, and the
minimum off-time requirement ensures that VO(MAX) is slightly
less than VIN. Figure 4 shows a method using a zener diode,
Z1, and zener limiting resistor, RZ, to limit output voltage to
the reverse breakdown voltage of Z1 plus 200 mV. The zener
diode reverse breakdown voltage, VZ, must be greater than
the maximum combined VF of all LEDs in the array. The max-
imum recommended value for RZ is 1 kΩ.
The output stage (SW and VOUT pins) of the LM3406/06HV
is capable of withstanding VO(MAX) indefinitely as long as the
output capacitor is rated to handle the full input voltage. When
an LED fails open-circuit and there is no output capacitor
present the surge in output voltage due to the collapsing mag-
netic field in the output inductor can exceed VIN and can
damage the LM3406/06HV IC. As an alternative to the zener
clamp method described previously, a diode can be connect-
ed from the output to the input of the regulator circuit that will
clamp the inductive surge to one VD above VIN.
Regardless of which protection method is used a resistance
in series with the VOUT pin, ROUT, is recommended to limit
the current in the event the VOUT pin is pulled below ground
when the LED circuit is reconnected. This can occur frequent-
ly if the lead lengths to the LEDs are long and the inductance
is significant. A resistor between 1 kΩ and 10 kΩ is recom-
mended.
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LM3406/LM3406HV
30020311
FIGURE 4. Two Methods of Output Open Circuit Protection
LOW POWER SHUTDOWN
The LM3406/06HV can be placed into a low power state (IIN-
SD = 240 µA) by grounding the RON pin with a signal-level
MOSFET as shown in Figure 5 . Low power MOSFETs like
the 2N7000, 2N3904, or equivalent are recommended de-
vices for putting the LM3406/06HV into low power shutdown.
Logic gates can also be used to shut down the LM3406/06HV
as long as the logic low voltage is below the over temperature
minimum threshold of 0.3V. Noise filter circuitry on the RON
pin can cause a few pulses with longer on-times than normal
after RON is grounded or released. In these cases the OVP/
OCP comparator will ensure that the peak inductor or LED
current does not exceed 300 mV / RSNS.
30020312
FIGURE 5. Low Power Shutdown
THERMAL SHUTDOWN
Internal thermal shutdown circuitry is provided to protect the
IC in the event that the maximum junction temperature is ex-
ceeded. The threshold for thermal shutdown is 165°C with a
25°C hysteresis (both values typical). During thermal shut-
down the MOSFET and driver are disabled.
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LM3406/LM3406HV
Design Considerations
SWITCHING FREQUENCY
Switching frequency is selected based on the trade-offs be-
tween efficiency (better at low frequency), solution size/cost
(smaller at high frequency), and the range of output voltage
that can be regulated (wider at lower frequency.) Many appli-
cations place limits on switching frequency due to EMI sen-
sitivity. The on-time of the LM3406/06HV can be programmed
for switching frequencies ranging from the 10’s of kHz to over
1 MHz. This on-time varies in proportion to both VIN and VO
in order to maintain first-order control over switching frequen-
cy, however in practice the switching frequency will shift in
response to large swings in VIN or VO. The maximum switch-
ing frequency is limited only by the minimum on-time and
minimum off-time requirements.
LED RIPPLE CURRENT
Selection of the ripple current, ΔiF, through the LED array is
similar to the selection of output ripple voltage in a standard
voltage regulator. Where the output ripple in a voltage regu-
lator is commonly ±1% to ±5% of the DC output voltage, LED
manufacturers generally recommend values for ΔiF ranging
from ±5% to ±20% of IF. Higher LED ripple current allows the
use of smaller inductors, smaller output capacitors, or no out-
put capacitors at all. Lower ripple current requires more output
inductance, higher switching frequency, or additional output
capacitance, and may be necessary for applications that are
not intended for human eyes, such as machine vision or in-
dustrial inspection.
BUCK CONVERTERS WITHOUT OUTPUT CAPACITORS
The buck converter is unique among non-isolated topologies
because of the direct connection of the inductor to the load
during the entire switching cycle. By definition an inductor will
control the rate of change of current that flows through it, and
this control over current ripple forms the basis for component
selection in both voltage regulators and current regulators. A
current regulator such as the LED driver for which the
LM3406/06HV was designed focuses on the control of the
current through the load, not the voltage across it. A constant
current regulator is free of load current transients, and has no
need of output capacitance to supply the load and maintain
output voltage. Referring to the Typical Application circuit on
the front page of this datasheet, the inductor and LED can
form a single series chain, sharing the same current. When
no output capacitor is used, the same equations that govern
inductor ripple current, ΔiL, also apply to the LED ripple cur-
rent, ΔiF. For a controlled on-time converter such as
LM3406/06HV the ripple current is described by the following
expression:
The triangle-wave inductor current ripple flows through RSNS
and produces a triangle-wave voltage at the CS pin. To pro-
vide good signal to noise ratio (SNR) the amplitude of CS pin
ripple voltage, ΔvCS, should be at least 25 mVP-P. ΔvCS is de-
scribed by the following:
ΔvCS = ΔiF x RSNS
BUCK CONVERTERS WITH OUTPUT CAPACITORS
A capacitor placed in parallel with the LED(s) can be used to
reduce the LED current ripple while keeping the same aver-
age current through both the inductor and the LED array. With
an output capacitor the output inductance can be lowered,
making the magnetics smaller and less expensive. Alterna-
tively, the circuit could be run at lower frequency but keep the
same inductor value, improving the power efficiency. Both the
peak current limit and the OVP/OCP comparator still monitor
peak inductor current, placing a limit on how large ΔiL can be
even if ΔiF is made very small. Adding a capacitor that re-
duces ΔiF to well below the target provides headroom for
changes in inductance or VIN that might otherwise push the
peak LED ripple current too high.
Figure 6 shows the equivalent impedances presented to the
inductor current ripple when an output capacitor, CO, and its
equivalent series resistance (ESR) are placed in parallel with
the LED array. Note that ceramic capacitors have so little ESR
that it can be ignored. The entire inductor ripple current still
flows through RSNS to provide the required 25 mV of ripple
voltage for proper operation of the CS comparator.
30020314
FIGURE 6. LED and CO Ripple Current
To calculate the respective ripple currents the LED array is
represented as a dynamic resistance, rD. LED dynamic resis-
tance is not always specified on the manufacturer’s
datasheet, but it can be calculated as the inverse slope of the
LED’s VF vs. IF curve. Note that dividing VF by IF will give an
incorrect value that is 5x to 10x too high. Total dynamic re-
sistance for a string of n LEDs connected in series can be
calculated as the rD of one device multiplied by n. Inductor
ripple current is still calculated with the expression from Buck
Regulators without Output Capacitors. The following equa-
tions can then be used to estimate ΔiF when using a parallel
capacitor:
The calculation for ZC assumes that the shape of the inductor
ripple current is approximately sinusoidal.
Small values of CO that do not significantly reduce ΔiF can
also be used to control EMI generated by the switching action
of the LM3406/06HV. EMI reduction becomes more important
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LM3406/LM3406HV
as the length of the connections between the LED and the
rest of the circuit increase.
INPUT CAPACITORS
Input capacitors at the VIN pin of the LM3406/06HV are se-
lected using requirements for minimum capacitance and rms
ripple current. The input capacitors supply pulses of current
approximately equal to IF while the power MOSFET is on, and
are charged up by the input voltage while the power MOSFET
is off. All switching regulators have a negative input
impedance due to the decrease in input current as input volt-
age increases. This inverse proportionality of input current to
input voltage can cause oscillations (sometimes called ‘power
supply interaction’) if the magnitude of the negative input
impedance is greater the the input filter impedance. Minimum
capacitance can be selected by comparing the input
impedance to the converter’s negative resistance; however
this requires accurate calculation of the input voltage source
inductance and resistance, quantities which can be difficult to
determine. An alternative method to select the minimum input
capacitance, CIN(MIN), is to select the maximum input voltage
ripple which can be tolerated. This value, ΔvIN(MAX), is equal
to the change in voltage across CIN during the converter on-
time, when CIN supplies the load current. CIN(MIN) can be
selected with the following:
A good starting point for selection of CIN is to use an input
voltage ripple of 5% to 10% of VIN. A minimum input capaci-
tance of 2x the CIN(MIN) value is recommended for all
LM3406/06HV circuits. To determine the rms current rating,
the following formula can be used:
Ceramic capacitors are the best choice for the input to the
LM3406/06HV due to their high ripple current rating, low ESR,
low cost, and small size compared to other types. When se-
lecting a ceramic capacitor, special attention must be paid to
the operating conditions of the application. Ceramic capaci-
tors can lose one-half or more of their capacitance at their
rated DC voltage bias and also lose capacitance with ex-
tremes in temperature. A DC voltage rating equal to twice the
expected maximum input voltage is recommended. In addi-
tion, the minimum quality dielectric which is suitable for
switching power supply inputs is X5R, while X7R or better is
preferred.
RECIRCULATING DIODE
The LM3406/06HV is a non-synchronous buck regulator that
requires a recirculating diode D1 (see the Typical Application
circuit) to carrying the inductor current during the MOSFET
off-time. The most efficient choice for D1 is a Schottky diode
due to low forward drop and near-zero reverse recovery time.
D1 must be rated to handle the maximum input voltage plus
any switching node ringing when the MOSFET is on. In prac-
tice all switching converters have some ringing at the switch-
ing node due to the diode parasitic capacitance and the lead
inductance. D1 must also be rated to handle the average cur-
rent, ID, calculated as:
ID = (1 – D) x IF
This calculation should be done at the maximum expected
input voltage. The overall converter efficiency becomes more
dependent on the selection of D1 at low duty cycles, where
the recirculating diode carries the load current for an increas-
ing percentage of the time. This power dissipation can be
calculating by checking the typical diode forward voltage,
VD, from the I-V curve on the product datasheet and then
multiplying it by ID. Diode datasheets will also provide a typical
junction-to-ambient thermal resistance, θJA, which can be
used to estimate the operating die temperature of the device.
Multiplying the power dissipation (PD = ID x VD) by θJA gives
the temperature rise. The diode case size can then be se-
lected to maintain the Schottky diode temperature below the
operational maximum.
Transient Protection
Considerations
Considerations need to be made when external sources,
loads or connections are made to the switching converter cir-
cuit due to the possibility of Electrostatic Discharge (ESD) or
Electric Over Stress (EOS) events occurring and damaging
the integrated circuit (IC) device. All IC device pins contain
zener based clamping structures that are meant to clamp
ESD. ESD events are very low energy events, typically less
than 5µJ (microjoules). Any event that transfers more energy
than this may damage the ESD structure. Damage is typically
represented as a short from the pin to ground as the extreme
localized heat of the ESD / EOS event causes the aluminum
metal on the chip to melt, causing the short. This situation is
common to all integrated
CS PIN PROTECTION
When hot swapping in a load (e.g. test points, load boards,
LED stack), any residual charge on the load will be immedi-
ately transferred through the output capacitor to the CS pin,
which is then damaged as shown in Figure 7 below. The EOS
event due to the residual charge from the load is represented
as VTRANSIENT.
From measurements, we know that the 8V ESD structure on
the CS pin can typically withstand 25mA of direct current
(DC). Adding a 1kΩ resistor in series with the CS pin, shown
in Figure 8, results in the majority of the transient energy to
pass through the discrete sense resistor rather than the de-
vice. The series resistor limits the peak current that can flow
during a transient event, thus protecting the CS pin. With the
1kΩ resistor shown, a 33V, 49A transient on the LED return
connector terminal could be absorbed as calculated by:
V = 25mA * 1kΩ + 8V = 33V
I = 33V / 0.67Ω = 49A
This is an extremely high energy event, so the protection
measures previously described should be adequate to solve
this issue.
13 www.national.com
LM3406/LM3406HV
30020370
FIGURE 7. CS Pin, Transient Path
30020365
FIGURE 8. CS Pin, Transient Path with Protection
Adding a resistor in series with the CS pin causes the ob-
served output LED current to shift very slightly. The reason
for this is twofold: (1) the CS pin has about 20pF of inherent
capacitance inside it which causes a slight delay (20ns for a
1kΩ series resistor), and (2) the comparator that is watching
the voltage at the CS pin uses a pnp bipolar transistor at its
input. The base current of this pnp transistor is approximately
100nA which will cause a 0.1mV change in the 200mV thresh-
old. These are both very minor changes and are well under-
stood. The shift in current can either be neglected or taken
into consideration by changing the current sense resistance
slightly.
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LM3406/LM3406HV
CS PIN PROTECTION WITH OVP
When designing output overvoltage protection into the switch-
ing converter circuit using a zener diode, transient protection
on the CS pin requires additional consideration. As shown in
Figure 9, adding a zener diode from the output to the CS pin
(with the series resistor) for output overvoltage protection will
now again allow the transient energy to be passed through
the CS pin’s ESD structure thereby damaging it.
Adding an additional series resistor to the CS pin as shown
in Figure 10 will result in the majority of the transient energy
to pass through the sense resistor thereby protecting the
LM340X device.
30020366
FIGURE 9. CS Pin with OVP, Transient Path
30020367
FIGURE 10. CS Pin with OVP, Transient Path with Protection
15 www.national.com
LM3406/LM3406HV
VIN PIN PROTECTION
The VIN pin also has an ESD structure from the pin to GND
with a breakdown voltage of approximately 80V. Any transient
that exceeds this voltage may damage the device. Although
transient absorption is usually present at the front end of a
switching converter circuit, damage to the VIN pin can still
occur.
When VIN is hot swapped in, the current that rushes in to
charge CIN up to the VIN value also charges (energizes) the
circuit board trace inductance as shown in Figure 11. The ex-
cited trace inductance then resonates with the input capaci-
tance (similar to an under-damped LC tank circuit) and
causes voltages at the VIN pin to rise well in excess of both
VIN and the voltage at the module input connector as clamped
by the input TVS. If the resonating voltage at the VIN pin ex-
ceeds the 80V breakdown voltage of the ESD structure, the
ESD structure will activate and then “snap-back” to a lower
voltage due to its inherent design. If this lower snap-back
voltage is less than the applied nominal VIN voltage, then sig-
nificant current will flow through the ESD structure resulting
in the IC being damaged.
An additional TVS or small zener diode should be placed as
close as possible to the VIN pins of each IC on the board, in
parallel with the input capacitor as shown in Figure 12. A mi-
nor amount of series resistance in the input line would also
help, but would lower overall conversion efficiency. For this
reason, NTC resistors are often used as inrush limiters in-
stead.
30020368
FIGURE 11. VIN Pin with Typical Input Protection
www.national.com 16
LM3406/LM3406HV
30020369
FIGURE 12. VIN Pin with Additional Input Protection
GENERAL COMMENTS REGARDING OTHER PINS
Any pin that goes “off-board” through a connector should have
series resistance of at least 1kΩ to 10kΩ in series with it to
protect it from ESD or other transients. These series resistors
limit the peak current that can flow (or cause a voltage drop)
during a transient event, thus protecting the pin and the de-
vice. Pins that are not used should not be left floating. They
should instead be tied to GND or to an appropriate voltage
through resistance.
Design Example 1
The first example circuit uses the LM3406 to create a flexible
LED driver capable of driving anywhere from one to five white
series-connected LEDs at a current of 1.5A ±5% from a reg-
ulated DC voltage input of 24V ±10%. In addition to the ±5%
tolerance specified for the average output current, the LED
ripple current must be controlled to 10%P-P of the DC value,
or 150 mAP-P. The typical forward voltage of each individual
LED at 1.5A is 3.9V, hence the output voltage ranges from
4.1V to 19.7V, adding in the 0.2V drop for current sensing. A
complete bill of materials can be found in Table 1 at the end
of this datasheet.
30020318
FIGURE 13. Schematic for Design Example 1
17 www.national.com
LM3406/LM3406HV
RON and tON
A moderate switching frequency of 500 kHz will balance the
requirements of inductor size and overall power efficiency.
The LM3406 will allow some shift in switching frequency when
VO changes due to the number of LEDs in series, so the cal-
culation for RON is done at the mid-point of three LEDs in
series, where VO = 11.8V. Note that the actual RON calculation
is done with the high accuracy expression listed in the Ap-
pendix.
RON = 144 kΩ
The closest 1% tolerance resistor is 143 kΩ. The switching
frequency and on-time of the circuit should be checked for
one, three and five LEDs using the equations relating RON and
tON to fSW. As with the RON calculation, the actual fSW and
tON values have been calculated using the high accuracy ex-
pressions listed in the Appendix.
fSW(1 LED) = 362 kHz
fSW(3 LEDs) = 504 kHz
fSW(5 LEDs) = 555 kHz
tON(1 LED) = 528 ns
tON(3 LEDs) = 1014 ns
tON(5 LEDs) = 1512 ns
OUTPUT INDUCTOR
Since an output capacitor will be used to filter some of the AC
ripple current, the inductor ripple current can be set higher
than the LED ripple current. A value of 40%P-P is typical in
many buck converters:
ΔiL = 0.4 x 1.5 = 0.6AP-P
With the target ripple current determined the inductance can
be chosen:
LMIN = [(24 – 11.8) x 1.01 x 10-6] / (0.6) = 20.5 µH
The closest standard inductor value is 22 µH. The average
current rating should be greater than 1.5A to prevent over-
heating in the inductor. Inductor current ripple should be
calculated for one, three and five LEDs:
ΔiL(1 LED) = [(24 - 4.1) x 5.28 x 10-7] / 22 x 10-6
= 478 mAP-P
ΔiL(3 LEDs) = [(24 - 11.8) x 1.01 x 10-6] / 22 x 10-6
= 560 mAP-P
ΔiL(5 LEDs) = [(24 - 19.7) x 1.51 x 10-6] / 22 x 10-6
= 295 mAP-P
The peak LED/inductor current is then estimated. This calcu-
lation uses the worst-case ripple current which occurs with
three LEDs.
IL(PEAK) = IL + 0.5 x ΔiL(MAX)
IL(PEAK) = 1.5 + 0.5 x 0.56 = 1.78A
In order to prevent inductor saturation the inductor’s peak
current rating must be above 1.8A. A 22 µH off-the shelf in-
ductor rated to 2.1A (peak) and 1.9A (average) with a DCR of
59 mΩ will be used.
USING AN OUTPUT CAPACITOR
This application does not require high frequency PWM dim-
ming, allowing the use of an output capacitor to reduce the
size and cost of the output inductor while still meeting the 10%
P-P target for LED ripple current. To select the proper output
capacitor the equation from Buck Regulators with Output Ca-
pacitors is re-arranged to yield the following:
The dynamic resistance, rD,of one LED can be calculated by
taking the tangent line to the VF vs. IF curve in the LED
datasheet. Figure 14 shows an example rD calculation.
www.national.com 18
LM3406/LM3406HV
30020324
FIGURE 14. Calculating rD from the VF vs. IF Curve
Extending the tangent line to the ends of the plot yields values
for ΔVF and ΔIF of 0.7V and 2000 mA, respectively. Dynamic
resistance is then:
rD = ΔVF / ΔIF = 0.5V / 2A = 0.25Ω
The most filtering (and therefore the highest output capaci-
tance) is needed when rD is lowest, which is when there is
only one LED. Inductor ripple current with one LED is 478
mAP-P. The required impedance of CO is calculated:
ZC = [0.15 / (0.478 - 0.15] x 0.35 = 0.114Ω
A ceramic capacitor will be used and the required capacitance
is selected based on the impedance at 362 kHz:
CO = 1/(2 x π x 0.16 x 3.62 x 105) = 3.9 µF
This calculation assumes that CO will be a ceramic capacitor,
and therefore impedance due to the equivalent series resis-
tance (ESR) and equivalent series inductance (ESL) of of the
device is negligible. The closest 10% tolerance capacitor val-
ue is 4.7 µF. The capacitor used should be rated to 25V or
more and have an X7R dielectric. Several manufacturers pro-
duce ceramic capacitors with these specifications in the 1206
case size. A typical value for ESR of 3 mΩ can be read from
the curve of impedance vs. frequency in the product
datasheet.
RSNS
Using the expression for RSNS:
RSNS = 0.2 / IF
RSNS = 0.2 / 1.5 = 0.133Ω
Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.13Ω device is the closest value, and a 0.33W, 1210
size device will handle the power dissipation of 290 mW. With
the resistance selected, the average value of LED current is
re-calculated to ensure that current is within the ±5% toler-
ance requirement. From the expression for average LED
current:
IF = 0.2 / 0.13 = 1.54A, 3% above the target current
INPUT CAPACITOR
Following the calculations from the Input Capacitor section,
ΔvIN(MAX) will be 24V x 2%P-P = 480 mV. The minimum re-
quired capacitance is calculated for the largest tON, corre-
sponding to five LEDs:
CIN(MIN) = (1.5 x 1.5 x 10-6) / 0.48 = 4.7 µF
As with the output capacitor, this required value is low enough
to use a ceramic capacitor, and again the effective capaci-
tance will be lower than the rated value with 24V across CIN.
Reviewing plots of %C vs. DC Bias for several capacitors re-
veals that a 4.7 µF, 1812-size capacitor in X7R rated to 50V
loses about 40% of its rated capacitance at 24V, hence two
such caps are needed.
Input rms current is high in buck regulators, and the worst-
case is when the duty cycle is 50%. Duty cycle in a buck
regulator can be estimated as D = VO / VIN, and when this
converter drives three LEDs the duty cycle will be nearly 50%.
IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA
Ripple current ratings for 1812 size ceramic capacitors are
typically higher than 2A, so two of them in parallel can tolerate
more than enough for this design.
RECIRCULATING DIODE
The input voltage of 24V ±5% requires Schottky diodes with
a reverse voltage rating greater than 30V. The next highest
standard voltage rating is 40V. Selecting a 40V rated diode
provides a large safety margin for the ringing of the switch
node and also makes cross-referencing of diodes from differ-
ent vendors easier.
The next parameters to be determined are the forward current
rating and case size. The lower the duty cycle the more ther-
mal stress is placed on the recirculating diode. When driving
one LED the duty cycle can be estimated as:
D = 4.1 / 24 = 0.17
The estimated average diode current is then:
ID = (1 - 0.17) x 1.54 = 1.28A
A 2A-rated diode will be used. To determine the proper case
size, the dissipation and temperature rise in D1 can be cal-
culated as shown in the Design Considerations section. VD
for a case size such as SMB in a 40V, 2A Schottky diode at
1.5A is approximately 0.4V and the θJA is 75°C/W. Power dis-
sipation and temperature rise can be calculated as:
PD = 1.28 x 0.4 = 512 mW
TRISE = 0.51 x 75 = 38°C
19 www.national.com
LM3406/LM3406HV
CB, CC AND CF
The bootstrap capacitor CB should always be a 22 nF ceramic
capacitors with X7R dielectric. A 25V rating is appropriate for
all application circuits. The COMP pin capacitor CC and the
linear regulator filter capacitor CF should always be 100 nF
ceramic capacitors, also with X7R dielectric and a 25V rat-
ings.
EFFICIENCY
To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. Electrical efficiency, η, should not be confused
with the optical efficacy of the circuit, which depends upon the
LEDs themselves. One calculation will be detailed for three
LEDs in series, where VO = 11.8V, and these calculations can
be repeated for other numbers of LEDs.
Total output power, PO, is calculated as:
PO = IF x VO = 1.54 x 11.8 = 18.2W
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (1.542 x 0.75) x 0.5 = 890 mW
Gate charging and VCC loss, PG, in the gate drive and linear
regulator:
PG = (IIN-OP + fSW x QG) x VIN
PG = (600 x 10-6 + 5 x 105 x 9 x 10-9) x 24 = 122 mW
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW
PS = 0.5 x 24 x 1.54 x 40 x 10-9 x 5 x 105 = 370 mW
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = 0.752 0.003 = 2 mW (negligible)
DCR loss, PL, in the inductor
PL = IF2 x DCR = 1.542 x 0.06 = 142 mW
Recirculating diode loss, PD = (1 - 0.5) x 1.54 x 0.4 = 300 mW
Current Sense Resistor Loss, PSNS = 293 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) =
18.2 / (18.2 + 2.1) = 89%
Temperature Rise in the LM3406 IC is calculated as:
TLM3406 = (PC + PG + PS) x θJA = (0.89 + 0.122 + 0.37) x 50 =
69°C
Design Example 2
The second example circuit uses the LM3406 to drive a single
white LED at 1.5A ±10% with a ripple current of 20%P-P in a
typical 12V automotive electrical system. The two-wire dim-
ming function will be employed in order to take advantage of
the legacy 'theater dimming' method which dims and bright-
ens the interior lights of automobiles by chopping the input
voltage with a 200Hz PWM signal. As with the previous ex-
ample, the typical VF of a white LED is 3.9V, and with the
current sense voltage of 0.2V the total output voltage will be
4.1V. The LED driver must operate to specifications over an
input range of 9V to 16V as well as operating without suffering
damage at 28V for two minutes (the 'double battery jump-
start' test) and for 300 ms at 40V (the 'load-dump' test). The
LED driver must also be able to operate without suffering
damage at inputs as low as 6V to satisfy the 'cold crank' tests.
A complete bill of materials can be found in Table 2 at the end
of this datasheet.
30020325
FIGURE 15. Schematic for Design Example 2
www.national.com 20
LM3406/LM3406HV
RON and tON
A switching frequency of 450 kHz helps balance the require-
ments of inductor size and overall power efficiency, but more
importantly keeps the switching frequency below 530 kHz,
where the AM radio band begins. This design will concentrate
on meeting the switching frequency and LED current require-
ments over the nominal input range of 9V to 16V, and will then
check to ensure that the transient conditions do not cause the
LM3406 to overheat. The LM3406 will allow a small shift in
switching frequency when VIN changes, so the calculation for
RON is done at the typical expected condition where VIN =
13.8V and VO = 4.1V. The actual RON calculation uses the
high accuracy equation listed in the Appendix.
RON = 124 kΩ
The closest 1% tolerance resistor is 124 kΩ. The switching
frequency and on-time of the circuit should be checked at
VIN-MIN and VIN-MAX which are 9V and 16V, respectively. The
actual fSW and tON values have been calculated with the high
accuracy equations in the Appendix.
fSW(VMIN) = 463 kHz
fSW(VMAX) = 440 kHz
tON(VMIN) = 1090 ns
tON(VMAX) = 650 ns
OUTPUT INDUCTOR
Since an output capacitor will be used to filter some of the
LED ripple current, the inductor ripple current can be set high-
er than the LED ripple current. A value of 40%P-P is typical in
many buck converters:
ΔiL = 0.4 x 1.5 = 0.6AP-P
The minimum inductance required to ensure a ripple current
of 600 mAP-P or less is calculated at VIN-MAX:
LMIN = [(16 – 4.1) x 6.5 x 10-7] / (0.6) = 12.9 µH
The closest standard inductor value is 15 µH. The average
current rating should be greater than 1.5A to prevent over-
heating in the inductor. Inductor current ripple should be
calculated for VIN-MIN and VIN-MAX:
ΔiL(VMIN) = [(9 - 4.1) x 6.5 x 10-7] / 15 x 10-6
= 357 mAP-P
ΔiL(VMAX) = [(16 - 4.1) x 1.09 x 10-6] / 15 x 10-6
= 516 mAP-P
The peak LED/inductor current is then estimated. This calcu-
lation uses the worst-case ripple current which occurs at VIN-
MAX.
IL(PEAK) = IL + 0.5 x ΔiL(MAX)
IL(PEAK) = 1.5 + 0.5 x 0.516 = 1.76A
In order to prevent inductor saturation the inductor’s peak
current rating must be above 1.8A. A 15 µH off-the shelf in-
ductor rated to 2.4A (peak) and 2.2A (average) with a DCR of
47 mΩ will be used.
USING AN OUTPUT CAPACITOR
This application does not require high frequency PWM dim-
ming, allowing the use of an output capacitor to reduce the
size and cost of the output inductor while still meeting the 20%
P-P (300 mA) target for LED ripple current. To select the proper
output capacitor the equation from Buck Regulators with Out-
put Capacitors is re-arranged to yield the following:
The dynamic resistance, rD,of one LED can be calculated by
taking the tangent line to the VF vs. IF curve in the LED
datasheet. Figure 14 shows an example rD calculation.
30020324
FIGURE 16. Calculating rD from the VF vs. IF Curve
Extending the tangent line to the ends of the plot yields values
for ΔVF and ΔIF of 0.7V and 2000 mA, respectively. Dynamic
resistance is then:
21 www.national.com
LM3406/LM3406HV
rD = ΔVF / ΔIF = 0.5V / 2A = 0.25Ω
The most filtering (and therefore the highest output capaci-
tance) is needed when ΔIL is highest, which occurs at VIN-
MAX. Inductor ripple current with one LED is 516 mAP-P. The
required impedance of CO is calculated:
ZC = [0.3 / (0.516 - 0.3] x 0.35 = 0.35Ω
A ceramic capacitor will be used and the required capacitance
is selected based on the impedance at 440 kHz:
CO = 1/(2 x π x 0.49 x 4.4 x 105) = 1.03 µF
This calculation assumes that CO will be a ceramic capacitor,
and therefore impedance due to the equivalent series resis-
tance (ESR) and equivalent series inductance (ESL) of of the
device is negligible. The closest 10% tolerance capacitor val-
ue is 1.5 µF. The capacitor used should have an X7R dielec-
tric and should be rated to 50V. The high voltage rating
ensures that CO will not be damaged if the LED fails open
circuit and a load dump occurs. Several manufacturers pro-
duce ceramic capacitors with these specifications in the 1206
case size. With only 4V of DC bias a 50V rated ceramic ca-
pacitor will have better than 90% of it's rated capacitance,
which is more than enough for this design.
RSNS
Using the expression for RSNS:
RSNS = 0.2 / IF
RSNS = 0.2 / 1.5 = 0.133Ω
Sub-1Ω resistors are available in both 1% and 5% tolerance.
A 1%, 0.13Ω device is the closest value, and a 0.33W, 1210
size device will handle the power dissipation of 290 mW. With
the resistance selected, the average value of LED current is
re-calculated to ensure that current is within the ±5% toler-
ance requirement. From the expression for average LED
current:
IF = 0.2 / 0.13 = 1.54A, 3% above the target current
INPUT CAPACITOR
Controlling input ripple current and voltage is critical in auto-
motive applications where stringent conducted electromag-
netic interference tests are required. ΔvIN(MAX) will be limited
to 300 mVP-P or less. The minimum required capacitance is
calculated for the largest tON, 1090 ns, which occurs at the
minimum input voltage. Using the equations from the Input
Capacitors section:
CIN(MIN) = (1.5 x 1.09 x 10-6) / 0.3 = 5.5 µF
As with the output capacitor, this required value is low enough
to use a ceramic capacitor, and again the effective capaci-
tance will be lower than the rated value with 16V across CIN.
Reviewing plots of %C vs. DC Bias for several capacitors re-
veals that a 3.3 µF, 1210-size capacitor in X7R rated to 50V
loses about 22% of its rated capacitance at 16V, hence two
such caps are needed.
Input rms current is high in buck regulators, and the worst-
case is when the duty cycle is 50%. Duty cycle in a buck
regulator can be estimated as D = VO / VIN, and when VIN
drops to 9V the duty cycle will be nearly 50%.
IIN-RMS = 1.5 x Sqrt(0.5 x 0.5) = 750 mA
Ripple current ratings for 1210 size ceramic capacitors are
typically higher than 2A, so two of them in parallel can tolerate
more than enough for this design.
RECIRCULATING DIODE
To survive an input voltage transient of 40V the Schottky
diode must be rated to a higher voltage. The next highest
standard voltage rating is 60V. Selecting a 60V rated diode
provides a large safety margin for the ringing of the switch
node and also makes cross-referencing of diodes from differ-
ent vendors easier.
The next parameters to be determined are the forward current
rating and case size. The lower the duty cycle the more ther-
mal stress is placed on the recirculating diode. When driving
one LED the duty cycle can be estimated as:
D = 4.1 / 13.8 = 0.3
The estimated average diode current is then:
ID = (1 - 0.3) x 1.54 = 1.1A
A 2A-rated diode will be used. To determine the proper case
size, the dissipation and temperature rise in D1 can be cal-
culated as shown in the Design Considerations section. VD
for a case size such as SMB in a 60V, 2A Schottky diode at
1.5A is approximately 0.4V and the θJA is 75°C/W. Power dis-
sipation and temperature rise can be calculated as:
PD = 1.1 x 0.4 = 440 mW
TRISE = 0.44 x 75 = 33°C
CB, CC AND CF
The bootstrap capacitor CB should always be a 22 nF ceramic
capacitors with X7R dielectric. A 25V rating is appropriate for
all application circuits. The COMP pin capacitor CC and the
linear regulator filter capacitor CF should always be 100 nF
ceramic capacitors, also with X7R dielectric and a 25V rat-
ings.
EFFICIENCY
To estimate the electrical efficiency of this example the power
dissipation in each current carrying element can be calculated
and summed. One calculation will be detailed for the nominal
input voltage of 13.8V, and these calculations can be repeat-
ed for other numbers of LEDs.
Total output power, PO, is calculated as:
PO = IF x VO = 1.54 x 4.1 = 6.3W
Conduction loss, PC, in the internal MOSFET:
PC = (IF2 x RDSON) x D = (1.542 x 0.75) x 0.3 = 530 mW
www.national.com 22
LM3406/LM3406HV
Gate charging and VCC loss, PG, in the gate drive and linear
regulator:
PG = (IIN-OP + fSW x QG) x VIN
PG = (600 x 10-6 + 4.5 x 105 x 9 x 10-9) x 13.8 = 64 mW
Switching loss, PS, in the internal MOSFET:
PS = 0.5 x VIN x IF x (tR + tF) x fSW
PS = 0.5 x 13.8 x 1.54 x 40 x 10-9 x 4.5 x 105 = 190 mW
AC rms current loss, PCIN, in the input capacitor:
PCIN = IIN(rms)2 x ESR = 0.752 0.003 = 2 mW (negligible)
DCR loss, PL, in the inductor
PL = IF2 x DCR = 1.542 x 0.05 = 120 mW
Recirculating diode loss, PD = (1 - 0.3) x 1.54 x 0.4 = 430 mW
Current Sense Resistor Loss, PSNS = 293 mW
Electrical efficiency, η = PO / (PO + Sum of all loss terms) =
6.3 / (6.3 + 1.6) = 80%
Temperature Rise in the LM3406 IC is calculated as:
TLM3406 = (PC + PG + PS) x θJA = (0.53 + 0.06 + 0.19) x 50 =
39°C
Thermal Considerations During
Input Transients
The error amplifier of the LM3406 ensures that average LED
current is controlled even at the transient load-dump voltage
of 40V, leaving thermal considerations as a primary design
consideration during high voltage transients. A review of the
operating conditions at an input of 40V is still useful to make
sure that the LM3406 die temperature is not exceeded.
Switching frequency drops to 325 kHz, the on-time drops to
350 ns, and the duty cycle drops to 0.12. Repeating the cal-
culations for conduction, gate charging and switching loss
leads to a total internal loss of 731 mW, and hence a die tem-
perature rise of 37°C. The LM3406 should operate properly
even if the ambient temperature is as high a 85°C.
Layout Considerations
The performance of any switching converter depends as
much upon the layout of the PCB as the component selection.
The following guidelines will help the user design a circuit with
maximum rejection of outside EMI and minimum generation
of unwanted EMI.
COMPACT LAYOUT
Parasitic inductance can be reduced by keeping the power
path components close together and keeping the area of the
loops that high currents travel small. Short, thick traces or
copper pours (shapes) are best. In particular, the switch node
(where L1, D1, and the SW pin connect) should be just large
enough to connect all three components without excessive
heating from the current it carries. The LM3406/06HV oper-
ates in two distinct cycles whose high current paths are shown
in Figure 17:
30020326
FIGURE 17. Buck Converter Current Loops
The dark grey, inner loop represents the high current path
during the MOSFET on-time. The light grey, outer loop rep-
resents the high current path during the off-time.
GROUND PLANE AND SHAPE ROUTING
The diagram of Figure 17 is also useful for analyzing the flow
of continuous current vs. the flow of pulsating currents. The
circuit paths with current flow during both the on-time and off-
time are considered to be continuous current, while those that
carry current during the on-time or off-time only are pulsating
currents. Preference in routing should be given to the pulsat-
ing current paths, as these are the portions of the circuit most
likely to emit EMI. The ground plane of a PCB is a conductor
and return path, and it is susceptible to noise injection just as
any other circuit path. The continuous current paths on the
ground net can be routed on the system ground plane with
less risk of injecting noise into other circuits. The path be-
tween the input source and the input capacitor and the path
between the recirculating diode and the LEDs/current sense
resistor are examples of continuous current paths. In contrast,
the path between the recirculating diode and the input capac-
itor carries a large pulsating current. This path should be
routed with a short, thick shape, preferably on the component
side of the PCB. Do not place any vias near the anode of
Schottky diode. Instead, multiple vias in parallel should be
used right at the pad of the input capacitor to connect the
component side shapes to the ground plane. A second pul-
sating current loop that is often ignored is the gate drive loop
formed by the SW and BOOT pins and capacitor CB. To min-
imize this loop and the EMI it generates, keep CB close to the
SW and BOOT pins.
CURRENT SENSING
The CS pin is a high-impedance input, and the loop created
by RSNS, RZ (if used), the CS pin and ground should be made
as small as possible to maximize noise rejection. RSNS should
therefore be placed as close as possible to the CS and GND
pins of the IC.
REMOTE LED ARRAYS
In some applications the LED or LED array can be far away
(several inches or more) from the LM3406/06HV, or on a sep-
arate PCB connected by a wiring harness. When an output
capacitor is used and the LED array is large or separated from
the rest of the converter, the output capacitor should be
placed close to the LEDs to reduce the effects of parasitic
inductance on the AC impedance of the capacitor. The current
sense resistor should remain on the same PCB, close to the
LM3406/06HV.
Remote LED arrays and high speed dimming with a parallel
FET must be treated with special care. The parallel dimming
FET should be placed on the same board and/or heatsink as
the LEDs to minimize the loop area between them, as the
23 www.national.com
LM3406/LM3406HV
switching of output current by the parallel FET produces a
pulsating current just like the switching action of the LM3406's
internal power FET and the Schottky diode. Figure 18 shows
the path that the inductor current takes through the LED or
through the dimming FET. To minimize the EMI from parallel
FET dimming the parasitic inductance of the loop formed by
the LED and the dimming FET (where only the dark grey ar-
rows appear) should be reduced as much as possible. Para-
sitic inductance of a loop is mostly controlled by the loop area,
hence making this loop as physically small (short) as possible
will reduce the inductance. 30020328
FIGURE 18. Parallel FET Dimming Current Loops
www.national.com 24
LM3406/LM3406HV
TABLE 1. BOM for Design Example 1
ID Part Number Type Size Parameters Qty Vendor
U1 LM3406 LED Driver eTSSOP-14 42V, 2A 1 NSC
L1 SLF10145T-220M1R-PF Inductor 10 x 10 x 4.5mm 22 µH, 1.9A, 59 mΩ1 TDK
D1 CMSH2-40 Schottky Diode SMB 40V, 2A 1 Central Semi
Cc, Cf VJ0603Y104KXXAT Capacitor 0603 100 nF 10% 2 Vishay
Cb VJ0603Y223KXXAT Capacitor 0603 22 nF 10% 1 Vishay
Cin1
Cin2
C4532X7R1H475M Capacitor 1812 4.7 µF, 50V 2 TDK
Co C2012X7R1E105M Capacitor 0805 1.0 µF, 25V 1 TDK
Rsns ERJ14RQFR13V Resistor 1210 0.13Ω 1% 1 Panasonic
Ron CRCW08051433F Resistor 0805 143 kΩ 1% 1 Vishay
TABLE 2. BOM for Design Example 2
ID Part Number Type Size Parameters Qty Vendor
U1 LM3406 LED Driver eTSSOP-14 42V, 2A 1 NSC
L1 SLF10145T-150M2R2-P Inductor 10 x 10 x 4.5mm 15 µH, 2.2A, 47 mΩ1 TDK
D1 CMSH2-60 Schottky Diode SMB 60V, 2A 1 Central Semi
Cc, Cf VJ0603Y104KXXAT Capacitor 0603 100 nF 10% 2 Vishay
Cb VJ0603Y223KXXAT Capacitor 0603 22 nF 10% 1 Vishay
Cin1
Cin2
C3225X7R1H335M Capacitor 1210 3.3 µF, 50V 2 TDK
Co C3216X7R1H105M Capacitor 1206 0.15 µF, 50V 1 TDK
Rsns ERJ14RQFR13V Resistor 1210 0.13Ω 1% 1 Panasonic
Ron CRCW08051243F Resistor 0805 124 kΩ 1% 1 Vishay
Rpd CRCW08051002F Resistor 0805 10 kΩ 1% 1 Vishay
TABLE 3. Bill of Materials for Efficiency Curves
ID Part Number Type Size Parameters Qty Vendor
U1 LM3406 Buck LED
Driver
eTSSOP-14 42V, 1.5A 1 NSC
Q1 Si3458DV N-MOSFET SOT23-6 60V, 2.8A 1 Vishay
D1 CMSH2-60M Schottky Diode SMA 60V, 2A 1 Central Semi
L1 VLF10045T-330M2R3 Inductor 10 x 10 x 4.5mm 33 µH, 2.3A, 70 mΩ1 TDK
Cin1 Cin2 C4532X7R1H685M Capacitor 1812 6.8 µF, 50V 2 TDK
Co C3216X7R1H474M Capacitor 1206 470 nF, 50V 1 TDK
Cf ,Cc VJ0603Y104KXXAT Capacitor 0603 100 nF 10% 2 Vishay
Cb VJ0603Y223KXXAT Capacitor 0603 22 nF 10% 1 Vishay
R3.5 ERJ6RQFR56V Resistor 0805 0.56Ω 1% 1 Panasonic
R.7 ERJ6RQFR62V Resistor 0805 0.62Ω 1% 1 Panasonic
R1 ERJ6RQFR30V Resistor 0805 0.3Ω 1% 1 Panasonic
R1.5 ERJ6RQFR16V Resistor 0805 0.16Ω 1% 1 Panasonic
Ron CRCW08051433F Resistor 0805 143kΩ 1% 1 Vishay
Rpd Rout CRCW06031002F Resistor 0603 10 kΩ 1% 2 Vishay
OFF*
DIM1
DIM2
160-1512 Terminal 0.062" 3 Cambion
VIN GND
CS/LED-
Vo/LED+
160-1026 Terminal 0.094" 2 Cambion
25 www.national.com
LM3406/LM3406HV
Appendix
The following expressions provide the best accuracy for users
who wish to create computer-based simulations or circuit cal-
culators:
www.national.com 26
LM3406/LM3406HV
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27 www.national.com
LM3406/LM3406HV
Physical Dimensions inches (millimeters) unless otherwise noted
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NS Package Number MXA14A
www.national.com 28
LM3406/LM3406HV
Notes
29 www.national.com
LM3406/LM3406HV
Notes
LM3406/06HV 1.5A Constant Current Buck Regulator for Driving High Power LEDs
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