MAX16809EVKIT+ - Maxim Integrated Products, Inc.

General Description

The MAX16809 evaluation kit (EV kit) is a 16-channel,

constant-current LED driver, capable of driving 40mA

each to 16 LED strings with a total forward voltage of

up to 32V. The MAX16809 EV kit is based on the

MAX16809 device, which has 16 constant-current-sink-

ing outputs with sink current settable using a single

resistor and a high-performance, current-mode pulse-

width-modulator (PWM) controller, for implementing a

DC-DC converter that generates the supply voltage to

drive the LED strings.

The MAX16809 EV kit operates at supply voltages

between 9V to 16V and temperatures ranging from

0°C to +70°C. It features a PWM dimming control,

adaptive control of the LED supply voltage, which

depends upon the operating voltage of the LED strings,

a built-in clock generator, and a low-current shutdown.

The MAX16809 EV kit is a fully assembled and tested

board.

Features

♦9V to 16V Supply Voltage Range

♦40mA LED Current (Per Each LED String)

♦Single-Resistor Current Adjust for 16 Channels

♦Up to 32V LED String Voltage

♦Boost Converter to Generate LED Supply Voltage

♦Adaptive LED Supply Voltage Control Increases

Efficiency

♦PWM Dimming Control

♦Output-Voltage-Spike Protection for Inductive-

Output Lines

♦Proven PCB Layout

Evaluates: MAX16809

MAX16809 Evaluation Kit

________________________________________________________________

Maxim Integrated Products

19-0821; Rev 0; 5/07

Component List

For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at

1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

Ordering Information

Denotes a lead-free and RoHS-compliant EV kit.

This limited temperature range applies to the EV kit PCB only.

The MAX16809 IC temperature range is -40°C to +125°C.

†

EP = Exposed paddle.

PART TEMP RANGE IC PACKAGE

MAX16809EVKIT+ 0°C to +70°C* 38 TQFN-EP†

DESIGNATION QTY DESCRIPTION

C1–C4, C6, C7,

C8, C10, C11,

C19–C23, C25,

C26

1nF ±10%, 50V X7R capacitors

(0603)

Murata GRM188R71H103KA01D

KEMET C0603C103K5RACTU

C5, C24, C30,

C31, C33 5

0.1µF ± 10%, 50V X7R capacitors

(0603)

Murata GRM188R71H104KA93D

TDK C1608X7R1H104K

C9 1

1µF ±20%, 16V X7R capacitor

(0805)

Murata GRM21BR71C105KA01L

TDK C2012X7R1C105K

C12, C13, C14 3

22µF ±20% 35V electrolytic

capacitors

Panasonic EEEFK1V220R

C15 1

1µF ±20%, 50V X7R capacitor

(1210)

KEMET C3225X7R1H105M

Murata GRM32ER71H105KA01L

DESIGNATION QTY DESCRIPTION

C16, C17, C18 3

22µF ±20%, 50V electrolytic

capacitors

Panasonic EEEFK1H220P

C27 1

560pF ±10%, 50V C0G capacitor

(0603)

KEMET C0603C561K5RACTU

Murata GRM188R71H561KA01D

C28 1

10pF ±10%, 50V C0G capacitor

(0603)

TDK C1608C0G1H00DB

Murata GRM1885C1H100JA01D

C29 1

220pF ±10%, 50V C0G capacitor

(0603)

KEMET C0603C221K5RACTU

Murata GRM188R71H221KA01D

C32 1

100pF ±10%, 50V C0G capacitor

(0603)

KEMET C0603C101K5RACTU

Murata GRM188R71H101KA01D

Evaluates: MAX16809

MAX16809 Evaluation Kit

2 _______________________________________________________________________________________

Component Suppliers

Note: Indicate that you are using the MAX16809 when contacting these component suppliers.

Component List (continued)

DESIGNATION QTY DESCRIPTION

C34 1

10nF ±10%, 50V X7R capacitor

(0603)

KEMET C0603C103K5RACTU

Murata GRM188R71H103KA01D

C35 0 Not installed, capacitor

D1 1 33V zener diode (SOD323)

Diodes Inc. MMSZ5257BS-7

D2 1

1A, 40V S chottky d i od e ( S M A)

Diodes Inc. CMSH1-40M

C entr al S em i cond uctor C M S H 1- 40M

D3–D10 8 15V dual zener diodes (SOT323)

Diodes AZ23C15-7-F

D11 1 20mA switching diode (SOD323)

Diodes Inc. 1N4148WS-7

D12 1

40V small-signal Schottky diode

(SOD523)

Diodes Inc. SDM03U40

GND, GND,

PWM, SHDN,

VBIAS, VIN

6 Wire loops

J1 1 0.1in 20-pin header

L1 1 27µH, 3.2A inductor

Coilcraft MSS1260-273ML

Q1 1 Switching transistor (SOT523)

Diodes Inc. MMBT2222AT-7-F

DESIGNATION QTY DESCRIPTION

Q2 1 60V, 5.5A N-channel MOSFET

Vishay Si4450DY

R1 1 430Ω ±1% resistor (0603)

R2 1 330kΩ ±1% resistor (0603)

R3 1 8.45kΩ ±1% resistor (0603)

R4, R6, R10 3 22kΩ ±1% resistors (0603)

R5 1 180kΩ ±1% resistor (0603)

R7 1 10.5kΩ ±1% resistor (0603)

R8 1 10Ω ±1% resistor (0603)

R9 1 1.2kΩ ±1% resistor (0603)

R10 1 22kΩ ±1% resistor (0603)

R11 1 50kΩ ±1% resistor (0603)

R12 1 75mΩ ±1% resistor (0603)

R13 0 Not installed, resistor

R14 1 2.2kΩ ±1% resistor (0603)

R15 1 10kΩ ±1% resistor (0603)

U1 1 MAX16809ATU+ (38-pin TQFN,

5mm x 7mm)

U2 1 Digital pnp transistor

ROHM DTA114WKA

U3 1 Digital npn transistor

ROHM DTC114WKA

U4 1

Dual inverter with hysteresis

Texas Instruments

SN74LVC2G14DCKR

— 1 PCB: MAX16809 Evaluation Kit+

SUPPLIER PHONE FAX WEBSITE

Central Semiconductor Corp. 631-435-1110 631-435-3388 www.centralsemi.com

Coilcraft, Inc. 847-639-6400 847-639-1469 www.coilcraft.com

Diodes Inc. 805-446-4800 805-446-4850 www.diodes.com

KEMET Corp. 978-658-1663 978-658-1790 www.kemet.com

Murata Mfg. Co., Ltd. 770-436-1300 770-436-3030 www.murata.com

ROHM Co., Ltd. 858-625-3630 858-625-3670 www.rohm.com

TDK Corp. 847-390-4373 847-390-4428 www.component.tdk.com

Texas Instruments Inc. — — www.ti.com

Vishay 402-563-6866 402-563-6296 www.vishay.com

Evaluates: MAX16809

MAX16809 Evaluation Kit

_______________________________________________________________________________________ 3

Quick Start

Recommended Equipment

•One 16V, 5A adjustable power supply

•One 5V power supply

•16 LED strings with a total forward voltage ≤32V

•One multimeter

•One PWM signal generator (optional)

Procedure

The MAX16809 EV kit is fully assembled and tested.

Follow the steps below to verify operation. Caution: Do

not turn on the power supply until all connections

are completed.

1) Connect LED strings with operating voltage of

approximately 32V between VLED (pins 1-4 of J1)

and OUT0–OUT15 (pins 5-20 of J1). All 16 channels

should have an LED string load connected of the

same type.

2) Connect the DC power supply (16V, 5A) to VIN and

GND.

3) Connect a DC power supply (0 to 5V) to VBIAS and

GND.

4) Turn on the power supplies and apply 10V to VIN

and 3V to 5V to VBIAS. Connect SHDN and PWM to

3V to 5V. All of the LEDs should turn on. Measure

the current through any LED string, which should be

40mA ±7%.

5) Increase the supply voltage to 16V and the LED

currents will be stable. Measure the current through

any LED string, which should be 40mA ±7%.

6) Apply a PWM signal with amplitude of 3V to 5V and

a frequency between 100Hz and 2kHz to the PWM

input. The LED brightness should increase as the

PWM duty cycle increases and viceversa.

7) Connect SHDN to GND and all LEDs should turn off.

Detailed Description

The MAX16809 EV kit is a 16-channel, constant-current

LED driver capable of driving 40mA each to 16 LED

strings, with a total forward voltage of up to 32V. The

MAX16809 EV kit can drive a total of 160 white LEDs in

16 strings, with operating current up to 40mA. The

MAX16809 EV kit can operate at input supply voltages

between 9V and 16V.

The MAX16809 EV kit evaluates the MAX16809 IC, which

has two major sections. The first section consists of 16

constant-current LED drivers capable of sinking up to

55mA when on and blocking up to 36V when off. The sec-

ond section is a high-performance current-mode PWM

controller that can control a DC-DC converter to generate

the supply voltage for driving the LED strings. The

MAX16809 EV kit uses the PWM controller to drive a

boost converter, which takes a 9V to 16V input and gen-

erates a 33V LED supply voltage. To drive a constant cur-

rent into an LED string, connect the LED string between

the 33V output and any of the 16 constant-current-sink

outputs. The resistor (R1) from the SET pin to ground pro-

grams the sink current of each output. The sink current of

any output can be up to 55mA and the amplitude is the

same value for all the outputs. The difference between the

total forward voltage and the LED supply voltage drops

between the constant-current-sink output and ground,

and is dissipated as power in the device.

The LED supply voltage generated by the boost con-

verter in the MAX16809 EV kit is adaptive. The LED

string with the highest total forward voltage dominates

the control loop, and the boost-converter voltage is

adjusted so that the driver associated with that string

receives just enough voltage required for current drive.

All the other strings, with lower total forward voltages,

will have excess supply voltage, which is then dropped

in the associated driver. This feedback mechanism

ensures that the linear-current-control circuit dissipates

the minimum possible power. An on-board inverter

(U4A) is configured to generate the clock input for the

MAX16809. The constant-current output-driver circuits

and U4 need a 3.3V to 5V input, which should be sup-

plied externally. If 5V is not available, it can be generat-

ed using an emitter-follower buffer from the REF output

of MAX16809.

Boost Converter

The boost converter that generates the 33V LED supply

voltage operates at a switching frequency of 350kHz in

continuous-conduction mode (CCM). The current-mode

PWM controller in the MAX16809 drives the external

MOSFET (Q2) to control the boost converter. The

MOSFET is turned on at the beginning of every switching

cycle and turned off when the current through the induc-

tor (L1) reaches the peak value set by the error-amplifier-

output voltage. Inductor current is sensed from the volt-

age across the ground-referenced current-sense resis-

tor (parallel combination of R12 and R13). This current-

sense information is passed on to the current-sense

comparators in the MAX16809 through the CS pin.

During the on period of the MOSFET, the inductor

stores energy from the input supply. When the switch is

turned off, the inductor generates sufficient voltage in

reverse direction to discharge the stored energy to

VLED. This generated voltage forms a source, in series

with the input supply voltage, and drives VLED through

the rectifier diode (D2).

Evaluates: MAX16809

MAX16809 Evaluation Kit

4 _______________________________________________________________________________________

As the boost converter is operated in CCM, only part of

the stored energy in the inductor is discharged to VLED.

The advantages of CCM include reduced input and out-

put filtering, reduced EMI due to lower peak currents,

and higher converter efficiency. However, these advan-

tages come at the cost of a right-half-plane zero in the

converter-transfer function. Compensating this zero

requires reducing the system bandwidth, which affects

the converter-dynamic response. As the 16-channel,

constant-current-sink outputs control the current through

the LEDs, slower control of VLED does not affect the

LED operation. Compensation of the feedback circuit is

explained in the

Feedback Compensation

section.

An internal comparator turns off the gate pulse to the

external MOSFET if the voltage at the CS pin exceeds

0.3V. The current through the inductor that produces

0.3V at the CS pin is the maximum inductor current

possible (the actual current can be a little higher than

this limit due to the 60ns propagation delay from the CS

pin to the MOSFET drive output). This condition can

happen when the feedback loop is broken, when the

output capacitor charges during power-up, or when

there is an overload at the output. This feature protects

the MOSFET by limiting the maximum current passing

through it during such conditions.

The RC filter, consisting of R9 and C10, removes the

voltage spike across the current-sense resistors pro-

duced by the turn-on gate current of the MOSFET and

the reverse-recovery current of D2. Without filtering,

these current spikes can cause sense comparators to

falsely trigger and turn off the gate pulse prematurely.

The filter time constant should not be higher than

required (the MAX16809 EV kit uses a 120ns time con-

stant), as a higher time constant adds additional delay

to the current-sense voltage, effectively increasing the

current limit.

During normal operating conditions, the feedback loop

controls the peak current. The error amplifier compares

a scaled-down version of the LED supply voltage

(VLED) with a highly accurate 2.5V reference. The error

amplifier and compensation network then amplify the

error signal, and the current comparator compares this

signal to the sensed-current voltage to create a PWM

drive output.

Power-Circuit Design

Initially, decide the input supply voltage range, output

voltage VLED (the sum of the maximum LED total for-

ward voltage and 1V bias voltage for the constant-cur-

rent-sink output), and the output current IOUT (the sum

of all the LED string currents).

Calculate maximum duty cycle DMAX using the following

equation:

where VDis the forward drop of the rectifier diode D2

(~0.6V), VINMIN is the minimum input supply voltage (in

this case, 9V), and VFET is the average drain-to-source

voltage of the MOSFET Q2 when it is on.

Select the switching frequency FSW based on the

space, noise, dynamic response, and efficiency con-

straints. Select the maximum peak-to-peak ripple on

the inductor current ILPP. For the MAX16809 EV kit,

FSW is 350kHz and ILPP is ±30% of the average induc-

tor current. Use the following equations to calculate the

maximum average-inductor current ILAVG and peak

inductor current ILPEAK:

Since ILPP is ±30% of the average-inductor current

ILAVG:

Calculate the minimum inductance value LMIN with the

inductor current ripple set to the maximum value:

Choose an inductor that has a minimum inductance

greater than this calculated value.

Calculate the current-sense resistor (R12 in parallel

with R13) using the equation below:

where 0.3V is the maximum current-sense signal volt-

age. The factor 0.75 is for compensating the reduction

of maximum current-sense voltage due to the addition

of slope compensation. Check this factor and adjust

after the slope compensation is calculated. See the

Slope Compensation

section for more information.

RIL

CS PEAK

=×03 075..

LVIN V D

FIL

MIN MIN FET MAX

SW PP

=−×

()

IL IL IL

PEAK AVG PP

IL IL

PP AVG

=××03 2.

IL I

AVG OUT

MAX

=−1

DVLED V VIN

VLED V V

MAX DMIN

D FET

=+−

+−

Evaluates: MAX16809

MAX16809 Evaluation Kit

_______________________________________________________________________________________ 5

The saturation current limit of the selected inductor

(ILSAT) should be greater than the value given by the

equation below. Selecting an inductor with 10% higher

ILSAT rating is a good choice:

Calculate the output capacitor COUT (parallel combina-

tion of C16, C17, C18, and C24) using the following

equation:

where VLEDPP is the peak-to-peak ripple in the LED

supply voltage. The value of the calculated output

capacitance will be much lower than what is actually

necessary for feedback loop compensation. See the

Feedback Compensation

section to calculate the out-

put capacitance based on the compensation require-

ments.

Calculate the input capacitor CIN (parallel combination

of C12, C13, C14, and C5) using the following equation:

where VINPP is the peak-to-peak input ripple voltage.

This equation assumes that input capacitors supply

most of the input ripple current.

Selection of Power Semiconductors

The switching MOSFET (Q2) should have a voltage rat-

ing sufficient to withstand the maximum output voltage,

together with the diode drop of D2, and any possible

overshoot due to ringing caused by parasitic induc-

tances and capacitances. Use a MOSFET with voltage

rating higher than that calculated by the following

equation:

The factor of 1.3 provides a 30% safety margin.

The continuous drain-current rating of the selected

MOSFET when the case temperature is at +70°C should

be greater than that calculated by the following equation.

The MOSFET must be mounted on a board, as per

manufacturer specifications, to dissipate the heat:

The MOSFET dissipates power due to both switching

losses, as well as conduction losses. Use the following

equation to calculate the conduction losses in the

MOSFET:

where RDSON is the on-state drain-source resistance of

the MOSFET with an assumed junction temperature of

100°C.

Use the following equation to calculate the switching

losses in the MOSFET:

where IGON and IGOFF are the gate currents of the

MOSFET (with VGS equal to the threshold voltage)

when it is turned on and turned off, respectively, and

CGD is the gate-to-drain MOSFET capacitance. Choose

a MOSFET that has a higher power rating than that cal-

culated by the following equation when the MOSFET

case temperature is at +70°C:

The MAX16809 EV kit uses a Schottky diode as the

boost-converter rectifier (D2). A Schottky rectifier diode

produces less forward drop and puts the least burden

on the MOSFET during reverse recovery. If a diode with

considerable reverse-recovery time is used, it should be

considered in the MOSFET switching-loss calculation.

The Schottky diode selected should have a voltage rat-

ing 20% above the maximum boost-converter output

voltage. The current rating of the diode should be

greater than IDin the following equation:

IIL

DAVG

MAX

=−

⎛

⎝

⎜

⎞

⎠

⎟

⎟×

112.

PP P

TOT COND SW

PIL VLED C F

SW AVG GD SW

GON GOFF

=×××

×+

⎛

⎝

⎜

⎞

⎠

⎟

PIL

DRDS

COND AVG

MAX ON

=×

ID IL

RMS AVG

MAX

⎛

⎝

⎜

⎞

⎠

⎟

⎟×

13.

V VLED V

DS D

()

×13.

CIL

F VIN

IN PP

SW PP

=××8

CDI

VLED F

OUT MAX OUT

PP SW

=×

IL IL

SAT PEAK

=×11.

Evaluates: MAX16809

MAX16809 Evaluation Kit

6 _______________________________________________________________________________________

Slope Compensation

When the boost converter operates in CCM with more

than 50% duty cycle, subharmonic oscillations occur if

slope compensation is not implemented. Subharmonic

oscillations do not allow the PWM duty cycle to settle to

a peak current value set by the voltage-feedback loop.

The duty cycle oscillates back and forth about the

required value, usually at half the switching frequency.

Subharmonic oscillations die out if a sufficient negative

slope is added to the inductor peak current. This

means that for any peak current set by the feedback

loop, the output pulse terminates sooner than normally

expected. The minimum slope compensation that

should be added to stabilize the current loop is half of

the worst-case (max) falling slope of inductor current.

Adding a ramp to the current-sense signal, with posi-

tive slope in sync with the switching frequency, can

produce the desired function. The greater the duty

cycle, the greater the added voltage, and the greater

the difference between the set current and the actual

inductor current. In the MAX16809 EV kit, the oscillator

ramp signal is buffered using Q1 and added to the cur-

rent-sense signal with proper scaling to implement the

slope compensation. Follow the steps below to calcu-

late the component values for slope compensation.

Calculate the worst-case falling slope of the inductor

current using the following equation:

From the inductor current falling slope, find its equiva-

lent voltage slope across the current-sense resistor RCS

(R12 parallel with R13) using the following equation:

The minimum voltage slope that should be added to

the current-sense waveform is half of VSLOPE for ensur-

ing stability up to 100% duty cycle. As the maximum

continuous duty cycle used is less than 100%, the mini-

mum required compensation slope becomes:

The factor 1.1 provides a 10% margin. Resistors R9

and R10 determine the attenuation of the buffered volt-

age slope from the emitter of Q1. The forward drop of

signal diode D11, together with the VBE of Q1, almost

cancel the 1.1V offset of the ramp waveform. Calculate

the approximate slope of the oscillator ramp using the

following equation:

where 1.7V is the ramp amplitude and FSW is the

switching frequency.

Select the value of R9 such that the input bias current of

the current-sense comparators does not add consider-

able error to the current-sense signal. The value of R10

for the slope compensation is given by the equation:

LED Driver

The MAX16809 features a 16-channel, constant-current

LED driver, with each channel capable of sinking up to

55mA of LED current. The LED strings are connected

between VLED and the constant-current-sink outputs to

drive regulated current through LED strings. The cur-

rent through all 16 channels is controlled through resis-

tor (R1) from the SET pin to ground. The MAX16809 EV

kit sets the current through each string at 40mA and the

maximum LED supply voltage to 33V. The MAX16809

EV kit drives LED strings with a total forward voltage of

up to 32V.

A 4-wire serial interface with four inputs (DIN, CLK, LE,

and OE) individually control the constant-current out-

puts. In the MAX16809 EV kit, a 50kHz clock signal,

generated by U4A, clocks 16 1s into the internal shift

circuit can be avoided if a microcontroller provides the

function.

The output enable (OE) can provide PWM dimming. An

inverted PWM signal, generated by the inverter U4B, is

necessary to drive the OE pin. When the PWM signal is

low (LED drivers off), it also influences the feedback

with the network formed by R6 and D12. See the

Adaptive LED Supply Voltage Control

section for more

details.

If an inverted PWM signal is available, use the circuit

shown in Figure1 to drive the OE input and feedback

network.

RVR

VC R

SLOPE

10 1 9=−

⎛

⎝

⎜

⎞

⎠

⎟×

VR F

SLOPE SW

=×17.

VC VD

SLOPE SLOPE MAX

MAX

=×−×().2111

VILR

SLOPE SLOPE CS

=×

IL VLED V VIN

SLOPE MAX D MIN

MIN

=+−()

Evaluates: MAX16809

MAX16809 Evaluation Kit

_______________________________________________________________________________________ 7

Output Current Setting

The amplitude of the output sink currents for all 16

channels is set to the same value by the resistor (R1)

from the SET pin to ground. The minimum allowed

value of RSET is 311Ω, which sets the output currents to

55mA. The maximum allowed value of RSET is 5kΩ. The

MAX16809 EV kit uses 430Ωfor RSET, which sets the

output current to 40mA. To set a different output cur-

rent, use the following equation:

where RSET is the current-setting resistor (R1) value in

ohms and IOUT is the desired output current in milliamps.

Adaptive LED Supply Voltage Control

To reduce power dissipation in the IC, the MAX16809

EV kit features adaptive control of VLED based on the

operating voltage of the LED strings. The constant-cur-

rent-sink outputs can sink stable currents with output

voltages as low as 0.8V. The voltage at each of the 16

outputs will be the difference between VLED and the

total forward voltage of the LED string connected to

that output. The MAX16809 EV kit implements a feed-

back mechanism to sense the voltage at each of the 16

constant-current-sink outputs. Using dual zener diodes

(D3–D10), the MAX16809 EV kit selects the lowest dri-

ver voltage (with the greatest LED string voltage) to

regulate. The boost-converter PWM then adjusts so that

VLED is high enough for this sink output to settle to

approximately 0.8V. All the other strings have sufficient

voltage, as their total forward voltages are lower. The

feedback mechanism ensures that the IC dissipates the

minimum possible power. For adaptive control to func-

tion efficiently connect LED strings to all 16 channels

and use an equal number of LEDs from the same bin in

each string. If some of the 16 channels are not used,

then the zener diodes (D3–D10) should be removed

from the unused channels.

Use the equation below to calculate the value of R2 to

get the required minimum voltage at the sink outputs:

where 2.5V is the feedback reference, VDZ is the for-

ward drop of the ORing diode (D3–D10), VS= 0.5V is

the required sink-output voltage, and VFLED is the nom-

inal total forward voltage of the LED strings. Select the

value of R2 such that R7 is approximately 10kΩ.

The zener diodes (D3–D10) also provide output over-

voltage protection. If an LED string gets partially or fully

shorted, making the sink-output voltage go high, the

15V zener diode connected to that output conducts in

reverse direction, and limits the VLED voltage. Under

this condition, the other LED strings might not turn on.

When the outputs are off, the LED drivers are at high

impedance and the feedback network now combines

R6 and D12 to provide a path for the feedback current

and to control VLED. Use the following equation to

R2 =+− ×

−−

(.)

VV R

FLED S

DZ S

25 7

RSET =17100

IOUT

VLED

COMP

VBIAS

PWM

OEB

U4B

33V

C28

10pF

C29

220pF

C30

0.1μF

OUT0 OUT1 OUT14 OUT15

12 12

AZ23C15-7

D12

SDM03U40

SN74LVC2G14

3D10

AZ23C15-7

330kΩ

22kΩ

R15

10kΩ

180kΩ

R11

50kΩ

10.5kΩ

Figure 1. Inverting PWM Drive Circuit

Evaluates: MAX16809

MAX16809 Evaluation Kit

8 _______________________________________________________________________________________

calculate the value of R6 to get the required LED sup-

ply voltage during PWM off time:

where 2.5V is the feedback-reference voltage, 0.4V is

the total voltage dropped by D4 and PWM input, and

VLEDOFF is the desired LED supply voltage during

PWM off time. VLEDOFF should be set to the worst-case

LED string voltage plus some additional headroom for

the LED drivers (0.8V), as well as a reserve voltage

(approximately 1V). The reserve voltage allows the

MAX16809 to provide current for very short PWM dim-

ming on-pulses. With pulses as low as 2µs, the VLED

control loop is not able to react, and the output capaci-

tors provide all the current. For longer PWM dimming

pulses, the control loop reacts and the supply operates

at the adaptive voltage level.

During an open LED condition, the 33V zener diode

(D1) limits the maximum LED supply voltage to 35.5V. If

VLED attempts to increase beyond this level, D1 con-

ducts in reverse direction and pulls the FB pin high,

which causes the boost regulator to cut back on the

PWM signal and reduce the output voltage.

PWM Dimming

The PWM dimming controls the LED brightness by

adjusting the duty cycle of the PWM input signal. A

high voltage at the PWM input enables the output cur-

rent; a low voltage turns off the output current. Connect

a signal with peak amplitude of 3V to 5V and with fre-

quency from 100Hz to 2kHz to the PWM input and vary

the duty cycle to adjust the LED brightness. The LED

brightness increases when the duty cycle increases

and vice versa. If an inverted PWM signal is available,

use that signal to implement PWM dimming, as shown

in Figure 1.

Feedback Compensation

Like any other circuit with feedback, the boost convert-

er that generates the supply voltage for the LED strings

needs to be compensated for stable control of its out-

put voltage. As the boost converter is operated in con-

tinuous-conduction mode, there exists a right-half-

plane (RHP) zero in the power-circuit transfer function.

This zero adds a 20dB/decade gain together with a 90-

degree phase lag, which is difficult to compensate. The

easiest way to avoid this zero is to roll off the loop gain

to 0dB at a frequency less than half of the RHP zero fre-

quency with a -20dB/decade slope. For a boost con-

verter, the worst-case RHP zero frequency (FZRHP) is

given by the following equation:

where DMAX is the maximum duty cycle, L is the induc-

tance of the inductor, and IOis the output current,

which is the sum of all the LED string currents.

The boost converter used in the MAX16809 EV kit is

operated with current-mode control. There are two

feedback loops within a current-mode-controlled con-

verter: an inner loop that controls the inductor current

and an outer loop that controls the output voltage. The

amplified voltage error produced by the outer voltage

loop is the input to the inner current loop that controls

the peak inductor current.

The internal current loop converts the double-pole 2nd-

order system, formed by the inductor and the output

capacitor COUT, to a 1st-order system having a single

pole consisting of the output filter capacitor and the out-

put load. As the output load is a constant current (i.e.,

very high Thevenin impedance), this pole is located near

the origin (0Hz). The phase lag created by the output

pole for any frequency will be 90 degrees. Since the

power-circuit DC gain is limited by other factors, the gain

starts falling at -20dB/decade from a non-zero frequency

before which the power-circuit gain stabilizes.

Total gain of the feedback loop at DC is given by the

following equation:

where GPis the power-circuit DC gain, and GEA is the

error-amplifier open-loop DC gain, typically 100dB. GFB

is the gain of the feedback network for adaptive control

of the VLED, which is seen from VLED to the error-

amplifier input (FB pin). The adaptive control senses

the voltages at the 16 constant-current-sink outputs

and adjusts the feedback to control these voltages to a

minimum value (Figure 2). As the LEDs carry constant

current, the voltage across the LEDs does not change

with variations in VLED. Any change in VLED directly

reflects to the constant-current-sink outputs and to the

error-amplifier input, making GFB equal to unity.

GGGG

TOT P EA FB

=× ×

ZRHP MAX

=−

××

VLED

2LI

()12

R6 =×

()

R2 2.5 - 0.4

VLED -2.5

OFF

Evaluates: MAX16809

MAX16809 Evaluation Kit

_______________________________________________________________________________________ 9

The DC gain of the power circuit is expressed as the

change in the output voltage, with respect to the

change in error-amplifier output voltage. As the boost

converter in the MAX16809 EV kit drives a constant-

current load, the power-circuit DC gain is calculated

based on a constant-current load:

Calculate the power-circuit DC gain using the following

equation:

where RCS is the current-sense resistor, FSW is the

switching frequency, and the factor 3 is to account for

the attenuation of error-amp output before it is fed to

the current-sense comparator.

The power-circuit gain is lowest at the minimum input

supply voltage and highest at the maximum input sup-

ply voltage. Any input supply voltage between 9V and

16V can be used for power-circuit gain calculation, as

the final compensation values obtained are the same.

Calculate the frequency FP2, at which the power-circuit

gain starts falling, at -20dB/decade using the following

equation:

where COUT is the output filter capacitor, which is the

parallel combination of C16, C17, C18, and C24. Adjust

the output capacitance so that the product of FP2 and

GPis below FZRHP / 6. The value of output capacitance

obtained this way will be much greater than the value

obtained using the maximum output voltage ripple

specification.

The compensation strategy is as follows. The gain-fre-

quency response of the feedback loop should cross 0dB

at or below half of the RHP zero frequency, with a slope of

-20dB/decade for the feedback to be stable and have

sufficient phase margin. The compensation network from

COMP pin to FB pin of the MAX16809 (formed by R5,

C28, C29, and R11) offers one dominant pole (P1), a zero

(Z1), and a high-frequency pole (P3). There are two very

low frequency poles and a zero in the loop before the

crossover frequency. The function of the zero (Z1) is to

compensate for the output pole and to reduce the slope

of the loop gain from -40dB/decade to -20dB/decade,

and also to reduce the phase lag by 90 degrees.

Choose the crossover frequency to be half of the worst-

case RHP zero frequency:

Place the zero (Z1) at one-third of the crossover fre-

quency, so that the phase margin starts improving from

a sufficiently lower frequency:

Use the following equation to calculate the dominant

pole location, so that the loop gain crosses 0dB at FC:

Since the open-loop gain of the error amplifier can have

variations, the dominant pole location can also vary

from device to device. In the MAX16809 EV kit, the

dominant pole location is decided by the error-amplifier

gain, so the combined effect is a constant-gain-band-

width product.

Select the value of R11 such that the input bias current

of the error amplifier does not cause considerable drop

across it. The effective AC impedance seen from the

FB pin is the sum of R11 and R7. It is preferable to

keep R7 much lower, compared to R11, to have better

control on the AC impedance. Find C29 using the fol-

lowing equation:

The location of the zero (Z1) decided by R5 and C29 is

given by the following equation:

Place the high-frequency pole (P3), formed by C28,

C29, and R5, at half the switching frequency to provide

further attenuation to any high-frequency signal propa-

gating through the system. The location of the high-fre-

quency pole (FP3) is given by the following equation,

and should be used to calculate the value of C28:

RCC

P3 1

×× +

⎛

⎝

⎜⎞

⎠

⎟

−

2π

FRC

529

=××

2π

CGRRF

EA P

29 1

2117

=××+×

π()

FFF

PZRHP Z

TOT P

=×

××

1= 3

CZRHP

= 2

CRG

PMAX

OUT CS P

=−

××××

()

L F VLED

IN CS

×× × +

⎛

⎝

⎜

⎞

⎠

⎟××

IN2

GVLED

POUT

=Δ

Evaluates: MAX16809

MAX16809 Evaluation Kit

10 ______________________________________________________________________________________

The MAX16809 EV kit uses electrolytic capacitors at the

output for filtering, so the zero produced by the ESR of

the capacitors can be low enough to be within or near

the crossover frequency. This zero should be compen-

sated using an additional pole (P4) placed at the ESR

zero location. The ESR zero frequency is calculated

using the following equation:

Use the following equation to calculate the value of C35

to place the pole (P4) at the ESR zero frequency:

If ceramic capacitors are used at the output for filtering,

the frequency of zero produced by the ESR and the

capacitance will be above the crossover frequency

(0dB gain frequency) of the feedback loop and need

not be considered in the compensation design.

Layout Considerations

LED driver circuits based on the MAX16809 device use

a high-frequency switching converter to generate the

supply voltage for LED strings. Proper care must be

taken while laying out the circuit to ensure proper opera-

tion. The switching-converter part of the circuit has

nodes with very fast voltage changes, producing high-

frequency electric fields, and branches with fast current

changes, producing high-frequency magnetic fields. As

the circuit converts power, the amplitude of these fields

will be high and can easily couple to sensitive parts of

the circuit, creating undesirable effects. Follow the

guidelines below to reduce noise as much as possible:

1) Connect the bypass capacitors from REF and VCC

as close as possible to the device and connect the

capacitor grounds to the analog ground plane

using vias close to the capacitor terminals. Connect

the AGND pin of the device to the analog ground

plane using a via close to the pin. Lay the analog

ground plane on the inner layer, preferably next to

the top layer. Use the analog ground plane to cover

the entire area under critical-signal components for

the power converter.

2) Keep the oscillator timing capacitor and resistor

very close to the RTCT pin and make the connec-

tion as short as possible. Connect the ground of the

timing capacitor to the analog ground plane using a

via close to the capacitor terminal. Make sure that

no switching node is present near the RTCT node

and keep the area of the copper connected to the

pin small. Keep the REF connection to the timing

resistor short and away from any switching node.

3) Have a power ground plane for the switching-con-

verter power circuit under the power components

(input filter capacitor, output filter capacitor, inductor,

MOSFET, rectifier diode, and current-sense resistor).

Connect all the ground connections to the power

ground plane using vias close to the terminals.

4) There are two loops in the power circuit that carry

high-frequency switching currents. One loop is

when the MOSFET is on (from the input filter capac-

itor positive terminal, through the inductor, the

MOSFET, and the current-sense resistor, to the

input capacitor negative terminal). The other loop is

when the MOSFET is off (from the input capacitor

positive terminal, through the inductor, the rectifier

diode, and the output filter capacitor, to the input

capacitor negative terminal). Analyze these two

loops and make the loop areas as small as possi-

ble. Wherever possible, have a return path on the

power ground plane for the switching currents on

the top-layer copper tracks, or through power com-

ponents. This reduces the loop area considerably

and provide a low inductance path for the switching

currents. Reducing the loop area also reduces radi-

ation during switching.

5) The gate-drive current of the MOSFET is another

high-frequency switching current to consider. There

are two major loops: one during the MOSFET turn-on

edge and the second during the turn-off edge. The

MOSFET turn-on loop is from the VCC bypass

capacitor positive terminal, through the MOSFET dri-

ver in the device, the gate-drive resistor, the MOS-

FET gate to source (CGS and CGD), and the current-

sense resistor to the VCC bypass capacitor negative

terminal. There is no direct path for the current from

the current-sense resistor to return to the VCC

bypass capacitor through the ground plane, as the

VCC bypass capacitor is connected to the analog

ground plane and the current-sense resistor is con-

nected to the power ground plane. The best solution

is to connect the analog ground plane to the power

ground plane directly under the MOSFET gate-drive

track. This ensures that the turn-off current also has a

return path on the ground plane.

6) The drain node of the MOSFET is a switching node.

Keep this node area small to reduce radiation and

capacitive coupling to other sensitive parts of the

circuit. However, the track should be wide enough

to carry the large switching currents.

7) Keep the node area and track length on the FB pin

small to reduce any noise pickup.

CFR

ZESR

35 1

=××π

FESR C

ZESR OUT

=××

2π

8) Connect the power ground plane for the constant-

current LED driver part of the circuit to the boost-

converter output filter capacitor negative terminal.

Power Dissipation

The MAX16809 dissipates power during normal operat-

ing conditions. The heat transferred to the exposed pad

from the die should be properly dissipated to the board

to prevent the device from entering into thermal shut-

down. The exposed pad land area on the top layer

should be of the same size as that of the exposed pad.

Thermal vias are used to carry the heat from the

exposed pad to other layers of the board and spread it

across the board area through copper planes. Thermal

vias should have a 0.4mm hole size and should be

placed at a distance of 1mm from center to center. For

a four-layer board, these vias should be connected to

the bottom ground plane and to one internal ground

plane. Do not use thermal relief for the thermal vias;

instead, use solid copper to get the minimum thermal

resistance.

Use the following equation to calculate the total power

dissipated in the MAX16809 device during normal

operation:

where IN is the LED current in channel N, VS is the

operating voltage of each of the LED driver outputs with

respect to GND pins, IBis the input bias current of

MAX16809 including the average of MOSFET drive cur-

rent, and VIN is the input supply voltage. To dissipate

1W of power, the exposed pad of the device should be

connected to a minimum of two square inches of cop-

per ground plane with 70µ copper thickness.

PVSINIV

NBIN

=×

∑+×

=− 015

Evaluates: MAX16809

MAX16809 Evaluation Kit

______________________________________________________________________________________ 11

Evaluates: MAX16809

MAX16809 Evaluation Kit

12 ______________________________________________________________________________________

CLK

VLED

CMSH1-40M

EEEFK1H220PEEEFK1V220R

VBIAS

VLED

VIN

OEB

CLK

VBIAS

OUT15

OUT14

OUT13

OUT12

OUT11

OUT10

OUT9

OUT8

OUT15

OUT14

OUT13

OUT12

OUT11

OUT10

OUT9

OUT8

COMP

N.C.

OUT

GND

DTA114WKA

DTC114WKA

OUT

GND

430Ω

G1 G1

CLK DIN PGND PGND V+

18 17 16 15 14 13

32 33 34 35 36 37 38

SET DOUT

N.C. AGND OUT VCC N.C. REF N.C.

VIN

PWM

SHDN

GND

VBIAS

33V

C28

10pF

C20

1nF

C21

1nF

C25

1nF

C26

1nF

C19

1nF

C22

1nF

C23

1nF

C10

1nF

C11

1nF

C29

220pF

0.1μF

C15

1μF

50V

C24

0.1μF

50V

C16

22μF

50V

C33

0.1μF

C34

10nF

C30

0.1μF

C31

0.1μF

1μF

12 12

AZ23C15-7

D12

SDM03U40

3D4

AZ23C15-7

330kΩ

22kΩ

180kΩ

1% R11

50kΩ

10.5kΩ

1% VLED

VLED

J1A

OUT0

OUT1

OUT2

OUT3

OUT4

OUT5

OUT6

OUT7

OUT0

OUT1

OUT2

OUT3

OUT4

OUT5

OUT6

OUT7

RTCT

N.C.

OUT7

OUT6

OUT5

OUT4

OUT3

OUT2

OUT1

OUT0

OUT15

OUT14

OUT13

OUT12

OUT11

OUT10

OUT9

OUT8

OUT15

OUT14

OUT13

OUT12

OUT11

OUT10

OUT9

OUT8

OUT7

OUT6

OUT5

OUT4

OUT3

OUT2

OUT1

OUT0

C35

OPEN

C32

100pF

8.45kΩ

22kΩ

C27

560pF

3D5

AZ23C15-7

1111

2222

3D6

AZ23C15-7

D10

U4B

SN74LVC2G14

U4A

D11

1N4148

27μH

MSS1278

Si4450DY

7865

SN74LVC2G14

MMBT2222

R15

10kΩ

R14

2.2kΩ

R10

22kΩ

R13

OPEN

R12

75mΩ

10Ω

1.2kΩ

C17

22μF

50V

C18

22μF

50V

C12

22μF

35V

C13

22μF

35V

C14

22μF

35V

MAX16809

Figure 2. MAX16809 EV Kit Schematic

Evaluates: MAX16809

MAX16809 Evaluation Kit

______________________________________________________________________________________ 13

Figure 3. MAX16809 EV Kit Component Placement Guide—Component Side

Evaluates: MAX16809

MAX16809 Evaluation Kit

14 ______________________________________________________________________________________

Figure 4. MAX16809 EV Kit PCB Layout—Component Side

Evaluates: MAX16809

MAX16809 Evaluation Kit

______________________________________________________________________________________ 15

Figure 5. MAX16809 EV Kit PCB Layout—Inner Layer 1

Evaluates: MAX16809

MAX16809 Evaluation Kit

16 ______________________________________________________________________________________

Figure 6. MAX16809 EV Kit PCB Layout—Inner Layer 2

Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are

implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________

Evaluates: MAX16809

MAX16809 Evaluation Kit

Figure 7. MAX16809 EV Kit PCB Layout—Solder Side