General Description
The MAX1951A high-efficiency, DC-DC step-down
switching regulator delivers up to 2A of output current.
The device operates from an input voltage range of 2.6V
to 5.5V and provides an adjustable output voltage from
0.8V to VIN, making the MAX1951A ideal for on-board
postregulation applications. The MAX1951A total output
error is less than ±1.5% over load, line, and temperature.
The MAX1951A operates at a fixed frequency of 1MHz
with an efficiency of up to 94%. The high operating fre-
quency minimizes the size of external components.
Internal soft-start control circuitry reduces inrush current.
Short-circuit and thermal-overload protection improve
design reliability.
The MAX1951A can start up safely with a prebiased or
without a preexisting output. This feature simplifies track-
ing supply designs for core and I/O applications and
redundant supply designs.
The MAX1951A is available in an 8-pin SO package
and operates over the -40°C to +85°C extended tem-
perature range.
Applications
ASIC/DSP/µP/FPGA Core and I/O Voltages
Set-Top Boxes
Networking and Telecommunications
Servers
TVs
Features
oCompact 0.385in2Circuit Footprint
o10µF Ceramic Input and Output Capacitors, 2µH
Inductor for 2A Output
oEfficiency Up to 94%
o1.5% Output Accuracy Over Load, Line, and
Temperature
oGuaranteed 2A Output Current
oOperates from 2.6V to 5.5V Supply
oAdjustable Output from 0.8V to VIN
oInternal Digital Soft-Soft
oShort-Circuit and Thermal-Overload Protection
o1MHz Switching Frequency Reduces Component
Size
oEnable Input Audio Shutdown for Reducing
Power Consumption
oSafe Startup into Prebiased Output
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
________________________________________________________________
Maxim Integrated Products
1
Ordering Information
19-4689; Rev 0; 7/09
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
EVALUATION KIT
AVAILABLE
PART TEMP RANGE PIN-PACKAGE
MAX1951AESA+ -40°C to +85°C 8 SO
MAX1951A
IN LX
COMP
OUTPUT
0.8V TO VIN, UP TO 2A
INPUT
2.6V TO 5.5V
ON
OFF
EN
FB
PGND GND
VCC
Typical Operating Circuit
Pin Configuration
PGND
COMPFB
1
+
2
8
7
IN
LXEN
GND
VCC
SO
TOP VIEW
3
4
6
5
MAX1951A
+
Denotes a lead(Pb)-free/RoHS-compliant package.
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(VIN = VCC = VEN = 3.3V, VPGND = VGND = 0V, FB in regulation, TA= -40°C to +85°C, unless otherwise noted. Typical values are at
TA= +25°C.) (Note 3)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
PARAMETER CONDITIONS MIN TYP MAX UNITS
IN AND VCC
IN Voltage Range 2.6 5.5 V
Supply Current Switching with no load, LX floating VIN = 5.5V 7 10 mA
Shutdown Current EN = GND 0.1 0.4 mA
VCC rising 2.19 2.32
VCC Undervoltage Lockout
Threshold When LX starts/stops switching VCC falling 1.92 2.07 V
COMP
COMP Transconductance From FB to COMP, VCOMP = 0.8V 40 50 80 µS
COMP Clamp Voltage, Low VIN = 2.6V to 5.5V, VFB = 0.9V 0.6 1 1.45 V
COMP Clamp Voltage, High VIN = 2.6V to 5.5V, VFB = 0.7V 1.97 2.13 2.28 V
FB
Output Voltage Range When using external feedback resistors to drive FB 0.8 VIN V
TA = 0°C to +85°C 0.789 0.796 0.804
FB Regulation Voltage
(Error Amplifier Only)
IOUT = 0A to 1.5A, VIN = 2.6V to
5.5V TA = -40°C to +85°C 0.786 0.804 V
FB Input Bias Current PNP input stage -0.1 +0.1 µA
LX
VIN = 5V 119
VIN = 3.3V 145 266
LX On-Resistance, PMOS ILX = -180mA
VIN = 2.6V 171
mΩ
VIN = 5V 122
VIN = 3.3V 133 246LX On-Resistance, NMOS ILX = 180mA
VIN = 2.6V 142
mΩ
IN, VCC to GND ........................................................-0.3V to +6V
COMP, FB, EN to GND...............................-0.3V to (VCC + 0.3V)
LX Current (Note 1).............................................................±4.5A
PGND to GND..............................................Internally connected
Continuous Power Dissipation (TA= +70°C)
8-Pin SO (derate 12.2mW/°C above +70°C).................976mW
Junction-to-Case Thermal Resistance (θJC) (Note 2)
8-Pin SO ........................................................................32°C/W
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature Range ............................-40°C to +150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Note 1: LX has internal clamp diodes to PGND and IN. Applications that forward bias these diodes should take care not to exceed
the IC’s package power dissipation limits.
Note 2: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-
layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.
EFFICIENCY vs. OUTPUT CURRENT
(VCC = VIN = 5V)
MAX1951A toc01
OUTPUT CURRENT (A)
EFFICIENCY (%)
1.61.20.80.4
10
20
30
40
50
60
70
80
90
100
0
02.0
VOUT = 3.3V VOUT = 2.5V
VOUT = 1.5V
EFFICIENCY vs. OUTPUT CURRENT
(VCC = VIN = 3.3V)
MAX1951A toc02
OUTPUT CURRENT (A)
EFFICIENCY (%)
1.61.20.80.4
10
20
30
40
50
60
70
80
90
100
0
02.0
VOUT = 2.5V
VOUT = 1.8V
VOUT = 1.5V
VOUT = 1.0V
SWITCHING FREQUENCY
vs. INPUT VOLTAGE
MAX1951A toc03
INPUT VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
5.04.54.03.53.0
925
950
975
1000
1025
1050
900
2.55.5
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VCC = VEN = 3.3V, VPGND = VGND = 0V, FB in regulation, TA= -40°C to +85°C, unless otherwise noted. Typical values are at
TA= +25°C.) (Note 3)
PARAMETER CONDITIONS MIN TYP MAX UNITS
LX Current-Sense
Transimpedance From LX to COMP, VIN = 2.6V to 5.5V 0.16 0.24 0.35 Ω
High side 2.2 3.1 4.5
LX Current-Limit Threshold Duty = 100%, VIN = 2.6V to 5.5V Low side -0.3 A
VLX = 5.5V 10
LX Leakage Current VIN = 5.5V VLX = 0V -10 µA
LX Switching Frequency VIN = 2.6V to 5.5V 0.8 0.96 1.1 MHz
LX Maximum Duty Cycle VCOMP = 1.5V, LX = Hi-Z, VIN = 2.6V to 5.5V 100 %
LX Minimum Duty Cycle VCOMP = 1V, IN = 2.6V to 5.5V 15 %
THERMAL
TJ rising 165
Thermal Shutdown Threshold When LX starts/stops switching TJ falling 155 °C
EN
Enable Low Threshold (VIL) 0.8 V
Enable High Threshold (VIH) 2.0 V
EN Input Current 1µA
Typical Operating Characteristics
(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA= +25°C, unless otherwise noted. See Figure 2.)
Note 3: Specifications to TA= -40°C are guaranteed by design and not production tested.
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
4 _______________________________________________________________________________________
LOAD REGULATION
MAX1951A toc04
OUTPUT CURRENT (A)
OUTPUT VOLTAGE DEVIATION (%)
1.61.20.80.4
-0.40
-0.30
-0.20
-0.10
0
0.10
0.20
0.30
0.40
0.50
-0.50
02.0
VOUT = 2.5V
VOUT = 1.8V
VOUT = 1.5V
LOAD TRANSIENT (50% TRANSIENT)
MAX1951A toc05
VOUT
(AC-COUPLED)
500mV/div
VOUT = 2.5V
IOUT
1A/div
40Fs/div
LOAD TRANSIENT (90% TRANSIENT)
MAX1951A toc06
VOUT
(AC-COUPLED)
200mV/div
IOUT
1A/div
40Fs/div
SOFT-START WAVEFORMS
(VIN = 3.3V, VOUT = 0.8V)
MAX1951A toc09
VOUT
500mV/div
EN
2V/div
1ms/div
STARTUP INTO PREBIASED
OUTPUT
MAX1951A toc10
LX
5V/div
VOUT
2V/div
VOUT = 2.5V
VOUT = 1.5V
EN
5V/div
1ms/div
SWITCHING WAVEFORMS
(VIN = 3.3V, VOUT = 1.8V, RL = 1I)
MAX1951A toc07
LX
2V/div
VOUT
(AC-COUPLED)
10mV/div
ILX
500mA/div
400ns/div
SOFT-START WAVEFORMS
(VIN = 3.3V, VOUT = 1.8V)
MAX1951A toc08
VOUT
1V/div
EN
2V/div
1ms/div
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA= +25°C, unless otherwise noted. See Figure 2.)
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
_______________________________________________________________________________________
5
STARTUP INTO PREBIASED
OUTPUT
MAX1951A toc11
LX
5V/div
VOUT
2V/div
VOUT = 2.5V
VOUT = 3.3V
EN
5V/div
1ms/div
SHUTDOWN WAVEFORMS
(VIN = 3.3V, VOUT = 2.5V, RL = 1.5I)
MAX1951A toc12
LX
2V/div
VOUT
2V/div
EN
2V/div
20Fs/div
SUPPLY CURRENT vs. INPUT VOLTAGE
MAX1951A toc13
INPUT VOLTAGE (V)
SUPPLY CURRENT (mA)
5.04.53.0 3.5 4.0
1
2
3
4
5
6
7
8
0
2.55.5
Typical Operating Characteristics (continued)
(Typical values are at VIN = VCC = 5V, VOUT = 1.5V, IOUT = 1.5A, and TA= +25°C, unless otherwise noted. See Figure 2.)
FEEDBACK VOLTAGE
vs. TEMPERATURE
MAX1951A toc14
TEMPERATURE (NC)
FEEDBACK VOLTAGE (mV)
603510-15
797
799
801
803
805
795
-40 85
CASE TEMPERATURE
vs. AMBIENT TEMPERATURE
MAX1951A toc15
AMBIENT TEMPERATURE (NC)
CASE TEMPERATURE (NC)
603510-15
-20
0
20
40
60
80
100
120
140
160
180
-40
-40 85
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
6 _______________________________________________________________________________________
Detailed Description
The MAX1951A high-efficiency switching regulator is a
small, simple, current-mode DC-DC step-down convert-
er capable of delivering up to 2A of output current. The
device operates in pulse-width modulation (PWM) at a
fixed frequency of 1MHz from a 2.6V to 5.5V input volt-
age and provides an output voltage from 0.8V to VIN,
making the MAX1951A ideal for on-board postregula-
tion applications. The high switching frequency allows
for the use of smaller external components, and an
internal synchronous rectifier improves efficiency and
eliminates the typical Schottky free-wheeling diode.
Using the on-resistance of the internal high-side MOS-
FET to sense switching currents eliminates current-
sense resistors, further improving efficiency and cost.
The MAX1951A total output error over load, line, and
temperature (-40°C to +85°C) is less than 1.5%.
Controller Block Function
The MAX1951A step-down converter uses a PWM cur-
rent-mode control scheme. An open-loop comparator
compares the integrated voltage-feedback signal against
the sum of the amplified current-sense signal and the
slope compensation ramp. At each rising edge of the
internal clock, the internal high-side MOSFET turns on
until the PWM comparator trips. During this on-time, cur-
rent ramps up through the inductor, sourcing current to
the output and storing energy in the inductor. The current-
mode feedback system regulates the peak inductor cur-
rent as a function of the output-voltage error signal. Since
the average inductor current is nearly the same as the
peak inductor current (< 30% ripple current), the circuit
acts as a switch-mode transconductance amplifier. To
preserve inner-loop stability and eliminate inductor stair-
casing, a slope-compensation ramp is summed into the
main PWM comparator. During the second half of the
cycle, the internal high-side p-channel MOSFET turns off,
and the internal low-side n-channel MOSFET turns on.
The inductor releases the stored energy as its current
ramps down while still providing current to the output. The
output capacitor stores charge when the inductor current
exceeds the load current, and discharges when the
inductor current is lower, smoothing the voltage across
the load. Under overload conditions, when the inductor
current exceeds the current limit (see the
Current Limit
section), the high-side MOSFET does not turn on at the
rising edge of the clock and the low-side MOSFET
remains on to let the inductor current ramp down.
Current Sense
An internal current-sense amplifier produces a current
signal proportional to the voltage generated by the
high-side MOSFET on-resistance and the inductor cur-
rent (RDS(ON) x ILX). The amplified current-sense signal
and the internal slope compensation signal are
summed together into the comparator’s inverting input.
The PWM comparator turns off the internal high-side
MOSFET when this sum exceeds the output from the
voltage-error amplifier.
Current Limit
The internal high-side MOSFET has a current limit of 3.1A
(typ). If the current flowing out of LX exceeds this limit, the
high-side MOSFET turns off and the synchronous rectifier
turns on. This lowers the duty cycle and causes the output
voltage to droop until the current limit is no longer exceed-
ed. A synchronous rectifier current limit of -0.6A (typ) pro-
tects the device from current flowing into LX. If the
negative current limit is exceeded, the synchronous recti-
fier turns off, forcing the inductor current to flow through
the high-side MOSFET body diode, back to the input, until
the beginning of the next cycle or until the inductor cur-
rent drops to zero. The MAX1951A utilizes a pulse-skip
mode to prevent overheating during short-circuit output
conditions. The device enters pulse-skip mode when the
FB voltage drops below 300mV, limiting the current to 3A
(typ) and reducing power dissipation. Normal operation
resumes upon removal of the short-circuit condition.
Pin Description
PIN NAME FUNCTION
1V
CC Supply Voltage. Bypass with a 0.1µF capacitor
to ground and a 10Ω resistor to IN.
2EN
Enable Input. Connect to VCC for normal
operation. Connect to GND to disable the
MAX1951A.
3 GND Signal Ground
4FB
Feedback Input. Connect an external resistor-
divider from the output to FB and GND to set
the output to a voltage between 0.8V and VIN.
5 COMP Regulator Compensation. Connect series RC
network to GND.
6 PGND
Power Ground. Internally connected to GND.
Keep power ground and signal ground planes
separate.
7LX
Inductor Connection. Connect an inductor
between LX and the regulator output.
8IN
Power-Supply Voltage. Input voltage range
from 2.6V to 5.5V. Bypass with a 10µF (min)
ceramic capacitor to GND and a 10Ω resistor
to VCC.
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
_______________________________________________________________________________________ 7
VCC Decoupling
Due to the high switching frequency and tight output
tolerance (1.5%), decouple VCC with a 0.1µF capacitor
connected from VCC to GND, and a 10Ωresistor con-
nected from VCC to IN. Place the capacitor as close as
possible to VCC.
Soft-Start
The MAX1951A employs digital soft-start circuitry to
reduce supply inrush current during startup conditions.
When the device exits undervoltage lockout (UVLO)
shutdown mode, or restarts following a thermal-over-
load event, or EN is driven high, the digital soft-start cir-
cuitry slowly ramps up the voltage to the error-amplifier
noninverting input.
Undervoltage Lockout
If VCC drops below 2.07V, the UVLO circuit inhibits
switching. Once VCC rises above 2.19V, the UVLO
clear and the soft-start sequence activates.
Shutdown Mode
Use the enable input, EN, to turn on or off the MAX1951A.
Connect EN to VCC for normal operation. Connect EN to
GND to place the device in shutdown. Shutdown causes
the internal switches to stop switching and forces LX into
a high-impedance state. In shutdown, the MAX1951A
draws 500µA of supply current. The device initiates a
soft-start sequence when brought out of shutdown.
Thermal-Overload Protection
Thermal-overload protection limits total power dissipation
in the device. When the junction temperature exceeds
TJ= +165°C, a thermal sensor forces the device into
shutdown, allowing the die to cool. The thermal sensor
turns the device on again after the junction temperature
cools by 9°C, resulting in a pulsed output during continu-
ous overload conditions. Following a thermal-shutdown
condition, the soft-start sequence begins.
Safe Startup into Prebiased Output
The MAX1951A can start up safely even with a prebi-
ased output. A zero crossover detection (ZCD) circuit
turns on the switches only after the soft-start ramping
voltage equals the prebiased output voltage. If the pre-
biased output voltage is greater than the set voltage,
the ZCD circuit turns on the low-side switch (after the
soft-start period is over) to discharge the output capac-
itor until its voltage equals the set voltage.
MAX1951A
VCC
COMP
EN
CURRENT SENSE
SLOPE
COMP
ERROR
SIGNAL
CLOCK
POSITIVE AND NEGATIVE CURRENT LIMITS
GND
FB
PGND
LX
IN
DAC
gm
SOFT-START/
UVLO
PWM
CONTROL
PREBIAS
THERMAL
SHUTDOWN
BANDGAP
REF
1.25V
ZERO-
CROSSING
DETECTOR
CLAMP
OSC
RAMP GEN
Figure 1. Functional Diagram
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
8 _______________________________________________________________________________________
Design Procedure
Adjustable Output Voltage
The MAX1951A provides an adjustable output voltage
between 0.8V and VIN. Connect FB to output for 0.8V
output. To set the output voltage of the MAX1951A to a
voltage greater than VFB (0.8V typ), connect the output
to FB and GND using a resistive divider, as shown in
Figure 2. Choose R2 between 2kΩand 20kΩ, and set
R3 according to the following equation:
R3 = R2 x [(VOUT/VFB) - 1]
The MAX1951A PWM circuitry is capable of a stable
minimum duty cycle of 18%. This limits the minimum
output voltage that can be generated to 0.18 VIN with
an absolute minimum of 0.8V. Instability may result for
VIN/VOUT ratios below 0.18.
Output Inductor Design
Use a 2µH inductor with a minimum 2A-rated DC cur-
rent for most applications. For best efficiency, use an
inductor with a DC resistance of less than 20mΩand a
saturation current greater than 3A (min). See Table 2
for recommended inductors and manufacturers. For
most designs, derive a reasonable inductor value
(LINIT) from the following equation:
LINIT = VOUT x (VIN - VOUT)/(VIN x LIR x IOUT(MAX) x fSW)
where fSW is the switching frequency (1MHz typ) of the
oscillator. Keep the inductor current ripple percentage
LIR between 20% and 40% of the maximum load cur-
rent for the best compromise of cost, size, and perfor-
mance. Calculate the maximum inductor current as:
IL(MAX) = (1 + LIR/2) x IOUT(MAX)
Check the final values of the inductor with the output
ripple voltage requirement. The output ripple voltage is
given by:
VRIPPLE = VOUT x (VIN - VOUT) x ESR/(VIN x LFINAL x fSW)
where ESR is the equivalent series resistance of the
output capacitors.
Input Capacitor Design
The input filter capacitor reduces peak currents drawn
from the power source and reduces noise and voltage
ripple on the input caused by the circuit’s switching.
The input capacitor must meet the ripple current
requirement (IRMS) imposed by the switching currents
defined by the following equation:
For duty ratios less than 0.5, the input capacitor RMS
current is higher than the calculated current. Therefore,
use a +20% margin when calculating the RMS current
at lower duty cycles. Use ceramic capacitors for their
low ESR and equivalent series inductance (ESL).
Choose a capacitor that exhibits less than 10°C tem-
perature rise at the maximum operating RMS current for
optimum long-term reliability.
After determining the input capacitor, check the input
ripple voltage due to capacitor discharge when the
high-side MOSFET turns on. Calculate the input ripple
voltage as follows:
VIN_RIPPLE = (IOUT x VOUT)/(fSW x VIN x CIN)
Keep the input ripple voltage less than 3% of the input
voltage.
Output Capacitor Design
The key selection parameters for the output capacitor
are capacitance, ESR, ESL, and the voltage rating
requirements. These affect the overall stability, output
ripple voltage, and transient response of the DC-DC
converter. The output ripple occurs due to variations in
the charge stored in the output capacitor, the voltage
drop due to the capacitor’s ESR, and the voltage drop
due to the capacitor’s ESL. Calculate the output voltage
ripple due to the output capacitance, ESR, and ESL as:
VRIPPLE = VRIPPLE(C) + VRIPPLE(ESR) + VRIPPLE(ESL)
where the output ripple due to output capacitance,
ESR, and ESL is:
VRIPPLE(C) = IP-P/(8 x COUT x fSW)
VRIPPLE(ESR) = IP-P x ESR
VRIPPLE(ESL) = (IP-P/tON) x ESL or (IP-P/tOFF) x ESL,
whichever is greater
and IP-P the peak-to-peak inductor current is:
IP-P = [(VIN – VOUT )/fSW x L)] x VOUT/VIN
Use these equations for initial capacitor selection, but
determine final values by testing a prototype or evalua-
tion circuit. As a rule, a smaller ripple current results in
less output-voltage ripple. Since the inductor ripple
current is a factor of the inductor value, the output-
voltage ripple decreases with larger inductance. Use
ceramic capacitors for their low ESR and ESL at the
switching frequency of the converter. The low ESL of
ceramic capacitors makes ripple voltages negligible.
Load-transient response depends on the selected
output capacitor. During a load transient, the output
instantly changes by ESR x ∆ILOAD. Before the con-
troller can respond, the output deviates further,
depending on the inductor and output capacitor val-
ues. After a short time (see the Load Transient graph in
the
Typical Operating Characteristic
s), the controller
responds by regulating the output voltage back to its
IVIVVV
RMS IN OUT OUT IN OUT
=× ××−()( ( ))12
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
_______________________________________________________________________________________ 9
nominal state. The controller response time depends on
the closed-loop bandwidth. A higher bandwidth yields
a faster response time, thus preventing the output from
deviating further from its regulating value.
Compensation Design
The double pole formed by the inductor and output
capacitor of most voltage-mode controllers introduces a
large phase shift that requires an elaborate compensation
network to stabilize the control loop. The MAX1951A uti-
lizes a current-mode control scheme that regulates the
output voltage by forcing the required current through the
external inductor, eliminating the double pole caused by
the inductor and output capacitor, and greatly simplifying
the compensation network. A simple type 1 compensa-
tion with single compensation resistor (R1) and compen-
sation capacitor (C2) in Figure 2 creates a stable and
high-bandwidth loop.
An internal transconductance error amplifier compen-
sates the control loop. Connect a series resistor and
capacitor between COMP (the output of the error ampli-
fier) and GND to form a pole-zero pair. The external
inductor, internal current-sensing circuitry, output
capacitor, and the external compensation circuit deter-
mine the loop system stability. Choose the inductor and
output capacitor based on performance, size, and cost.
Additionally, select the compensation resistor and
capacitor to optimize control-loop stability. The compo-
nent values shown in the typical application circuit
(Figure 2) yield stable operation over a broad range of
input-to-output voltages.
The basic regulator loop consists of a power modulator,
an output feedback divider, and an error amplifier. The
power modulator has DC gain set by gmc x RLOAD, with
a pole-zero pair set by RLOAD, the output capacitor
(COUT), and its ESR. The following equations define the
power modulator:
Modulator gain:
GMOD = ∆VOUT/∆VCOMP = gmc x RLOAD
Modulator pole frequency:
fpMOD = 1/(2 x πx COUT x (RLOAD + ESR))
Modulator zero frequency:
fzESR = 1/(2 x πx COUT x ESR)
where RLOAD = VOUT/IOUT(MAX) and gmc = 4.2S.
The feedback divider has a gain of GFB = VFB/VOUT,
where VFB is equal to 0.8V. The transconductance error
amplifier has a DC gain, GEA(DC), of 70dB. The com-
pensation capacitor, C2, and the output resistance of
the error amplifier, ROEA (20MΩ), set the dominant
pole. C2and R1 set a compensation zero. Calculate the
dominant pole frequency as:
fpEA = 1/(2π x C2 x ROEA)
Determine the compensation zero frequency as:
fzEA = 1/(2πx C2 x R1)
For best stability and response performance, set the
closed-loop unity-gain frequency much higher than the
modulator pole frequency. In addition, set the closed-
loop crossover unity-gain frequency less than, or equal
to 1/5 of the switching frequency. However, set the
maximum zero crossing frequency to less than 1/3 of
the zero frequency set by the output capacitance and
its ESR when using POSCAP, SPCAP, OSCON, or other
electrolytic capacitors. The loop-gain equation at the
unity-gain frequency is:
GEA(fc) x GMOD(fc) x VFB/VOUT = 1
where GEA(fc
)
= gmEA x R1, and GMOD(fc) = gmc x
RLOAD x fpMOD/fC, where gmEA = 60µS.
R1calculated as:
R1= VOUT x K/(gmEA x VFB x GMOD(fc))
where K is the correction factor due to the extra phase
introduced by the current loop at high frequencies
(>100kHz). K is related to the value of the output
capacitance (see Table 1 for values of K vs. C). Set the
error-amplifier compensation zero formed by R1and C2
at the modulator pole frequency at maximum load. C2
is calculated as follows:
C2= (2 x VOUT x COUT/(R1 x IOUT(MAX))
As the load current decreases, the modulator pole also
decreases; however, the modulator gain increases
accordingly, resulting in a constant closed-loop unity-
gain frequency. Use the following numerical example to
calculate R1and C2values of the typical application
circuit of Figure 2.
VOUT = 1.5V
IOUT(MAX) = 2A
COUT = 10µF
RESR = 0.010Ω
gmEA = 60µS
gmc = 4.2S
fSWITCH = 1MHz
DESCRIPTION
COUT (µF) K
10 0.55
22 0.47
V al ues ar e for outp ut i nd uctance fr om 1.2µH
to 2.2µH . D o not use outp ut i nd uctor s l ar g er
than 2.2µH . U se fC
= 200kH z to cal cul ate R1
.
Table 1. K Value
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
10 ______________________________________________________________________________________
RLOAD = VOUT/IOUT(MAX) = 1.5V/2A = 0.75Ω
fpMOD = [1/(2πx COUT x (RLOAD + RESR)]
= [1/(2 x πx10 x10-6 x (0.75 + 0.01)] = 20.9Hz.
fzESR = [1/(2πx COUT x RESR)]
= [1/(2 x πx 10 x10-6 x 0.01)] = 1.59MHz.
For a 2µH output inductor, pick the closed-loop unity-
gain crossover frequency (fC) at 200kHz. Determine the
power modulator gain at fC:
GMOD(fc
)
= gmc x RLOAD x fpMOD/fC= 4.2 x 0.75 x
20.9kHz/200kHz = 0.33
then:
R1= VOx K/(gmEA x VFB x GMOD(fC)) = (1.5 x
0.55)/(60 x 10-6 x 0.8 x 0.33) ≈52.3kΩ(1%)
C2= (2 x VOUT x COUT)/R1x IOUT(MAX)
= (2 x 1.5 x 10 x 10-6)/(52.3kΩx 2)
≈143pF, choose 150pF, 10%
Applications Information
PCB Layout Considerations
Careful PCB layout is critical to achieve clean and sta-
ble operation. The switching power stage requires par-
ticular attention. Follow these guidelines for good PCB
layout:
1) Place decoupling capacitors as close as possible to
the IC. Keep the power ground plane (connected to
PGND) and signal ground plane (connected to
GND) separate.
2) Connect input and output capacitors to the power
ground plane; connect all other capacitors to the
signal ground plane.
3) Keep the high-current paths as short and wide as
possible. Keep the path of switching current (C1 to IN
and C1 to PGND) short. Avoid vias in the switching
paths.
4) If possible, connect IN, LX, and PGND separately to
a large copper area to help cool the IC to further
improve efficiency and long-term reliability.
5) Ensure all feedback connections are short and
direct. Place the feedback resistors as close as pos-
sible to the IC.
6) Route high-speed switching nodes away from sensi-
tive analog areas (FB, COMP).
Thermal Considerations
See the MAX1951A Evaluation Kit for an optimized lay-
out example. Thermal performance can be further
improved with one of the following options:
1) Increase the copper areas connected to GND, LX,
and IN.
2) Provide thermal vias next to GND and IN, to the
ground plane and power plane on the back side of
PCB with openings in the solder mask next to the
vias to provide better thermal conduction.
3) Provide forced-air cooling to further reduce case
temperature.
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
______________________________________________________________________________________ 11
MAX1951A
IN
ON
R1
51.1kΩ
R4
10Ω
C4
0.1µF
C2
220pF
R2
15.0kΩ
1%
R3
13.0kΩ
1%
L1
2µH
C3
10µF
C1
10µF
OFF
LX
COMP
1.5V AT 2A
2.6V TO 5.5V
FB
GND PGND
VCC
COMPONENT VALUES
OUTPUT
VOLTAGE (V)
0.8
1.5
2.5
3.3
R1 (kΩ)
10
52.3
86.6
115
R2 (kΩ)
OPEN
15
15
15
R3 (kΩ)
SHORT
13
31.6
46.4
C2 (pF)
470
150
150
150
EN
Figure 2. MAX1951A Adjustable Output Typical Application Circuit
MAX1951A
1MHz, 2A, 2.6V to 5.5V Input, PWM DC-DC
Step-Down Regulator with Enable
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
12
____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2009 Maxim Integrated Products Maxim is a registered trademark of Maxim Integrated Products, Inc.
Table 2. External Components List
C O M PO N EN T ( F I G U R E 2 ) FUNCTION DESCRIPTION
L1 Output inductor
2µH ±20% inductor
Sumida CDRH4D28-1R8 or
TOKO A915AY-2R0M
C1 Input filtering capacitor
10µF ±20%, 6.3V X5R capacitor
Taiyo Yuden JMK316BJ106ML or
TDK C3216X5R0J106MT
C2 Compensation capacitor
220pF ±10%, 50V capacitor
Murata GRM1885C1HZZ1JA01 or
Taiyo Yuden UMK107CH221KZ
C3 Output filtering capacitor
10µF ±20%, 6.3V X5R capacitor
Taiyo Yuden JMK316BJ106ML or
TDK C3216X5R0J106MT
C4 VCC bypass capacitor
0.1µF ±20%, 16V X7R capacitor
Taiyo Yuden EMK107BJ104MA,
TDK C1608X7R1C104K, or
Murata GRM188R171C104KA01
R1 Loop compensation resistor Figure 2
R2 Feedback resistor Figure 2
R3 Feedback resistor Figure 2
R4 Bypass resistor 10Ω ±5% resistor
Table 3. Component Suppliers
MANUFACTURER WEBSITE
Murata Electronics North America, Inc. www.murata-northamerica.com
Sumida Corp. www.sumida.com
Taiyo Yuden www.t-yuden.com
TDK Corp. www.component.tdk.com
TOKO America, Inc. www.tokoam.com
Chip Information
PROCESS: BiCMOS
Package Information
For the latest package outline information and land patterns, go
to www.maxim-ic.com/packages.
PACKAGE TYPE PACKAGE CODE DOCUMENT NO.
8 SO S8-6F 21-0041
