LM2731
LM2731 0.6/1.6 MHz Boost Converters With 22V Internal FET Switch in SOT-23
Literature Number: SNVS217E
LM2731
April 29, 2010
0.6/1.6 MHz Boost Converters With 22V Internal FET Switch
in SOT-23
General Description
The LM2731 switching regulators are current-mode boost
converters operating at fixed frequencies of 1.6 MHz (“X” op-
tion) and 600 kHz (“Y” option).
The use of SOT-23 package, made possible by the minimal
power loss of the internal 1.8A switch, and use of small in-
ductors and capacitors result in the industry's highest power
density. The 22V internal switch makes these solutions per-
fect for boosting to voltages up to 20V.
These parts have a logic-level shutdown pin that can be used
to reduce quiescent current and extend battery life.
Protection is provided through cycle-by-cycle current limiting
and thermal shutdown. Internal compensation simplifies de-
sign and reduces component count.
Switch Frequency
X Y
1.6 MHz 0.6 MHz
Features
■22V DMOS FET switch
■1.6 MHz (“X”), 0.6 MHz (“Y”) switching frequency
■Low RDS(ON) DMOS FET
■Switch current up to 1.8A
■Wide input voltage range (2.7V–14V)
■Low shutdown current (<1 µA)
■5-Lead SOT-23 package
■Uses tiny capacitors and inductors
■Cycle-by-cycle current limiting
■Internally compensated
Applications
■White LED Current Source
■PDA’s and Palm-Top Computers
■Digital Cameras
■Portable Phones and Games
■Local Boost Regulator
Typical Application Circuit
20059110
20059130
© 2010 National Semiconductor Corporation 200591 www.national.com
LM2731 0.6/1.6 MHz Boost Converters With 22V Internal FET Switch in SOT-23
20059153
20059156
20059155
White LED Flash Application
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LM2731
Connection Diagram
Top View
20059111
5-Lead SOT-23 Package
See NS Package Number MF05A
Ordering Information
Order Number Package Type Package Drawing Supplied As Package ID
LM2731XMF
SOT23-5 MF05A
1K Tape and Reel S51A
LM2731XMFX 3K Tape and Reel S51A
LM2731YMF 1K Tape and Reel S51B
LM2731YMFX 3K Tape and Reel S51B
Pin Descriptions
Pin Name Function
1 SW Drain of the internal FET switch.
2 GND Analog and power ground.
3 FB Feedback point that connects to external resistive divider.
4 SHDN Shutdown control input. Connect to Vin if the feature is not used.
5 VIN Analog and power input.
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LM2731
Absolute Maximum Ratings (Note 1)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Storage Temperature Range −65°C to +150°C
Operating Junction
Temperature Range −40°C to +125°C
Lead Temp. (Soldering, 5 sec.) 300°C
Power Dissipation (Note 2) Internally Limited
FB Pin Voltage −0.4V to +6V
SW Pin Voltage −0.4V to +22V
Input Supply Voltage −0.4V to +14.5V
SHDN Pin Voltage −0.4V to VIN + 0.3V
θJ-A (SOT23-5) 265°C/W
ESD Rating (Note 3)
Human Body Model 2 kV
Electrical Characteristics
Limits in standard typeface are for TJ = 25°C, and limits in boldface type apply over the full operating temperature range
(−40°C ≤ TJ ≤ +125°C). Unless otherwise specified: VIN = 5V, VSHDN = 5V, IL = 0A.
Symbol Parameter Conditions Min
(Note 4)
Typical
(Note 5)
Max
(Note 4)Units
VIN Input Voltage 2.7 14 V
VOUT (MIN) Minimum Output Voltage
Under Load
RL = 43Ω
X Option
(Note 8)
VIN = 2.7V 5.4 7 V
VIN = 3.3V 810
VIN = 5V 13 17
RL = 43Ω
Y Option
(Note 8)
VIN = 2.7V 8.25 10
VIN = 3.3V 10.5 12
VIN = 5V 14 16
RL = 15Ω
X Option
(Note 8)
VIN = 2.7V 3.75 5
VIN = 3.3V 56.5
VIN = 5V 8.75 11
RL = 15Ω
Y Option
(Note 8)
VIN = 2.7V 56
VIN = 3.3V 5.5 7.5
VIN = 5V 911
ISW Switch Current Limit (Note 6) 1.8
1.4
2 A
RDS(ON) Switch ON Resistance ISW = 100 mA
Vin = 5V
260 400
500
mΩ
ISW = 100 mA
Vin = 3.3V
300 450
550
SHDNTH Shutdown Threshold Device ON 1.5 V
Device OFF 0.50
ISHDN Shutdown Pin Bias
Current
VSHDN = 0 0 µA
VSHDN = 5V 0 2
VFB Feedback Pin Reference
Voltage
VIN = 3V 1.205 1.230 1.255 V
IFB Feedback Pin Bias
Current
VFB = 1.23V 60 500 nA
IQQuiescent Current VSHDN = 5V, Switching "X" 2 3.0 mA
VSHDN = 5V, Switching "Y" 1.0 2
VSHDN = 5V, Not Switching 400 500 µA
VSHDN = 0 0.024 1
FB Voltage Line
Regulation
2.7V ≤ VIN ≤ 14V 0.02 %/V
FSW Switching Frequency
(Note 7)
“X” Option 11.6 1.85 MHz
“Y” Option 0.40 0.60 0.8
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LM2731
Symbol Parameter Conditions Min
(Note 4)
Typical
(Note 5)
Max
(Note 4)Units
DMAX Maximum Duty Cycle
(Note 7)
“X” Option 86 93 %
“Y” Option 92 96
ILSwitch Leakage Not Switching VSW = 5V 1 µA
Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the
device outside of the limits set forth under the operating ratings which specify the intended range of operating conditions.
Note 2: The maximum power dissipation which can be safely dissipated for any application is a function of the maximum junction temperature, TJ(MAX) = 125°
C, the junction-to-ambient thermal resistance for the SOT-23 package, θJ-A = 265°C/W, and the ambient temperature, TA. The maximum allowable power
dissipation at any ambient temperature for designs using this device can be calculated using the formula:
If power dissipation exceeds the maximum specified above, the internal thermal protection circuitry will protect the device by reducing the output voltage as
required to maintain a safe junction temperature.
Note 3: The human body model is a 100 pF capacitor discharged through a 1.5 kΩ resistor into each pin.
Note 4: Limits are guaranteed by testing, statistical correlation, or design.
Note 5: Typical values are derived from the mean value of a large quantity of samples tested during characterization and represent the most likely expected value
of the parameter at room temperature.
Note 6: Switch current limit is dependent on duty cycle (see Typical Performance Characteristics).
Note 7: Guaranteed limits are the same for Vin = 3.3V input.
Note 8: L = 10 µH, COUT = 4.7 µF, duty cycle = maximum
Typical Performance Characteristics Unless otherwise specified: VIN = 5V, SHDN pin tied to VIN.
Iq Vin (Active) vs Temperature - "X"
20059102
Iq Vin (Active) vs Temperature - "Y"
20059104
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LM2731
Oscillator Frequency vs Temperature - "X"
20059105
Oscillator Frequency vs Temperature - "Y"
20059101
Max. Duty Cycle vs Temperature - "X"
20059107
Max. Duty Cycle vs Temperature - "Y"
20059106
Iq Vin (Idle) vs Temperature
20059125
Feedback Bias Current vs Temperature
20059126
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LM2731
Feedback Voltage vs Temperature
20059127
RDS(ON) vs Temperature
20059128
Current Limit vs Temperature
20059129
RDS(ON) vs VIN
20059152
Efficiency vs Load Current - "X"
VIN = 2.7V, VOUT = 5V
20059135
Efficiency vs Load Current - "X"
VIN = 3.3V, VOUT = 5V
20059136
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LM2731
Efficiency vs Load Current - "X"
VIN = 4.2V, VOUT = 5V
20059137
Efficiency vs Load Current - "X"
VIN = 2.7V, VOUT = 12V
20059138
Efficiency vs Load Current - "X"
VIN = 3.3V, VOUT = 12V
20059139
Efficiency vs Load Current - "X"
VIN = 5V, VOUT = 12V
20059140
Efficiency vs Load Current - "X"
VIN = 5V, VOUT = 18V
20059141
Efficiency vs Load Current - "Y"
VIN = 2.7V, VOUT = 5V
20059142
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LM2731
Efficiency vs Load Current - "Y"
VIN = 3.3V, VOUT = 5V
20059143
Efficiency vs Load Current - "Y"
VIN = 4.2V, VOUT = 5V
20059144
Efficiency vs Load Current - "Y"
VIN = 2.7V, VOUT = 12V
20059145
Efficiency vs Load Current - "Y"
VIN = 3.3V, VOUT = 12V
20059146
Efficiency vs Load Current - "Y"
VIN = 5V, VOUT = 12V
20059147
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LM2731
Block Diagram
20059112
Theory of Operation
The LM2731 is a switching converter IC that operates at a
fixed frequency (0.6 or 1.6 MHz) for fast transient response
over a wide input voltage range and incorporates pulse-by-
pulse current limiting protection. Because this is current mode
control, a 33 mΩ sense resistor in series with the switch FET
is used to provide a voltage (which is proportional to the FET
current) to both the input of the pulse width modulation (PWM)
comparator and the current limit amplifier.
At the beginning of each cycle, the S-R latch turns on the FET.
As the current through the FET increases, a voltage (propor-
tional to this current) is summed with the ramp coming from
the ramp generator and then fed into the input of the PWM
comparator. When this voltage exceeds the voltage on the
other input (coming from the Gm amplifier), the latch resets
and turns the FET off. Since the signal coming from the Gm
amplifier is derived from the feedback (which samples the
voltage at the output), the action of the PWM comparator
constantly sets the correct peak current through the FET to
keep the output voltage in regulation.
Q1 and Q2 along with R3 - R6 form a bandgap voltage refer-
ence used by the IC to hold the output in regulation. The
currents flowing through Q1 and Q2 will be equal, and the
feedback loop will adjust the regulated output to maintain this.
Because of this, the regulated output is always maintained at
a voltage level equal to the voltage at the FB node "multiplied
up" by the ratio of the output resistive divider.
The current limit comparator feeds directly into the flip-flop
that drives the switch FET. If the FET current reaches the limit
threshold, the FET is turned off and the cycle terminated until
the next clock pulse. The current limit input terminates the
pulse regardless of the status of the output of the PWM com-
parator.
Application Hints
SELECTING THE EXTERNAL CAPACITORS
The best capacitors for use with the LM2731 are multi-layer
ceramic capacitors. They have the lowest ESR (equivalent
series resistance) and highest resonance frequency which
makes them optimum for use with high frequency switching
converters.
When selecting a ceramic capacitor, only X5R and X7R di-
electric types should be used. Other types such as Z5U and
Y5F have such severe loss of capacitance due to effects of
temperature variation and applied voltage, they may provide
as little as 20% of rated capacitance in many typical applica-
tions. Always consult capacitor manufacturer’s data curves
before selecting a capacitor. High-quality ceramic capacitors
can be obtained from Taiyo-Yuden, AVX, and Murata.
SELECTING THE OUTPUT CAPACITOR
A single ceramic capacitor of value 4.7 µF to 10 µF will provide
sufficient output capacitance for most applications. If larger
amounts of capacitance are desired for improved line support
and transient response, tantalum capacitors can be used.
Aluminum electrolytics with ultra low ESR such as Sanyo Os-
con can be used, but are usually prohibitively expensive.
Typical AI electrolytic capacitors are not suitable for switching
frequencies above 500 kHz due to significant ringing and
temperature rise due to self-heating from ripple current. An
output capacitor with excessive ESR can also reduce phase
margin and cause instability.
In general, if electrolytics are used, it is recommended that
they be paralleled with ceramic capacitors to reduce ringing,
switching losses, and output voltage ripple.
SELECTING THE INPUT CAPACITOR
An input capacitor is required to serve as an energy reservoir
for the current which must flow into the coil each time the
switch turns ON. This capacitor must have extremely low
ESR, so ceramic is the best choice. We recommend a nomi-
nal value of 2.2 µF, but larger values can be used. Since this
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LM2731
capacitor reduces the amount of voltage ripple seen at the
input pin, it also reduces the amount of EMI passed back
along that line to other circuitry.
FEED-FORWARD COMPENSATION
Although internally compensated, the feed-forward capacitor
Cf is required for stability (see Basic Application Circuit).
Adding this capacitor puts a zero in the loop response of the
converter. The recommended frequency for the zero fz should
be approximately 6 kHz. Cf can be calculated using the for-
mula:
Cf = 1 / (2 X π X R1 X fz)
SELECTING DIODES
The external diode used in the typical application should be
a Schottky diode. A 20V diode such as the MBR0520 is rec-
ommended.
The MBR05XX series of diodes are designed to handle a
maximum average current of 0.5A. For applications exceed-
ing 0.5A average but less than 1A, a Microsemi UPS5817 can
be used.
LAYOUT HINTS
High frequency switching regulators require very careful lay-
out of components in order to get stable operation and low
noise. All components must be as close as possible to the
LM2731 device. It is recommended that a 4-layer PCB be
used so that internal ground planes are available.
As an example, a recommended layout of components is
shown:
20059116
Recommended PCB Component Layout
Some additional guidelines to be observed:
1. Keep the path between L1, D1, and C2 extremely short.
Parasitic trace inductance in series with D1 and C2 will
increase noise and ringing.
2. The feedback components R1, R2 and CF must be kept
close to the FB pin of U1 to prevent noise injection on the
FB pin trace.
3. If internal ground planes are available (recommended)
use vias to connect directly to ground at pin 2 of U1, as
well as the negative sides of capacitors C1 and C2.
SETTING THE OUTPUT VOLTAGE
The output voltage is set using the external resistors R1 and
R2 (see Basic Application Circuit). A value of approximately
13.3 kΩ is recommended for R2 to establish a divider current
of approximately 92 µA. R1 is calculated using the formula:
R1 = R2 X (VOUT/1.23 − 1)
SWITCHING FREQUENCY
The LM2731 is provided with two switching frequencies: the
“X” version is typically 1.6 MHz, while the “Y” version is typi-
cally 600 kHz. The best frequency for a specific application
must be determined based on the trade-offs involved:
Higher switching frequency means the inductors and capac-
itors can be made smaller and cheaper for a given output
voltage and current. The down side is that efficiency is slightly
lower because the fixed switching losses occur more fre-
quently and become a larger percentage of total power loss.
EMI is typically worse at higher switching frequencies be-
cause more EMI energy will be seen in the higher frequency
spectrum where most circuits are more sensitive to such in-
terference.
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LM2731
20059117
Basic Application Circuit
DUTY CYCLE
The maximum duty cycle of the switching regulator deter-
mines the maximum boost ratio of output-to-input voltage that
the converter can attain in continuous mode of operation. The
duty cycle for a given boost application is defined as:
This applies for continuous mode operation.
INDUCTANCE VALUE
The first question we are usually asked is: “How small can I
make the inductor?” (because they are the largest sized com-
ponent and usually the most costly). The answer is not simple
and involves trade-offs in performance. Larger inductors
mean less inductor ripple current, which typically means less
output voltage ripple (for a given size of output capacitor).
Larger inductors also mean more load power can be delivered
because the energy stored during each switching cycle is:
E = L/2 X (lp)2
Where “lp” is the peak inductor current. An important point to
observe is that the LM2731 will limit its switch current based
on peak current. This means that since lp(max) is fixed, in-
creasing L will increase the maximum amount of power avail-
able to the load. Conversely, using too little inductance may
limit the amount of load current which can be drawn from the
output.
Best performance is usually obtained when the converter is
operated in “continuous” mode at the load current range of
interest, typically giving better load regulation and less output
ripple. Continuous operation is defined as not allowing the in-
ductor current to drop to zero during the cycle. It should be
noted that all boost converters shift over to discontinuous op-
eration as the output load is reduced far enough, but a larger
inductor stays “continuous” over a wider load current range.
To better understand these trade-offs, a typical application
circuit (5V to 12V boost with a 10 µH inductor) will be ana-
lyzed. We will assume:
VIN = 5V, VOUT = 12V, VDIODE = 0.5V, VSW = 0.5V
Since the frequency is 1.6 MHz (nominal), the period is ap-
proximately 0.625 µs. The duty cycle will be 62.5%, which
means the ON time of the switch is 0.390 µs. It should be
noted that when the switch is ON, the voltage across the in-
ductor is approximately 4.5V.
Using the equation:
V = L (di/dt)
We can then calculate the di/dt rate of the inductor which is
found to be 0.45 A/µs during the ON time. Using these facts,
we can then show what the inductor current will look like dur-
ing operation:
20059118
10 µH Inductor Current,
5V–12V Boost (LM2731X)
During the 0.390 µs ON time, the inductor current ramps up
0.176A and ramps down an equal amount during the OFF
time. This is defined as the inductor “ripple current”. It can also
be seen that if the load current drops to about 33 mA, the
inductor current will begin touching the zero axis which means
it will be in discontinuous mode. A similar analysis can be
performed on any boost converter, to make sure the ripple
current is reasonable and continuous operation will be main-
tained at the typical load current values.
MAXIMUM SWITCH CURRENT
The maximum FET switch current available before the current
limiter cuts in is dependent on duty cycle of the application.
This is illustrated in the graphs below which show typical val-
ues of switch current for both the "X" and "Y" versions as a
function of effective (actual) duty cycle:
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LM2731
20059150
Switch Current Limit vs Duty Cycle - "X"
20059151
Switch Current Limit vs Duty Cycle - "Y"
CALCULATING LOAD CURRENT
As shown in the figure which depicts inductor current, the load
current is related to the average inductor current by the rela-
tion:
ILOAD = IIND(AVG) x (1 - DC)
Where "DC" is the duty cycle of the application. The switch
current can be found by:
ISW = IIND(AVG) + ½ (IRIPPLE)
Inductor ripple current is dependent on inductance, duty cy-
cle, input voltage and frequency:
IRIPPLE = DC x (VIN-VSW) / (f x L)
combining all terms, we can develop an expression which al-
lows the maximum available load current to be calculated:
The equation shown to calculate maximum load current takes
into account the losses in the inductor or turn-OFF switching
losses of the FET and diode. For actual load current in typical
applications, we took bench data for various input and output
voltages for both the "X" and "Y" versions of the LM2731 and
displayed the maximum load current available for a typical
device in graph form:
20059148
Max. Load Current (typ) vs VIN - "X"
20059149
Max. Load Current (typ) vs VIN - "Y"
DESIGN PARAMETERS VSW AND ISW
The value of the FET "ON" voltage (referred to as VSW in the
equations) is dependent on load current. A good approxima-
tion can be obtained by multiplying the "ON Resistance" of
the FET times the average inductor current.
FET on resistance increases at VIN values below 5V, since
the internal N-FET has less gate voltage in this input voltage
range (see Typical performance Characteristics curves).
Above VIN = 5V, the FET gate voltage is internally clamped to
5V.
The maximum peak switch current the device can deliver is
dependent on duty cycle. For higher duty cycles, see Typical
performance Characteristics curves.
THERMAL CONSIDERATIONS
At higher duty cycles, the increased ON time of the FET
means the maximum output current will be determined by
power dissipation within the LM2731 FET switch. The switch
power dissipation from ON-state conduction is calculated by:
P(SW) = DC x IIND(AVE)2 x RDS(ON)
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LM2731
There will be some switching losses as well, so some derating
needs to be applied when calculating IC power dissipation.
INDUCTOR SUPPLIERS
Recommended suppliers of inductors for this product include,
but are not limited to Sumida, Coilcraft, Panasonic, TDK and
Murata. When selecting an inductor, make certain that the
continuous current rating is high enough to avoid saturation
at peak currents. A suitable core type must be used to mini-
mize core (switching) losses, and wire power losses must be
considered when selecting the current rating.
SHUTDOWN PIN OPERATION
The device is turned off by pulling the shutdown pin low. If this
function is not going to be used, the pin should be tied directly
to VIN. If the SHDN function will be needed, a pull-up resistor
must be used to VIN (approximately 50k-100kΩ recommend-
ed). The SHDN pin must not be left unterminated.
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LM2731
Physical Dimensions inches (millimeters) unless otherwise noted
5-Lead SOT-23 Package
Order Number LM2731XMF, LM2731XMFX, LM2731YMF or LM2731YMFX
NS Package Number MF05A
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LM2731
Notes
LM2731 0.6/1.6 MHz Boost Converters With 22V Internal FET Switch in SOT-23
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