General Description
The MAX8545/MAX8546/MAX8548 are voltage-mode
pulse-width-modulated (PWM), step-down DC-DC con-
trollers ideal for a variety of cost-sensitive applications.
They drive low-cost n-channel MOSFETs for both the
high-side switch and synchronous rectifier, and require
no external current-sense resistor. These devices can
supply output voltages as low as 0.8V.
The MAX8545/MAX8546/MAX8548 have a wide 2.7V to
28V input range, and do not need any additional bias
voltage. The output voltage can be precisely regulated
from 0.8V to 0.83 x VIN. These devices can provide effi-
ciency up to 95%. Lossless short-circuit and current-limit
protection is provided by monitoring the RDS(ON) of the
low-side MOSFET. The MAX8545 and MAX8548 have a
current-limit threshold of 320mV, while the MAX8546 has
a current-limit threshold of 165mV. All devices feature
foldback-current capability to minimize power dissipation
under short-circuit condition. Pulling the COMP/EN pin
low with an open-collector or low-capacitance, open-
drain device can shut down all devices.
The MAX8545/MAX8546 operate at 300kHz and the
MAX8548 operates at 100kHz. The MAX8545/MAX8546/
MAX8548 are compatible with low-cost aluminum elec-
trolytic capacitors. Input undervoltage lockout prevents
proper operation under power-sag operations to prevent
external MOSFETs from overheating. Internal soft-start is
included to reduce inrush current. These devices are
offered in space-saving 10-pin µMAX®packages.
Applications
Features
♦2.7V to 28V Input Range
♦Foldback Short-Circuit Protection
♦No Additional Bias Supply Needed
♦0.8V to 0.83 x VIN Output
♦Up to 95% Efficiency
♦Low-Cost External Components
♦No Current-Sense Resistor
♦All n-Channel MOSFET Design
♦Adaptive Gate Drivers Eliminate Shoot-Through
♦Lossless Overcurrent and Short-Circuit
Protection
♦300kHz Switching Frequency
(MAX8545/MAX8546)
♦100kHz Switching Frequency (MAX8548)
♦Pin-Compatible with the MAX1967
♦Thermal Shutdown
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
________________________________________________________________ Maxim Integrated Products 1
MAX8545
MAX8546
MAX8548
ON
OFF
OPTIONAL
BSTVIN
COMP/
EN
VLVCC
DH
LX
DL
GND
FB
INPUT
2.7V TO 28V
OUTPUT
0.8V TO
0.9 x VIN
UP TO 6A
Typical Operating Circuit
Ordering Information
Pin Configuration appears at end of data sheet.
19-2795; Rev 2; 6/07
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
PART
TEMP RANGE
PIN-
PACKAGE
PKG
CODE
MAX8545EUB
-40°C to +85°C
10 µMAX U10-2
MAX8545EUB+
-40°C to +85°C
10 µMAX U10-2
MAX8546EUB
-40°C to +85°C
10 µMAX U10-2
MAX8546EUB+
-40°C to +85°C
10 µMAX U10-2
MAX8548EUB
-40°C to +85°C
10 µMAX U10-2
MAX8548EUB+
-40°C to +85°C
10 µMAX U10-2
Selector Guide
PART
SWITCHING
FREQUENCY
(kHz)
CURRENT-LIMIT
THRESHOLD
(mV)
MAX8545 300 -320
MAX8546 300 -165
MAX8548 100 -320
Set-Top Boxes
Graphic Card and Video
Supplies
Desktops and Desknotes
PCI Express Power
Supplies
Telecom Power Supplies
Notebook Docking
Station Supplies
Cable Modems and
Routers
Networking Power
Supplies
+Denotes a lead-free package.
µMAX is a registered trademark of Maxim Integrated Products, Inc.
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(VIN = VL = VCC = 5V, TA= -40°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
(All voltages referenced to GND unless otherwise noted.)
VIN to GND ............................................................-0.3V to +30V
VCC to GND .............................-0.3V, lower of 6V or (VL + 0.3V)
FB to GND ................................................................-0.3V to +6V
BST to GND ............................................................-0.3V to +36V
VL, DL, COMP to GND ..............................-0.3V to (VCC + 0.3V)
BST to LX..................................................................-0.3V to +6V
DH to LX....................................................-0.3V to (VBST + 0.3V)
VL Short to GND ......................................................................5s
LX to GND ................................................................-1V to +30V
Input Current (any pin) .....................................................±50mA
Continuous Power Dissipation (TA= +70°C)
10-Pin µMAX (derate 5.6mW/°C above +70°C) ..........444mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
PARAMETER
SYMBOL
CONDITIONS
MIN TYP MAX
UNITS
VCC = VL, VIN separate from VCC 4.9
28.0
VIN Operating Range VIN VIN = VL = VCC 2.7 5.5 V
VIN Undervoltage Lockout
(UVLO) Trip Level Rising and falling edge, hysteresis = 2%
2.35 2.50 2.66
V
VIN Operating Supply Current VFB = 0.88V (no switching) 0.7 1.2 mA
VL Output Voltage 5.5V < VIN < 28V, VCC = VL,
1mA < ILOAD < 25mA 4.7 5 5.3 V
Thermal Shutdown Rising temperature, typical
hysteresis = 10°C (Note 1)
+160
°C
OSCILLATOR
MAX8545, MAX8546
250 300
360
Frequency fOSC MAX8548 80
100
120 kHz
DH output, MAX8545, MAX8546 5
Minimum Duty Cycle DCMIN MAX8548 10 %
DH output, MAX8545, MAX8546 83 86
Maximum Duty Cycle
DCMAX
MAX8548 90 95 %
SOFT-START
MAX8545, MAX8546 6.6
Digital Ramp Period MAX8548
10.2
ms
MAX8545, MAX8546 VOUT / 64
Soft-Start Levels MAX8548 VOUT / 32 V
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VL = VCC = 5V, TA= -40°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)
PARAMETER
SYMBOL
CONDITIONS
MIN
TYP
MAX
UNITS
ERROR AMPLIFIER
2.7V < VCC < 5.5V, 0°C to +85°C
0.787 0.800 0.815
FB Regulation Voltage 2.7V < VCC < 5.5V, -40°C to +85°C
0.782 0.800 0.815
V
FB to COMP/EN Gain
4000
V / V
FB to COMP/EN
Transconductance -5µA < ICOMP/EN < +5µA 70
108
160 µS
FB Input Bias Current VFB = 0.88V 1 2 µA
COMP/EN Source Current VCOMP/EN = 0V 15 46 100 µA
LX to GND, MAX8545, MAX8548,
VFB = 0.8V
-355 -320 -280
Current-Limit Threshold Voltage
(Across Low-Side MOSFET) LX to GND MAX8546, VFB = 0.8V
-185 -165 -140
mV
LX to GND, VFB = 0V,
MAX8545, MAX8548
-105
-75 -45
Foldback Current-Limit Threshold
Voltage (Across Low-Side
MOSFET) When Output is Short MAX8546, LX to GND, VFB = 0 -53 -38 -22
mV
MOSFET DRIVERS
Rising edge, DH going low to DL going high
96
Break-Before-Make Time
Falling edge, DL going low to DH going high
28 ns
DH On-Resistance in Low State 1.6 4 Ω
DH On-Resistance in High State 2.5 5.5 Ω
DL On-Resistance in Low State 1.1 2.5 Ω
DL On-Resistance in High State 2.5 5.5 Ω
BST Leakage Current VBST = 33V, VLX = 28V, VFB = 0.88V 0 50 µA
LX Leakage Current VBST = 33V, VLX = 28V, VFB = 0.88V 33 100 µA
Note 1: Thermal shutdown disables the buck regulator when the die reaches this temperature. Soft-start is reset but the VL regulator
remains on.
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
4 _______________________________________________________________________________________
Typical Operating Characteristics
(VIN = VL = VCC = 5V, typical values are at TA = +25°C, unless otherwise noted.)
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc01
LOAD CURRENT (A)
(CIRCUIT OF FIGURE 1; TABLE 1a)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VIN = 3.3V
VOUT = 1.2V
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc02
LOAD CURRENT (A)
(CIRCUIT OF FIGURE 1; TABLE 1a)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VIN = 5V
VOUT = 1.8V
VOUT = 3.3V
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc03
LOAD CURRENT (A)
(CIRCUIT OF FIGURE 2; TABLE 2a)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VOUT = 1.8V
VOUT = 3.3V
VIN = 12V
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc04
LOAD CURRENT (A)
(CIRCUIT OF FIGURE 2; TABLE 2a)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VIN = 17V
VOUT = 3.3V
VOUT = 1.8V
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc05
LOAD CURRENT (A)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VOUT = 1.8V
VOUT = 1.2V
VIN = 3.3V
(CIRCUIT OF FIGURE 1; TABLE 1b)
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc06
LOAD CURRENT (A)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VOUT = 3.3V
VOUT = 2.5V
VOUT = 1.8V
VOUT = 1.2V
VIN = 5V
(CIRCUIT OF FIGURE 1; TABLE 1b)
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc07
LOAD CURRENT (A)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VOUT = 3.3V
VOUT = 2.5V
VOUT = 1.8V
VOUT = 1.2V
VIN = 12V
(CIRCUIT OF FIGURE 2; TABLE 2b)
EFFICIENCY vs. LOAD CURRENT
MAX8545/46/48 toc08
LOAD CURRENT (A)
EFFICIENCY (%)
10.1
10
20
30
40
50
60
70
80
90
100
0
0.01 10
VOUT = 3.3V
VOUT = 2.5V VOUT = 1.8V
VOUT = 1.2V
VIN = 17V
(CIRCUIT OF FIGURE 2; TABLE 2b)
CHANGE IN OUTPUT VOLTAGE
vs. LOAD CURRENT
MAX8545/46/48 toc09
LOAD CURRENT (A)
OUTPUT VOLTAGE (V)
5432
2.485
2.490
2.495
2.500
2.505
2.510
2.515
2.520
2.480
16
VOUT = 2.5V
VIN = 12V
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
_______________________________________________________________________________________ 5
CHANGE IN OUTPUT VOLTAGE
vs. INPUT VOLTAGE
MAX8545/46/48 toc10
INPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
4.54.03.53.0
1.79
1.80
1.81
1.82
1.83
1.84
1.78
2.5
ILOAD = 6A
CHANGE IN OUTPUT VOLTAGE
vs. INPUT VOLTAGE
MAX8545/46/48 toc11
INPUT VOLTAGE (V)
OUTPUT VOLTAGE (V)
18 20 22 24161412
2.49
2.50
2.51
2.52
2.48
10
ILOAD = 6A
FREQUENCY vs. INPUT VOLTAGE
MAX8545/46/48 toc12
INPUT VOLTAGE (V)
FREQUENCY (kHz)
22.9417.8812.827.76
294
298
302
306
310
290
2.70 28.00
VOUT = 2.5V
NO LOAD
MAX8545/
MAX8546
FREQUENCY vs. TEMPERATURE
MAX8545/46/48 toc13
TEMPERATURE (°C)
FREQUENCY (kHz)
60.0035.0010.00-15.00
294
298
302
306
310
290
-40.00 85.00
VIN = 12V
VOUT = 2.5V
NO LOAD
MAX8545/
MAX8546
0
VOUT
AC COUPLED
100mV/div
IOUT
5A/div
40μs/div
LOAD TRANSIENT RESPONSE
MAX8545 toc14
VIN = 17V
VOUT = 2.5V
Typical Operating Characteristics (continued)
(VIN = VL = VCC = 5V, typical values are at TA = +25°C, unless otherwise noted.)
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
6 _______________________________________________________________________________________
Typical Operating Characteristics (continued)
(VIN = VL = VCC = 5V, typical values are at TA = +25°C, unless otherwise noted.)
VOUT
1V/div
VIN
5V/div
INDUCTOR
CURRENT
5A/div
1ms/div
STARTUP WAVEFORM
MAX8545 toc15
ILOAD = 3A
VOUT
1V/div
VIN
5V/div
INDUCTOR
CURRENT
2A/div
2ms/div
SHUTDOWN WAVEFORM
MAX8545 toc16
ILOAD = 3A
VOUT
2V/div
VIN
5V/div
IIN
10A/div
IOUT
5A/div
SHORT-CIRCUIT WAVEFORM
MAX8545 toc18
VOUT
2V/div
VIN
20V/div
IIN
2A/div
INDUCTOR
CURRENT
5A/div
1ms/div
0
0
0
0
SHORT-CIRCUIT WAVEFORM
MAX8545 toc17
GAIN AND PHASE vs. FREQUENCY
MAX8545/46/48 toc19
FREQUENCY (kHz)
GAIN (dB)
PHASE (DEGREES)
101
-120
-100
-80
-60
-40
-20
0
20
40
60
-140
0.1 100
180
150
120
90
60
30
VIN = 17V, VOUT = 2.5V
ILOAD = 6A
GAIN
PHASE
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
_______________________________________________________________________________________ 7
Pin Description
PIN NAME FUNCTION
1
COMP/EN
Compensation Input. Pull COMP/EN low with an open-collector or open-drain device to turn
off the output.
2 FB Feedback Input. Connect a resistive-divider network to set VOUT. FB threshold is 0.8V.
3V
CC Internal Chip Supply. Connect VCC to VL through a 10Ω resistor. Bypass VCC to GND with at
least a 0.1µF ceramic capacitor.
4V
IN Power Supply for LDO Regulator for VIN > 5.5V, and Chip Supply for VIN < 5.5V. Bypass VIN with
at least a 1µF ceramic capacitor to GND.
5VL
Output of Internal 5V LDO. Connect VL to VIN for VIN < 5.5V. Bypass VL with at least a 1µF ceramic
capacitor to GND.
6 DL Low-Side External MOSFET Gate-Driver Output. DL swings from VL to GND.
7 GND Ground and Negative Current-Sense Input
8 LX Inductor Switching Node. LX is used for both current limit and the return supply of the DH driver.
9 DH High-Side External MOSFET Gate-Driver Output. DH swings from BST to LX.
10 BST Positive Supply of DH Driver. Connect a 0.1µF ceramic capacitor between BST and LX.
VIN
VL
FB
BST
DH
LX
VCC
COMP/EN
5V LINEAR
REGULATOR
RAMP
GENERATOR
MAX8545
MAX8546
MAX8548
1V
PWM COMP
ERROR
AMPLIFIER
800mV
REF SOFT-START
INTERNAL
CHIP SUPPLY
100kHz/
300kHz*
CLOCK
GENERATOR
*SEE SELECTOR GUIDE
CONTROL
LOGIC
TEMPERATURE
SHUTDOWN
DL
GND
FOLD-
BACK
FB
CURRENT-LIMIT
COMPARATOR
Functional Diagram
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
8 _______________________________________________________________________________________
Detailed Description
The MAX8545/MAX8546/MAX8548 are BiCMOS switch-
mode power-supply controllers designed to implement
simple, buck-topology regulators in cost-sensitive
applications. The main power-switching circuit consists
of two n-channel MOSFETs, an inductor, and input/out-
put filter capacitors. An all n-channel synchronous-rec-
tified design provides high efficiency at reduced cost.
These devices have an internal 5V linear regulator that
steps down the input voltage to supply the IC and the
gate drivers. The low-side-switch gate driver is directly
powered from the 5V regulator (VL), while the high-
side-switch gate driver is indirectly powered from VL
plus an external diode-capacitor boost circuit.
Current-Limit and
Short-Circuit Protection
The MAX8545/MAX8546/MAX8548 employ a valley cur-
rent-sensing algorithm that uses the RDS(ON) of the low-
side n-channel MOSFET to sense the current. This
eliminates the need for an external sense resistor usually
placed in series with the output. The voltage measured
across the low-side MOSFET’s RDS(ON) is compared to
a fixed -320mV reference for the MAX8545/MAX8548
and a fixed -165mV reference for the MAX8546. The cur-
rent limit is given by the equations below:
Aside from current limiting, these devices feature fold-
back short-circuit protection. This feature is designed
to reduce the current limit by 80% as the output voltage
drops to 0V.
MOSFET Gate Drivers
The DH and DL drivers are optimized for driving n-chan-
nel MOSFETs with low gate charge. An adaptive dead-
time circuit monitors the DL output and prevents the
high-side MOSFET from turning on until the low-side
MOSFET is fully off. There must be a low-resistance,
low-inductance connection from the DL driver to the
MOSFET gate for the adaptive dead-time circuit to work
properly. Otherwise, the sense circuitry in the MAX8545/
MAX8546/MAX8548 may detect the MOSFET gate as off
while there is actually charge left on the gate. Use very
short, wide traces measuring no less than 50mils to 100
mils wide if the MOSFET is 1in away from the MAX8545/
MAX8546/MAX8548. The same type of adaptive dead-
time circuit monitors the DH off edge. The same recom-
mendations apply for the gate connection of the
high-side MOSFET.
The internal pulldown transistor that drives DL low is
robust, with a 1.1Ω(typ) on-resistance. This helps pre-
vent DL from being pulled up due to capacitive cou-
pling from the drain to the gate of the low-side
synchronous-rectifier MOSFET during the fast rise time
of the LX node.
Soft-Start
The MAX8545/MAX8546/MAX8548 feature an internally
set soft-start function that limits inrush current. It accom-
plishes this by ramping the internal reference input to the
controller’s transconductance error amplifier from 0 to
the 0.8V reference voltage. The ramp time is 1024 oscil-
lator cycles for the MAX8548 and 2048 oscillator cycles
for the MAX8545/MAX8546. At the nominal 100kHz and
300kHz switching rate, the soft-start ramp is approxi-
mately 10.2ms and 6.8ms, respectively.
High-Side Gate-Drive Supply (BST)
A flying-capacitor boost circuit generates gate-drive volt-
age for the high-side n-channel MOSFET. The flying
capacitor is connected between the BST and LX nodes.
On startup, the synchronous rectifier (low-side MOSFET)
forces LX to ground and charges the boost capacitor to
VL. On the second half-cycle, the MAX8545/MAX8546/
MAX8548 turn on the high-side MOSFET by closing an
internal switch between BST and DH. This provides the
necessary gate-to-source voltage to drive the high-side
MOSFET gate above its source at the input voltage.
Internal 5V Linear Regulator
All MAX8545/MAX8546/MAX8548 functions are internally
powered from an on-chip, low-dropout 5V regulator (VL).
These devices have a maximum input voltage (VIN) of
28V. Connect VCC to VL through a 10Ωresistor and
bypass VCC to GND with a 0.1µF ceramic capacitor. The
VIN-to-VL dropout voltage is typically 140mV, so when VIN
is less than 5.5V, VL is typically VIN - 140mV.
The internal linear regulator can source a minimum of
25mA and a maximum of approximately 40mA to supply
power to the IC low-side and high-side MOSFET drivers.
Duty-Cycle Limitations for
Low VOUT/VIN Ratios
The MAX8545/MAX8546/MAX8548s’ output voltage is
adjustable down to 0.8V. However, the minimum duty
cycle can limit the ability to supply low-voltage outputs
ImV
RMAX
LIMIT DS ON
=165 8546
()
()
ImV
RMAX MAX
LIMIT DS ON
=320 8545 8548
()
(/)
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
_______________________________________________________________________________________ 9
from high-voltage inputs. With high input voltages, the
required duty factor is approximately:
where RDS(ON) x ILOAD is the voltage drop across the
synchronous rectifier. Therefore, the maximum input
voltage (VIN(DFMAX)) that can supply a given output
voltage is:
If the circuit cannot attain the required duty cycle dic-
tated by the input and output voltages, the output volt-
age still remains in regulation. However, there may be
intermittent or continuous half-frequency operation as
the controller attempts to lower the average duty cycle
by deleting pulses. This can increase output voltage
ripple and inductor current ripple, which increases
noise and reduces efficiency. Furthermore, circuit sta-
bility is not guaranteed.
Applications Information
Design Procedures
1) Input Voltage Range. The maximum value
(VIN(MAX)) must accommodate the worst-case high
input voltage. The minimum value (VIN(MIN)) must
account for the lowest input voltage after drops due
to connectors, fuses, and switches are considered.
In general, lower input voltages provide the best
efficiency.
2) Maximum Load Current. There are two current
values to consider. Peak load current (ILOAD(MAX))
determines the instantaneous component stresses
and filtering requirements and is key in determining
output capacitor requirements. ILOAD(MAX) also
determines the required inductor saturation rating.
Continuous load current (ILOAD) determines the
thermal stresses, input capacitor, and MOSFETs,
as well as the RMS ratings of other heat-contribut-
ing components such as the inductor.
3) Inductor Value. This choice provides tradeoffs
between size, transient response, and efficiency.
Higher inductance value results in lower inductor
ripple current, lower peak current, lower switching
losses, and, therefore, higher efficiency at the cost
of slower transient response and larger size. Lower
inductance values result in large ripple currents,
smaller size, and poor efficiency, while also provid-
ing faster transient response.
Setting the Output Voltage
An output voltage between 0.8V and (0.83 x VIN) can
be configured by connecting FB to a resistive divider
between the output and GND (see Figures 1 and 2).
Select resistor R4 in the 1kΩto 10kΩrange. R3 is then
given by:
where VFB = +0.8V.
Inductor Selection
Determine an appropriate inductor value with the fol-
lowing equation:
where LIR is the ratio of inductor ripple current to aver-
age continuous maximum load current. Choosing LIR
between 20% to 40% results in a good compromise
between efficiency and economy. Choose a low-core-
loss inductor with the lowest possible DC resistance.
Ferrite-core-type inductors are often the best choice for
performance; however, the MAX8548’s low switching
frequency also allows the use of powdered iron core
inductors in ultra-low-cost applications where efficiency
is not critical. With any core material, the core must be
large enough not to saturate at the peak inductor cur-
rent (IPEAK):
Setting the Current Limit
The MAX8545/MAX8546/MAX8548 provide valley cur-
rent limit by sensing the voltage across the external
low-side MOSFET. The minimum current-limit threshold
voltage is -280mV for the MAX8545/MAX8548 and
-140mV for the MAX8546. The MOSFET on-resistance
required to allow a given peak inductor current is:
where IVALLEY = ILOAD(MAX) x (1 - LIR / 2), and
RDS(ON)MAX is the maximum on-resistance of the low-
side MOSFET at the maximum operating junction
temperature.
RV
Ifor the MAX
DS ON MAX VALLEY
()
. ( )≤014 8546
RV
Ifor the MAX MAX
DS ON MAX VALLEY
()
. ( / )≤028 8545 8548
II LIR I
PEAK LOAD MAX LOAD MAX
=+
⎛
⎝
⎜⎞
⎠
⎟×
() ()
2
LV VV
V f LIR I
OUT IN OUT
IN OSC LOAD MAX
=× −
()
××× ()
RR
V
V
OUT
FB
34 1=−
⎡
⎣
⎢⎤
⎦
⎥
VDC VR I
IN DFMAX MIN OUT DS ON LOAD() ()
≤+×
()
()
1
VR I
V
OUT DS ON LOAD
IN
+×()
()
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
10 ______________________________________________________________________________________
MAX8545*
MAX8546
MAX8548*
BSTVCC
COMP/
EN
VLVIN
DH
LX
DL
GND
FB
+2.7V TO
+5.5V INPUT
OUT
1.8V/3A OR 6A
D2 R2
C10 C11
C1
C5
D1 Q1
L1
R4
R3
C6
C3 C2C4
C7 C8
*FOLLOW THE DATA SHEET DESIGN PROCEDURE TO SELECT THE
EXTERNAL COMPONENTS FOR THE MAX8545/MAX8548.
R1
C9
Figure 1. Typical Application Circuit (2.7V to 5V) Input (see Tables 1a, 1b)
MAX8545*
MAX8546
MAX8548*
BST
VIN
COMP/
EN
VLVCC
DH
LX
DL
GND
FB
+10V TO
+24V INPUT
OUT
2.5V/3A OR 6A
D2 R2
C10 C11
C1
C12
C9
C5
D1 Q1
R1
L1
R4
R3
C6
C3 C2C4
C7 C8
*FOLLOW THE DATA SHEET DESIGN PROCEDURE TO SELECT THE
EXTERNAL COMPONENTS FOR THE MAX8545/MAX8548.
Figure 2. Typical Application Circuit (10V to 24V) Input (see Tables 2a, 2b)
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
______________________________________________________________________________________ 11
A limitation of sensing current across a MOSFET’s on-
resistance is that the current-limit threshold is not accu-
rate since MOSFET RDS(ON) specifications are not
precise. This type of current limit provides a coarse level
of fault protection. It is especially suited when the input
source is already current-limited or otherwise protected.
Power MOSFET Selection
The MAX8545/MAX8546/MAX8548 drive two external,
logic-level, n-channel MOSFETs as the circuit switching
elements. The key selection parameters are:
1) On-resistance (RDS(ON)): the lower, the better.
2) Maximum drain-to-source voltage (VDSS) should be
at least 10% higher than the input supply rail at the
high-side MOSFET’s drain.
3) Gate charges (QG, QGD, QGS): the lower, the better.
Choose the MOSFETs with rated RDS(ON) at VGS = 4.5V
for an input voltage greater than 5V, and at VGS = 2.5V
for an input voltage less than 5.5V. For a good compro-
mise between efficiency and cost, choose the high-side
MOSFET (N1) that has conduction losses equal to the
switching losses at nominal input voltage and maximum
output current. For N2, make sure it does not spuriously
turn on due to a dV/dt caused by N1 turning on as this
would result in shoot-through current degrading the
efficiency. MOSFETs with a lower QGD/QGS ratio have
higher immunity to dV/dt.
MOSFET Power Dissipation
For proper thermal-management design, the power dis-
sipation must be calculated at the desired maximum
operating junction temperature, maximum output cur-
rent, and worst-case input voltage (for the low-side
MOSFET (N2) the worst case is at VIN(MAX), for the high-
side MOSFET (N1) the worst case can be either at
VIN(MIN) or VIN(MAX)). N1 and N2 have different loss
components due to the circuit operation. N2 operates as
a zero-voltage switch; therefore, the major losses are:
the channel conduction loss (PN2CC), the body-diode
conduction loss (PN2DC), and the gate-drive loss
(PN2DR):
Use RDS(ON) at TJ(MAX):
where VFis the body-diode forward voltage drop, tdt is
the dead time between N1 and N2 switching transitions
(which is 30ns), and fSis the switching frequency.
Because of zero-voltage switch operation, the N2 gate-
drive losses are due to charging and discharging the
input capacitor, CISS. These losses are distributed
between the average DL gate driver’s pullup and pull-
down resistors and the internal gate resistance. The
RDL is typically 1.8Ω, and the internal gate resistance
(RGATE) of the MOSFET is typically 2Ω. The drive
power dissipated in N2 is given by:
N1 operates as a duty-cycle control switch and has the
following major losses: the channel conduction loss
(PN1CC), the voltage and current overlapping switching
loss (PN1SW), and the drive loss (PN1DR). N1 does not
have a body-diode conduction loss because the diode
never conducts current:
Use RDS(ON) at TJ(MAX):
where IGATE is the average DH high driver output-cur-
rent capability determined by:
where RDH is the high-side MOSFET driver’s average
on-resistance (2.05Ωtyp) and RGATE is the internal
gate resistance of the MOSFET (2Ωtyp):
where VGS ~ VL.
In addition to the losses above, allow about 20% more
for additional losses due to MOSFET output capaci-
tance and N2 body-diode reverse recovery charge dis-
sipated in N1. Refer to the MOSFET data sheet for
thermal resistance specifications to calculate the PC
board area needed. This information is essential to
maintain the desired maximum operating junction tem-
perature with the above calculated power dissipation.
To reduce EMI caused by switching noise, add a 0.1µF
ceramic capacitor from the high-side MOSFET drain to
the low-side MOSFET source or add resistors in series
PQVfR
RR
NDR GS GS S GATE
DH GATE
1=×××+
IVL
RR
GATE ON DH GATE
()
=× +
1
2
PVI f
QQ
I
N SW IN LOAD S GS GD
GATE
1=× ×× +
PV
VIR
NCC OUT
IN LOAD DS ON12
=⎛
⎝
⎜⎞
⎠
⎟×
()
×()
PCVf
R
RR
N DR ISS GS S GATE
GATE DL
22
=×
()
×× +
PIVtf
N DC LOAD F dt S22=× × × ×
PV
VIR
NCC OUT
IN
LOAD DS ON22
1=−
⎛
⎝
⎜⎞
⎠
⎟××
()
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
12 ______________________________________________________________________________________
with DH and DL to slow down the switching transitions.
However, adding series resistors increases the power
dissipation of the MOSFET, so ensure temperature rat-
ings of the MOSFET are not exceeded.
Input-Capacitor Selection
The input capacitors (C2 and C3 in Figure 1) reduce
noise injection and current peaks drawn from the input
supply. The input capacitor must meet the ripple-cur-
rent requirement (IRMS) imposed by the switching cur-
rents. The RMS input ripple current is given by:
For optimal circuit reliability, choose a capacitor that
has less than 10°C temperature rise at the RMS current.
IRMS is maximum when the input voltage equals 2 x
VOUT, where IRMS = 1/2 ILOAD.
Output Capacitor Selection
The key parameters for the output capacitor are the
actual capacitance value, the equivalent series resis-
tance (ESR), the equivalent series inductance (ESL),
and the voltage-rating requirements. All these parame-
ters affect the overall stability, output ripple voltage,
and transient response.
The output ripple has three components: variations in the
charge stored in the output capacitor, the voltage drop
across the ESR, and the voltage drop across the ESL.
VRIPPLE = VRIPPLE(ESR) + VRIPPLE(C) + VRIPPLE(ESL)
The output voltage ripple as a consequence of the ESR
and output capacitance is:
where IP-P is the peak-to-peak inductor current (see the
Inductor Selection section).
While these equations are suitable for initial capacitor
selection to meet the ripple requirement, final values
may also depend on the relationship between the LC
double-pole frequency and the capacitor ESR-zero fre-
quency. Generally, the ESR zero is higher than the LC
double pole; however, it is preferable to keep the ESR
zero close to the LC double pole when possible to
negate the sharp phase shift of the typically high-Q
double LC pole (see the Compensation Design sec-
tion). Aluminum electrolytic or POS capacitors are rec-
ommended. Higher output current requires multiple
capacitors to meet the output ripple voltage.
The MAX8545/MAX8546/MAX8548s’ response to a load
transient depends on the selected output capacitor. After
a load transient, the output instantly changes by (ESR x
ΔILOAD) + (ESL x dI/dt). Before the controller can
respond, the output deviates further depending on the
inductor and output capacitor values. After a short period
of time (see the Typical Operating Characteristics), the
controller responds by regulating the output voltage back
to its nominal state. The controller response time
depends on the closed-loop bandwidth. Higher band-
width results in faster response time, preventing the out-
put voltage from further deviation. Do not exceed the
capacitor’s voltage or ripple-current ratings.
Boost Diode and Capacitor Selection
A low-current Schottky diode, such as the CMPSH-3
from Central Semiconductor, works well for most appli-
cations. Do not use large power diodes since higher
junction capacitance can charge up BST to LX voltage
that could exceed the device rating of 6V. The boost
capacitor should be in the range of 0.1µF to 0.47µF,
depending on the specific input and output voltages
and the external components and PCB layout. The
boost capacitance needs to be as large as possible to
prevent it from charging to excessive voltage, but small
enough to adequately charge during the minimum low-
side MOSFET conduction time, which happens at the
maximum operating duty cycle (this occurs at the mini-
mum input voltage). In addition, ensure the boost
capacitor does not discharge to below the minimum
gate-to-source voltage required to keep the high-side
MOSFET fully enhanced for lowest on-resistance. This
minimum gate-to-source voltage VGS(MIN) is deter-
mined by:
where QGis the total gate charge of the high-side
MOSFET and CBOOST is the boost capacitor value.
Compensation Design
The MAX8545/MAX8546/MAX8548 use a voltage-mode
control scheme that regulates the output voltage. This is
done by comparing the error amplifier’s output (COMP) to
a fixed internal ramp. The inductor and output capacitor
create a double pole at the resonant frequency, which
VV
Q
C
GS MIN L G
BOOST
()
=−
V I ESR
VI
Cf
VV ESL
L ESL
IVV
fL
V
V
RIPPLE ESR P P
RIPPLE C PP
OUT SW
RIPPLE ESL IN
PP IN OUT
SW
OUT
IN
()
()
()
=×
=××
=×
+
=−
×
⎛
⎝
⎜⎞
⎠
⎟⎛
⎝
⎜⎞
⎠
⎟
−
−
−
8
II VVV
V
RMS LOAD OUT IN OUT
IN
=× ×−
()
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
______________________________________________________________________________________ 13
has a gain drop of 40dB per decade, and a phase shift
of 180°. The error amplifier must compensate for this
gain drop and phase shift to achieve a stable high-
bandwidth, closed-loop system.
The basic regulator loop consists of a power modulator
(Figure 3), an output feedback divider, and an error
amplifier. The power modulator has DC gain set by
VIN/VRAMP, with a double pole set by the inductor and
output capacitor, and a single zero set by the output
capacitor (COUT) and its equivalent series resistance
(ESR). Below are equations that define the power mod-
ulator:
The DC gain of the power modulator is:
where VRAMP = 1V.
The pole frequency due to the inductor and output
capacitor is:
The zero frequency due to the output capacitor’s ESR is:
The output capacitor is usually comprised of several
same capacitors connected in parallel. With n capaci-
tors in parallel, the output capacitance is:
COUT = n x CEACH
The total ESR is:
The ESR zero (fZESR) for a parallel combination of
capacitors is the same as for an individual capacitor.
The feedback divider has a gain of GFB = VFB/VOUT,
where VFB is 0.8V.
The transconductance error amplifier has DC gain
GEA(dc) of 72dB. A dominant pole (fDPEA) is set by the
compensation capacitor (CC), the amplifier output
resistance (RO) equals 37MΩ, and the compensation
resistor (RC):
The compensation resistor and the compensation
capacitor set a zero:
The total closed-loop gain must equal unity at the
crossover frequency. The crossover frequency should
be higher than fZESR, so that the -1 slope is used to
cross over at unity gain. Also, the crossover frequency
should be less than or equal to 1/5 the switching fre-
quency (fSW) of the controller:
The loop-gain equation at the crossover frequency is:
VFB/VOUT x GEA(fC)x GMOD(fC)= 1
where GEA(fC)= gmEA ×RC, and GMOD(fC)=
GMOD(DC) ×(fPMOD)2 / (fZESR ×fC).
The compensation resistor, RC, is calculated from:
RC= VOUT/ gmEA x VFB x GMOD(fC)
where gmEA = 108µS.
Due to the underdamped (Q > 1) nature of the output
LC double pole, the error-amplifier compensation zero
should be approximately 0.2 fPMOD to provide good
phase boost. CCis calculated from:
A small capacitor, CF, can also be added from COMP to
GND to provide high-frequency decoupling. CFadds
another high-frequency pole, fPHF, to the error-amplifier
response. This pole should be greater than 100 times the
error-amplifier zero frequency to have negligible impact
on the phase margin. This pole should also be less than
1/2 the switching frequency for effective decoupling:
100 fZEA < fPHF < 0.5 fsw
Select a value for fPHF in the range given above, then
solve for CFusing the following equation:
PCB Layout Guidelines
Careful PCB layout is critical to achieve low switching
losses and stable operation. If possible, mount all the
power components on the top side of the board with their
CRf
FC PHF
=××
1
2π
CRf
CC PMOD
=××
5
2π
ff
f
ZESR C SW
<≤5
fCR
ZEA CC
=××
1
2π
fCRR
DPEA COC
=×× +
()
1
2π
ESR ESR
n
EACH
=
fESR C
ZESR OUT
=××
1
2π
fLC
PMOD
OUT
=1
2π
GV
V
MOD DC IN
RAMP
()
=
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
14 ______________________________________________________________________________________
ground terminals flush against one another. Follow these
guidelines for good PCB layout:
1) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for sta-
ble, jitter-free operation.
2) Connect the power and analog grounds close to
the IC pin 7.
3) Keep the power traces and load connections short.
This practice is essential for high efficiency. Using
thick copper PCBs (2oz vs. 1oz) can enhance full-
load efficiency by 1% or more. Correctly routing
PCB traces is a difficult task that must be
approached in terms of fractions of centimeters,
where a few milohms of excess trace resistance
cause a measurable efficiency penalty.
4) LX and GND connections to the low-side MOSFET
for current sensing must be made using Kelvin
sense connections to guarantee the current-limit
accuracy. With 8-pin MOSFETs, this is best done
by routing power to the MOSFETs from outside
using the top copper layer, while connecting LX
and GND inside (underneath) the 8-pin package.
5) When tradeoffs in trace lengths must be made, it is
preferable to allow the inductor charging current
path to be longer than the discharge path. For
example, it is better to allow some extra distance
between the inductor and the low-side MOSFET or
between the inductor and the output filter capacitor.
6) Ensure that the connection between the inductor
and C3 is short and direct.
7) Route switching nodes (BST, LX, DH, and DL) away
from sensitive analog areas (COMP and FB).
Ensure the C1 ceramic bypass capacitor is immediately
adjacent to the pins and as close to the device as possi-
ble. Furthermore, the VIN and GND pins of MAX8545/
MAX8546/MAX8548 must terminate at the two ends of
C1 before connecting to the power switches and C2.
MAX8545
MAX8546
MAX8548
N
N
RAMP
GENERATOR
PWM
COMP/EN
R2
C10
COUT
VOUT
R3
R4
L
VIN
0.8V
ERROR
AMPLIFIER
DH
LX
DL
FB
Figure 3. Compensation Scheme
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
______________________________________________________________________________________ 15
COMPONENT
QTY
DESCRIPTION
C1, C4
2
1µF, 10V X7R ceramic capacitors
Taiyo Yuden LMK212BJ105MG
C2
0
Not installed
C3
1
1200µF, 10V, 44mΩ, 1.25A aluminum
electrolytic capacitor
SANYO 10MV1200AX
(10 x 16 case size)
C5, C8, C9
3
0.1µF, 10V X7R ceramic capacitors
Kemet C0603C104M8RAC
C6, C7
2
1000µF, 6.3V , 69m Ω, 0.8A al um i num
el ectr ol yti c cap aci tor s
S AN Y O 6.3M V 1000AX ( 8 x 20 case si ze)
C10
1
1.5nF, 10V X7R ceramic capacitor
Kemet C0603C152M8RAC
C11
0
Not installed
D1, D2
2
30V, 100mA Schottky diodes
Central Semiconductor CMPSH-3
L1
1
4.7µH, 5.7A, 18mΩ inductor
Sumida CDRH124-4R7
Q1
1
20V/30V, 35mΩ dual n-channel,
8-pin SO
V i shay S i 4966D Y ( f o r 2.7 V t o 3.6 VIN
)
Fair chil d FD S 6912A ( f or 4 .5 V t o 5 .5 VIN
)
R1
1
10Ω ±5% resistor
R2
1
150kΩ ±5% resistor
R3
1
5.11kΩ ±1% resistor
R4
1
4.02kΩ ±1% resistor
COMPONENT QTY DESCRIPTION
C1, C4 2 1µF, 10V X7R ceramic capacitors
Taiyo Yuden LMK212BJ105MG
C2, C3 2
1200µF, 10V, 44mΩ, 1.25A aluminum
electrolytic capacitors
S ANY O 10MV1200AX
(10 x 16 case size)
C5, C8, C9 3 0.1µF, 10V X7R ceramic capacitors
Kemet C0603C104M8RAC
C6, C7 2
1500µF, 6.3V, 44m Ω, 1.25A al um i num
el ectr ol ytic cap acitor s
S ANY O 6.3M V1500AX (10 x 20 case si ze)
C10 1 1.5nF, 10V X7R ceramic capacitor
Kemet C0603C152M8RAC
C11 0 Not installed
D1, D2 2 30V, 100mA Schottky diodes
Central Semiconductor CMPSH-3
L1 1 2.1µH, 8A, 11.6mΩ inductor
Sumida CEP122-2R1
Q1 1
20V, 18m Ω d ual n- channel, 8-p i n S O
Fair chil d FD S 6898A ( f or 2 .7 V t o 3 .6 VIN
)
Fair chil d FD S 6890A ( f or 4 .5 V t o 5 .5 VIN
)
R1 1 10Ω ±5% resistor
R2 1 110kΩ ±5% resistor
R3 1 5.11kΩ ±1% resistor
R4 1 4.02kΩ ±1% resistor
Table 1a. Component Selection for
Standard Applications for VIN = 2.7V to
5.5V, VOUT = 1.8V / 3A (Figure 1)
(MAX8546 Only)
Table 1b. Component Selection for
Standard Applications for VIN = 2.7V to
5.5V, VOUT = 1.8V / 6A (Figure 1)
(MAX8546 Only)
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
16 ______________________________________________________________________________________
COMPONENT
QTY
DESCRIPTION
C1
1
1µF, 10V X7R ceramic capacitor
Taiyo Yuden LMK212BJ105MG
C2
0
Not installed
C3
1
470µF, 35V, 39mΩ, 1.45A aluminum
electrolytic capacitor
SANYO 35MV470AX
(10 x 22 case size)
C4, C12
2
1µF, 35V X7R ceramic capacitors
Taiyo Yuden GMK316BJ105ML
C5, C8, C9
3
0.1µF, 10V X7R ceramic capacitors
Kemet C0603C104M8RAC
C6, C7
2
1000µF, 6.3V, 69mΩ, 0.8A aluminum
electrolytic capacitors
SANYO 6.3MV1000AX
(8 x 20 case size)
C10
1
6.8nF, 10V X7R ceramic capacitor
Kemet C0603C6822M8RAC
C11
0
Not installed
D1, D2
2
30V, 100mA Schottky diodes
Central Semiconductor CMPSH-3
L1
1
8.2µH, 5.8A, 9.5mΩ inductor
Sumida CEP125-8R2
Q1
1
30V, 35mΩ, dual n-channel, 8-pin SO
Fairchild FDS6912A
R1
1
10Ω ±5% resistor
R2
1
82kΩ ±5% resistor
R3
1
8.66kΩ ±1% resistor
R4
1
4.02kΩ ±1% resistor
COMPONENT QTY DESCRIPTION
C1 1 1µF, 10V X7R ceramic capacitor
Taiyo Yuden LMK212BJ105MG
C2, C3 2
470µF, 35V, 39mΩ, 1.45A aluminum
electrolytic capacitors
SANYO 35MV470AX
(10 x 22 case size)
C4, C12 2 1µF, 35V X7R ceramic capacitors
Taiyo Yuden GMK316BJ105ML
C5, C8, C9 3 0.1µF, 10V X7R ceramic capacitors
Kemet C0603C104M8RAC
C6, C7 2
1500µF, 6.3V, 44mΩ, 1.25A aluminum
electrolytic capacitors
SANYO 6.3MV1500AX
(10 x 20 case size)
C10 1 6.8nF, 10V X7R ceramic capacitor
Kemet C0603C682M8RAC
C11 0 Not installed
D1, D2 2 30V, 100mA Schottky diodes
Central Semiconductor CMPSH-3
L1 1 4µH, 8.3A, 6.6mΩ inductor
Sumida CEP125-4R0
Q1 1
30V, 18mΩ (LSFET)/35mΩ (HSFET),
dual n-channel, 8-pin SO
Fairchild FDS6982
R1 1 10Ω ±5% resistor
R2 1 68kΩ ±5% resistor
R3 1 8.66kΩ ±1% resistor
R4 1 4.02kΩ ±1% resistor
Table 2a. Component Selection for
Standard Applications for VIN = 10V to
24V, VOUT = 2.5V / 3A (Figure 2)
(MAX8546 Only)
Table 2b. Component Selection for
Standard Applications for VIN = 10V to
24V, VOUT = 2.5V / 6A (Figure 2)
(MAX8546 Only)
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
______________________________________________________________________________________ 17
56
LX
GNDVIN
1
2
10
9
BST
DHFB
VCC
COMP/EN
μMAX
TOP VIEW
3
4
8
7
DLVL
MAX8545
MAX8546
MAX8548
Pin Configuration Chip Information
TRANSISTOR COUNT: 3351
PROCESS: BiCMOS
MAX8545/MAX8546/MAX8548
Low-Cost, Wide Input Range, Step-Down
Controllers with Foldback Current Limit
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
18 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2007 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information
go to www.maxim-ic.com/packages.)
10LUMAX.EPS
PACKAGE OUTLINE, 10L uMAX/uSOP
1
1
21-0061 REV.DOCUMENT CONTROL NO.APPROVAL
PROPRIETARY INFORMATION
TITLE:
TOP VIEW
FRONT VIEW
1
0.498 REF
0.0196 REF
S6°
SIDE VIEW
α
BOTTOM VIEW
0° 0° 6°
0.037 REF
0.0078
MAX
0.006
0.043
0.118
0.120
0.199
0.0275
0.118
0.0106
0.120
0.0197 BSC
INCHES
1
10
L1
0.0035
0.007
e
c
b
0.187
0.0157
0.114
H
L
E2
DIM
0.116
0.114
0.116
0.002
D2
E1
A1
D1
MIN
-A
0.940 REF
0.500 BSC
0.090
0.177
4.75
2.89
0.40
0.200
0.270
5.05
0.70
3.00
MILLIMETERS
0.05
2.89
2.95
2.95
-
MIN
3.00
3.05
0.15
3.05
MAX
1.10
10
0.6±0.1
0.6±0.1
Ø0.50±0.1
H
4X S
e
D2
D1
b
A2 A
E2
E1 L
L1
c
α
GAGE PLANE
A2 0.030 0.037 0.75 0.95
A1
Revision History
Pages changed at Rev 2: 1, 2, 8, 11, 15, 16, 18
