October 1994 - Linear Technology

OCTOBER 1994 VOLUME IV NUMBER 3

The LT1372 500kHz

Switching Regulator

by Bob Essaff

IN THIS ISSUE . . .

COVER ARTICLE

The LT1372 500kHz

Switching Regulator ....... 1

Bob Essaff

Editor's Page ................... 2

Richard Markell

LTC  in the News............. 2

DESIGN FEATURES

PCMCIA Card Socket

Power Management ......... 3

Doug La Porte

An Ultra-Selective Bandpass

Filter with Adjustable Gain

........................................ 6

Philip Karantzalis

New LT1182/1183 Dual

CCFL/LCD Contrast

Switching Regulator ....... 8

Anthony Bonte

The LT1366 Family:

Precision, Rail-to-Rail

Bipolar Amplifiers ......... 13

William Jett and Sean Gold

LTC1325 Battery

Management System Offers

Unparalleled Flexibility..17

Anthony Ng, Teo Yang Long, and Robert Reay

DESIGN INFORMATION

Simple Thermal Analysis —

A Real Cool Subject for LTC

Regulators..................... 23

Alan Rich

DESIGN IDEAS

................................. 25–29

(complete list on page 25)

New Device Cameos ....... 30

Design Tools .................. 32

Sales Offices ................. 32

Introduction

The LT1372 is a 500kHz, constant-

frequency, current-mode bipolar

switching regulator with an on-chip

power switch. The LT1372 is similar

to the popular LT1172 but offers

higher efficiency, and, due to the in-

creased switching frequency, uses

smaller external components, which

save on board space, weight, and

cost. An entire DC-to-DC converter

based on the LT1372 consumes only

0.5 square inches. Packaged in an 8-

lead SOIC or DIP, it operates over a

2.5V to 35V input supply range and

its current-limited switch can carry

up to 1.5A with 0.45Ω on resistance.

New design techniques increase

the LT1372’s flexibility while main-

taining ease of use. Switching is easily

synchronized to an external logic-

level source. A logic low on the

shutdown pin reduces supply cur-

rent to 15µA. Unique error-amplifier

circuitry can regulate positive or nega-

tive output voltages while maintaining

simple frequency-compensation tech-

niques. Nonlinear error-amplifier

transconductance reduces output

overshoot on start-up or overload re-

covery. Oscillator frequency shifting

protects external components during

overload conditions.

The LT1372 uses a new, high-speed

process that is tailored to high-fre-

quency switching regulators. Though

slow when compared with state-of-

the-art digital technologies, this

process is a compromise between

speed and breakdown voltage, which

is important for switching regulator

design. Smaller device geometries al-

low for faster circuits at lower

operating currents, increased circuit

complexity, and lower parasitic resis-

tance for a given die area.

continued on page 20

Figure 1. LT1372 Block diagram

–

1372_1.eps

1.24V

REF

GND SENSE GND

0.1Ω

SWITCH

5:1 FREQUENCY

SHIFT

SHUTDOWN

DELAY & RESET

500kHz

OSC

SYNC

SD/SYNC

NFB

50k

LOW DROPOUT

2.3V REG ANTI-SAT &

DRIVE BOOST

NFB

100k

EA IA

≈ 6

VIN

LOGIC DRIVER

COMP

and LTC are registered trademarks and LT is a trademark of Linear Technology Corporation.

LINEAR TECHNOLOGY

Linear Technology Magazine • October 1994

EDITOR'S PAGE

On Growing and Growth

Another feature article examines the

LTC1325 battery management system

IC. This part offers the designer a com-

plete solution to the problem of charging

a wide variety of batteries, including

NiMH, NiCad, lead-acid, and other

battery chemistries. The part allows

easy communications to a micropro-

cessor by means of a four-wire interface.

LT brings the future of switching regu-

lators to the forefront by introducing

the new LT1372 constant-frequency,

current-mode switching regulator.

The LT1372 saves board space, weight,

and cost by combining a 500kHz

switching frequency with an on-chip

power switch (0.45 ohms) that can

carry 1.5 amps.

Linear Technology has entered the

PCMCIA market with a variety of so-

lutions. The LT1312 and LT1313 are

micropower, single and dual 120mA

PCMCIA VPP regulators/drivers to

power and protect the VPP pins of

PCMCIA slots. Also featured in the

article on PCMCIA are the LTC1314

and LTC1315 single and dual VPP

switching-matrix ICs with internal

N-channel gate drivers. The

LTC1164-8 merits a feature article as

it is the first completely integrated,

high-Q, bandpass filter offered by Lin-

ear Technology. The part requires only

an external operational amplifier and

two resistors to implement very selec-

tive narrow bandpass filters with gains

up to one thousand.

We also present some timely infor-

mation on thermal analysis. The

article “Simple Thermal Analysis—A

Real Cool Subject for LTC Regulators”

gives thermal information and meth-

ods of calculating heat dissipation,

heat sinking, and temperature rise.

The Design Ideas section is full, as

usual. The LT1585 regulator is fea-

tured in a circuit developed to power

the new Pentium

class of micropro-

cessor. A novel circuit detailed in this

section directly senses the negative

output in a positive-to-negative con-

verter. Additionally, we have many

other new design ideas.

Every year, things grow at their

own pace: the beans grow up the

trellis in August and the peppers and

tomatoes ripen in September; the

Great Pumpkin may not be ready

until October.

Linear Technology is different. The

applications group has grown about

sixfold in only six years. The IC-

design group has grown by a factor of

more than three. We’ve got stacks and

piles of scratched notes on customer

problems on each applications

engineer’s desk. Each and every de-

signer is scratching his or her head

trying to pack the most functionality

into the smallest die size. There are

now design centers in Boston and

Singapore, in addition to the corpo-

rate headquarters in Milpitas. Analog

circuit designers are difficult to find

anywhere in the world, let alone where

you’d like to find them.

Growing a company is not like

growing a single Jwala pepper, but

more like growing the whole crop for

salsa. Design and applications need

fertilization (new parts) and new ideas

to produce growth; more fabrication

facilities must be built, packaging

needs to be done, and these facilities

must be provided on a worldwide ba-

sis in order to attract the type of

talent required to grow the business

in the direction that we’d all like.

This issue is filled with morsels for

the engineering palate. The LT1366

family is our first series of dual and

quad bipolar operational amplifiers

that combine rail-to-rail input and

output specifications with precision

offset-voltage specifications. This fam-

ily maintains precision specifications

over a wide range of operating condi-

tions and capacitive loads. A feature

article highlights (and backlights) the

LT1182 and LT1183 dual CCFL/LCD

contrast switching regulators. These

parts are dual switching regulators

that drive both the cold cathode fluo-

rescent lamp and the liquid crystal

display panels in portable computers

and hand-held instruments.

LTC in the News...

Linear Technology’s Annual

Sales Cross $200 Million

Thanks to the support of all of

you who rely on the quality and

performance of Linear Technology’s

products, the company’s annual

sales crossed the $200 million mark

in fiscal 1994. Net sales for the

fiscal year ended July 3, 1994 were

a record $200,538,000, an increase

of 33% over the previous year. The

Company also reported record net

income for the year of $56,827,000

or $1.51 per share, an increase of

56% over the $36,435,000 reported

for fiscal 1993.

Linear Technology has added Leo

T. McCarthy and Richard M. Moley

to its Board of Directors. Mr.

McCarthy and Mr. Moley fill exist-

ing vacancies on the board.

McCarthy is currently Lieutenant

Governor of the State of California

and has announced plans to retire

from office at the end of 1994. Rich-

ard M. Moley is Chairman of the

Board, President, and CEO of

StrataCom, Inc., a telecommuni-

cations company.

In its June 21 issue, Financial

World magazine included Linear

Technology on its list of “America’s

50 Best Mid-Cap Companies,” rank-

ing the company 21st.

In a very exclusive selection,

WORTH magazine picked Linear

Technology for its list of “50 New

Blue Chip Stocks.” More than a

thousand companies were consid-

ered before the final selection was

made. The only other semiconduc-

tor company to make the list was

Intel.

LTC also made another presti-

gious current list: Financial World’s

“America’s Best 200 Growth Com-

panies.” We made this list last year,

too, but this year LTC was 13th

compared to 22nd on the list in

1993.

In June the Entrepreneur of the

Year Institute named Linear Tech-

nology President and CEO Robert

H. Swanson, Jr. “Northern Califor-

nia Entrepreneur of the Year in

Electronics.”

by Richard Markell

Pentium is a trademark of Intel Corporaton

Linear Technology Magazine • October 1994

PCMCIA Card Socket

Power Management

Introduction

A variety of products are designed

with sockets to accept PCMCIA cards,

such as extended memories, fax mo-

dems, network interfaces, wireless

communicators, and a wide assort-

ment of other devices. These products

include personal digital assistants

(PDAs), bar-code readers, palmtop

computers, and portable medical

equipment. The Personal Computer

Memory Card International Asso-

ciation (PCMCIA) has released

specifications that outline the gen-

eral power requirements for these

cards. The specification calls for

an unusual amount of voltage

switching.

Host power delivery to the PC card

socket flows through two paths: the

main V

supply pins and the VPP

programming pins. Both supplies are

switchable to different voltages to

accommodate a wide range of card

types. The V

supply is the main

supply and must be capable of pro-

viding up to 1A at either 3.3V or 5V, as

well as a high impedance state. The

1A rating is an absolute maximum

derived from the contact rating of

500mA per pin for both V

pins. One

of the most stringent actual require-

ments is during hard disk drive

spin-up. Present hard drives require

5V at 500–600mA for a short dura-

Table 1. LT1312 truth table

EN0 EN1 SENSE VPPOUT VALID

00 X 0V 1

1 0 X 12V 0

0 1 3V–3.6V 3.3V 1

0 1 4.5V–5.5V 5V 1

1 1 X Hi-Z 1

X = Don’t care

Figure 1. Typical single-socket power-management system

tion during spin-up. Current draw

drops to 300–420mA during read and

write operations. The VPP supply must

source 12V at up to 120mA, 3.3V or

5V at lesser currents, 0V, and realize

a high impedance state. The VPP sup-

ply is intended solely for flash-memory

programming. The 120mA current

requirement allows erasing two flash

devices and writing to two devices

simultaneously. These diverse, spe-

cific requirements call for specialized

solutions.

There are two general approaches

to providing VPP power to PCMCIA

card sockets: linear driver/regula-

tors and a VPP switch matrix. The

linear regulator approach utilizes a

regulator that incorporates resistors

and switches to program the output

voltage to the appropriate level. This

solution requires a voltage of about

13V or higher. At first this may seem

undesirable but, as will be shown

later, when designed as part of a por-

table system’s power supply from the

beginning, this higher voltage is de-

rived at little cost. The linear regulator

also provides current and thermal

limiting for short-circuit protection.

The switch-matrix approach consists

of several MOSFET switches arranged

to route the input voltages to the VPP

pins. This approach requires that all

voltages be present at the matrix in-

puts. The switch approach has the

advantages of simplicity and effi-

ciency, but lacks built-in current

limiting and short-circuit protection.

LT1312: Linear VPP

Driver/Regulator

Figure 1 shows a space-efficient,

cost-effective power-management

solution for PCMCIA sockets. Power

to the two VPP pins, usually tied

together to reduce cost, is provided

by the LT1312 single VPP driver/

regulator.

The LT1312 is a linear regulator

with programmable output, designed

specifically for PCMCIA VPP drive

applications. Figure 2 shows the cir-

cuit block diagram. The LT1312 takes

a raw, unregulated 13–20V input

supply and produces a clean, regu-

lated, selectable output voltage in

conformance with the PCMCIA stan-

dard. The VPP pins are programmable

to provide either 0V, 3.3V, 5V, 12V, or

a high-impedance state. Two enable

inputs (EN0 and EN1) and the V

sense inputs select among these five

states, as shown in Table 1. When a

voltage is selected, a comparator

in the LT1312 automatically switches

the VPP

OUT

pin to 5V or 3.3V, depend-

ing on the voltage present at the

SENSE pin. The SENSE pin connects

to the PCMCIA socket V

pin.

A second comparator monitors the

output voltage when the VPP pin re-

quires 12V. This comparator drives

DESIGN FEATURES

13V TO 20V

EN0

51k

1µF

10µF

VPP

OUT

VPP1

VPP2 PCMCIA

CARD

SOCKET

LT1312EN1

VALID SENSE

GND

Q1

1/2 Si9956DY

Si9956DY

3.3V

PCMCIA_1.eps

IN1 OUT1

LTC1165

PCMCIA

CONTROLLER

IN2 OUT2

IN3 OUT3

GND

LOGIC

A_VPP_PGM

A_VPP_V

A_V

A_VPP_VALID

by Doug La Porte

Linear Technology Magazine • October 1994

A dual VPP linear regulator, the

LT1313, is also available.

Three N-channel MOSFETs, driven

by an LTC1165 triple inverting MOS-

FET gate driver, provide V

pin power

switching. P-channel MOSFETs can

be used in place of the driver and N-

channel parts, at the expense of

significantly higher on-resistance.

This is of greater significance as V

goes lower, as in the case of the two

3.3V switches. These very low-thresh-

old P-channel parts are also much

more costly and more difficult to

obtain than the more common N-

channel parts. The LTC1165 provides

a natural break-before-make action

and smooth transitions due to the

asymmetrical turn-on and turn-off

the open collector VALID pin low when

VPP is greater than 11V. Many

PCMCIA controllers require this sig-

nal for verification that flash memory

programming can proceed safely.

When switching the VPP signal to

12V, the output is clean and of mod-

erate rise time to prevent voltage

overshoot and subsequent damage to

flash memory parts.

The LT1312 also has a 200mA

nominal output current capability

with a 250mA short-circuit current

limit and thermal shutdown. These

protection features can be very im-

portant when considering the overall

reliability and robustness of the main

system. An additional feature of the

LT1312 is its low 25µA quiescent cur-

rent in 0V or high-impedance modes.

Figure 2. LT1312 Block diagram

Figure 3. Deriving 14V power from a 3.3V auxiliary winding

DESIGN FEATURES

–

LOW DROPOUT

LINEAR

REGULATOR

VOLTAGE

CONTROL

LOGIC

11V

–

VPPOUT

EN1

EN0

VCC SENSE

PCMCIA_2.eps

VALID

PCMCIA_3.eps

5.5V TO 14.5V

C1

22µF

C2

1000pF

C6

100µF

= AVX 22µF, 25V

TPSD226M025R0200

= AVX 100µF, 10V

TPSD107M010R0100

= MBRS140

= MBRS1100

C1,C3, C5

C6, C7

D1

= Si9430DY

= Si9410DY

= DALE LPE-6582-A086

DALE (605) 665-9301

Q1

Q2

C7

100µF

HC86

C3

22µF

Q3

VN7002

R2

100Ω

R3

12k

R1

100Ω

R4

22Ω

C4

1000pF

D1 D3

18V

14V

AUXILIARY

SUPPLY

3.3V

OUTPUT

T1

22µH

3.38T

D2 C5

22µF

1µF

TO SOCKET

VPP PINS

FROM SOCKET

PINS

PDRIVE

NDRIVE

SENSE+

LTC1148-3.3

OR

LTC1142

(3.3V REG)

VPP

OUT

SENSE

EN0

EN1

VALID

GND

LT1312

PCMCIA

CONTROLLER

SENSE–

R5

0.040Ω

of the MOSFET’s. A noninverting ver-

sion, the LTC1163, is available for

controllers with high asserted logic.

Auxiliary Winding

Power Supplies for Use

with the LT1312

Because the LT1312 provides ex-

cellent output regulation, the input

voltage may come from a loosely regu-

lated source. One convenient and

economic source of power is an aux-

iliary winding on the main switching

regulator inductor of the system power

supply.

LTC1142 or LTC1148-3.3

Auxiliary Winding Power

Supply for Low Input

Voltages

An auxiliary winding to the 3.3V

inductor of an LTC1142 or LTC1148-

3.3-based 3.3V power supply creates

a loosely regulated 14V power sup-

ply, as shown in Figure 3. Diode D2

rectifies the 11V output from this

additional winding, adding it to the

main 3.3V output. (Note the phasing

of the auxiliary winding, as shown in

the figure.) Referencing the auxiliary

winding to the main 3.3V output pro-

vides DC current feedback from the

auxiliary supply to the main 3.3V

section. Returning the lead of C5 to

the 3.3V output as shown improves

the AC transient response.

A TTL logic high on the enable line

(EN0) activates the 12V output. This

will force the 3.3V section of the

LTC1142 (or LTC1148-3.3) into the

continuous mode of operation. A re-

sistor divider, composed of R2, R3,

and switch Q3, forces an offset to

counteract the internal offset at the

−SENSE input of the part. Burst

Mode

operation ceases when this

external offset cancels the device’s

built-in 25mV offset, forcing the

switching regulator into continuous-

mode operation. (See the LTC1142

and LTC1148 data sheets for further

detail.) In this mode, the LT1312 out-

put can be loaded without regard to

loading on the 3.3V output of the

regulator.

Burst Mode is a trademark of Linear Technology

Corporaton

Linear Technology Magazine • October 1994

Table 2. LTC1314 truth table

EN0 EN1 VCC0 VCC1 VPPOUT DRV3 DRV5

00XXGNDXX

01XXV

CC IN XX

10XXVPP

IN XX

1 1 X X Hi-Z X X

XX 1 0 X 1 0

XX 0 1 X 0 1

XX 0 0 X 0 0

XX 1 1 X 0 0

X = Don’t care

LTC1314: V

switch driver

and VPP switch matrix

Figure 4 shows another approach

that is very space and power efficient.

Here the LTC1314 PCMCIA switch

matrix, used in conjunction with the

LT1301 DC-DC converter, provides

complete power management for a

PCMCIA card slot. The LTC1314 and

LT1301 combination provide a highly

efficient, minimal parts count solu-

tion. This circuit is especially useful

for designers who are adding PCMCIA

sockets to existing systems that cur-

rently have 5V or 3.3V available. Table

2 shows the truth table for the

LTC1314. A dual version of this part,

the LTC1315, is available in the 24-

lead SSOP package.

The LTC1314 directly drives three

inexpensive N-channel MOSFETs pro-

viding V

pin power switching. An

on-chip charge pump provides the

necessary voltage to fully enhance

the MOSFETs. With the built in charge

pump, the MOSFET drive is available

without the need for a constant 12V

supply. The LT1301 switching regu-

lator is normally in shutdown mode

and consumes only 10µA, except

during the brief flash memory-pro-

gramming cycles where VPP0 is set to

12V. This method results in low on-

resistance V

switching while main-

taining maximum efficiency.

The VPP switching is accomplished

by a combination of the LTC1314 and

LT1301 DC/DC converter. As stated

above, the LT1301 DC/DC converter

is in shutdown mode to conserve

power until the VPP pins request 12V.

When VPP pins require 12V,

the LT1301 is activated and the

LTC1314’s internal switches route

the VPP

pin to the VPP

OUT

pin. The

LT1301 is capable of delivering 12V

at 120mA while maintaining high

efficiency. The LTC1314’s break-

before-make and slope controlled

switching will control the output volt-

age transition to be smooth, of

moderate slope and without over-

shoot. This is critical for flash memory

products to prevent damaging parts

from overshoot and ringing exceeding

the 14V part limit.

The LTC1314 with the

LT1121 Linear Regulator

Often, systems have supply volt-

ages greater than 12V available. The

LTC1314, in conjunction with the

LT1121 linear regulator, supplies the

PC card socket with all necessary

voltages. Figure 5 shows this circuit,

where Q1 is used to invert the shut-

down logic signal. Any spare TTL

inverter can also be used. The auxil-

iary winding technique shown earlier

Figure 5. LTC1314 with the LT1121 linear regulator

continued on page 19

DESIGN FEATURES

Figure 4. LTC1314 switch matrix with the LT1301 boost regulator

VPP

OUT

VPP1

VPP2

SHDN VPP

CC(IN)

LTC1314

PC CARD

SOCKET

EN0

EN1 DRV5

DRV3

GND

CC0

PCMCIA_4.eps

CC1

3.3V

10µF

Q2A

Q2B

0.1µF

Q1A

1/2 Si9956DY

Si9956DY

SELECT SENSE

LT1301

L1

22µH

D1

MBRS130LT3

SHDN NCI

LIM

PGND GND

C2

33µF

C1

47µF

L1 = SUMIDA CD75-220K

C1 = AVX TPSD476M016R0150

C2 = AVX TPSD336M020R0200

SUMIDA (708) 956-0666

PCMCIA

CONTROLLER

PCMCIA_5.eps

VPP

OUT

VPP1

VPP2

SHDN VPP

2N7002

CC(IN)

LTC1314

PC CARD

SOCKET

EN0

EN1 DRV5

DRV3

GND

CC0

CC1

3.3V

10µF

Q2A

Q2B

0.1µF

Q1A

1/2 Si9956DY

Si9956DY

OUT

LT1121

SHDN

56.2k

121k

LIM

PGND GND

1µF

200pF

10µF

100k

13V TO 20V

(MAY BE FROM

AUXILIARY

WINDING)

PCMCIA

CONTROLLER

Linear Technology Magazine • October 1994

An Ultra-Selective Bandpass Filter with

Adjustable Gain

tions. One signal-detection applica-

tion occurs when two signals are very

closely spaced in the frequency spec-

trum and only one of the signals has

useful information. The LTC1164-8

can extract the signal of interest and

suppress its unwanted neighbor. For

example, a small 1kHz, 10mV

RMS

sig-

nal is combined with an unwanted

950Hz, 40mV

RMS

signal. The two sig-

nals differ in frequency by only five

center frequency. A capacitor, C

across resistor R

reduces clock

feedthrough and provides a smooth

sine-wave output.

Signal Detection

in a Hostile Environment

An outstanding feature of the

LTC1164-8 is its ultra-selectivity. A

bandpass filter with ultra-selectivity

is ideal for signal detection applica-

Figure 2. LTC1164-8 ultra-narrow, 1kHz bandpass filter with gain (gain = 340k/R

, 1/2πR

10 f

CENTER

Introduction

The LTC1164-8 is a monolithic,

ultra-selective, eighth order, elliptic

bandpass filter. The passband of the

LTC1164-8 is tuned with an external

clock; the clock-to-center-frequency

ratio is 100:1. The stopband attenu-

ation of the LTC1164-8 is greater

than 50dB for input frequencies out-

side a narrow band defined as ±4% of

the center frequency of the filter (see

Figure 1).

One Op Amp and Two

Resistors Build an

Ultra-Selective Filter

The LTC1164-8 requires an exter-

nal op amp and two external resistors.

The filter’s gain at its center frequency

is equal to 3.4R

. For optimum

dynamic range with a gain equal to

one, the external resistor R

should

be 90.9k and the external resistor R

should be 340k. For gains other than

one, R

= 340k/gain. Gains of up to

1000 are possible. The complete con-

figuration is shown in Figure 2. Note

that programming the filter’s gain

with input resistor R

is equivalent

to providing the LTC1164-8 with

noiseless preamplification, since the

filter’s internal noise is not amplified.

The wideband noise of the LTC1164-

8 measures 400µV

RMS

at ±5V and is

independent of the filter’s gain and

DESIGN FEATURES



FREQUENCY (kHz)

–80

–90

–70

–50

–60

–40

–20

–30

–10

GAIN (dB)

11648_1.eps

0.90 0.95 1.00 1.101.05

11648_2.eps

1

2

3

4

5

6

7



5V 100kHz

–5V

IN

340k

14

13

12

11

10

9

8



LTC1164-8



90.9k

F

160pF

–5V

–

LT1006 V

OUT

by Philip Karantzalis

Figure 1. LTC1164-8 Gain vs. frequency

response Figure 3. Narrow-band signal extraction showing input to and output from the LTC1164-8

filter. Filter center frequency set to 1kHz with gain = 100.

Linear Technology Magazine • October 1994

Figure 4. Signal detection in the presence of noise example showing input to and output from

the LTC1164-8 filter. Filter center frequency set to 1kHz with gain = 100.

percent and the 950Hz signal is four

times larger than the 1kHz signal. To

detect the 1kHz signal, the LTC1164-

8 is set to a gain of 100 and the clock

frequency is set to 100kHz. At the

filtered output of the LTC1164-8 the

following signals will be present: an

extracted 1kHz, 1V

RMS

signal and a

rejected 950Hz, 2.7mV

RMS

signal, as

shown in Figure 3. In a narrow-band

signal separation and extraction ap-

plication, as described previously, the

LTC1164-8 provides a simple and

reliable detection circuit solution.

A second signal-detection applica-

tion occurs when a small signal is to

be detected in the presence of noise.

For example, a 1kHz, 10mV

RMS

signal

is mixed with a wideband noise signal

that measures 5mV

RMS

in a 400Hz

frequency band. The signal-to-noise

ratio is just 6dB. With the LTC1164-

8 set for a center frequency of 1kHz

CLK

is equal to 100kHz) and a gain of

100, the 1kHz, 10mV

RMS

signal will

be detected and amplified. The wide-

band noise will be band-limited by

the very narrow band gain response

of the LTC1164-8. At the output of a

LTC1164-8, the 1kHz signal will be

RMS

, as shown in Figure 4. The

total band-limited noise will be

70mV

RMS

, with a signal-to-noise ra-

tio of more than 20dB, as shown in

Figure 5. In applications of signal

detection in the presence of noise, the

LTC1164-8 provides asynchronous

detection. Signal detection circuits

such as synchronous demodulators

and lock-in amplifiers require the

presence of a reference or carrier sig-

nal to provide phase and frequency

information of the signal to be de-

tected. With an LTC1164-8, signal

detection is accomplished by select-

ing a very narrow signal detection

band around the frequency of the

desired signal, which is defined as

CLK

divided by 100 (f

CLK

is the clock

frequency of the LTC1164-8), and by

selecting the filter gain by choosing

the value of a resistor. Figure 5. Wideband noise input to LTC1164-8 filter. Plots show input to and output from the

filter. Filter center frequency set to 1kHz with gain = 100.

DESIGN FEATURES

Linear Technology Magazine • October 1994

New LT1182/1183 Dual CCFL/LCD

Contrast Switching Regulator

by Anthony Bonte

Introduction

The current generation of portable

computers and instruments use back-

lit liquid-crystal displays (LCDs).

These displays also appear in other

applications, including medical

equipment, automobiles, gas pumps,

and retail terminals. Cold-cathode

fluorescent lamps (CCFLs) provide

the highest available efficiency for

backlighting the display. These lamps

require high voltage AC to

operate, mandating an efficient, high-

voltage DC–AC converter. In addition

to good efficiency, the converter

should deliver the lamp drive in the

form of a sine wave. This is desirable

to minimize RF emissions. Such emis-

sions can cause interference with

other devices, and can also degrade

overall operating efficiency. The sine-

wave excitation also provides optimal

current-to-light conversion in the

lamp. The circuit should also permit

lamp intensity control from zero to

full brightness with no hysteresis or

“pop-on.”

The LCD also requires a bias sup-

ply for contrast control. The supply’s

output should be regulated, and vari-

able over a considerable range.

Manufacturers now offer an array of

either positive or negative contrast-

voltage displays. These displays differ

in operating-voltage range, contrast-

adjust range and power consumption.

The small size and battery-pow-

ered operation associated with

LCD-equipped apparatus mandate

low component count and high effi-

ciency. Size constraints place severe

limitations on circuit architecture,

and long battery life is usually a pri-

ority. Laptop and hand-held portable

computers offer an excellent example.

The CCFL and its power supply are

responsible for almost 50% of the

battery drain. Displays found in newer

color machines can have contrast

power-supply battery drains as high

as 20%. Additionally, all components,

including PC board and hardware,

usually must fit within the LCD

enclosure with a height restriction

of 0.25".

New LTC Parts

Linear Technology has invested an

enormous amount of time, resources,

and technical capability in providing

the premier backlight/contrast solu-

tions for system designers. In

continuing this commitment, Linear

Technology introduces the LT1182/

LT1183. These new devices achieve a

high level of system integration for a

backlight/contrast solution. In addi-

tion, they reduce system power

dissipation, maintain high efficiency

operation, require fewer external com-

ponents, and reduce overall system

cost.

The LT1182/LT1183 are

dual, fixed-frequency, current-mode

switching regulators that provide the

control function for cold-cathode fluo-

rescent lighting and liquid-crystal

display contrast. Two high-current,

high-efficiency switches are included

on the die, along with an oscillator,

reference, output-drive logic, control

blocks, and protection circuitry. Sepa-

rate analog and power grounds for

the regulators minimize interaction.

Combining both control functions

onto one die and maximizing perfor-

mance allows the IC to fit into a

narrow-body 16-pin SOIC. Board

space and insertion cost are reduced

in comparison with devices that re-

quire two independent regulators.

Also, external components required

with previous solutions have been

integrated onto the IC, providing a

more economical solution. The

LT1184 is also available and provides

only the CCFL function.

The LT1182/LT1183 operate with

supply voltages from 3V to 30V and

draw only 9mA quiescent current. An

active low shutdown pin reduces total

supply current to less than 50µA for

standby operation. A 200kHz switch-

ing frequency minimizes the size of

required magnetic components. The

use of current-mode switching tech-

niques with cycle-by-cycle limiting gives

high reliability and simple loop fre-

quency compensation.

The CCFL switching regulator typi-

cally drives an inductor that acts as a

switched-mode current source for a

current-driven Royer-class converter

with efficiencies as high as 90%. The

control loop forces the regulator to

pulse-width modulate the inductor’s

average current to maintain constant

current in the lamp. The constant

current’s value, and thus lamp inten-

sity, is programmable. This drive

technique provides a wide range of

intensity control. A unique lamp-

current programming block permits

either grounded-lamp or floating-

lamp configurations. Grounded-lamp

circuits directly control one-half of ac-

tual lamp current. Floating-lamp

circuits directly control the Royer’s

primary-side converter current.

Floating-lamp circuits provide differ-

ential drive to the lamp and reduce the

loss from stray lamp-to-frame capaci-

tance, extending illumination range.

The LCD contrast switching regula-

tor is typically configured as a flyback

converter and generates a bias supply

for contrast control. The supply’s vari-

able output permits adjustment of

contrast for the majority of available

displays. Newer types of displays re-

quire a constant supply voltage and

provide contrast adjustment through a

separate digital control pin. A unique,

dual-polarity error amplifier and the

selection of a flyback converter topol-

ogy allow either positive or negative

LCD-contrast voltages to be generated

with minor circuit changes. The LT1182

and LT1183 differ in their pinouts for

the LCD-contrast error amplifier. The

LT1182 brings out the individual

DESIGN FEATURES

Linear Technology Magazine • October 1994

Figure 1. LT1182/LT1183 CCFL/LCD contrast top-level block diagram

error-amplifier inputs for setting up

positive- and negative-polarity con-

trast capability, whereas the LT1183

ties the error-amplifier inputs to-

gether and brings out an internal

reference. The reference may be used

in generating negative contrast volt-

ages or in programming lamp current.

Operation

Fixed-frequency, current-mode

switchers control switch duty cycle

directly by switch current rather than

by output voltage. Referring to the

block diagram for the LT1182/LT1183

in Figure 1, the switch for each regu-

lator is turned ON at the start of each

oscillator cycle. The switches are

turned OFF when switch current

reaches a predetermined level. The

control of output lamp current is

obtained by using the output of a

unique programming block to set

current trip level. The contrast volt-

age is controlled by the output of a

dual-input-stage error amplifier,

which sets current trip level. The cur-

rent-mode switching technique has

several advantages. First, it provides

excellent rejection of input voltage

variations. Second, it reduces the 90°

phase shift at mid-frequencies in

the energy storage inductor. This

simplifies closed-loop frequency com-

pensation under widely varying

input-voltage or output-load condi-

tions. Finally, it allows simple,

pulse-by-pulse current limiting to

provide maximum switch protection

under output overload or short-cir-

cuit conditions.

The LT1182/LT1183 incorporate

a low-dropout internal regulator that

provides a 2.4V supply for most of the

internal circuitry. This low-dropout

design allows input voltage to vary

from 3V to 30V with little change in

quiescent current. An active low shut-

down pin reduces total supply current

to less than 50µA and locks out

switching action for standby opera-

tion. The LT1182/LT1183 incorporate

undervoltage protection by sensing

regulator dropout and locking out

switching below about 2.5V. The regu-

lator also provides thermal-shutdown

protection and locks out switching in

the presence of excessive junction

temperatures.

A 200kHz oscillator is the basic

clock for all internal timing. The oscil-

lator turns on each output switch by

means of its own logic and driver

circuitry. Adaptive anti-saturation

circuitry detects the onset of satura-

tion in each power switch and adjusts

base-drive current instantaneously

to limit switch saturation. This mini-

mizes driver dissipation and provides

rapid turn-off of the switch. The CCFL

power switch is guaranteed to pro-

vide a minimum of 1.25A and the LCD

power switch is guaranteed to pro-

vide a minimum of 0.625A. The

anti-saturation circuitry provides a

DESIGN FEATURES

FBN

–

LIM



AMP2

COMP2 COMP1

UNDER-

VOLTAGE

LOCKOUT

THERMAL

SHUTDOWN

2.4V

REGULATOR

SHUTDOWN

200kHz

OSC

DRIVE 2

SHUTDOWN

LCD

PGND LCD

VC FBP ICCFL AGND DIO BULB CCFL

VC CCFL

PGND

CCFL

ROYERBAT

LOGIC 2

ANTI-

SAT2

LOGIC 1 DRIVE 1

LCD

GAIN = 4.4

Q3

2 ×

Q5

1 ×

Q4

5 × Q8

1 ×

Q10

2 ×

R1

0.125Ω

R4

0.1Ω

LT1183: FBP AND FBN ARE TIED TOGETHER TO PIN 10

REFERENCE IS BROUGHT OUT TO PIN 11



R2

0.25Ω

Q6

2 ×

R3

D2

Q11

Q7

9 ×

Q9

3 ×

8711 25

15 4

V2

1.24V V1

0.45V

–

CCFL

0µA TO 100µA

–10mV

–+

––

–

LIM



AMP1

ANTI-

SAT1

1182_1.eps

–

Linear Technology Magazine • October 1994

ratio of switch current to driver cur-

rent of about 40:1.

Simplified Lamp-

Current Programming

A programming block in the

LT1182/LT1183 controls lamp

current, permitting either grounded-

lamp or floating-lamp configurations.

Grounded configurations control

lamp current by directly controlling

one-half of actual lamp current and

converting it to a feedback signal to

close a control loop. Floating configu-

rations control lamp current by

directly controlling the Royer’s pri-

mary-side converter current and

generating a feedback signal to close

a control loop.

Previous backlighting solutions

have used a traditional error ampli-

fier in the control loop to regulate

lamp current. This approach con-

verted an RMS current into a DC

voltage for the input of the error am-

plifier. This approach used several

time constants to provide stable loop

frequency compensation. This com-

pensation scheme meant that the loop

had to be fairly slow and that output

overshoot with start-up or overload

conditions had to be carefully evalu-

ated in terms of transformer stress

and breakdown voltage requirements.

The LT1182/LT1183 eliminate the

error-amplifier concept entirely and

replace it with a novel lamp-current

programming block. This block pro-

vides an easy-to-use interface to

program lamp current. The program-

mer circuit also reduces the number

of time constants in the control loop

by combining the error-signal con-

version scheme and frequency

compensation into a single capacitor.

The control loop thus exhibits the

response of a single-pole system, al-

lows for faster loop transient response,

and virtually eliminates overshoot

under start-up or overload conditions.

Finally, these parts include open-

lamp protection circuitry, with user

control by means of a simple, exter-

nal RC network. This significantly

eases the breakdown requirements

for the transformer and lowers the

cost associated with winding a high-

voltage transformer.

Lamp current is programmed at

the input of the programmer block,

the I

CCFL

pin. This pin is internally

regulated to 450mV and accepts a DC

input current signal of 0–100µA. This

input signal is converted to a 0–500µA

source current at the CCFL V

pin.

By regulating the I

CCFL

pin, the input

programming current can be set with

DAC, PWM, or potentiometer control.

In a grounded-lamp configuration,

the low-voltage side of the lamp con-

nects directly to the LT1182/LT1183

DIO pin. This pin is the common

connection between the cathode and

anode of two internal diodes. In pre-

vious grounded-lamp solutions, these

diodes were discrete units; in the

LT1182/LT1183, they are integrated

onto the IC. Bidirectional lamp cur-

rent flows in the DIO pin and thus the

diodes conduct on alternate half

cycles. Lamp current is controlled by

monitoring one-half of the lamp cur-

rent. The diode conducting on

negative half cycles has one-tenth of

its current diverted to the CCFL pin

and nulls against the source current

provided by the lamp-current pro-

grammer circuit. The compensation

capacitor on the CCFL V

pin pro-

vides both loop compensation and an

averaging function to the rectified

sinusoidal lamp current. Therefore,

input programming current is related

to one-half of average lamp current. If

a floating-lamp configuration is used,

the DIO pin is grounded.

In a floating-lamp configuration,

the lamp is fully floating with no

galvanic connection to ground. This

allows the transformer to provide sym-

metric, differential drive to the lamp.

Balanced drive eliminates the field

imbalance associated with parasitic

lamp-to-frame capacitance and re-

duces “thermometering” (uneven

lamp intensity along the lamp length)

at low lamp currents. Display de-

signs should be carefully evaluated

in relation to construction shape,

materials, and asymmetric lamp

wiring, as energy leakage terms de-

grading efficiency up to 20% have

been noted in practice. Maintaining

closed-loop control of lamp current

now necessitates deriving a feedback

signal from the primary side of the

Royer transformer. Previous solutions

have used an external precision shunt

and high-side sense amplifier con-

figuration. This approach has been

integrated onto the LT1182/LT1183

for simplicity of design and ease of

use. Primary-side Royer-converter

current is related to lamp current by

the turns ratio of the transformer

and its reflected impedances. The

Royer-converter current is monitored

across an internal 0.1Ω resistor,

which is connected across the input

terminals of a high-side sense amp-

lifier. The input terminals are

represented by the BAT and Royer

pins. A 0–1A Royer primary-side, cen-

ter-tap current is translated to a

0–500µA sink current at the CCFL V

pin to null against the current source

provided by the programmer circuit.

Once again, the compensation ca-

pacitor on the CCFL V

pin provides

both loop compensation and an aver-

aging function to the error sink

current; therefore, input-program-

ming current is related to average

Royer-converter current. Floating-

lamp circuits operate similarly to

grounded-lamp circuits, except for

the derivation of the feedback signal.

However, floating-lamp circuits

permit the lamp to operate

over a 40:1 intensity range with-

out “thermometering,” whereas

grounded-lamp circuits are usually

limited to a 10:1 range.

Open-lamp protection is provided

by an internal, 7V-threshold com-

parator connected between the BAT

and Lamp pins. This circuit sets a

maximum voltage level across the

primary side of the Royer converter

and limits the maximum transformer

output voltage under start-up or

open-lamp conditions. This eases

transformer voltage-rating require-

ments. The Lamp pin is connected to

the junction of an external resistor

divider network. The divider network

connects from the center tap of the

Royer transformer to the top side of

the Royer inductor. A capacitor across

the top of the divider network filters

DESIGN FEATURES

Linear Technology Magazine • October 1994

C11

2.2µF

35V

BAT

8V TO 28V

POSCON

NEGCON

1182_2.eps

EITHER NEGCON OR POSCON 

MUST BE GROUNDED.

GROUNDING NEGCON GIVES

15V FIXED. GROUNDING

POSCON GIVES VARIABLE

NEGATIVE CONTRAST FROM

–10V TO –30V.

1

2

3

4

5

6

7

16

15

14

13

12

11

10

CCFL



DIO

CCFL VC

AGND

SHDN

LCD VC

CCFL V



BULB

BAT

ROYER



FBP

FBN

LCD V

CCFL

PGND

LCD

PGND

LT1182

LAMP

UP TO 6mA

10 6

R1

750Ω

L2

100µH

1235 4

ALUMINUM ELECTROLYTIC IS RECOMMENDED FOR C3B WITH AN 

ESR ≥ 0.5Ω TO PREVENT LT1182 HIGH-SIDE SENSE RESISTOR 

DAMAGE DUE TO SURGE CURRENTS AT TURN-ON.



C7, 1µF

C1*

0.033µF

D1

1N5818 D2

1N914

D4

1N914

V (CCFL)

0V TO 5V kHz PWM

C6

4.7µF

R4

34k

SHUTDOWN

C8

1µF

R7, 1k

R5, 40.2k, 1%

C5

1000pF R2

220k

R3

100k

C3A

2.2µF

35V

C10

10µF

35V

R11

54.9k

R12

4.99k

V (CONTRAST)

0V TO 5V

R10, 120k, 1%

R9, 20k, 1%

C4

2.2µF

IN

≥ 3V

D3

1N5934A

24V

C3B

2.2µF

35V

C2

27pF

3kV

C1 MUST BE A LOW LOSS CAPACITOR, C1 = WIMA MKP-20

Q1, Q2 = ZETEX ZTX849 OR ROHM 2SC5001

L1 = SUMIDA EPS-207 OR COILTRONICS

CTX110605. PIN NUMBERS SHOWN FOR 

COILTRONICS UNIT (C1 VALUE MAY REQUIRE 

ADJUSTMENT WITH COILTRONICS).

L2 = COILTRONICS CTX100-4

L3 = COILTRONICS CTX02-12403

*DO NOT SUBSTITUTE COMPONENTS

COILTRONICS (407) 241-7876

SUMIDA (708) 956-0666

Q2* Q1*

N = 1:2



NOTE: SET V (CONTRAST) TO 5V IF ONLY POSITIVE CONTRAST VOLTAGES ARE GENERATED.

0µA TO 50µA I

CCFL



CURRENT GIVES ≈

0mA TO 6mA BULB

CURRENT.

out switching ripple and sets a time

constant that determines how quickly

the clamp activates. When the com-

parator activates, current is generated

to pull the CCFL V

pin down. This

action transfers the entire regulator

loop from current-mode operation

into voltage-mode operation.

Dual-Polarity

Contrast Capability

The LCD contrast regulator can be

operated in many standard switching

configurations. A dual-input-stage

error amplifier can regulate either

positive or negative contrast voltages.

In fact, if a flyback configuration is

chosen, divider networks can be

implemented for both polarities of

contrast. Output polarity is then de-

termined by which side of the

transformer secondary the output

connector grounds. If only positive

contrast voltages are to be generated,

one can also use a boost converter

topology for higher efficiency

operation. The FBN pin is the nonin-

verting terminal for the negative

Figure 2. 90% Efficient floating CCFL configuration with dual-polarity LCD contrast

contrast-control error amplifier. The

inverting terminal is offset from

ground by −10mV to define the error-

amplifier output state under startup

conditions. The FBN pin acts as a

summing junction for the resistor

divider network. FBN input bias cur-

rent is typically −1µA. The FBP pin is

the inverting terminal for the positive

contrast-control error amplifier. The

noninverting terminal is tied to an

internal 1.24V reference. FBP input

bias current is typically 0.5µA. The

LCD V

pin is the high-impedance

current output (gm) for the contrast

error amplifier. A series RC network

to ground typically performs loop fre-

quency compensation.

Floating CCFL with

Variable-Negative/Fixed-

Positive Contrast

Figure 2 is a complete floating CCFL

circuit with variable-negative/fixed-

positive contrast voltage generation

based on the LT1182. Q1, Q2, L1, and

C1 form the core of the Royer con-

verter. The LT1182 CCFL V

pin and

L2 form the switch-mode current

source that drives the Royer. The

200kHz switching frequency allows

the use of only 100µH for L2. Hookup

for a floating-lamp configuration is

as simple as placing the high-side

sense resistor in series with the cen-

ter tap of the Royer transformer and

floating the lamp on the transformer

secondary.

The feedback signal generated by

the Royer-converter current is con-

verted to an error current at the CCFL

pin. This current is then averaged

by compensation capacitor C7 to null

against the programming current

source. C7’s value of 1µF provides

stable loop compensation and excel-

lent transient response under startup

conditions. It is important to note

that the catch diode D1 is returned to

the BAT side of the high-side sense

resistor. This connection maintains

current flow in L2 when the switch is

off, improves operating efficiency by

returning current to the input sup-

ply, and ensures that the feedback

signal is generated only by the Royer

converter current. The transfer func-

tion between primary-side converter

current and programming current

DESIGN FEATURES

Linear Technology Magazine • October 1994

DESIGN FEATURES

continued on page 22

C11

2.2µF

35V

BAT

8V TO 28V

POSCON

NEGCON

1182_3.eps

1

2

3

4

5

6

7

16

15

14

13

12

11

10

CCFL



DIO

CCFL VC

AGND

SHDN

LCD VC

CCFL V



BULB

BAT

ROYER



FBP

FBN

LCD V

CCFL

PGND

LCD

PGND

LT1182

LAMP

UP TO 6mA

210

R1

750Ω

L2

100µH

12354

C7

1µF

C1*

0.033µF

D1

1N5818 D2

1N914

D4

1N914

N = 1:2



V (CCFL)

0V TO 5V

SHUTDOWN

C8

1µF

R7, 1k

R5

84.5k

C5

1000pF R2

220k

R3

100k

C2

27pF

3kV

C10

10µF

35V

R11

54.9k

R12

4.99k

V (CONTRAST)

0V TO 5V R10

120k

R9

20k

C4

2.2µF

IN

≥ 3V

D3

1N5934A

24V

C3

4.7µF

35V

MUST BE A LOW LOSS CAPACITOR

WIMA MKP-20

ZETEX ZTX849 OR ROHM 2SC5001

SUMIDA EPS-207 OR

COILTRONICS CTX110602

PIN NUMBERS SHOWN FOR 

COILTRONICS UNIT

CTX100-4

COILTRONICS CTX02-12403

C1 

C1 = 

Q1, Q2 = 

L1 = 

 



L2 = 

L3 =

Q2* Q1*

NOTE: SET V (CONTRAST) TO 5V IF ONLY POSITIVE CONTRAST

VOLTAGES ARE GENERATED.

0µA TO 50µA I

CCFL



CURRENT GIVES ≈

0mA TO 6mA BULB

CURRENT.

EITHER NEGCON OR POSCON

MUST BE GROUNDED.

GROUNDING NEGCON GIVES

15V FIXED. GROUNDING

POSCON GIVES VARIABLE

NEGATIVE CONTRAST FROM

–10V TO –30V.

*DO NOT SUBSTITUTE COMPONENTS

COILTRONICS (407) 241-7876

SUMIDA (708) 956-0666

THE I

CCFL

CURRENT REQUIRED FOR A

GIVEN RMS BULB CURRENT IS:

CCFL

≈ (9 × 10

–3

)(I

BULB

)

must be empirically determined and

is dependent upon a myriad of factors

such as lamp characteristics, display

construction (which dominates radi-

ated loss terms), transformer turns

ratio, and the tuning of the Royer

oscillator.

The method used to generate the

input programming current is shown

in Figure 2. A 1kHz, 0–5V pulse-

width-modulated logic signal is

converted to DC by the R4/C6 filter

network. R5 then converts the DC

voltage into programming current at

the I

CCFL

pin. 0µA to 50µA I

CCFL

cur-

rent produces 0mA to 6mA lamp

current.

R2, R3, and C5 form the external

divider network for the open-lamp

protection circuit. The divider moni-

tors the voltage across the Royer

converter and compares it with the

internal clamp voltage. In this ex-

ample, the divider is set to limit the

maximum Royer primary voltage to

about 10V. Capacitor C5 filters out

ripple at the emitters of Q1 and Q2,

which have a ripple component of

twice the Royer frequency.

Efficiency with typical CCFL lamps

ranges from 85–90% at full load, with

lamp current being variable from

200µA to 6mA. Higher electrical effi-

ciencies are obtainable by increasing

the harmonic content in the wave-

forms (lowering the value of C1) and

increasing the magnetics size to de-

crease copper loss. The component

values chosen were an interactive

compromise in maximizing photomet-

ric output versus input power. Base

drive for Q1 and Q2 is provided by

the tickler winding and the value of

R1. R1’s value must be chosen to

guarantee sufficient base drive with

minimum betas for Q1 and Q2. The

battery supply range can extend from

8V to 28V, with a nominal battery

voltage of 12V. The LT1182 is typi-

cally powered from a 5V or 3.3V logic

supply. The LT1182 can also be

powered from the battery supply, but

Figure 3. 90% Efficient grounded CCFL configuration with dual-polarity LCD contrast

Linear Technology Magazine • October 1994

Introduction

The LT1366 family is Linear

Technology’s first series of dual and

quad bipolar operational amplifiers

to combine rail-to-rail input and out-

put specifications with precision V

specs. The LT1366 family maintains

precision specifications over a wide

range of operating conditions. The

devices will operate with supply volt-

ages as low as +1.8V, and are fully

specified for +3V, +5V, and ±15V op-

eration. Offset voltage is typically

150µV when operating from a single

5V supply. Supply rejection is 115dB.

Open loop gain, A

VOL

, is 2 million

driving a 2k load. Common-mode re-

jection ratio is typically 86dB over the

full rail-to-rail input range. The com-

bination of precision specifications

and rail-to-rail operation makes the

LT1366 series versatile amplifiers,

suitable for signal processing tasks

that demand the widest possible

common-mode range.

The amplifiers are available in two

versions, which differ in their ability

to drive capacitive loads. The LT1366

dual and LT1367 quad have conven-

put stages. Figure 1 shows a simpli-

fied schematic of the LT1366. The

input stage consists of two differen-

tial amplifiers, a PNP stage Q1–Q2

and an NPN stage Q3–Q4, which are

active over different portions of

the input common-mode range. Lat-

eral devices are used in both input

stages, eliminating the need

for clamps across the input pins.

Each input stage is trimmed for offset

voltage. A complementary output con-

figuration (Q23–26) is employed to

create an output stage with rail-to-

rail swing. The device is fabricated on

Linear Technology’s proprietary

complementary bipolar process,

which ensures very similar DC and

AC characteristics for the output de-

vices Q24 and Q26.

Since two separate input stages

are used to obtain the rail-to-rail in-

put range, there are also two sets of

input specifications for the device.

Linear Technology fully specifies the

input characteristics for each stage

and the difference in each parameter

between stages. The specification for

tional compensation, and are stable

with load capacitances of 1000pF or

less. For use in low-frequency or DC

applications, the LT1368 dual and

LT1369 quad are compensated for

use with a 0.1µF capacitor at the

output. In a noisy environment, the

large output capacitor cleans up the

output signal by improving the sup-

ply rejection and providing a low

output impedance at high

frequencies.

The LT1366/LT1368 dual

amplifiers are available with indus-

try-standard pin-out in either 8-pin

SO or 8-pin mini-DIP packages. The

LT1367/LT1369 quad amplifiers are

available in the 16-pin narrow (150

mil width) SO16. The pinout for the

SO16 is the same as the industry

standard, but with two end pins (8

and 9) left open.

Obtaining

Rail-to-Rail Operation

As might be expected, the LT1366

differs from a conventional op amp in

the design of both the input and out-

Figure 1. LT1366 Simplified schematic diagram

DESIGN FEATURES

1366_1.eps

S

–300mV)

IN–

V–

IN+V–

V–V–

Q12

Q4 Q7

Q13 Q18 Q19

Q20 Q25

Q26

Q24

Q10

V+

Q3 Q15Q14

Q2Q1

Q5 Q16

Q17

Q23

OUT

Q22

Q21

Q11

D4 D5

D6 D7

V–

V+V+

V+

The LT1366 Family: Precision,

Rail-to-Rail Bipolar Amplifiers by William Jett

and Sean Gold

Linear Technology Magazine • October 1994

Two circuits prevent the output

from reversing polarity when the

input voltage exceeds the common-

mode range. When the noninverting

input exceeds the positive supply by

approximately 300mV, device Q12

turns on, pulling the output of the

second stage low, which forces the

output high. For inputs below the

negative supply, diodes D1–D2 turn

on, overcoming the saturation of the

input pair Q1–Q2.

Improved Supply

Rejection—the LT1368

The LT1368 is a variation of the

LT1366 offering greater supply rejec-

tion and lower high-frequency output

impedance. The LT1368 requires a

0.1µF load capacitance for compen-

sation. The output capacitance forms

a filter, which reduces pickup from

the supply and lowers the output

impedance. This additional filtering

is helpful in mixed analog/digital

systems with common supplies, or in

systems employing switching

supplies.

Figures 2a and 2b show the out-

puts of the LT1366 and the LT1368

with a 200mV

P–P

, 100kHz square wave

added to the positive supply. Note

that the signal on the output of the

LT1368 is only 20mV

P–P

Performance

Table 1 summarizes the guaran-

teed DC performance of the LT1366.

As mentioned before, the device is

fully tested and guaranteed for +3V,

+5V, and ±15V operation. In addition

to the traditional op amp specs, Lin-

ear Technology also guarantees the

input bias-current shift properly ac-

counts for the opposite directions of

the input current in the PNP and NPN

stages.

First, looking at the input stage,

Q5 switches the current from current

source I1 between the two input

stages. When the input common mode

voltage V

is near the negative sup-

ply, Q5 is reverse biased, so the

current from I1 becomes the tail cur-

rent for the PNP differential pair

Q1–Q2. At the other extreme, when

is near the positive supply, the

PNPs Q1–Q2 are biased off. The cur-

rent from I1 then flows through Q5 to

the current mirror D3–Q6, furnish-

ing the tail current for the NPN

differential pair Q3–Q4. The switch-

over point between stages occurs when

is equal to the base voltage of Q5,

which is biased approximately 1.3V

below the positive supply.

The collector currents of the two

input pairs are combined in the sec-

ond stage, consisting of Q7–Q11. Most

of the voltage gain in the amplifier is

contained in this stage. Differential

amplifier Q14–Q15 buffers the out-

put of the second stage, converting

the output voltage to differential cur-

rents. The differential currents pass

through current mirrors D4–Q16 and

D5–Q17, and are converted to differ-

ential voltages by Q18 and Q19. These

voltages are also buffered and ap-

plied to the output Darlington pairs

Q23–24 and Q25–26. Capacitors C1

and C2 form local feedback loops

around the output devices, lowering

the output impedance at high

frequencies.

DESIGN FEATURES

Figure 2a. LT1366 Power-supply rejection test Figure 2b. LT1368 power-supply rejection test

maximum shift in offset voltage and

input currents between the NPN and

PNP input stages.

Applications

The LT1366 series opens up a new

range of possibilities for low-power

and rail-to-rail applications. The fol-

lowing circuits demonstrate the

LT1366’s versatility and precision in

solving problems ranging from simple

buffers to continuous-time active

filters.

Potentiometer Buffer

It is often attractive in automotive

and position-sensing applications to

connect a potentiometer directly

across a reference supply. Buffering

a potentiometer connected in this way

requires an op amp with rail-to-rail

input and output swing (Figure 3).

Despite the circuit’s simplicity, care

must be taken in component selec-

tion. The potentiometer should

present a load commensurate with

the op amp’s supply current, yet be

small enough to assure negligible

input-bias-current induced offset er-

rors. In the example shown, a 10kΩ

potentiometer draws 500µA from the

input supply and causes less than

50µV of error due to bias current. The

offset-voltage error changes with the

potentiometer setting, with the maxi-

mum error at mid-scale and the

minimum error near the supply rails.

When the op amp’s output satu-

rates, the offset-voltage error becomes

the difference between the input sig-

nal and the saturation voltage. The

saturation voltage as a function of

Figure 3. Rail-to-rail potentiometer buffer

–

1366_3.eps

P

10k

L



1/2 LT1366

Linear Technology Magazine • October 1994

Table 1. LT1366 guaranteed DC performance—25°C

VS = 3V VS = 5V VS = ±15V

Max offset voltage VCM = VCC 450µV 450µV 650µV

= VEE 450µV 450µV 650µV

Max input bias current VCM = VCC 35nA 35nA 35nA

= VEE 35nA 35nA 35nA

Max input offset current VCM = VCC 6nA 6nA 6nA

= VEE 6nA 6nA 6nA

Max offset voltage shift over CMR 400µV 400µV 500µV

Max input bias current shift 70nA 70nA 70nA

Max input offset current shift 6nA 6nA 6nA

Min open-loop gain (RL = 2k) 500k 500k 2000k

Min channel separation 120dB

Max output voltage—LOW no load 12mV 12mV 12mV

ISINK = 2.5mA 200mV 200mV 200mV

Max output voltage—HIGH no load 8mV 8mV 8mV

ISOURCE = 2.5mA 250mV 250mV 250mV

Min output current ±10mA ±15mA ±30mA

Max supply current per amp 500µA 500µA 550µA

load current is shown in Figure 4. For

light loads, such as an A/D converter,

the op amp can swing to within 5mV

of each rail. At 1mA of load current,

the offset voltage error is less than

75mv; at 10mA of load current, it is

less than 500mV.

Topside Current Source

The circuit shown in Figure 5 takes

advantage of the LT1366’s rail-to-rail

input range to form a wide-compli-

ance current source. The LT1366

adjusts Q1’s gate voltage to force the

voltage across the sense resistor

SENSE

) to equal the voltage from the

DESIGN FEATURES

In this example, the circuit deliv-

ers 1A at 200mV of sense voltage.

With a 5V input supply the power

dissipation is 5W. For operation at

70°C ambient temperature, the

MOSFET’s heat sink must have a

thermal resistance of:

= θ

JA SYSTEM

- θ

JC FET

= 55°C/5W - 1.25°C/W

= 9.75°C/W.

This is easily achievable with a

small heat sink. When input voltages

are greater than 5V, the use of a

larger heat sink or derating of the

output current is necessary.

The circuit’s supply regulation is

about 0.03%/V. The output imped-

ance is equal to the MOSFET’s output

impedance multiplied by the op amp’s

open-loop gain. Degradations in cur-

rent-source compliance occur when

the voltage across the MOSFET’s on-

resistance and the sense resistor

drops below the voltage required to

maintain the desired output current.

This condition occurs when V

− V

OUT

< I

LOAD

× [R

SENSE

+ R

High Side Current-

Sense Amplifier

In power control, it is sometimes

necessary to sense load current at

low loss near the input supply. The

current-sense amplifier shown in Fig-

ure 6 amplifies the voltage across a

small value sense resistor by the ratio

of the current-source resistors (R2/

R1). The LT1366 forces the low-power

MOSFET’s gate voltage such that the

sense voltage appears across a cur-

rent-source resistor R1. The resulting

current in Q1’s drain is converted to

Figure 6. High side current-sense amplifier

–

VIR

OINS

=









=•

20Ω

1366_6.eps

S

0.2Ω



200Ω



TPO610L

1/2 LT1366



20k

supply to the potentiometer’s wiper. A

rail-to-rail op amp is needed because

the voltage across the sense resistor

must drop to zero when the divided

reference voltage is set to zero. Q2

acts as a constant current sink to

minimize error in the reference volt-

age when the supply voltage varies.

The circuit can operate over a wide

supply range (5V < V

< 30V). At low

input voltage, circuit operation is lim-

ited by the MOSFET’s gate-drive

requirements. At high input voltage,

circuit operation is limited by the

LT1366’s absolute maximum ratings

and the output power requirements.

–

1366_5.eps

Q2

2N4340

5V < V

< 30V

1A > I

LOAD

> 160mA



10k

40k

LT1004

-1.2

SENSE



0.2Ω

100Ω

1/2 LT1366

0.0033µF

Q1

MTP23P06

OUT

Figure 4. LT1366 V

SAT

vs. load current

LOAD CURRENT (mA)

SATURATION VOLTAGE (mV)

100

0.001 0.1 1 10

1366_09.eps

10.01

1000

TOP SIDE

BOTTOM

SIDE

Figure 5. Topside current source

Linear Technology Magazine • October 1994

–

3.3V

OUT

R1

29.5k* R2

8.6k*

29.5k*

20k 13k

* = 1% RESISTORS

1µF

A1

1/4 LT1367

C1

10,000pF

11.8k*

A2

1/4 LT1367

C2

10,000pF

21.5k*

11.8k*

A3

1/4 LT1367

10,000pF

SECTION 2SECTION 1

A4

1/4 LT1367

10,000pF

1366_7.eps

DESIGN FEATURES

a ground-referred voltage at R2. (V

[R2/R1])

The circuit takes advantage of the

LT1366’s ability to sense signals up

to the supply rail, which permits the

use of small-value, low-loss sense

resistors. The LT1366 and the gain

setting resistors are also biased at

low current to reduce losses in the

current sense.

Single-Supply, 1kHz, Fourth-

Order Butterworth Filter

The circuit shown in Figure 7 takes

advantage of all four op amps in the

LT1367 to form a fourth-order But-

terworth filter. The filter is a simplified

state-variable architecture consist-

ing of two cascaded second-order

sections. Each section uses the 360

degree phase shift around the two op

amp loop to create a negative sum-

ming junction at A1’s positive input.

The circuit has two-thirds the power

dissipation and component count as

the classic three op amp biquad,

yet

it has the same low component sensi-

tivities for center frequency, w

and Q.

For cutoff frequencies other than

the 1kHz example shown, use the

following formula for each section:

= 1/(R1 C1 R2 C2),

where R1 = 1/(w

Q C1) and R2 = Q/(w

C2)

Figure 7. 1kHz fourth-order Butterworth filter

Figure 8. Frequency response of fourth-order Butterworth filter

The DC bias applied to A2 and A4

for single-supply operation is not

needed when split supplies are avail-

able. The circuit’s output can swing

rail-to-rail and displays the maxi-

mally flat amplitude response with a

1kHz cutoff frequency with 80dB/

decade rolloff (Figure 8).

References:

1. Hahn, James. 1982. State Variable Filter Trims

Predecessor’s Component Count. Electronics,

April 21, 1982.

2. Thomas, L.C. 1971. The Biquad: Part I—Some

Practical Design Considerations. IEEE Transac-

tions on Circuit Theory, 3:350–357, May 1971.

Linear Technology Magazine • October 1994

LTC1325 Battery Management System

Offers Unparalleled Flexibility

all the functional blocks needed to

implement a simple but sophisticated

battery charger (see Figure 1). The

main features of the LTC1325 may be

summarized as follows:

❏It has all the functional blocks

needed to build a charger,

including a 10-bit ADC, fault-

detection circuitry, a switching

buck regulator controller, a P-

channel MOSFET driver, a timer,

a 3V regulator for powering

external temperature sensors,

and a programmable battery

divider.

❏The functional blocks are placed

under the control of an external

microprocessor for maximum

flexibility and adaptability to

different battery chemistries or

charge rates.

❏Communication with the micro-

processor is via an easy-to-use,

four-wire serial interface.

❏It has stand-alone fault detection

circuitry to protect the battery

against temperature or voltage

extremes.

❏In addition to charging, the chip

can discharge batteries for

battery conditioning purposes

and the part includes an accu-

rate capacity monitoring function

(gas gauge).

❏It charges batteries using a

switching buck regulator for

higher efficiency and lower power

dissipation.

❏The supply range is 4.5 to 16V,

so that the LTC1325 can be

powered from the charging

supply for up to ten cells.

❏A shutdown mode drops the

supply current to 50µA

Charging Circuit

Unlike most other charger ICs,

which employ linear regulators, the

LTC1325 charges batteries using a

switching buck regulator. This ap-

proach improves efficiency and

minimizes power dissipation, espe-

cially when charging high-capacity

batteries. The only external compo-

nents required are an inductor, a

P-channel MOSFET switch, a sense

resistor, and a catch diode (see Figure

2). A programmable battery divider

that accommodates one to sixteen

cells removes the need for an external

divider. All the circuitry for control-

ling the loop is integrated on-chip; no

external ICs are needed.

The LTC1325 operates from 4.5V

to 16V, so that it can be powered

directly from the charging supply.

The wide supply range makes it pos-

sible to charge up to ten cells without

the need for an external regulator to

drop the charging supply down to

power the LTC1325. When charging

is completed and the charging supply

is removed, the chip does not load

down other system supplies, because

the microprocessor can program the

LTC1325 into shutdown mode, in

which the quiescent current drops to

Introduction

NiCad and NiMH batteries may be

classified according to how quickly

they are designed to be charged. Fast

charge rate batteries can be charged

in as little as fifteen minutes, com-

pared to an overnight wait for standard

rate batteries. However, nothing

comes for free. With fast batteries,

overcharging must be limited or bat-

tery life will be adversely affected. The

requirement to terminate charge com-

plicates the design of fast chargers.

There are basically two ways to

limit overcharging: a battery can be

charged in several stages, i.e., with

the charging current reduced in later

stages, or the proper charge-termina-

tion technique can be used. The

complete charging algorithm depends

on the battery type and takes into

account the battery manufacturer’s

specifications and recommendations.

The charger circuit should also pro-

vide protection against faults (such

as excessively high battery tempera-

ture or voltage) to safeguard the

battery when normal termination

fails.

The LTC1325 is new fast charger

IC designed to provide the user with

by Anthony Ng, Teo Yang Long, and Robert Reay

Figure 1. LTC 1325 fast-charger circuit

DESIGN FEATURES

1325_1.eps

PGATE

DIS

BATT

BAT

REG

4.7µF

SENSE

HTF

MCV

LTF

CLK

OUT

REG

FILTER

p1.3

p1.2

p1.4

MPU

(e.g. 8051)

GND

SENSE

THERM 2 R

DIS

IRFZ94

TRK

L1

50µH

D1

1N5818

(4.5V-16V)

P1

IRF9Z30

THERM 1

R5 R6



1µF



10µF

Linear Technology Magazine • October 1994

comparators that monitor T

BAT

and

CELL

to detect low temperature faults

(LTF), high temperature faults (HTF),

low cell voltages (BATR) and high cell

voltages (MCV). The LTF, HTF, and

MCV thresholds are set by an exter-

nal resistor divider to maximize

flexibility. The LTC1325 also includes

a 32-bit counter that permits the

microprocessor to limit the maximum

charging time to one of eight time-out

values (5, 10, 20, 40, 80, 160, 320

minutes or no time-out). It is possible

to disable timer faults by selecting no

time-out.

Battery Conditioning

Under some operating or storage

conditions, NiCad and NiMH batter-

ies may lose full capacity. It is often

necessary to subject such batteries

to repeated deep discharge and charge

cycles to restore them to full capacity.

The LTC1325 can be programmed

into discharge mode, in which it au-

tomatically discharges V

CELL

to 0.9V.

This voltage is defined as the end of

discharge voltage (EDV). Fault pro-

tection is also active in discharge

mode to protect the battery against

temperature extremes (LTF, HTF) and

measure T

BAT

, T

AMB

and V

CELL

. By

keeping track of elapsed time, the

microprocessor will have the means

to calculate all existing termination

techniques (including dT

BAT

/dt and

BAT/dt

) and perform averaging to

reduce the probability of false termi-

nation. This flexibility also means

that a single circuit can charge both

NiCad and NiMH batteries. The

LTC1325 has an on-chip, 10-bit, suc-

cessive-approximation ADC with a

five-channel input multiplexer. Three

channels are dedicated to T

BAT

, V

CELL

and the gas gauge (see the section on

capacity monitoring); the other two

channels can be used for other pur-

poses, such as sensing T

AMB

another external sensor. The LTC1325

can be programmed into idle mode, in

which the charge loop is turned off.

This permits measurements to be

made without the switching noise

present when charging.

Fault Protection

The LTC1325 monitors battery

temperature, cell voltage, and elapsed

time for faults, and prevents the ini-

tiation or the continuation of charging

should a fault arise. The fault detec-

tion circuit (see Figure 3) consists of

50µA. In shutdown mode, the digital

inputs stay alive to await the wake-

up signal from the microprocessor.

The buck-regulator control circuit

maintains the average voltage across

the sense resistor (R

SENSE

) at V

DAC

. In

addition, a programmable duty cycle

modulates the P-channel MOSFET

driver output P

GATE

to reduce average

charging current. The average charg-

ing current is given by:

CHARGE

= V

DAC

× (duty cycle)/R

SENSE

The microprocessor can set V

DAC

one of four values(150mV, 50mV,

30mV or 15mV) and the duty cycle to

one of five values (1/16, 1/8, 1/4, 1/2,

1) giving 20 possible I

CHARGE

values

with a single R

SENSE

resistor.

Charge Termination

A plethora of charge termination

techniques are used in the LTC1325.

These are based on battery tempera-

ture (T

BAT

), cell voltage (V

CELL

), time

(t), ambient temperature (T

AMB

) or a

combination of these parameters.

Unlike other fast charging ICs, the

LTC1325 does not lock the user into

a particular termination technique

and its shortcomings. Instead, it pro-

vides the microprocessor a means to

Figure 2. LTC1325 charge, discharge, and gas-gauge circuit

DESIGN FEATURES

1325_2.eps

DUTY RATIO

GENERATOR

CHARGE

DISCHARGE

TO ADC MUX

111kHz

OSCILLATOR

ONE SHOT

DR0-DR2 3

–

DAC

VR0, VR1 GAS-GAUGE (GG) CHIP BOUNDARY

REG (3.072V)

C1

16pF

SENSE

R1

500k

DAC

S3 S4

PGATE

DIS

L1

50µH

D1

1N5818

(4.5V-16V)

SENSE

BATTERY

FILTER

R2

125k RF

DIS

TRK

P1

IRF9Z30

N1

IRFZ94



GG

0



VR1

0

1



VR0

0

1

0

1

DAC

VOLTAGE

15mV

30mV

60mV

160mV

0mV

Linear Technology Magazine • October 1994

to detect the EDV discharge termina-

tion point.

Capacity Monitoring

The LTC1325 may be programmed

into gas gauge mode (GG = 1 in Figure

2). In this mode, the sense resistor

senses the battery load current. The

sense voltage is filtered by an 1kΩ ×

lowpass filter. Amplifier A1 is con-

figured as an inverting amplifier with

a gain of four. The output of A1 is the

average sense voltage and is con-

verted by the ADC when the gas-gauge

channel is selected by the micropro-

cessor. By accumulating gas-gauge

measurements over time, the micro-

processor can determine how much

charge has left the battery and what

capacity remains.

Conclusions

By leaving all decisions (except fault

detection) to the microprocessor, the

LTC1325 does not lock the user into

any charge algorithm or charge ter-

mination technique. Multistage

charging algorithms or several charg-

ing algorithms can be implemented

in software, so that a wide range of

battery types can be charged with the

same circuit. The design of the exter-

nal charging circuit is made as simple

as possible by incorporating all the

functional blocks needed and mini-

mizing external component count. In

addition to charging batteries, the

LTC1325 has provisions for condi-

tioning batteries and measuring

battery capacity. Each of these three

functions is under the full control of

the microprocessor.

shutdown protection at some cost in

efficiency. This small loss in efficiency

can be balanced by considering the

system efficiency and utilizing an

auxiliary winding on the inductor of

the primary power supply. The switch-

matrix approach is the ultimate in

simplicity but it requires all voltages

to be present the inputs, since no

voltages are generated. The best

method depends on the application.

Conclusion

The PCMCIA specification requires

a significant amount of voltage switch-

ing on both the primary supply (V

)

and the flash-memory programming

supply pins (VPP). The complexity of

these requirements mandates spe-

cific integrated solutions. Each of the

two approaches for VPP switching,

linear regulation and switch matrix,

has its merits. Linear regulation pro-

vides current limiting and thermal-

Figure 3. Fault-detection circuitry

DESIGN FEATURES

PCMCIA, continued from page 5

with the LT1312 can be used to pro-

vide a rough 13V input to the LT1121

for an efficient system design. The

LTC1314 will enable the LT1121 lin-

ear regulator only during flash

memory programming intervals. In

all other modes the LT1121 is in

shutdown mode and consumes only

16µA. This LT1121 also provides ther-

mal- and current-limiting features to

increase the robustness of the socket.

1325_3.eps

FLTF

FHTF

BATR

FEDO

FMCV

–

BATP

1.6V

–

LTF

HTF

BAT

MCV

BATT

REG

SENSE

100mV

900mV

REG

BATTERY

DIVIDER

3.072V

LINEAR

REGULATOR

–

Linear Technology Magazine • October 1994

the switch is turned off and the drive

current is reduced to the idle level for

the duration of the cycle. At T4, the

cycle is complete and repeats from

T0.

Figure 3 is a simplified diagram of

the switch circuitry used when the

switch is being turned off. The pri-

mary path to turn the switch off is

current source I

and transistor Q1.

turns on Q1, which turns the switch

off by pulling down on its base. Turn-

off time is determined by the current

Q1 pulls from the switch’s base. Q1’s

collector current is limited to I

times

Q1’s h

. To increase Q1’s collector

current by simply increasing I

would

be a waste of power. This is because

an increase in Q1’s collector current

is beneficial only during the switch

turn-off transition time and not the

entire time the switch is off.

In order to reduce switch turn-off

time, a turn-off enhancement net-

work is used. Referring to Figure 3,

the network comprises C1, R1, D1,

and D2. When the switch voltage be-

gins to rise, its dV/dt feeds an

additional current, I

, into Q1’s base,

via R1 and D1. This increases Q1’s

collector current and reduces switch

turn-off time. This action is sustained

through the entire switch turn-off

transition time. R1 limits the amount

of current fed back to Q1, and D2

discharges C1 when the switch is

turned on.

Figure 2. Switch drive current

stray capacitance, turn-on time, turn-

off time, and control of drive current

all contribute to AC losses. Although

there is an inherent benefit with a

high-speed process, there are circuit

design techniques that can further

decrease AC losses.

Switch turn-on time can be re-

duced by increasing the switch

drive-current level, but having exces-

sive drive current after the switch has

turned on is a waste of power. The

LT1372 boosts drive current at switch

turn-on time, senses when turn-on

has occurred, and then reduces drive

current to a lower level. Figure 2

shows the LT1372’s control of switch

drive current. At T0, a new cycle be-

gins with the start of oscillator

“deadtime” (the time when the switch

is guaranteed to be off). Deadtime

provides a “look ahead” prior to switch

turn-on time and allows the drive

current to reach the boosted level

before the switch is turned on. At T1,

the drive current has attained the

boosted level and the switch turn-on

signal is set. At T2, the switch has

turned on. The switch drive circuitry

senses this condition and reduces

drive current to the lower level for the

duration of switch on time. The lower

level of drive current more closely

approaches the minimum current

required to keep the switch on. At T3,

LT1372, continued from page 1

Block Diagram

A block diagram of the LT1372 is

shown in Figure 1 (see page 1). The

circuit includes an internal, low-drop-

out regulator, trimmed oscillator,

trimmed reference, error amplifier,

current amplifier, current compara-

tor, and bipolar switch. Other unique

LT1372 features shown in the dia-

gram include a combined shutdown

and synchronization function, 5:1 os-

cillator frequency shifting network,

negative regulation feedback ampli-

fier, and switch-drive boost circuitry.

The Switch

In order to operate effectively at

high frequencies, the switch must

have low AC switching losses. Switch

DESIGN FEATURES

1372_2.eps

OSCILLATOR

VOLTAGE

DEAD-TIME

BOOSTED

NOMINAL

SWITCH ON

IDLEIDLE

T0 T1 T2 T3 T4

SWITCH OFF

3-STATE

SWITCH

DRIVE

CURRENT

SWITCH

VOLTAGE

Figure 3. Simplified switch diagram

1372_3.eps

SWITCH

Figure 4. Schematic diagram: LT1372 boost

converter

1372_4.eps

S/S

C1

47µF



2.7V TO 11V

OUT



12V

NFB

C2

0.047µF

N/C 3

OFF

R1

*SUMIDA CD73-100KC

SUMIDA (708) 956-0666

R3

1.24k

C3

33µF

R2

10.7k

D1

MBRS130T3

LT1372

L1*

10µH

1 6, 7

GND

Linear Technology Magazine • October 1994

Figure 6. LT1372 Positive-to-negative converter with direct feedback

Boost Converter

The boost converter in Figure 4

shows a typical LT1372 application.

This circuit converts an input volt-

age, which can vary from 2.7V to 11V,

into a regulated 12V output. Using all

surface-mount components, the en-

tire boost converter consumes only

0.5 square inches of board space.

Figure 5 shows the circuit’s efficiency,

which can reach 87% on a 5V input.

The reference voltage on the FB pin

is trimmed to 1.24V, and the output

voltage is set by the R2/R3 resistor

divider ratio (V

OUT

= V

REF

× (R2/R3 +

1). R1 and C2 frequency compensate

the circuit.

Positive-to-Negative Flyback

with Direct Feedback

A unique feature of the LT1372 is

its ability to directly regulate nega-

tive output voltages. As shown in the

positive-to-negative flyback converter

in Figure 6, only two resistors are

required to set the output voltage.

The reference voltage on the NFB pin

is −2V

REF

, making V

OUT

= −2

VREF

(R2/R3 + 1). Efficiency for this circuit

reaches 72% on a 5V input.

Dual-Output Flyback with

Overvoltage Protection

Multiple-output flyback convert-

ers offer an economical means of

producing multiple output voltages,

but the power-supply designer must

be aware of cross-regulation issues,

which can cause electrical overstress

on the supply and loads. Figure 7 is a

dual-output flyback converter with

overvoltage protection. Typically, in

multiple-output flyback designs, only

one output is voltage sensed and regu-

lated. The remaining outputs are

“quasi-regulated” by the turn ratios

of the transformer secondary. Cross

regulation is a function of the trans-

former used, and is a measure of how

well the quasi-regulated outputs

maintain regulation under varying

load conditions. For evenly loaded

outputs, as shown in Figure 8, cross

regulation can be quite good, but

when the loads differ greatly, as in

Figure 7. LT1372 Dual-output flyback converter with over-voltage protection

DESIGN FEATURES

OUTPUT CURRENT (mA)

EFFICIENCY (%)

1000

1372_5.eps

10 100

OUT

= 12V V

= 9V

= 5V

= 3V

1372_6.eps

S/S

FBN/C

C1

22µF



2.7V TO 16V

D1

MBRS130LT3

C2

0.047µF

OFF

R1

R2

2.49k

R3

2.49k

T1 = COILTRONICS CTX10-2P

COILTRONICS (407) 241-7876

C3

47µF

–V

OUT



–5V

D2

P6KE-15A

D3

MUR110

NFB

LT1372

1 6, 7

GND I

OUT



0.3A

0.5A

0.75A



3V

5V

IOUT (mA)

–30

–5

–10

–15

–20

–25

VOUT (V)

100

1372_9.eps

110

IN = 5V

VOUT

–VOUT

IOUT (mA)

–30

–5

–10

–15

–20

–25

VOUT (V)

100

1372_8.eps

110

IN = 5V

VOUT

–VOUT

1372_7.eps

S/S

100µF



2.7V TO 13V

1.24k 13.3k

MBRS140T3

2, 3

6, 7

C2

0.047µF

OFF

R1

12.1k

2.49k

T1 = DALE LPE-4841-A038

DALE (605) 665-9301

47µF

–V

OUT



–15V

OUT



15V

P6KE-20A

1N4148

NFB

LT1372

1 6, 7

GND

Figure 5. Efficiency of boost converter shown

in Figure 4

Figure 9. Cross regulation of Figure 7’s

circuit. −V

OUT

unloaded; only +V

OUT

voltage

sensed.

Figure 8. Cross regulation of Figure 7’s

circuit. +V

OUT

and −V

OUT

evenly loaded.

Linear Technology Magazine • October 1994

to isolate high-current paths from

low-current signal paths. We recom-

mend that separate power traces and

ground traces be provided for each

regulator to provide further isolation.

High-current ground paths for each

switch should meet only at the ground

star. The compensation components

for each regulator V

pin and the

ground-referred feedback divider net-

works for the LCD contrast converter

should connect directly to the analog

ground pin for best regulation.

Grounded CCFL with

Variable-Negative/Fixed-

Positive Contrast

Figure 3 (see page 12) is a complete

grounded CCFL circuit with

variable-negative/fixed-positive con-

trast-voltage generation based on the

LT1182. The circuit is almost identi-

cal to that of Figure 2, except that the

lamp connects directly to the LT1182

DIO pin, thereby closing a feedback

loop at the CCFL V

pin. Also, the BAT

and ROYER pins have been tied to-

gether. This ensures that the high-side

sense amplifier is disabled and does

not generate any feedback signal.

Performance is similar to that of the

floating circuit, with efficiencies

ranging from 85–90% for typical

CCFL lamps at full load. However,

the lamp-current range is limited to

1–6mA. Thermometering becomes

pronounced at currents below about

1mA. The battery supply range can

extend from 8V to 28V, with a nomi-

nal battery voltage of 12V. The LT1182

is typically powered from a 5V or 3.3V

logic supply. Programming of lamp

current in this example has been

simplified to a resistor and a variable

0–5V supply.

with a corresponding loss in operat-

ing efficiency of about 3%. Lamp

intensity is smoothly variable from

minimum to full intensity.

The LCD contrast regulator is con-

figured as a flyback converter with

divider networks set up to control

either positive or negative contrast

voltages. Grounding the negative side

of the secondary produces a fixed

+15V output. Grounding the positive

side of the secondary produces a vari-

able −10V to −30V output range at

30mA maximum load current. Out-

put voltage varies in direct proportion

to changes in V

(CONTRAST)

. LCD con-

trast efficiency is about 80% at full

power. The flyback converter runs in

discontinuous mode to minimize the

size of the magnetics.

The LT1182 has separate power

grounds for each switch and an ana-

log ground in order to minimize

interaction between the regulators.

Pay special attention to power and

ground traces on the PC board layout

CCFL, continued from page 12

in Figure 7 eliminates this require-

ment. This circuit senses both the

+15V and −15V outputs and prevents

either from going beyond its regulat-

ing value. Figure 10 shows the

unloaded −15V output being held con-

stant. The circuit’s efficiency, which

can reach 79% on a 5V input, is

shown in Figure 11.

the case of a load disconnect, there

may be trouble. Figure 9 shows that

when only the +15V output is voltage

sensed, the −15V quasi-regulated

output exceeds −25V when unloaded.

This can cause electrical overstress

on the output capacitor, output di-

ode, and the load when reconnected.

Adding output-voltage clamps is one

way to fix the problem, but the circuit

Conclusion

The LT1372 does not exhibit the

usual trade-offs between increased

switching frequency and efficiency.

In Figure 12, the efficiency of a

LT1372 is compared to an LT1172,

switching at 100kHz. The LT1372

has comparable efficiencies and

uses less board space.

DESIGN FEATURES

References:

1. Williams, Jim. 1992. Illumination Circuitry for

Liquid Crystal Displays. Linear Technology Cor-

poration, Application Note 49.

2. Williams, Jim. 1993. Techniques for 92% Effi-

cient LCD Illumination. Linear Technology

Corporation, Application Note 55.

1372_12.eps

LOAD

10mA

EFFICIENCY (%)

100

100mA 1A

= 5V

OUT

= 12V LT1372

LT1172

OUTPUT CURRENT (mA)

EFFICIENCY (%)

200

1372_11.eps

5 10 100

VOUT = ±15V VIN = 9V

VIN = 5V

VIN = 3V

OUT

(mA)

–30

–5

–10

–15

–20

–25

OUT

(V)

100

1372_10.eps

110

= 5V

OUT

–V

OUT

Figure 10. Cross regulation of Figure 7’s

circuit. −V

OUT

unloaded; both −V

OUT

and

OUT

sensed.

Figure 11. Efficiency of dual-output flyback

converter in Figure 7. Figure 12. LT1372 versus LT1172 efficiency

Linear Technology Magazine • October 1994

Simple Thermal Analysis — A Real

Cool Subject for LTC Regulators

Some typical LTC regulators and

their thermal characteristics are

shown in Table 1.

There are several other common

thermal resistance terms:

— thermal resistance from the

case of the package to the heat

sink.

— thermal resistance from a

heat sink surface to the ambient

temperature.

The last two terms are determined

by how a regulator is mounted to the

heat sink and by the properties of the

heat sink. Heat sinks are used to

decrease the thermal resistance and

therefore lower the temperature rise

of the regulator.

Temperature is a term with which

we are all very familiar. All thermal

calculations will use the Centigrade

scale or °C.

— temperature of the junction of

the regulator die.

— temperature of the case of the

regulator.

— ambient temperature.

The maximum operating junction

temperature, T

J MAX

for LTC regula-

tors is shown on the device data sheet.

What is Thermal Analysis?

The goal of any thermal analysis is

to determine the regulator junction

temperature, T

, to ensure that this

temperature is less than either the

regulator rating or a design specifica-

tion. In the simplest case, temperature

rise is calculated by multiplying the

power times the total of all thermal

resistance:

= P × θ

total

TOTAL

includes the thermal resis-

tance junction-to-case (θ

), thermal

resistance case-to-heat sink (θ

), and

thermal resistance heat sink-to-am-

bient (θ

represents the temperature rise

above the ambient temperature; there-

fore, to determine the actual junction

temperature of the regulator, the

ambient temperature must be added

to T

Regulator junction temperature =

Ambient Temperature + T

For example, consider a circuit

using an LT1129CT operating in a

50°C enclosure with an input voltage

of 8VDC, an output voltage of 5VDC,

and a load current of 1 ampere

The power dissipated by the

LT1129CT is:

P = (V

- V

OUT

) × I

LOAD

= (8V - 5V) × 1 amp

= 3 watts

The first question is, does this cir-

cuit need a heat sink?

Since we have assumed no heat

sink on the LT1129CT for the pur-

pose of this calculation, we must use

thermal resistance from junction to

ambient, θ

= 50°C/watt.

= P × θ

+ T

= 3 watts × 50°C/watt + 50°C

= 150°C + 50°C = 200°C

The junction temperature, T

that

we just calculated is greater than the

LT1129CT’s maximum junction tem-

perature specification of 125°C;

therefore this circuit must use a heat

sink.

Now the task at hand is to calcu-

late the correct heat sink to use. The

selected heat sink must hold the junc-

tion temperature at less than 125°C

for the LT1129CT.

As the temperatures go up...

so go the problems with voltage

regulators.

Introduction

Linear Technology Corporation

applications engineers get lots of calls

saying, “that $X%#@& voltage regu-

lator is so hot I can’t touch it!” The

purpose of the article is to show you,

the design engineer, how to perform

simple thermal calculations to deter-

mine regulator temperature and select

the proper package style and/or heat

sink. In addition, it will show an alter-

nate method of specifying thermal

parameters on LTC voltage regula-

tors.

Definition of Terms

Power dissipation is the param-

eter that causes a regulator to heat

up; the unit for power is watts. Power

is the product of the voltage across a

linear regulator times the load cur-

rent (see Figure 1).

Thermal resistance is a measure of

the flow of heat from one surface to

another surface; the unit of thermal

resistance is °C/watt. Common terms

for thermal resistance that show up

on most LTC data sheets are:

—thermal resistance from the

junction of the die to the case of

the package.

—thermal resistance from the

junction of the die to the ambient

temperature.

Figure 1. Typical linear regulator circuit

Table 1. θ

and θ

for three LTC regulators.

Device θJC (°C/watt) θJA (°C/watt)

LT1005CT 5.0

LT1083MK 1.6

LT1129CT 5.0 50

DESIGN INFORMATION

VOUT

VIN

ILOAD

PDISS = (VIN – VOUT) ILOAD (WATTS)

Cool_1.eps

by Alan Rich

Linear Technology Magazine • October 1994

= P × θ

TOTAL

+ T

125°C = 3 watts × θ

TOTAL

+ 50°C

TOTAL

= 25°C/watt and,

TOTAL

= θ

+ θ

For this configuration:

= 5°C/watt (LT1129CT data sheet)

= 0.2°C/watt (typical for heat sink

mounting)

= heat sink specification

Plugging in these numbers:

25°C/W = 5°C/W + 0.2°C/W + θ

= 19.8°C/watt

Therefore, the heat sink selected

must have a thermal resistance of

less than 19.8°C/watt to hold the

LT1129CT junction temperature at

less than 125°C. Obviously, the lower

the heat sink thermal resistance,

the lower the LT1129CT junction

temperature. A lower junction tem-

perature will increase reliability.

Now, let’s consider a circuit using

an LT1129CT operating in a 50°C

enclosure with an input voltage of

only 6VDC, an output voltage of 5VDC,

and a load current of 1 ampere.

The power dissipated by the LT1129CT

is:

P = (V

- V

OUT

) × I

LOAD

= (6V - 5V) × 1 amp

= 1 watt

Does this circuit need a heat sink?

Again, for the purposes of the cal-

culation, we must use thermal

resistance from junction to ambient,

= 50°C/watt for the LT1129CT.

= P × θ

+ T

= 1 watt × 50°C/watt + 50°C

= 50°C + 50°C = 100°C

The junction temperature, T

that

we just calculated is now less than

the LT1129CT’s maximum junction

temperature specification of 125°C.

Therefore this circuit does not need a

heat sink. This illustrates the advan-

tage of a low dropout regulator like

the LT1129CT.

An Alternative

Method for Specifying

Thermal Parameters

Linear Technology Corp. has intro-

duced an alternative method to specify

and calculate thermal parameters of

voltage regulators. Previous regula-

tors, with a single thermal resistance

junction-to-case (θ

), used an aver-

age of temperature rise of the control

and power sections. This could easily

allow excessive junction temperature

under certain conditions of ambient

temperature and heat sink thermal

resistance.

Several LTC voltage regulators

include thermal resistance and

maximum junction temperature

specifications for both the control and

power sections, as shown in

Table 2.

As an example, let’s calculate the

junction temperature for the same

application shown before, using an

LT1085CT instead of the LT1129CT.

Once again, we are operating in a

50°C enclosure; the input voltage is

8VDC, the output voltage is 5VDC,

and the load current is 1 ampere.

The power dissipated by the

LT1085CT is the same as before, 3

watts. We will assume we have se-

lected a heat sink with a thermal

resistance, θ

of 10°C/watt. First

calculate the control section of the

LT1085CT:

= 0.7°C/watt (LT1085CT data sheet)

= 0.2°C/watt (typical)

= 10°C/watt

TOTAL

= θ

+ θ

= 0.7°C/watt + 0.2°C/watt + 10°C/watt

= 10.9°C/watt

To determine the control section junc-

tion temperature:

= P × θ

TOTAL

+ T

= 3 watts × 10.9°C/watt + 50°C

= 82.7°C (T

J MAX

= 125°C)

To calculate the power section of the

LT1085CT:

= 3°C/watt (LT1085CT data sheet)

TOTAL

= θ

+ θ

= 3°C/W + 0.2°C/watt + 10°C/watt

= 13.2°C/watt

To determine the power section junc-

tion temperature:

= P × θ

TOTAL

+ T

= 3 watts × 13.2°C/watt + 50°C

= 89.6°C (T

J MAX

= 150°C)

In both cases, the junction tem-

perature is below the maximum rating

for the respective section; this en-

sures reliable operation.

Conclusion

This article is an introduction to

thermal analysis for voltage regula-

tors; however, the techniques

also apply to other devices, including

operational amplifiers, voltage refer-

ences, resistors, and the like. For the

more advanced student of thermal

analysis, it can be shown that there is

a direct analogy between electronic

circuit analysis and thermal analysis

as shown in Table 3.

All standard electronic network

analysis techniques (Kirchhoff’s laws,

Ohm’s law) and computer circuit

analysis programs (SPICE) can be

applied to complex thermal systems.

DESIGN INFORMATION

The LTC1192CT is guaranteed for 700mA, but

could be selected to output 1 Amp.

Thermal World Electrical World

Power Current

Temperature differences Voltage

Thermal resistance Resistance

Table 3. Analogy between thermal analysis

and electronic circuit analysis

Control Power

Device θJC TJ MAX θJC TJ MAX

LT1083MK 0.6°C/W 150°C 1.6°C/W 200°C

LT1085CT 0.7°C/W 125°C 3.0°C/W 150°C

Table 2. Two examples showing thermal

resistance of control and power sections of

LTC regulators.

Linear Technology Magazine • October 1994

Low-Noise Portable Communications

DC-to-DC Converter

DESIGN IDEAS

Portable communications products

pack plenty of parts into close prox-

imity. Digital clock noise must be

eliminated not only from the audio

sections, but also from the antenna,

which, by the very nature of the prod-

uct, is located only inches from active

circuitry. If a switching regulator is

used in the power supply, it becomes

another potential source of noise. The

LTC1174 stepdown converter is de-

signed specifically to eliminate noise

at audio frequencies while maintain-

ing high efficiency at low output

currents.

Figure 1 shows an all-surface-

mount solution for a 5V, 120mA

output derived from five to seven

NiCad or NiMH cells. Small input and

output capacitors are used to con-

serve space, without sacrificing

reliability. In applications where it is

desired, a shutdown feature is avail-

able; otherwise short this pin to V

The LTC1174’s internal switch,

which is connected between V

and

, is current controlled at a peak

threshold of approximately 340mA.

This low peak threshold is one of the

key features that allows the LTC1174

DESIGN IDEAS

Low-Noise Portable

Communications DC-to-DC

Converter....................... 25

Mitchell Lee

LT1585: New Linear

Regulator Solves Load Tran-

sients ............................ 26

Craig Varga

Luma Keying with the

LT1203 Video Multiplexer

...................................... 27

Frank Cox

A Tachless Motor-Speed

Regulator ...................... 28

Mitchell Lee

Figure 1. Low-noise, high-efficiency, stepdown regulator for personal communications devices

to minimize system noise compared

to other chips that carry significantly

higher peak currents, easing shield-

ing and filtering requirements and

decreasing component stress.

To conserve power and maintain

high efficiency at light loads, the

LTC1174 uses Burst Mode

opera-

tion. Unfortunately, this control

scheme can also generate audio-fre-

quency noise at both light and heavy

loads. In addition to electrical noise,

acoustical noise can emanate from

capacitors and coils under these con-

ditions. A feedforward capacitor (C3)

shifts the noise spectrum up out of

the audio band, eliminating these

problems. C3 also reduces peak-to-

peak output ripple, which measures

approximately 30mV over the entire

load range.

The interactions of load current,

efficiency, and operating frequency

are shown in Figure 2. High efficiency

is maintained at even low current

levels, dropping below 70% at around

800µA. No-load supply current is less

than 200µA, dropping to approxi-

mately 1µA in shutdown mode. The

operating frequency rises above the

telephony bandwidth of 3kHz at a

load of 1.2mA. Most products draw

such low load currents only in standby

mode with the audio circuits

squelched, when noise is not an

issue.

The frequency curve depicted in

Figure 2 was measured with a spec-

trum analyzer, not a counter. This

ensures that the lowest-frequency

noise peak is observed, rather than a

faster switching frequency compo-

nent. Any tendency to generate

subharmonic noise is quickly exposed

using this measurement method.

dI1174_1.eps

5-7 CELL

INPUT

SHUTDOWN

PGM

C1 = PANASONIC SP SERIES (201-348-4630)

C3 = AVX TPS SERIES (803-946-0690)

L1 = COILTRONICS OCTAPAK (407-241-7876)

MBR0520L C3

33µF/20V

L1

CTX33-1

OFF C2

6.8nF

5V/120mA

OUTPUT

LTC1174CS8

GND

C1

15µF/12.5V

R1

91k

R2

30k

OUTPUT CURRENT (mA)

0.1

100

FREQUENCY (kHz)

EFFICIENCY

100

dI1174_2.eps

0.1 1 10

by Mitchell Lee

Figure 2. Parameter interaction chart for

Figure 1’s circuit.

Linear Technology Magazine • October 1994

LT1585: New Linear Regulator

Solves Load Transients

and the equivalent series inductance

(ESL) from the output capacitor(s)

terminal(s) to the load connection.

Note that these contributions also

include the lead trace parasitics from

the capacitor(s) to the load.

The resistive component is simply

∆I × ESR. The droop to point A,

23.6mV, is the ESR contribution.

Calculating ESR:

23.6mV/3.8 Amps = 6.2mΩ

full load current. This is the inflection

point in the curve. Since the regulator

now measures the output voltage as

being too low, it overshoots the load

current and recharges the output

capacitors to the correct voltage.

Faster Regulator

Means Fewer Capacitors,

Less Board Space

The regulator has one major effect

on the system’s transient behavior.

The faster the regulator, the less bulk

capacitance is needed to keep the

droop from becoming excessive. It is

here that the advantage of the LT1585

shows up. The response time of the

LT1585 is about one-half that of the

last generation three-terminal

regulators.

The response in the first several

hundred nanoseconds is controlled

by the careful placement of bypass

capacitors. Figure 3 is a schematic

diagram of the circuit, but the layout

is critical, so consult the LTC factory

for circuit and layout information.

The latest hot new microproces-

sors have added a significant

complication to the design of the power

supplies that feed them. These de-

vices have the ability to switch from

consuming very little power to requir-

ing several amps in tens of

nanoseconds. To add a further

complication, they are extremely in-

tolerant of supply voltage variations.

Gone are the days of the popcorn

three-terminal regulator and the

0.1µF decoupling capacitor. The

LT1585 is the first low-dropout regu-

lator specifically designed for tight

output voltage tolerance (optimized

for the latest generation processors)

and fast transient response.

Figure 1 shows the kind of re-

sponse that can and must be achieved

if these microprocessors are to oper-

ate reliably. Figure 2 details the first

several microseconds of the transient

in Figure 1. The load change in this

case is 3.8 Amps in about 20ns. Two

parasitic elements dominate the tran-

sient performance of the system. Both

are controlled by the type, quantity,

and location of the decoupling ca-

pacitors in the system.

Anatomy of a

Load Transient.

The instantaneous droop at the

leading edge of the transient is the

result of the sum of the effects of the

equivalent series resistance (ESR)

Figure 3. Schematic Diagram: LT1585 responding to fast transients

Figure 2. Detailed sketch of first few microseconds of transient

The effects of inductance are pre-

dicted by the formula V = LdI/dt. The

voltage from point A to the bottom of

the trough is the inductive contribu-

tion (13.4mV). ESL is calculated to be

0.07nH. After the load current stops

rising the inductive effects end, bring-

ing the voltage to point B. At this

point the curve settles into a gentle

droop. The droop rate is dV/dt = I/C.

There is about 1300µF of useful ca-

pacitance on the board in this case

(see Figure 3). As the regulator out-

put current starts to approach the

new load current, the droop rate less-

ens until the regulator supplies the

DESIGN IDEAS

Figure 1. Transient response of 200mA to 4

Amp load step

dI1585_2.eps

23.6mV

13.4mV

“A”

NOMINAL V

“B”

dI1585_3.eps

OUT

C1 TO C2

220µF

10V

SANYO OSCON

× 2

GND

U1

LT1585CT

LOAD

C9 TO C18

1µF

SMD

× 10

C3 TO C8

220µF

10V

AVX TYPE TPS

× 6

by Craig Varga

Linear Technology Magazine • October 1994

Luma Keying with the

LT1203 Video Multiplexer

In video systems, the action of

switching between two or more active

video sources is referred to as a “wipe”

or a “key.” When the decision to switch

video sources is based on an attribute

of the active video itself, the action is

called keying. A wipe is controlled by

a non-video signal, such as a ramp.

The circuit presented in Figure 1 is

referred to as a “luma key” because it

switches between two sources when

the luminance (“luma”) of a mono-

chrome key signal reaches a set level.

It is also possible to key on the color

of the video source and this, not sur-

prisingly, this is called “chroma

keying.”

Figure 1’s operation is very straight-

forward. A monochrome video source

is used to generate the key signal. The

LT1363 is used as a buffer and may

not be needed in all applications. If

coherent with the color reference of

the sources. The key-source video

should be monochrome to prevent

the key comparator from switching

on the color subcarrier.

the key signal is to be used as one of

the switched signals, it is convenient

to “loop through” the input of this

buffer. The LT1016 comparator

switches when the video level exceeds

the DC reference on its inverting in-

put, which is controlled by the “key

sensitivity” control. The TTL key

signal controls an LT1203 video mul-

tiplexer. Any two video sources may

be connected to the inputs of the

LT1203, as long as they are gen locked

and within the common-mode range

(on ±5 volt supplies this is ±3 volts

over 0–70°C) of the multiplexer. The

LT1203’s fast switching speed, low

offset, and clean switching make it a

natural for an active video switching

application like this one. Composite

color signals can be used, but the

best results will be obtained if the

key signal’s horizontal sync is phase

Figure 1. Schematic diagram: luma keyer

DESIGN IDEAS

COMPOSITE BLANKING (TTL LEVELS)

dI1203_1.eps

–

KEY VIDEO

10k LT1363

LT1016

1000Ω

–5

15V

LT1004 - 2.5

INPUT

300Ω232pF36.5pF

10k

KEY SENSITIVITY

100k

SOURCE 1 IN1

GRD

IN2 NC

LT1203

75Ω

1000Ω

–

LT1363

675Ω

SOURCE 2 3

LOGIC

OUT

IN1

GRD

IN2 NC

LT1203

LOGIC

OUT

1000Ω

510Ω

VIDEO

OUT

510Ω

2.2k

COMPOSITE SYNC (TTL LEVELS) 300Ω

74AC04 9.3µH

by Frank Cox

Figure 2. Two-level image of IC designer

Linear Technology Magazine • October 1994

A common requirement in many

motor applications is a means of

maintaining constant speed with vari-

able loading or variable supply voltage.

Speed control is easily implemented

using tachometer feedback, but the

cost of a tach may be prohibitive in

many situations, and adds mechani-

cal complications to the product. A

lower-cost solution with no moving

parts is presented here.

Motor speed changes under condi-

tions of varying loads because of the

effects of series loss terms in the

motor. The effects of the predominant

contributors to loss, copper and

brush/commutator resistance (col-

lectively known as R

), are best

understood by considering the cir-

cuit model for a motor (see Figure 1).

A motor’s back-EMF (V

) is propor-

tional to speed (n), and the motor

current (I

) is proportional to the

load torque (T). The following equa-

tion predicts the speed of the motor,

for any given condition of loading:

where K

and K

are constants of

proportionality for rotational velocity

and torque. For a fixed terminal volt-

age, the speed of the motor must

decrease as increasing load torque is

applied to the shaft. For a fixed load,

the speed of the motor will also change

if the supply (terminal) voltage is

changed.

A voltage regulator fixes the prob-

lem of a varying terminal voltage, but

the only way to eliminate torque from

Equation (1) is by reducing R

Nonstandard video signals can be

used for the inputs to the LT1203.

For instance, it is possible to select

between two DC input levels to con-

struct a two-level image. Figure 2 is

an example of an image constructed

this way. A monochrome video signal

is sliced and used to key between

black (zero volts) and gray (approxi-

mately 0.5 volt) to generate this image

of a famous linear IC designer. An

image formed in this way is not a

standard video output until the blank-

ing and sync intervals are recon-

structed. The second LT1203 blanks

the video and an LT1363 circuit sums

composite sync to the video and drives

a cable. For more information on this

part of the circuit, see AN 57, page 7.

A clamp is not used, as the DC levels

are arbitrarily set by the inputs, but

one could be used, as in the figure on

page 7 of AN57, if the sources were

video. As another option, Figure 3

shows the same key signal used as

one of the inputs to the multiplexer.

Figure 3. Key signal used as input to the MUX

A Tachless Motor-Speed Regulator

zero. Physically this is impractical,

but an electrical solution exists.

If a motor is driven from a regu-

lated source whose output impedance

is opposite in sign and equal in mag-

nitude to R

(see Figure 1), the result

is a motor that runs at a constant

speed—regardless of loading and

power-source variations. Figure 2

shows a circuit that does it all. The

LT1170 is configured as a buck-boost

converter, which can take a wide-

ranging 3-to-20V input source and

produce a regulated output of, say,

6V. The circuit shown can deliver 1A

at 6V with a 5V input, adequate for

many small, permanent-magnet DC

motors.

To cancel the effects of the motor

resistance, a negative output imped-

ance is introduced with an op amp

and a current-sense resistor (R

). As

the motor current increases, the

LT1006 responds by increasing the

motor terminal voltage by an amount

equal to I

× R

. Depending on the

value of R3, the speed can be made to

increase, decrease, or stay the same

under load. If R3 is just right, the

Figure 1. On the right is shown an equivalent

circuit for a motor. On the left is the model

for a circuit which will stabilize the motor’s

speed against changes in supply voltage and

VTERMINAL RM

KVKT × KV

T ×

n =−(1)

by Mitchell Lee

DESIGN IDEAS

dImotr_1.eps

TERMINAL

MOTORSPEED

REGULATOR

= T/K

= K

•n

REF



–+

–











−•RR

Linear Technology Magazine • October 1994

motor speed will remain constant until

the LT1170 reaches full power and

the circuit runs out of steam.

Many small motors in the 1-to-10W

class are not well characterized. In

order to choose proper component

values for a given motor, figures for

and V

are necessary. Fortunately,

these are easily measured using a

DVM and a motor-characterization

test stand. If you don’t have a motor-

characterization test stand, it is also

possible to use a lathe or drill press to

do the job.

Chuck-up the candidate motor’s

shaft in a variable speed drill press or

lathe, which is set to run at the same

speed you’re intending to operate the

motor. Clamp down the motor frame

so it won’t spin. Turn on the big

machine, and measure the open-cir-

cuit motor terminals with a DVM.

This is the motor voltage, V

, as shown

in Figure 1. Switch the meter to mea-

sure the motor’s short circuit current,

. Motor resistance R

= V

With these figures the other compo-

nent values can be calculated:

MAX

= motor current at full load

REF

= 1.244V

R1 = series combination of 619Ω + 619Ω =

1238Ω

RS ≤ 1/I

MAX

(drops less than 1V at

maximum load)

R2 = (V

× R1/V

REF

) - R1 (2)

R3 = (R2 × R

)/(R

+ R

) (3)

loading from the motor. Check the

motor’s unloaded speed, and adjust

R2 if necessary to set it precisely.

With the motor driving a nominal

load, decrease R3 until the motor

commences “hunting.” R3 will be near

the nominal calculated value. This

threshold is very close to optimum

motor-resistance cancellation. R5

and C5 offer a convenient means of

compensating for frictional and iner-

tial effects in the mechanical system,

eliminating instabilities. System sta-

bility should be evaluated under a

variety of loading conditions. The ef-

fect of R5 is to reduce the negative

output impedance of the circuit at

high frequencies. Systems with a net

positive impedance are inherently

stable.

When the system stability is satis-

factory, a final adjustment of R3 can

be made to achieve the desired speed

regulation under conditions of vary-

ing loads. These final values can be

used in production. Note that R2 de-

fines the regulated speed value and

may be production trimmed in preci-

sion applications.

Figure 2. Tachless motor-speed regulator

The component values shown in

Figure 2 are for a small motor with

the following characteristics mea-

sured at 360RPM: V

= 7.8V, I

3.7A, R

= 2.1Ω, I

MAX

≈ 1A.

, a copper resistor, is either lo-

cated close to or wound around the

motor to assist in tracking changes in

armature resistance with tempera-

ture. Copper has a strong, 3930ppm/

°C temperature coefficient, matching

the TC of the motor winding.

Setup Procedure

Initial tests should be performed

with a potentiometer in place of, and

twice the value of, R3. R5 and C5

should be disconnected; remove all

DESIGN IDEAS

–

dImotr_2.eps

–

3V TO 20V

INPUT C1

1000µF

C3

1000µF

C2

330µF

C4

1µF

L1 = L2 = 50-2-52 COILTRONICS (407-241-7876). CAN BE COUPLED AS INDICATED BY PHASING DOTS.

LT1006 POWER SUPPLY PINS CONNECTED TO INPUT SUPPLY.

L1

50µH

LT1170

LT1006

MBRD340CT

R3

309Ω

GND

R2

6490Ω

R1a

619Ω

10k

R5

C5

2.2µF

R1b

619Ω

10k

R4

L2

50µH



100mΩ

Linear Technology Magazine • October 1994

New Device Cameos

NEW DEVICE CAMEOS

LTC1147L-3.3 and LTC1148L-

3.3: Extremely Low Dropout

Stepdown Controllers

The LTC1147L-3.3 and LTC1148L-

3.3 are the industry’s first 3.3V,

low-dropout, stepdown switching

regulator controllers. The LTC1148L-

3.3 is a synchronous switching

regulator controller available

in a compact 14-lead SOIC. The

LTC1147L-3.3 is a non-synchronous

switching regulator controller, avail-

able in a space-saving 8-lead SOIC.

Both controllers operate down to a

minimum input voltage of less than

3.6V, providing extremely low drop-

out operation. Battery life is extended

by providing high efficiencies at load

currents from milliamps (when the

device is in standby or sleep mode) to

amps (under full power conditions).

The 100% duty cycle characteristic in

dropout mode allows maximum en-

ergy to be extracted from the battery

pack, further extending operating

time. Both devices use current-mode

architecture with Burst Mode

op-

eration to provide extremely high

efficiency over the entire load range

demanded by the next generation of

notebook, pen-based, and handheld

computers.

LT1312/LT1313

Micropower Single and

Dual 120mA PCMCIA VPP

Regulators/Drivers

The LT1312 single and LT1313 dual

regulators power and protect the VPP

pins of standard PCMCIA card slots.

Built-in current limit and thermal

shutdown ensure that a short-circuit

condition on either side of a standard

PCMCIA socket will not damage the

host computer or PC card through

the VPP pins. Further, the LT1312

and LT1313 produce clean, tightly

regulated 0V, 3.3V, 5V, 12V, and

Hi-Z outputs from any available un-

regulated 13V to 20V supply, thus

eliminating the cost of an extra 12V

regulator to generate clean VPP power.

Direct connection with industry-

standard PCMCIA interface control-

lers is provided through compatible

digital inputs.

Both regulators boast micropower

operation, with the quiescent current

falling to just 30 microamps for the

LT1312 and 60 microamps for the

LT1313, when programmed to the 0V

or Hi-Z modes. A single 1 microfarad

capacitor is the only external compo-

nent required to ensure smooth

transitions between operating volt-

ages. Both parts have no overshoot

when supplying 12V to sensitive flash

memory cards.

Full 3.3V/5V compatibility is pro-

vided by an internal comparator that

continuously monitors the PC card

supply and adjusts the regulated

VPP output to match V

when the

VPP = V

mode is selected. Both

devices include VPP valid status out-

puts to indicate when the output

voltage is in regulation at 12V.

The combination of micropower

operation and full card and socket

protection make these parts ideal

solutions for battery powered appli-

cations. The LT1312 and LT1313 will

find application in palmtop, note-

book, and desktop computers,

instruments, handi-terminals, and

bar-code readers.

LTC1314/LTC1315 Single

and Dual PCMCIA VPP

Switching Matrix with Built-

in V

N-Channel Gate Drivers

The LTC1314 single and LTC1315

dual power controllers provide all the

switching necessary to control one or

two PCMCIA card slots. Both devices

boast 120mA VPP output current

capability and provide 0V, V

, 12V,

and Hi-Z output states. V

switching

between OFF, 3.3V, and 5V is accom-

plished with inexpensive low R

DS(ON)

N-channel switches driven by an in-

ternal voltage multiplier, which

generates 12V gate drive from the 5V

logic supply. Both the LTC1314 and

LTC1315 have logic-compatible in-

puts that interface directly with

industry standard PCMCIA card

controllers.

Both the LTC1314 and LTC1315

are designed for micropower opera-

tion and draw only 1 microamp when

programmed to the Hi-Z or 0V oper-

ating modes. Built-in charge pumps,

which drive both the internal and

external N-FET switches, eliminate

the need for continuous 12V power.

In fact, the external 12V supply is

only required when the VPP = 12V

mode is selected. When not needed,

the external 12V supply is shut down

by a logic-compatible signal from the

LTC1314, further reducing system

power consumption.

Internal break-before-make switch

action and ramped gate drive ensure

that the outputs of the LTC1314 and

LTC1315 move smoothly between

operating voltages, with no switch

overlap or VPP overshoot. Small by-

pass capacitors are the only external

components required.

The combination of efficient opera-

tion and full card and socket control

make the LTC1314/LTC1315 ideal

solutions for battery-powered appli-

cations. These parts will find

application in palmtop, notebook, and

desktop computers, instruments,

handi-terminals, and bar-code

readers.

LTC1390: Serial-Controlled,

Eight-channel Analog

Multiplexer

The LTC1390 is a low cost, eight-

channel analog multiplexer featuring

eight individual switches controlled

by a three-wire digital interface.

It has a bidirectional data retrans-

mission capability, allowing it to be

wired in series with a serial A/D con-

verter using only one serial port. The

interface also allows multi-input ex-

pansion with the connection of several

LTC1390s, using only a single digital

port.

The specifications for each analog

channel achieved at bipolar ±5V sup-

ply are full analog signal range, 75Ω

Linear Technology Magazine • October 1994

NEW DEVICE CAMEOS

maximum on-resistance (R

), ±5

nanoamp maximum ON and OFF

channel leakage current, and ±10

picocoulomb maximum charge injec-

tion. For dynamic switching between

channels, a guaranteed break-before-

make operation is ensured. All pins

can withstand repeated ±2kV ESD

strikes and 100mA fault currents

without latching-up. All digital in-

puts are TTL and CMOS compatible.

The LTC1390 is also provided with a

full set of specifications for operation

on a +3V supply. The LTC1390 is

available in 16-lead plastic DIP and

SOIC packages.

LTC1348 Ultra-low-power,

3.3V-Powered Three-

driver, Five-receiver

Transceiver with

True RS232 Output Levels

The LTC1348 is a new three-driver,

five-receiver RS232 interface trans-

ceiver that supplies true RS232

output-voltage levels when operated

from a single 3.3V power supply. The

integral charge-pump power genera-

tor uses three 0.1µF commutating

capacitors to generate ±8V power

supplies for the driver stages. The

charge pump provides sufficient out-

put current to drive a serial-port

mouse and 3kΩ2500pF loads at

data rates up to 120kbaud.

Power consumption is only 500µA

in normal operation. Driver-enable

and receiver-enable controls provide

flexible power management capabil-

ity to reduce power to as little as

0.2µA when the circuit is shut down.

Receiver and driver outputs are high

impedance when disabled or with

power off. Driver outputs and receiver

inputs are tolerant of the full ±25 volt

levels that can occur on RS232 data

lines.

Like all Linear Technology RS232

transceivers, the LTC1348 is protected

against ±10kV ESD strikes to the

RS232 inputs and outputs. This inte-

grated ESD resistance saves the

expense and space of external protec-

tion devices.

The circuit is available in 28-lead

DIP, SOIC, and SSOP packages. All

applications such as High-bit-rate

Digital Subscriber Line (HDSL).

Whereas the recently introduced

LTC1278-4’s 400ksps conversion rate

easily covers T1 data rates, the

LTC1279’s 600ksps is ideal for HDSL’s

faster E1 data rates. The parallel data

bus provides an excellent, yet flexible

and easy, interface to popular DSPs.

The analog-signal input range is 0V

to 5V when operating on a single 5V

power-supply voltage or ±2.5V when

operating on a split ±5V power-sup-

ply voltage.

The LTC1279’s higher speed is

achieved without sacrificing dynamic

conversion characteristics. The im-

proved S/(N+D) (SINAD) is a minimum

of 70dB at f

= 600ksps and f

100kHz. The high-performance con-

version characteristics ensure that

the bit-error rate is extremely low

(<<10

−12

Other key LTC1279 specifications

include differential and integral lin-

earity errors of ±1LSB (max), THD of

−78dB (max), and a typical 4MHz

full-power bandwidth, ideal for

undersampling applications.

Key features include low power dis-

sipation of just 75mW (f

= 600ksps,

+5V or ±5V supply voltage) during

operation and just 5mW when power

shutdown is active. The LTC1279 in-

cludes an internal voltage reference

and conversion clock with synchro-

nizing circuitry. The LTC1279 is

available in a small-footprint SOIC

package that results in efficient use

of circuit-board space.

package configurations use a flow-

through pinout to simplify PC board

layout.

LTC1261: Regulated

Switched-Capacitor Inverter

Designed to generate a regulated

negative supply from a low positive

voltage, the LTC1261 is ideal for pro-

viding bias voltages for GaAs FET RF

transmitters and other circuits that

require a precisely regulated negative

supply. The heart of the LTC1261 is a

charge pump that can both double

and invert the input voltage, allowing

it to generate a −5V output from a +3V

input. An on-board PWM regulation

loop keeps the output within 5% of

the set value at all times, with up to

15mA load current, even with varia-

tions in the input voltage. Output

ripple is typically below 5mV.

An open-drain output monitors the

regulation loop and pulls down to

indicate that the LTC1261 is in regu-

lation, allowing external circuitry to

monitor the output without additional

components. A shutdown pin brings

the output voltage to zero and cuts

the quiescent current to only 5 mi-

croamps. An on-board resistor string

allows the LTC1261CS14 to provide

−3.5V, −4V, −4.5 V and −5V outputs

with simple pin connections; other

output voltages can be set using ex-

ternal resistors. The LTC1261CS8 is

available with adjustable and fixed

−4V and −4.5V options.

The LTC1261CS14 is available in a

14-pin narrow SO package and re-

quires only one or two external 0.1µF

capacitors for operation, depending

on the input voltage; input and out-

put bypass capacitors are also

required. The LTC1261CS8 offers all

the same features with a standard

inverting-only charge pump in an SO8

package.

LTC1279: 12-Bit,

600ksps Sampling Analog-to-

Digital Converter

The LTC1279 is the second in a

series of high-speed, high-resolution

analog-to-digital converters optimized

for telecom digital-data transmission

Burst Mode

is a trademark of Linear

Technology Corporation.

Information furnished by Technology Cor-

poration is believed to be accurate and reliable.

However, Linear Technology makes no repre-

sentation that the circuits described herein

will not infringe on existing patent rights.

For further information on the

above or any of the other devices

mentioned in this issue of Linear

Technology, use the reader service

card or call the LTC literature ser-

vice number: 1-800-4-LINEAR. Ask

for the pertinent data sheets and

application notes.

Linear Technology Magazine • October 1994

AppleTalk



is a registered trademark of Apple Computer, Inc.

LINEAR TECHNOLOGY CORPORATION

1630 McCarthy Boulevard

Milpitas, CA 95035-7487

(408) 432-1900

Literature Department 1-800-4-LINEAR

DESIGN TOOLS

Applications on Disk

NOISE DISK

This IBM-PC (or compatible) progam allows the user to calculate circuit noise

using LTC op amps, determine the best LTC op amp for a low noise application,

display the noise data for LTC op amps, calculate resistor noise, and calculate

noise using specs for any op amp. Available at no charge.

SPICE MACROMODEL DISK

This IBM-PC (or compatible) high density diskette contains the library of LTC

op amp SPICE macromodels. The models can be used with any version of

SPICE for general analog circuit simulations. The diskette also contains

working circuit examples using the models, and a demonstration copy of

PSPICE

by MicroSim. Available at no charge.

Technical Books

1990 Linear Databook — This 1440 page collection of data sheets covers op

amps, voltage regulators, references, comparators, filters, PWMs, data con-

version and interface products (bipolar and CMOS), in both commercial and

military grades. The catalog features well over 300 devices. $10.00

1992 Linear Databook Supplement — This 1248 page supplement to the

1990 Linear Databook

is a collection of all products introduced since then.

The catalog contains full data sheets for over 140 devices. The

1992 Linear

Databook Supplement

is a companion to the

1990 Linear Databook

, which

should not be discarded. $10.00

1994 Linear Databook, Volume III — This 1826 page supplement to the

1990

Linear Databook

and

1992 Linear Databook Supplement

is a collection of

all products introduced since 1992. A total of 152 product data sheets are

included with updated selection guides. The

1994 Linear Databook Volume III

is a supplement to the 1990 and 1992 Databooks, which should not be

discarded. $10.00

Linear Applications Handbook — 928 pages full of application ideas

covered in depth by 40 Application Notes and 33 Design Notes. This catalog

covers a broad range of “real world” linear circuitry. In addition to detailed,

systems-oriented circuits, this handbook contains broad tutorial content

together with liberal use of schematics and scope photography. A special

feature in this edition includes a 22 page section on SPICE macromodels.

$20.00

1993 Linear Applications Handbook Volume II — Continues the stream

of “real world” linear circuitry initiated by the

1990 Handbook

. Similar in scope

to the 1990 edition, the new book covers Application Notes 41 through 54 and

Design Notes 33 through 69. Additionally, references and articles from non-

LTC publications that we have found useful are also included. $20.00

Interface Product Handbook

— This 336 page handbook features LTC’s

complete line of line driver and receiver products for RS232, RS485, RS423,

RS422 and AppleTalk



applications. Linear’s particular expertise in this

area involves low power consumption, high numbers of drivers and receivers

in one package, 10kV ESD protection of RS232 devices and surface mount

packages. Available at no charge.

Monolithic Filter Handbook — This 234 page book comes with a disk which

runs on PCs. Together, the book and disk assist in the selection, design and

implementation of the right switched capacitor filter circuit. The disk contains

standard filter responses as well as a custom mode. The handbook contains

over 20 data sheets, Design Notes and Application Notes. $40.00

SwitcherCAD Handbook — This 144 page manual, including disk, guides

the user through SwitcherCAD—a powerful PC software tool which aids in the

design and optimization of switching regulators. The program can cut days off

the design cycle by selecting topologies, calculating operating points and

specifying component values and manufacturer's part numbers. $20.00

1994 Power Solutions Brochure — This 52 page collection of circuits

contains real-life solutions for common power supply design problems. There

are over 45 circuits, including descriptions, graphs and performance specifi-

cations. Topics covered include micropower DC/DC, step-up and step-down

switching regulators, off-line switching regulators, linear regulators, switched

capacitor conversion and power management. Available at no charge.

FRANCE

Linear Technology S.A.R.L.

Immeuble “Le Quartz”

58 Chemin de la Justice

92290 Chatenay Malabry

France

Phone: 33-1-41079555

FAX: 33-1-46314613

GERMANY

Linear Techonolgy GmbH

Untere Hauptstr. 9

D-85386 Eching

Germany

Phone: 49-89-3197410

FAX: 49-89-3194821

JAPAN

Linear Technology KK

5F NAO Bldg.

1-14 Shin-Ogawa-Cho, Shinjuku-Ku

Tokyo, 162 Japan

Phone: 81-3-3267-7891

FAX: 81-3-3267-8010

KOREA

Linear Technology Korea Branch

Namsong Building, #505

Itaewon-Dong 260-199

Yongsan-Ku, Seoul

Korea

Phone: 82-2-792-1617

FAX: 82-2-792-1619

SINGAPORE

Linear Technology Pte. Ltd.

507 Yishun Industrial Park A

Singapore 2776

Phone: 65-753-2692

FAX: 65-541-4113

TAIWAN

Linear Technology Corporation

Rm. 801, No. 46, Sec. 2

Chung Shan N. Rd.

Taipei, Taiwan, R.O.C.

Phone: 886-2-521-7575

FAX: 886-2-562-2285

UNITED KINGDOM

Linear Technology (UK) Ltd.

The Coliseum, Riverside Way

Camberley, Surrey GU15 3YL

United Kingdom

Phone: 44-276-677676

FAX: 44-276-64851

Linear Technology Corporation

1630 McCarthy Boulevard

Milpitas, CA 95035-7487

Phone: (408) 432-1900

FAX: (408) 434-0507

International

Sales Offices

World Headquarters

U.S. Area

Sales Offices

CENTRAL REGION

Linear Technology Corporation

Chesapeake Square

229 Mitchell Court, Suite A-25

Addison, IL 60101

Phone: (708) 620-6910

FAX: (708) 620-6977

NORTHEAST REGION

Linear Technology Corporation

One Oxford Valley

2300 E. Lincoln Hwy.,Suite 306

Langhorne, PA 19047

Phone: (215) 757-8578

FAX: (215) 757-5631

Linear Technology Corporation

266 Lowell St., Suite B-8

Wilmington, MA 01887

Phone: (508) 658-3881

FAX: (508) 658-2701

NORTHWEST REGION

Linear Technology Corporation

782 Sycamore Dr.

Milpitas, CA 95035

Phone: (408) 428-2050

FAX: (408) 432-6331

SOUTHEAST REGION

Linear Technology Corporation

17060 Dallas Parkway

Suite 208

Dallas, TX 75248

Phone: (214) 733-3071

FAX: (214) 380-5138

SOUTHWEST REGION

Linear Technology Corporation

22141 Ventura Blvd.

Suite 206

Woodland Hills, CA 91364

Phone: (818) 703-0835

FAX: (818) 703-0517