19-0042; Rev 1; 4/93 MA MALAA Wideband Transconductance Amplifiers General Description The MAX435 and MAX436 are high-speed, wideband transconductance amplifiers (WTAs) with true differen- tial, high-impedance inputs. Their unique architecture provides accurate gain without negative feedback, elim- inating closed-loop phase shift a primary cause of circuit oscillation in conventional high-speed amplifiers. The output of the WTA is a current that is proportional to the applied differential input voltage, providing inherent short-circuit protection for the outputs. Circuit gain is set by the ratio of two impedances and an internally set current gain factor (K). The current output and the absence of negative feedback allow the differential output MAX435 to achieve a bandwidth of 275MHz, an 800V/us slew rate, and a 1% settling time of 18ns to a 0.5V step input. The single- ended output MAX436 achieves a bandwidth of 200MHz, an 850V/us slew rate, and a 1% settling time of 18ns to a 0.5V step input. Both amplifiers offer exceptional wideband common-mode rejection, with a CMRR of 53dB at 10MHz. 300V input offset voltage offers a level of DC precision rarely found in high-speed op amps. Unlike current-feedback amplifiers, the MAX435/MAX436 have fully symmetrical, high-impedance inputs that tol- erate wide differential input voltages without destruc- tive failure or amplifier saturation, virtually eliminating overload recovery time. The unique performance fea- tures of these WTAs suit them for a wide variety of applications, such as high-speed instrumentation am- plifiers and wideband, high-gain bandpass amplifiers. And, with its differential output, the MAX435 can be used in high-speed differential line driver and receiver applications. Applications High-Speed Instrumentation Amplifiers High-Speed Filters Wideband, High-Gain Bandpass Amplifiers Differential Line Receivers Differential Line Drivers Features 4 275MHz Bandwidth (MAX435) 4 850V/us Slew Rate 18ns Settling Time to 1% #53dB CMAR at 10MHz # Low Noise, 7nV/V/Hz at 1kHz @ No Feedback 4 True Differential High-Impedance Inputs # Shutdown Capability: 450A (MAX435) Ordering Information PART TEMP. RANGE PIN-PACKAGE MAX435CPD OC to +70C 14 Plastic DIP MAX435CSD 0C to +70C 14S0 MAX435C/D 0C to +70C Dice MAX435EPD -40C to 485C = 14 Plastic DIP MAX435ESD -40C to +85C 1450 MAX435MJD -55C to +125C 14 CERDIP Ordering information continued on last page. " Dice are specified at Ta = +25C, DC parameters only. Pin Configurations TOP VIEW 7 . <a + TL anaaa [4 V+ \+ To] ALAA Te) V+ INe (Ef MAX435 Fy toute ins ep MAX436 FF) tour 7+ @ fiz] V+ 7B V+ ne. [a] rss nc [) Ise ac av 7G v. IN- a lout- iN-EE fo} N.C y- [7 6] V- VE 6) V- DIP/SO Typical Operating Circuits Vout+ 8 7- ____I in- Vout Vout MAAXUM Maxim Integrated Products 1 For free samples & the latest literature: http:/)www.maxim-ic.com, or phone 1-800-998-8800 SEVXVW/SEPXVNMAX435/MAX436 Wideband Transconductance Amplifiers ABSOLUTE MAXIMUM RATINGS \ Total Supply Voltage (V+ toV-) . 0... cee eee 12V SQ (derate 8.33mW/"C above +70C) Positive Supply Voltage (V+ toGND)...............065 6.0V CERDIP (derate 9.09mW/"C above +70C) Negative Supply Voltage (V-toGND) ................ 6.0V Input Voltage (V- - 0.3V) to (V+ + 0.3V) Current Limit through Z+ Of Z- 0... cece 10mA Output Short-Circuit Duration ....... lser Short-Circuit Duration (to GND or Veg) Continuous Power Dissipation (Ta = +70C) Plastic DIP (derate 10.00mW/"C above +70C) Operating Temperature Ranges: MAX MAX43_E eee beeen Continuous MAX43_MJD oo. eee Leber eee 1 sec Storage Temperature Range Lead Temperature (soldering, 10 sec) bees 800mWw 667mW 727mW 0C to +70C -40C to +85C -65C to +125C -65C to + 160C +300C Stresses beyond those listed under Absolute Maximum Ratings" may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect device reliability. ELECTRICAL CHARACTERISTICS MAX435 (V+ = 5V, V- = -5V, -2.5V s IN+ $ 2.5V, -2.5V < IN- s 2.5V, 2.4 = Z. = 500, Z = 4000, Rset = 5.9kQ, Ta = +25C, Tj = +25C, unless otherwise noted. Specification is at +25C junction temperature due to requirements of high-speed automatic testing. Actual values at operating temperatures may exceed the value at Tj = +25C. No-load operating junction temperature may rise 30C to 50C above ambient.) PARAMETER SYMBOL CONDITIONS MIN TYP MAX | UNITS Operating Supply Voltage V+, V- -55C to +125C +4.75 +500 +5.25 v Supply Current Is -58C to +125C 30 35 41 mA Input Voltage Range IN+, IN- -55C to +125C -2.5 25 Vv Input Offset Voltage Vos 0.3 3.0 mV Input Offset-Voltage Drift AVos/AT 3.5 pv/'C Ta = 425C 1.0 3.0 input Bias Current lp OC to +70C 5 pA -40C to +85C 10 -55C to +125" 10 Input Resistance RIN 200 800 kQ DC Resistance Looking into 2+ or Z- Rz 0.15 1.50 Q Output Voltage Range VouT+, Vout. | Zt+ = ZL. = 00Q -3.5 3.5 Vv Output impedance Rout 2.0 3.5 kQ Output Current lout -10 10 mA Differential Output Offset Current los, DIFF Z+ and Z- are open, inputs are grounded | -140 0 140 pA Output Offset Current (either side) los Z+ and Z- are open, inputs are grounded | -100 = -20 100 pA oe ecurent Drift Alos/AT Z+ and Z- are open, inputs are grounded 23 pase Ta = +26C 3.90 400 4.10 Current Gain Ratio K O'C to +70C 3.80 415 | aia -40C to +85C 3.70 4.15 -55C to +125 3.60 4.20 Common-Mode Rejection Ratio CMRR AtDC 70 90 dB At 10MHz, Z = 200Q, Zi4 = Zc. = 250 53 Power-Supply Rejection Ratio PSRR At DC 50 77 dB Settling Time to 1% ts 21 = 2000, Zi+ = ZL. = 252 18 ns Slew Rate SR 2; shorted 800 Vius Z, shorted, Zi4 = Zi. = 250 550 -3dB Bandwidth BW Zt = 2000, Zi+ = ZL. = 250 276 MHz Input Noise-Voltage Density en At 1kHz, single-ended input, Z: = 500 7.0 nVNAz 2 MAAXLAAWideband Transconductance Amplifiers ELECTRICAL CHARACTERISTICS MAX436 (V+ = BV, V- = -SV, -2.5V S$ IN+ $ 2.5V, -2.5V $ IN- $ 2.5V, 214 = 500, 2; = 4000, Reet = 5.9kQ, Ta = +25C, Tj = +25C, unless otherwise noted. Specification is at +25C junction temperature due to requirements of high-speed automatic testing. Actual values at operating temperatures may exceed the value at Tj = +25C. No-load operating junction temperature may rise 30C to 50C above ambient.) PARAMETER SYMBOL CONDITIONS MIN. TYP MAX [UNITS Operating Supply Voltage V+, V- -55C to +125C +4.75 +5.00 +45.25 Vv Supply Current Is -58C to +125C 30 35 4 mA Input Voltage Range IN+, IN- -65C to +125C -25 25 Vv Input Offset Voltage Vos 0.3 3.0 mV Input Offset-Voltage Drift AVos/AT 3.5 nVv/C Ta = +26C 1.0 3.0 Input Bias Current lp OC to +70C 5 pA -40C to +85C : 10 -58C to +125" 10 Input Resistance Rin 200 700 kQ DC Resistance Looking into Z+ or Z- Rz 0.15 1.50 Output Voltage Range Vout, Vout. { Zi+ = 500Q -3.5 3.5 Vv Output Impedance Rout 2.0 3.3 Output Current lout -20 20 mA Output Offset Current los Z+ and Z- are open, inputs are grounded. | -100 6 100 pA Output Offset-Current Drift Alos/AT Z+ and Z- are open, inputs are grounded. 0.75 pAsC Ta = +25C 78 8.0 8.2 Current Gain Ratio K O'Cto+70C a7 83] AJA -40C to +85C 7.6 8.3 -58C to +125" 7.4 8.4 Common-Mode Rejection Ratio CMRR AtDC 70 84 dB At 10MH2z, Z; = 2000, ZL, = 250 53 Power-Supply Rejection Ratio PSRR At DC 40 50 dB Settling Time to 1% , ts 2 = 2000, ZiL+ = 252 18 ns Slew Rate SR 2: shorted 850 Vius 2 shorted, ZL+ = ZL. = 25 450 -3dB Bandwidth BW Zt = 2000, ZL4 = 250 200 MHz Input Noise-Voltage Density en At 1kHz, single-ended input, Z; = 50 7.0 nVNAz MA AXIAA 3 SEVXVW/SEPXVNMAX435/MAX436 Wideband Transconductance Amplifiers MAX435 COMMON-MODE REJECTION RATIO vs. FREQUENCY Rt =2000 RL=250 CMRR (48) 3 s-8 nN oS 0 1 10 100 1k 10k 100k 1M 10M 100M FREQUENCY (Hz) MAX435 OFFSET VOLTAGE vs. TEMPERATURE OFFSET VOLTAGE (mV) - IN- =OV 7- + Z+ 102- 60-40 -20 0 20 40 60 80 100 120140 TEMPERATURE (C) MAX435 OUTPUT OFFSET CURRENT vs. INPUT VOLTAGE =t5V =IN-=0V Z+AND (lout-) OUTPUT OFFSET CURRENT (aA) 3-200 (4 0 1 2 3 COMMON-MODE INPUT VOLTAGE (V) MAX435 MAX435. GAIN vs. FREQUENCY RL = 25 = & ft 1 10 100 1000 FREQUENCY (MHz) MAX435 OUTPUT OFFSET CURRENT vs. TEMPERATURE = z = 3 2 S = =IN- =0V 2 7+ AND Z- ARE OPEN = AVERAGE SOURCE CURRENT FROM ) AND (lour-) -60-40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) MAX435 Z TERMINAL INPUT IMPEDANCE. 200 vs. TEMPERATURE Vs= 178 125 0.75 INPUT IMPEDANCE (Q) 3 o 0.25 0 -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) CURRENT GAIN RATIO (K) DIFFERENTIAL OUTPUT OFFSET CURRENT(mA) BIAS CURRENT (uA) Typical Operating Characteristics MAX435 CURRENT GAIN RATIO (K) vs. TEMPERATURE . mn Vg = Rt= > a 405 4.00 3.95 3.90 -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) MAX435 DIFFERENTIAL OUTPUT OFFSET CURRENT vs. TEMPERATURE = IN- =0V + AND Z- ARE OPEN fos = (louts)0s - (louT-}os -60-40 -20 0 20 40 60 80 109 120 140 TEMPERATURE (C) MAX435 INPUT BIAS CURRENT vs. TEMPERATURE Vs = -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) AAAXIAAWideband Transconductance Amplifiers Typical Operating Characteristics (continued) MAX435 MAX435 MAX435 INPUT IMPEDANCE OUTPUT IMPEDANCE MAXIMUM OUTPUT CURRENT vs. TEMPERATURE vs. TEMPERATURE vs. TEMPERATURE es g 2 a B? 2 z e e =o 2 2 5 5 z E z 2 1 3 18 -60 -40 -20 0 20 40 60 80 100 120 140 ~-60 -40 -20 0 20 40 60 80 100 120 140 -60 -40-20 0 20 40 60 80 100 120 140 TEMPERATURE (C) TEMPERATURE (C) TEMPERATURE (C) MAX435 MAX435 SUPPLY CURRENT POWER-SUPPLY REJECTION RATIO vs. TEMPERATURE vs. TEMPERATURE = z = = = a = ~ a. = a =475VT0 1 =-4,75V TO 0 -60-40 -20 0 20 40 60 80 100 120140 ~-60 -40 -20 0 20 40 60 80 100 120140 TEMPERATURE (C) TEMPERATURE (C) MAAXMIAA 5 SEVXVW/SEPXVNMAX435/MAX436 Wideband Transconductance Amplifiers Typical Operating Characteristics (continued) MAX435 MAX435 MAX435 COMMON-MODE REJECTION MAXIMUM OUTPUT CURRENT SUPPLY CURRENT RATIO vs. TEMPERATURE vs. Rset vs. Reet Vs =i5V Vg =35V Rt = 40082 = 4002 RL = 50Q CMRR (dB) OUTPUT CURRENT (mA) SUPPLY CURRENT (mA) 0 -! 60-40 -20 0 20 40 60 80 100 120140 { 2 5 10 1 2 5 10 TEMPERATURE (C) ser (ke) RSet (kQ) MAX435 MAX435 POWER DISSIPATION MAXIMUM PACKAGE POWER vs. Rset TION vs. TEMPERATURE +5V 4000 POWER DISSIPATION (mW) MAXIMUM POWER DISSIPATION (mW) ~60 -40 -20 0 20 40 60 80 100 120 140 Rget (K2) TEMPERATURE (*C) 6 MAAXIAAWideband Transconductance Amplifiers MAX436 COMMON-MODE REJECTION RATIO vs. FREQUENCY Rt = 2000 RL = 250 60 CMRR (dB) GAIN (dB) 40 20 1 10 100 tk Ok 100k 1M 10M 100M FREQUENCY (Hz) MAX436 OFFSET VOLTAGE vs. TEMPERATURE =i5V _ _ \ =IN- =0V z z Z 3 = = 3 = 2 bh 5 aw E 3 2 = 3 1 60-40 -20 0 20 40 60 80 100 120140 TEMPERATURE (C) MAX436 Z TERMINAL INPUT IMPEDANCE vs. TEMPERATURE 175 g 150 3 & 125 3 =z > & 1.0 2 a zc = 3B 5 075 a 5 a oso my a on oO -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C} MAAXIAA Typical Operating Characteristics (continued) MAX436 MAX436 GAIN va. FREQUENCY Rt = 200, RL = 260 a4 10 100 1900 FREQUENCY (MHz) MAX436 OUTPUT OFFSET CURRENT vs. TEMPERATURE =t6V =0V Z- ARE -60-40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (*C) MAX436 INPUT BIAS CURRENT vs. TEMPERATURE Mm oO 4 -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) MAX436 CURRENT GAIN RATIO (K) vs. TEMPERATURE Rt SEVXVW/SEPXVUN 500 < Ss z= z 3 = @ 7 3 78 iT -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) MAX436 OUTPUT OFFSET CURRENT vs. INPUT VOLTAGE 3 Zz a 5 = i Ss = 3 i= te RL =500 3 2 40 4 COMMON-MODE INPUT VOLTAGE (V) MAX436 INPUT IMPEDANCE vs. TEMPERATURE 1 INPUT IMPEDANCE (kt) -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C)MAX435/MAX436 Wideband Transconductance Amplifiers Typical Operating Characteristics (continued) MAX436 MAX436 MAX436 OUTPUT IMPEDANCE MAXIMUM OUTPUT CURRENT SUPPLY CURRENT vs. TEMPERATURE vs. TEMPERATURE vs. TEMPERATURE 3 o Ss reg z 2 S = = = 5 3 a = Rser = 5,9kQ -60 -40 -20 0 20 40 60 80 100 120 140 60 -40 -20 0 20 40 60 a0 100 120 140 -60-40 -20 0 20 40 60 80 100 120140 TEMPERATURE (C} TEMPERATURE (C) TEMPERATURE (C) MAX436 MAX436 MAX436 POWER-SUPPLY REJECTION RATIO COMMON-MODE REJECTION MAXIMUM OUTPUT CURRENT vs. TEMPERATURE ' RATIO vs. TEMPERATURE vs. RSET V5 = 45V = 4002 RL = 500 z s z & g 3 a oO 5 Vs=25V 2 | CATSVTO IN =+2.5V 10-2.5V 2-4, 75V 10 60-40-20 0 20 40 60 80 100 120140 40 -20 0 20 40 60 a0 100 120 140 TEMPERATURE (*C} TEMPERATURE (C) Ret (ka) MAX436 MAX436 MAX436 SUPPLY CURRENT POWER DISSIPATION MAXIMUM PACKAGE POWER ve. vs. RSET qDISSIPATION vs. TEMPERATURE Vg 245V =45V _ = 4000 =400Q = RL =50Q _ = = = = S$ E 5 = S 2 B = = 2 a an o & 5 2 WY Oo = & = a = -60 -40 -20 0 20 40 60 80 100 120 140 TEMPERATURE (C) 8 MAAXIAAWideband Transconductance Amplifiers Pin Description PIN NAME FUNCTION MAX435 | MAX436 1, 12, 14] 1, 12,14] V+ Positive Power Supply (+5V) 2 2 IN+ | Noninverting Amplifier Input 3 3 Z+ Positive, Transconductance 4 4,9 N.C. | No Connect - Leave this pin open 5 5 Z- Negative Transconductance Terminal 6 6 IN- | inverting Amplifier Input Terminal 7,8,10 | 7,8, 10 V- Negative Power Supply (-5V) 9 - lout- | Inverting Differential Output 11 11 !ser | Supply-Current Set Terminal 13 - lout+ | Noninverting Differential Output ~ 13 lout | Amplifier Output Terminal Theory of Operation The MAX435/MAX436 are wideband transconductance amplifiers (WTA) that operate with no feedback. These amplifiers are intrinsically stable since closed-loop phase shift does not affect amplifier stability. Unlike conven- tional voltage-mode amplifiers, WTAs need no compen- sation from an internally set dominant pole. Voltage-mode amplifiers require this pole to roll off open- loop gain and prevent oscillation from high-frequency phase shift. 2 13 lout Nt Q___ + > K Vin rt Vin IN = O_ 6 9 lout- a 2 13 lout Ne O + _ VIN Kv Figure 1. a) MAX435 Operational Model. b) MAX436 Opera- tional Model. PAMXIAA The MAX435/MAX436's unique architecture allows single- ended (MAX436) or differential (MAX435) signal gain to be set by the ratio of two impedances: the user-selected transconductance element or network (Zt), and the output load impedance (ZL). The WTAs are essentially voltage- controlled current sources, as shown in Figure 1. The MAX435/MAX436's output is a current proportional to the differential input voltage, and inversely proportional to the impedance of the user-selected transconductance element or network (Zt). The current output provides inherent short-circuit protection for the output terminals. A differential input voltage (VIN) applied between the input terminals causes current to flow in the transconductance element (Zt). This current is equal to VIN/Zt. The current in the transconductance element is multiplied by the preset current gain (K) of the WTA, and appears at the output terminal(s) as a current equal to (K) x (VINV/Zt. This current flows through the load impedance to produce an output voltage according to the following equation: vur= (2m { Where: K = WTA Current-Gain Ratio ZL = Output Load Impedance Zt = Transconductance Element impedance VIN= Differential Input Voltage Unlike current-feedback amplifiers, the MAX435/MAX436 have symmetrical inputs with an input impedance of about 750kQ. This allows true differential input applica- tions to be realized, as shown in the MAX435 Typical Operating Circuit. And, because the input is symmetri- cal, opposite signal polarity can be obtained by swap- ping the input terminals. With proper selection of component values, it is possible to implement an amplifier circuit that accepts differential input voltages covering the WTA's entire input voltage range (-2.5V to +2.5V), without overloading its output stage. This characteristic makes the MAX435/MAX436 ideal for use in wideband instrumentation amplifiers, differential receivers, and settling-time measurement circuits. Design Procedure Setting the Circuit Gain The MAX435/MAX436 produce an output current by multi- plying a differential input voitage, Vin, by the transconduct- ance, K/Zt. The voltage gain (Av) is set by the impedance of the transconductance network (Zt) and the output load impedance (ZL), according to the following formula: =K(Ze fi SEPXVW/SEPXVWMAX435/MAX436 Wideband Transconductance Amplifiers The factor K in the gain equation refers to the WTA's current gain. K is factory trimmed to provide a low-drift, stable circuit gain for WTA applications. K is trimmed to 4.0 +2.5% for the MAX435, to 8.042.5% for the MAX436. The factor Zt in the gain equation is the impedance of a user-selected, 2-terminal transconductance elernent or network. This network is connected across the WTAs transconductance terminals, labeled as Z+ and Z-. The transconductance network should be selected, along with the output impedance, to provide the desired circuit gain and frequency shaping. To maintain linearity, the transcon- ductance network should also be selected so that the current flowing through it, equal to Vin/Zt, does not exceed 2.5mA under worst-case conditions of maximum input voltage and minimum transconductance element imped- ance (Zt). Output current should not exceed +10mA per output for the MAX435, or +20mA for the MAX436. Supply Current (iser)Control An extemal current source controls the WTA's intemal current. Connecting an external resistor (RSET) from the ISET pin to the negative power supply (V-) controls the current source. The typical value for this resistor is 5.9kQ, which provides an output-current drive capability of +10mA per output for the MAX435, +20mA for the MAX436. Connect a 0.22uF ceramic capacitor from the (SET pin to V+, to decouple the current source. A larger resistor value will reduce the devices supply current, power dissipation, and output-current drive ca- pability. A smaller resistor value may also be used to increase the output-current capability, but take care that the power dissipation rating for the device package type is not exceeded for the specific circuit operating condi- tions. !t is especially important to consider the maximum load current and ambient operating temperature of the circuit. Refer to the graphs in the Typical Operating Characteristics section for typical values of load current and power dissipation for different values of RSET. Shutdown Mode The WTA supply current varies with the value of RSET, but it is relatively independent of the differential input voltage and the output load. The WTAs achieve much of their high-speed performance by maintaining a con- stant level of current within the internal transistors, and steering some of this current to the output when a differential input voltage is applied. To reduce power dissipation when the output load does not need to be actively driven, the ISET pin can be used to place the 10 WTA into a low-power shutdown mode. If the ISET pin is disconnected from the RSET resistor and left open, the supply current of the MAX435 is reduced to approxi- mately 450A, while the supply current of the MAX436 is reduced to approximately 850nA. To reactivate the WTA, simply reconnect the ISET pin to the RSET resistor. Supply Current PIN Rset Typ, +25C) MAX435 MAX436 lset (11) | Open (00) 450 850 DC Accuracy The DC accuracy of circuits implemented with the MAX435/MAX436 is affected by several parameters, including: 1) input Offset Voltage 2) Output Offset Current 3) Accuracy of WTA Current Gain Factor (K) 4) Tolerance of External Transconductance and Load Impedances The input offset voltage of the WTA is caused by a mis- match of VBE voltages between the transistors in the am- plifier input stage - the same mechanism that produces input offset voltages in ordinary bipolar amplifiers. The input offset voltage is measured as the voltage between the transconductance terminals (from Z+ to Z-), with each of the inputs (IN+ and IN-) grounded and no transconduct- ance element in the circuit (2+ and Z- are open). In an actual application circuit, the input offset voltage causes a current to flow in the transconductance element when no differential voltage is applied at the input termi- nals. This current is then multiplied by K to produce an error current at the output terminal(s). Asecond source of DC error is the output offset current. This parameter is measured with no transconductance element in the circuit (Z+ and Z- are open) and the input terminals (IN+ and IN-) grounded. Output offset current is the current that flows from an output terminal of the WTA under these conditions. The output offset current is the result of imperfect matching of devices in the output current mirror(s) of the WTA. The output current caused by the input offset voltage, as discussed previously, is NOT included in the Output Offset Current specification, since that component of the total output current will vary with the value of the transconductance element. RAMAWideband Transconductance Amplifiers VIN i Vout RLeO = Rout | RL {e| Rout + Ri _ K__ Vf our) Py Vout = (vi) fk +Rz I Rout + Ri Rz= 0.150 Rout = 3.502 hy IS Z- TERMINAL INPUT IMPEDANCE. SINCE Rz IS TYPICALLY 0.152, IT CAN USUALLY BE IGNORED Figure 2. Finite Output-Impedance of WTA The total DC output voltage error (at each output for the MAX435) due to the input offset voltage and output offset current is calculated with the following formulas: MAX435: Vv VeRR+ = (RL+) [(los+) +(K) Gal v VerR- = (RL) [(los-) - (K) Ga! VERR(DIFF) = (VERR+) (VERR-) MAX436: VeRR = RL (los +K a) Where: VERR = Output-Voltage Error RL = Output Load Resistance Rt = Transconductance Element Resistance los = Output Offset Current Vos = Input Offset Voltage (from Z+ to Z-) K = WTACurrent Gain Variation of the current gain factor (K) between devices and over operating temperature is also a source of gain error in WTA applications. K for each of the devices is factory trimmed to an initial tolerance of +2.5%. The variation of K with operating temperature is listed in the Electrical Characteristics tables. MAAXLAA The finite output impedance also affects the amplifier gam. The output(s) are voltage-controlled current sources. An ideal current source would have an infinite output impedance, so that the output current would be independent of the load impedance. In practice, the MAX435/MAX436's output impedance is about 3.5kQ. As shown in Figure 2, the WTA output impedance (ROUT) is paralleled with the circuit load impedance (RL), reduc- ing the equivalent load impedance. After accounting for the finite WTA output impedance (RouT), the actual circuit gain is calculated with the following formula: Ay= Vout _/(K \} (Rout) (RL) VIN Rt || Rout + Ri The voltage gain error (A Ay) with respect to the theoret- ical gain (Av = K x RL/R}) is equal to: AAv RL Av RL+RouT Power-Supply Bypassing and Board Layout Although the WTA architecture eliminates closed-loop phase shift as a source of circuit oscillation, careful high-frequency circuit design methods should be used to optimize the performance of WTA circuits. Proper power-supply bypassing and board layout de- serve careful attention. It is recommended that a ground plane be used with the MAX435/MAX436. The ground plane should include the entire portion of the PC board that is not dedicated to a specific signal trace. Sockets are not recommended with the WTAs, because the additional pin-to-pin capacitance they introduce de- grades wideband performance. Keep the length of traces connecting to the WTA input terminals as short as possible to minimize signal reflections and/or inductive coupling of high-frequency signals to the WTA. if the input signals must travel more than a few inches, use controlled impedance lines or coaxial cables; all signals should be properly terminated. Minimize the PCB pad Figure 3. MAX435 Coaxial Cable Driving Circuit 11 SEVXVW/SEPXUNMAX435/MAX436 Wideband Transconductance Amplifiers INPUT 500mV/div +OUTPUT 200mV/div -QUTPUT 200mV/div TIME (ns} 5ns/div Rr = 400Q, Ri+ = RL- = 25 Figure 4. MAX435 Pulse Response Figure 5. MAX436 Coaxial Cable Driving Circuit INPUT 500mV/div QUTPUT 200mV/div TIME (ns) Sns/div t= E (ni Rt = 4000, RL =25Q Figure 6. MAX436 Pulse Response 12 area for input connections, to prevent capacitive cou- pling of stray high-frequency signais. Passive components used with the WTA should prefera- bly be surface mount, to minimize stray inductance. If surface-mount components are not used, component lead lengths should be kept to an absolute minimum. Bypass each power supply directly to the ground plane with a 0.22uF ceramic capacitor, placed as close to the supply pins as possible. Bypass the ISET pin with a0.22uF ceramic capacitor to the V+ pin. Keep capacitor lead lengths as short as possible to minimize series inductance; surface-mount (chip) capacitors are ideal for this application. Vout Figure 7. Summing Amplifier Capacitive Load Driving Since the WTA requires no feedback, phase shift due to capacitive loading of the output will not degrade the circuit stability. The primary effect of capacitive load- ing is a reduction in the output slew rate and bandwidth, which is limited by the. rate at which the WTA output current can charge the capacitive load. Avoid capac- itive coupling from the WTA output terminals to the input or transconductance terminals, since it introduces high- frequency feedback and possible oscillations. Application Circuits TheWTA's unique architecture allows the implementation of many unique application circuits, some of which have been included here. For the sake of clarity, bypass capacitors and the Rset resistor have not been shown in the following applications circuit schematics. The value of RseT is 5.9kQ for every application circuit in Figures 3-16. For every application circuit in this data sheet, each power supply was bypassed to GND with a 0.22uF ceramic capacitor. The IseT pin was bypassed to V+ with a 0.22u.F ceramic capacitor. MAAXLAAWideband Transconductance Amplifiers Vout POLE FREQUENCY = FP = RL 1 2m RLCL PASSBAND GAIN = (i 1 CORNER FREQUENCY = Fo =--_- my (RD OD PASSBAND GAIN = K fe) Figure 8. Lowpass Amplifier Figure 10. Highpass Amplifier GAIN (4B) -24 [FIGURE 8 R= 100Q Ri =25Q = 6.8nF tk 10k 100k 1M 10M 100M FREQUENCY (H2) 18 GAIN (dB) 10 CIRCUIT Ris 25a 30 = 6.8nF -% 1k 10k 100k 1M 10M 100M FREQUENCY (Hz) Figure 9. Lowpass Amplifier Gain vs. Frequency Coaxial Cable Drivers To maximize power transfer and minimize distortion of high- speed signals, transmission lines must be terminated at the source and receiving ends with the characteristic impedance of the line itself. A 500 coaxial cable requires an impedance of 50Q at each end for optimum performance. With voltage-mode amplifiers, the transmitting end of a coaxial cable is typically back-terminated with a resistor in series with the amplifier output, since the voltage-mode amplifiers output has a very low impedance. The WTA output is a current source, so its output imped- ance is relatively high (approximately 3.5kQ). Therefore, when driving coaxial cables with the WTA, the back-termi- nation resistor should be placed in parallel with the output impedance of the WTA, as shown in Figures 3 and 5. Note that back terminating cables in this manner will reduce the effective voltage gain of the circuit by a factor of 2. The pulse response of the circuit in Figure 3 is snown in Figure 4, while Figure 6 is the pulse response of Figure 5's circuit. MA AXIAA Figure 11. Highpass Amplifier Gain vs. Frequency Summing Amplifier Two or more signals can be summed together by simply tying the output terminals of WTAs together as shown in Figure 7. Lowpass Amplitier A parallel RC network at the amplifier output will result in the lowpass amplifier circuit of Figure 8, with a -3dB corner frequency of: 1 ~ (2m) (RL) (CL) The response of Figure 8's circuit with Rt = 100Q, CL = 6.8nF and RL = 2522 is plotted in Figure 9. Highpass Amplifier A series RC network between the transconductance terminals, as shown in the circuit of Figure 10, will produce a highpass amplifier. The response of this circuit is plotted in Figure 11, for Rt = 100, Ct =6.8nF and RL = 25. Fe 13 SEPXVWWSEPXUWMAX435/MAX436 Wideband Transconductance Amplifiers Vout 1 LOW CORNER FREQUENCY = FL ==- a FL= ap RD CO POLEF Y= Fp ==_ OLE FREQUENCY = Fe = Ea PASSBAND GAIN = (a 4 HIGH CORNER FREQUENCY = Fr = A De ALY (CD Q1S AFUNCTION OF PARASITICS OF Le AND Cr Figure 12. Bandpass Amplifier Figure 14, Tuned Amplifier GAIN (dB) OF 12 CIRCUIT: naw se 10k 400k 1M 10M 100M FREQUENCY (Hz} 40 30 14 CIRCUIT: Lt 20 10 0 GAIN (dB) -10 1M 40M 100M 1G FREQUENCY (Hz) Figure 13. Bandpass Amplifier Gain vs. Frequency Bandpass Amplifier Since there is no interaction between the transcon- ductance network impedance and the output load impedance, poles and zeros in the WTA transfer function can be independently set by the two imped- ance networks. The circuit in Figure 12 is a bandpass amplifier, with the low corner frequency set by the impedance of the transconductance net- work. The high corner frequency is set by the imped- ance of the RC network at the amplifier output. The passband gain is (K) x (RL/R). Figure 13 is a plot of the response of the circuit in Figure 12, with Rt = 100Q, Ct = 20nF, RL = 252, and CL = 395pF. 14 Figure 15. Tuned Amplifier Gain vs. Frequency Tuned Amplifier The circuit in Figure 14 is a tuned amplifier circuit, tuned to the resonant frequency of the LC transconductance network, or 1 Fos 2m Lt Ct The impedance of the transconductance network is a minimum at the resonant frequency, providing maximum amplifier gain at that frequency. The Q of the amplifier is a function of the parasitic components associated with the LC network. Figure 15 is the frequency response of the circuit in Figure 14, with Lt = 2.93nH and Ct = 9.9pF. MAAXIAAWideband Transconductance Amplifiers + Vin CRYSTAL alo lot. CENTER FREQUENCY = Fc = CRYSTAL FREQUENCY Figure 16. Crystal Tuned Amplifier GAIN (<8) 40 FREQUENCY (MHz) Figure 17. Crystal Tuned Amplifier Gain vs. Frequency The high-frequency gain of this circuit (beyond the reso- nant frequency) can be further reduced by increasing the load capacitance. Crystal Tuned Amplitier If a higher Q and more accurate control of the tuned amplifier frequency is required, a crystal can replace the tuned LC network as the WTA transconductance ele- ment, as shown in the "Crystal Tuned Amplifier circuit of Figure 16. The crystals impedance is a minimum at its resonant frequency, resulting in maximum gain for the amplifier circuit at that frequency. The frequency response of Figure 16's circuit with a 25MHz crystal is shown in Figure 17. Video Twisted-Pair Driver/Receiver Circuit For distances fess than about 5000 ft. (1500m), a single channel of baseband (composite) video can be trans- mitted with surprisingly high quality across twisted-pair cabling, saving significant cost over traditional coaxial cabling. When using twisted pair as a transmission medium, two things are of highest importance: balanced (differential) transmission, to minimize common-mode noise, and proper termination to minimize reflections. The MAX435/MAX436, with 53dB CMRR at 10MHz andhigh input and output impedance, are well suited for this application. Figure 18's circuit illustrates the MAX435 and MAX436 used as a driver/receiver circuit for twisted-pair video transmissions. One differential output MAX435 drives the balanced twisted pair from a ground-referred input sig- nal, eliminating the need for a balun transformer or two single-ended output drivers. +5V 0.22uF IN+ T 1000 a = Vv OT VIDEO MAxa6 | OUT ct Ve (OpF-500pF) 7- | 752 IN: 0.22uF rr 7 0.22uF a 47k 7 RSET VO EV Figure 18. Video Twisted Pair Driver/Receiver Circuit MAAXIAA 15 SEPXVWW/SEPXUWMAX435/MAX436 Wideband Transconductance Amplifiers The receiver circuit uses the single-ended output MAX436 for balanced to single-ended line conversion. The 1002 resistor from IN+ to IN- provides proper line termination. The transconductance network (from Z+ to Z-) performs the gain adjustment (+6dB), as well as line equalization. Line equalization is sometimes required to boost the high-frequency gain of the receiver to account for the limited bandwidth of the twisted pair. In Figure 18's circuit, compensation is achieved by adjust- ing R1 for proper brightness (to boost overall gain to compensate for ohmic losses), and C1 for best color (to add a zero/pole pair to extend the bandwidth). Since this equalization is done on the receiver end, the end user simply views the screen and adjusts the controls for the best picture. For many NTSC applications, this line equalization may not be necessary, since the NTSC monitor performs a fair amount of loss equalization by calibrating to the test pattems in the vertical interval test signal (VITS). ___ Ordering information (continued) PART TEMP. RANGE PIN-PACKAGE MAX436CPD O'C to +70C 14 Plastic DIP MAX436CSD 0C to +70C 14S0 MAX436C/D 0C to +70C Dice MAX436EPD -40C to +85C 14 Plastic DIP MAX436ESD -40C to +85C 14S0 MAX436MJD _-55C to + 125C 14 CERDIP * Dice are specified at Ta = +25C, DC parameters only. Chip Topography MAX496/MAX436 Ve TE Lod af Pir tour er iiiy pee 072" : | . eer (1.830 mm) z |! | | ! { * v- 2a a ~ V- tour ac.) .072 (1.830 mm) () are for MAX436 only. TRANSISTOR COUNT: 230; SUBSTRATE CONNECTED TO V-. Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied. Maxim reserves the right to change the circuitry and specifications without notice at any time. 16 Maxim integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 (408) 737-7600 1993 Maxim Integrated Products Printed USA AAAXMIAA is a registered trademark of Maxim Integrated Products.