PVIN SW
AGND
FB
PGOOD
EN
SS/TRK
AVIN
COMP
LM20146 L
RF
VCC
PGND
CIN
RC1
CC1
VIN
CSS
CVCC
COUT
VOUT
RFB2
RFB1
CFRT
RT
(optional)
LM20146
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SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
LM20146 6A, Adjustable Frequency Synchronous Buck Regulator
Check for Samples: LM20146
1FEATURES DESCRIPTION
The LM20146 is a full featured adjustable frequency
2• Input Voltage Range 2.95V to 5.5V synchronous buck regulator capable of delivering up
• Accurate Current Limit Minimizes Inductor to 6A of continuous output current. The current mode
Size control loop can be compensated to be stable with
• 97% Peak Efficiency virtually any type of output capacitor. For most cases,
compensating the device only requires two external
• Adjustable Switching Frequency (250 kHz to components, providing maximum flexibility and ease
750 kHz) of use. The device is optimized to work over the input
• 16mΩand 20mΩIntegrated FET Switches voltage range of 2.95V to 5.5V making it suitable for
• Starts up into Pre-biased Loads a wide variety of low voltage systems.
• Output Voltage Tracking The device features internal over voltage protection
(OVP) and over current protection (OCP) circuits for
• Peak Current Mode Control increased system reliability. A precision enable pin
• Adjustable Output Voltage Down to 0.8V and integrated UVLO allows the turn-on of the device
• Adjustable Soft-Start with External Capacitor to be tightly controlled and sequenced. Start-up
• Precision Enable Pin with Hysteresis inrush currents are limited by both an internally fixed
and externally adjustable Soft-Start circuit. Fault
• Integrated OVP, UVLO, Power Good and detection and supply sequencing are possible with
Thermal Shutdown the integrated power good circuit.
• 16-Pin HTSSOP Exposed Pad Package The LM20146 is designed to work well in multi-rail
power supply architectures. The output voltage of the
APPLICATIONS device can be configured to track a higher voltage rail
• Simple to Design, High Efficiency Point of using the SS/TRK pin. If the output of the LM20146 is
Load Regulation from a 5V or 3.3V Bus pre-biased at startup it will not pull the ouput low.
• High Performance DSPs, FPGAs, ASICs and The frequency of this device can be adjusted from
Microprocessors 250 kHz to 750 kHz by connecting an external
resistor from the RT pin to ground.
• Broadband, Networking and Optical
Communications Infrastructure The LM20146 is offered in a 16-pin HTSSOP
package with an exposed pad that can be soldered to
the PCB, eliminating the need for bulky heatsinks.
Typical Application Circuit
1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
2All trademarks are the property of their respective owners.
PRODUCTION DATA information is current as of publication date. Copyright © 2008–2013, Texas Instruments Incorporated
Products conform to specifications per the terms of the Texas
Instruments standard warranty. Production processing does not
necessarily include testing of all parameters.
FB
PGOOD
COMP
NC
PVIN
PVIN
SW
SS/TRK
AVIN
VCC
EN
PGND
PGND
SW
AGND
RT
EP
1
2
3
4
5
6
7
8
16
15
14
13
12
11
10
9
LM20146
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
www.ti.com
Connection Diagram
Figure 1. Top View
16-Pin HTSSOP
See PWP Package
PIN DESCRIPTIONS
Pin # Name Description
1 SS/TRK Soft-Start or Tracking control input. An internal 5 µA current source charges an external capacitor to set the
Soft-Start ramp rate. If driven by a external source less than 800 mV, this pin overrides the internal reference
that sets the output voltage. If left open, an internal 1ms Soft-Start ramp is activated.
2 FB Feedback input to the error amplifier from the regulated output. This pin is connected to the inverting input of
the internal transconductance error amplifier. An 800 mV reference connected to the non-inverting input of the
error amplifier sets the closed loop regulation voltage at the FB pin.
3 PGOOD Power good output signal. Open drain output indicating the output voltage is regulating within tolerance. A
pull-up resistor of 10 kΩto 100 kΩis recommend for most applications.
4 COMP External compensation pin. Connect a resistor and capacitor to this pin to compensate the device.
5 NC Connect this pin to GND to ensure proper operation
6,7 PVIN Input voltage to the power switches inside the device. These pins should be connected together at the device.
A low ESR capacitor should be placed near these pins to stabilize the input voltage.
8,9 SW Switch pin. The PWM output of the internal power switches.
10,11 PGND Power ground pin for the internal power switches.
12 EN Precision enable input for the device. An external voltage divider can be used to set the device turn-on
threshold. If not used the EN pin should be connected to PVIN.
13 VCC Internal 2.7V sub-regulator. This pin should be bypassed with a 1 µF ceramic capacitor.
14 AVIN Analog input supply that generates the internal bias. Must be connected to PVIN through a low pass RC filter.
15 AGND Quiet analog ground for the internal bias circuitry.
16 RT Frequency adjust pin. Connecting a resistor on this pin to ground will set the oscillator frequency.
EP Exposed Pad Exposed metal pad on the underside of the package with a weak electrical connection to ground. It is
recommended to connect this pad to the PC board ground plane in order to improve heat dissipation.
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
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Absolute Maximum Ratings(1)(2)
Voltages from the indicated pins to GND
AVIN, PVIN, EN, PGOOD, SS/TRK, COMP, FB, RT -0.3V to +6V
Storage Temperature -65°C to 150°C
Junction Temperature 150°C
Power Dissipation(3) 2.6W
Lead Temperature (Soldering, 10 sec) 260°C
Minimum ESD Rating(4) ±2kV
(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for
which the device is intended to be functional, but do not ensure specific performance limits. For ensured specifications and test
conditions, see the Electrical Characteristics.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
(3) The maximum allowable power dissipation is a function of the maximum junction temperature, TJ_MAX, the junctions-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD_MAX = (TJ_MAX – TA)/θJA. The maximum power dissipations of 2.6W is determined using TA= 25°C, θJA = 25°C/W, and TJ_MAX
= 125°C. The θJA specification of 25°C/W listed in the electrical characteristics table is measured with the part surface mounted to a 2" x
2" FR4 4 layer board. See Figure 36 for more detailed θJA information.
(4) The human body model is a 100 pF capacitor discharged through a 1.5 kΩresistor to each pin.
Operating Ratings
PVIN, AVIN to GND 2.95V to 5.5V
Junction Temperature −40°C to + 125°C
Electrical Characteristics
Unless otherwise stated, the following conditions apply: AVIN = PVIN = VIN = 5V. Limits in standard type are for TJ= 25°C
only, limits in bold face type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits
are specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at
TJ= 25°C, and are provided for reference purposes only.
Symbol Parameter Conditions Min Typ Max Unit
VFB Feedback pin voltage VIN = 2.95V to 5.5V 0.788 0.8 0.81 V
2
ΔVOUT/ΔIOUT Load Regulation IOUT = 100 mA to 6A 0.08 %/A
ICL Switch Current Limit Threshold VIN = 3.3V 7.35 8.5 9.35 A
RDS_ON High-Side Switch On Resistance ISW = 3.5A 20 27 mΩ
RDS_ON Low-Side Switch On Resistance ISW = 3.5A 16 23 mΩ
IQOperating Quiescent Current Non-switching, VFB = VCOMP 3.5 6mA
ISD Shutdown Quiescent current VEN = 0V 75 180 µA
VUVLO VIN Under Voltage Lockout Rising VIN 2.45 2.7 2.95 V
VUVLO_HYS VIN Under Voltage Lockout Hysteresis Falling VIN 45 100 mV
VVCC VCC Voltage IVCC = 0 µA 2.45 2.7 2.95 V
ISS Soft-Start Pin Source Current VSS/TRK = 0V 24.5 7µA
VTRACK SS/TRK Accuracy, VSS - VFB VSS/TRK = 0.4V -10 315 mV
Oscillator
FOSCH Oscillator Frequency RT= 49.9 kΩ675 750 825 kHz
FOSCL Oscillator Frequency RT= 249 kΩ225 260 290 kHz
DCMAX Maximum Duty Cycle ILOAD = 0A 85 %
TON_TIME Minimum On Time 100 ns
TCL_BLANK Current Sense Blanking Time After Rising VSW 80 ns
Error Amplifier and Modulator
IFB Feedback pin bias current VFB = 0.8V 1 100 nA
ICOMP_SRC COMP Output Source Current VFB = 0.6V, VCOMP = 0.5V 80 100 µA
ICOMP_SNK COMP Output Sink Current VFB = 1.0V, VCOMP = 0.6V 80 100 µA
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Electrical Characteristics (continued)
Unless otherwise stated, the following conditions apply: AVIN = PVIN = VIN = 5V. Limits in standard type are for TJ= 25°C
only, limits in bold face type apply over the junction temperature (TJ) range of -40°C to +125°C. Minimum and Maximum limits
are specified through test, design, or statistical correlation. Typical values represent the most likely parametric norm at
TJ= 25°C, and are provided for reference purposes only.
Symbol Parameter Conditions Min Typ Max Unit
Gm Error Amplifier Transconductance ICOMP = ± 50 µA 450 510 600 µmho
AVOL Error Amplifier Voltage Gain 2000 V/V
Power Good
VOVP Over Voltage Protection Rising Threshold With respect to VFB 105 108 111 %
VOVP_HYS Over Voltage Protection Hysteresis 2 3%
VPGTH PGOOD Rising Threshold With respect to VFB 92 94 96 %
VPGHYS PGOOD Falling Hysteresis 2 3%
TPGOOD PGOOD deglitch time 16 µs
IOL PGOOD Low Sink Current VPGOOD = 0.4V 0.6 1 mA
IOH PGOOD High Leakage Current VPGOOD = 5V 5 100 nA
Enable
VIH_EN EN Pin turn-on Threshold VEN Rising 1.08 1.18 1.28 V
VEN_HYS EN Pin Hysteresis 66 mV
Thermal Shutdown
TSD Thermal Shutdown 160 °C
TSD_HYS Thermal Shutdown Hysteresis 10 °C
Thermal Resistance
θJA Junction to Ambient See(1) 25 °C/W
(1) The maximum allowable power dissipation is a function of the maximum junction temperature, TJ_MAX, the junctions-to-ambient thermal
resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation at any ambient temperature is calculated
using: PD_MAX = (TJ_MAX – TA)/θJA. The maximum power dissipations of 2.6W is determined using TA= 25°C, θJA = 25°C/W, and TJ_MAX
= 125°C. The θJA specification of 25°C/W listed in the electrical characteristics table is measured with the part surface mounted to a 2" x
2" FR4 4 layer board. See Figure 36 for more detailed θJA information.
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VOUT = 1.2V
VOUT = 1.5V
VOUT = 2.5V
L=SPM6530T-1R0M120
VOUT = 1.2V
VOUT = 1.5V
VOUT = 2.5V
VOUT = 3.3V
L=SPM6530T-1R0M120
LM20146
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Typical Performance Characteristics
Unless otherwise specified: CIN = COUT = 100µF, L = 1.0µH (TDK SPM6530T-1R0M120), VIN = 5V, VOUT = 1.2V, RLOAD =
1.2Ω, fSW = 500 kHz, TA= 25°C for efficiency curves, loop gain plots and waveforms, and TJ= 25°C for all others.
Efficiency Efficiency
vs. vs.
Load Current (VIN = 5V) Load Current (VIN = 3.3V)
Figure 2. Figure 3.
High-Side FET Resistance Low-Side FET Resistance
vs. vs.
Temperature (TJ) Temperature (TJ)
Figure 4. Figure 5.
Error Amplifier Gain
vs.
Frequency Line Regulation
Figure 6. Figure 7.
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Typical Performance Characteristics (continued)
Unless otherwise specified: CIN = COUT = 100µF, L = 1.0µH (TDK SPM6530T-1R0M120), VIN = 5V, VOUT = 1.2V, RLOAD =
1.2Ω, fSW = 500 kHz, TA= 25°C for efficiency curves, loop gain plots and waveforms, and TJ= 25°C for all others.
Feedback Pin Voltage
vs.
Load Regulation Temperature (TJ)
Figure 8. Figure 9.
Switching Frequency Switching Frequency
vs. vs.
Temperature (TJ) RT
Figure 10. Figure 11.
Quiescent Current Shutdown Current
vs. vs.
VIN (Not Switching) Temperature (TJ)
Figure 12. Figure 13.
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VOUT (100 mV/DIV)
IOUT (2A/DIV)
TIME (100 és/DIV)
IOUT (600 mA to 6A)
LM20146
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SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
Typical Performance Characteristics (continued)
Unless otherwise specified: CIN = COUT = 100µF, L = 1.0µH (TDK SPM6530T-1R0M120), VIN = 5V, VOUT = 1.2V, RLOAD =
1.2Ω, fSW = 500 kHz, TA= 25°C for efficiency curves, loop gain plots and waveforms, and TJ= 25°C for all others.
Enable Threshold UVLO Threshold
vs. vs.
Temperature (TJ) Temperature (TJ)
Figure 14. Figure 15.
Peak Current Limit Peak Current Limit
vs. vs.
Temperature (TJ) VOUT
Figure 16. Figure 17.
Peak Current Limit
vs.
VIN Load Transient Response
Figure 18. Figure 19.
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VOUT (500 mV/DIV)
RLOAD = 1.2Ö
RFB1 = 0Ö
VSS/TRK (500 mV/DIV)
TIME (4 ms/DIV)
VEN (1V/DIV)
VOUT (500 mV/DIV)
RLOAD = 1.2Ö
TIME (200 és/DIV)
VOUT (500 mV/DIV)
RLOAD = 1.2Ö
VEN (5V/DIV)
CSS/TRK = None
CSS/TRK = 33 nF
CSS/TRK = 68 nF
TIME (2 ms/DIV)
VOUT (50 mV/DIV)
VIN (1V/DIV)
TIME (100 és/DIV)
VIN (3V to 5V)
LM20146
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
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Typical Performance Characteristics (continued)
Unless otherwise specified: CIN = COUT = 100µF, L = 1.0µH (TDK SPM6530T-1R0M120), VIN = 5V, VOUT = 1.2V, RLOAD =
1.2Ω, fSW = 500 kHz, TA= 25°C for efficiency curves, loop gain plots and waveforms, and TJ= 25°C for all others.
Line Transient Response Start-Up (Soft-Start)
Figure 20. Figure 21.
Start-Up (Tracking) Power Down
Figure 22. Figure 23.
Short Circuit Input Current PGOOD
vs. vs.
VIN IPGOOD
Figure 24. Figure 25.
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8.5
+
-
COMP
CONTROL
LOGIC
CURRENT
LIMIT
OVERVOLTAGE
UNDERVOLTAGE
ERROR AMP
PWM COMPARATOR
SS/TRK
PGOOD
+
-
+2.7V
REGULATOR
PVIN
PVIN
THERMAL
PROTECTION
PGND
SW
AVIN
CURRENT SENSE
VCC
1.18V
UVLO
2.7V
AGND
OSCILLATOR
+
752 mV
864 mV PG-L
PG-L
2.7V
5 PA
gm = 510 Pmho
800 mV
+
-
+
DISCHARGE
DISCHARGE
SLOPE COMP
DIODE
EMULATION
+
-
VREF
+
-
FB
EN
+
-
+
-
(50 Ps)
+
-
+
-
RT
PVIN
LM20146
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SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
Block Diagram
Figure 26.
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LM20146
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OPERATION DESCRIPTION
General
The LM20146 switching regulator features all of the functions necessary to implement an efficient low voltage
buck regulator using a minimum number of external components. This easy to use regulator features two
integrated switches and is capable of supplying up to 6A of continuous output current. The regulator utilizes peak
current mode control with nonlinear slope compensation to optimize stability and transient response over the
entire output voltage range. Peak current mode control also provides inherent line feed-forward, cycle-by-cycle
current limiting and easy loop compensation. The switching frequency can be varied from 250 kHz to 750 kHz.
The device can operate at high switching frequency allowing use of a small inductor while still achieving high
efficiency. The precision internal voltage reference allows the output to be set as low as 0.8V. Fault protection
features include: current limiting, thermal shutdown, over voltage protection, and shutdown capability. The device
is available in the HTSSOP package featuring an exposed pad to aid thermal dissipation. The LM20146 can be
used in numerous applications to efficiently step-down from a 5V or 3.3V bus. The typical application circuit for
the LM20146 is shown in Figure 29 in the design guide.
Precision Enable
The enable (EN) pin allows the output of the device to be enabled or disabled with an external control signal.
This pin is a precision analog input that enables the device when the voltage exceeds 1.18V (typical). The EN pin
has 66 mV of hysteresis and will disable the output when the enable voltage falls below 1.11V (typical). If the EN
pin is not used, it should be connected to VIN. Since the enable pin has a precise turn-on threshold it can be
used along with an external resistor divider network from VIN to configure the device to turn-on at a precise input
voltage. The precision enable circuitry will remain active even when the device is disabled.
Peak current Mode Control
In most cases, the peak current mode control architecture used in the LM20146 only requires two external
components to achieve a stable design. The compensation can be selected to accommodate any output
capacitor type or value. The external compensation also allows the user to set the crossover frequency and
optimize the transient performance of the device.
For duty cycles above 50% all current mode control buck converters require the addition of an artificial ramp to
avoid sub-harmonic oscillation. This artificial linear ramp is commonly referred to as slope compensation. What
makes the LM20146 unique is the amount of slope compensation will change depending on the output voltage.
When operating at high output voltages the device will have more slope compensation than when operating at
lower output voltages. This is accomplished in the LM20146 by using a non-linear parabolic ramp for the slope
compensation. The parabolic slope compensation of the LM20146 is much better than the traditional linear slope
compensation because it optimizes the stability of the device over the entire output voltage range.
Current Limit
The precise current limit of the LM20146 is set at the factory to be within 10% over the entire operating
temperature range. This enables the device to operate with smaller inductors that have lower saturation currents.
When the peak inductor current reaches the current limit threshold, an over current event is triggered and the
internal high-side FET turns off and the low-side FET turns on allowing the inductor current to ramp down until
the next switching cycle. For each sequential over-current event, the reference voltage is decremented and PWM
pulses are skipped resulting in a current limit that does not aggressively fold back for brief over-current events,
while at the same time providing frequency and voltage foldback protection during hard short circuit conditions.
Soft-Start and Voltage Tracking
The SS/TRK pin is a dual function pin that can be used to set the start up time or track an external voltage
source. The start up or Soft-Start time can be adjusted by connecting a capacitor from the SS/TRK pin to ground.
The Soft-Start feature allows the regulator output to gradually reach the steady state operating point, thus
reducing stresses on the input supply and controlling start up current. If no Soft-Start capacitor is used the device
defaults to the internal Soft-Start circuitry resulting in a start up time of approximately 1 ms. For applications that
require a monotonic start up or utilize the PGOOD pin, an external Soft-Start capacitor is recommended. The
SS/TRK pin can also be set to track an external voltage source. The tracking behavior can be adjusted by two
external resistors connected to the SS/TRK pin as shown in Figure 34. in the design guide.
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VSS
VFB
VENABLE
VPGOOD
VSWITCH
IL
OVP-
LOW SIDE ON UVP PRE-BIASED
STARTUP CONDITION CURRENT LIMIT
CURRENT LIMIT
DISABLE
VOVP
VUVP
TPGOOD
VFB FOLDBACK
0.8V
2.7V
VOVPHYS
VPGHYS
Soft Start Time
FSW FOLDBACK
`
0.8V
0.0V `
LM20146
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SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
Pre-Bias Start up Capability
The LM20146 is in a pre-biased state when the device starts up with an output voltage greater than zero. This
often occurs in many multi-rail applications such as when powering an FPGA, ASIC, or DSP. In these
applications the output can be pre-biased through parasitic conduction paths from one supply rail to another.
Even though the LM20146 is a synchronous converter it will not pull the output low when a prebias condition
exists. During start up the LM20146 will not sink current until the Soft-Start voltage exceeds the voltage on the
FB pin. Since the device can not sink current it protects the load from damage that might otherwise occur if
current is conducted through the parasitic paths of the load.
Power Good and Over Voltage Fault Handling
The LM20146 has built in under and over voltage comparators that control the power switches. Whenever there
is an excursion in output voltage above the set OVP threshold, the part will terminate the present on-pulse, turn-
on the low-side FET, and pull the PGOOD pin low. The low-side FET will remain on until either the FB voltage
falls back into regulation or the zero cross detection is triggered which in turn tri-states the FETs. If the output
reaches the UVP threshold the part will continue switching and the PGOOD pin will be asserted and go low.
Typical values for the PGOOD resistor are on the order of 100 kΩor less. To avoid false tripping during transient
glitches the PGOOD pin has 16 µs of built in deglitch time to both rising and falling edges. The powergood
behavior for fault conditions is illustrated in Figure 27.
Figure 27. Powergood Behavior
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UVLO
The LM20146 has a built-in under-voltage lockout protection circuit that keeps the device from switching until the
input voltage reaches 2.7V (typical). The UVLO threshold has 45 mV of hysteresis that keeps the device from
responding to power-on glitches during start up. If desired the turn-on point of the supply can be changed by
using the precision enable pin and a resistor divider network connected to VIN as shown in Figure 33. in the
design guide.
Thermal Protection
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event that the maximum
junction temperature is exceeded. When activated, typically at 160°C, the LM20146 tri-states the power FETs
and resets soft start. After the junction cools to approximately 150°C, the part starts up using the normal start up
routine. This feature is provided to prevent catastrophic failures from accidental device overheating.
Light Load Operation
The LM20146 offers increased efficiency when operating at light loads. Whenever the load current is reduced to
a point where the peak to peak inductor ripple current is greater than two times the load current, the part will
enter the diode emulation mode preventing significant negative inductor current. The point at which this occurs is
the critical conduction boundary and can be calculated by the following equation:
(1)
Several diagrams are shown in Figure 28 illustrating continuous conduction mode (CCM), discontinuous
conduction mode, and the boundary condition.
It can be seen that in diode emulation mode, whenever the inductor current reaches zero the SW node will
become high impedance. Ringing will occur on this pin as a result of the LC tank circuit formed by the inductor
and the parasitic capacitance at the node. If this ringing is of concern an additional RC snubber circuit can be
added from the switch node to ground.
At very light loads, usually below 100 mA, several pulses may be skipped in between switching cycles, effectively
reducing the switching frequency and further improving light-load efficiency.
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Time (s)
Discontinuous Conduction Mode (DCM)
IPeak
Time (s)
Time (s)
Time (s)
Time (s)
Discontinuous Conduction Mode (DCM)
DCM - CCM Boundary
Continuous Conduction Mode (CCM)
Continuous Conduction Mode (CCM)
Switchnode Voltage Switchnode VoltageInductor CurrentInductor CurrentInductor Current
VIN
IAVERAGE
IAVERAGE
VIN
LM20146
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Figure 28. Modes of Operation for LM20146
Design Guide
This section walks the designer through the steps necessary to select the external components to build a fully
functional power supply. As with any DC-DC converter numerous trade-offs are possible to optimize the design
for efficiency, size, or performance. These will be taken into account and highlighted throughout this discussion.
To facilitate component selection discussions the circuit shown in Figure 29 below may be used as a reference.
Unless otherwise indicated all formulas assume units of Amps (A) for current, Farads (F) for capacitance,
Henries (H) for inductance and Volts (V) for voltage.
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VIN
IL AVG = IOUT 'IL
Time
Time
IL
VSW
LMIN = (VIN - VOUT) x D
'iL x fSW
D = VOUT
VIN
CIN
PVIN SW
GND
FB
PGOOD
RFB1
RFB2
COUT
EN
CSS
SS/TRK
AVIN
CF
CC1
COMP
RC1
VIN LM20146 L
RF
VCC
CVCC
VOUT
PGND
VIN
RPG
VPG
RT
RT
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Figure 29. Typical Application Circuit
The first equation to calculate for any buck converter is duty-cycle. Ignoring conduction losses associated with
the FETs and parasitic resistances it can be approximated by:
(2)
Inductor Selection (L)
The inductor value is determined based on the operating frequency, load current, ripple current, and duty cycle.
The inductor selected should have a saturation current rating greater than the peak current limit of the device.
Keep in mind the specified current limit does not account for delay of the current limit comparator, therefore the
current limit in the application may be higher than the specified value. To optimize the performance and prevent
the device from entering current limit at maximum load, the inductance is typically selected such that the ripple
current, ΔiL, is less than 30% of the rated output current. Figure 30, shown below illustrates the switch and
inductor ripple current waveforms. Once the input voltage, output voltage, operating frequency, and desired ripple
current are known, the minimum value for the inductor can be calculated by the formula shown below:
(3)
Figure 30. Switch and Inductor Current Waveforms
If needed, slightly smaller value inductors can be used, however, the peak inductor current, IOUT +ΔiL/2, should
be kept below the peak current limit of the device. In general, the inductor ripple current, ΔiL, should be greater
than 10% of the rated output current to provide adequate current sense information for the current mode control
loop. If the ripple current in the inductor is too low, the control loop will not have sufficient current sense
information and can be prone to instability.
14 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: LM20146
IIN-RMS = IOUT D(1 - D)
VDROOP = 'IOUTSTEP x RESR + L x 'IOUTSTEP2
COUT x (VIN - VOUT)
'VOUT = 'iL x 1
8 x fSW x COUT
RESR +
LM20146
www.ti.com
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
Output Capacitor Selection (COUT)
The output capacitor, COUT, filters the inductor ripple current and provides a source of charge for transient load
conditions. A wide range of output capacitors may be used with the LM20146 that provide excellent performance.
The best performance is typically obtained using ceramic, SP, or OSCON type chemistries. Typical trade-offs are
that the ceramic capacitor provides extremely low ESR to reduce the output ripple voltage and noise spikes,
while the SP and OSCON capacitors provide a large bulk capacitance in a small volume for transient loading
conditions.
When selecting the value for the output capacitor the two performance characteristics to consider are the output
voltage ripple and transient response. The output voltage ripple can be approximated by using the formula shown
below.
where
•ΔVOUT (V) is the amount of peak to peak voltage ripple at the power supply output
• RESR (Ω) is the series resistance of the output capacitor
• fSW(Hz) is the switching frequency
• COUT (F) is the output capacitance used in the design (4)
The amount of output ripple that can be tolerated is application specific; however a general recommendation is to
keep the output ripple less than 1% of the rated output voltage. Keep in mind ceramic capacitors are sometimes
preferred because they have very low ESR; however, depending on package and voltage rating of the capacitor
the value of the capacitance can drop significantly with applied voltage. The output capacitor selection will also
affect the output voltage droop during a load transient. The peak droop on the output voltage during a load
transient is dependent on many factors; however, an approximation of the transient droop ignoring loop
bandwidth can be obtained using the following equation.
where
• COUT (F) is the minimum required output capacitance
• L (H) is the value of the inductor
• VDROOP (V) is the output voltage drop ignoring loop bandwidth considerations
•ΔIOUTSTEP (A) is the load step change
• RESR (Ω) is the output capacitor ESR
• VIN (V) is the input voltage
• VOUT (V) is the set regulator output voltage (5)
Both the tolerance and voltage coefficient of the capacitor needs to be examined when designing for a specific
output ripple or transient drop target.
Input Capacitor Selection (CIN)
Good quality input capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the
switch current during the on-time. In general it is recommended to use a ceramic capacitor for the input as they
provide both a low impedance and small footprint. One important note is to use a good dielectric for the ceramic
capacitor such as X5R or X7R. These provide better over temperature performance and also minimize the DC
voltage derating that occurs on Y5V capacitors. For most applications, a 22 µF, X5R, 6.3V input capacitor is
sufficient; however, additional capacitance may be required if the connection to the input supply is far from the
PVIN pins. The input capacitor should be placed as close as possible PVIN and PGND pins of the device.
Non-ceramic input capacitors should be selected for RMS current rating and minimum ripple voltage. A good
approximation for the required ripple current rating is given by the relationship:
(6)
Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 15
Product Folder Links: LM20146
RT = 78000
fSW - 55
RFB1 = - 1
VOUT
0.8 x RFB2
LM20146
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
www.ti.com
As indicated by the RMS ripple current equation, highest requirement for RMS current rating occurs at 50% duty
cycle. For this case, the RMS ripple current rating of the input capacitor should be greater than half the output
current. For best performance, low ESR ceramic capacitors should be placed in parallel with higher capacitance
capacitors to provide the best input filtering for the device.
Setting the output Voltage (RFB1, RFB2)
The resistors RFB1 and RFB2 are selected to set the output voltage for the device. Table 1, shown below, provides
suggestions for RFB1 and RFB2 for common output voltages.
Table 1. Suggested Values for RFB1 and RFB2
RFB1(kΩ) RFB2(kΩ) VOUT
short open 0.8
4.99 10 1.2
8.87 10.2 1.5
12.7 10.2 1.8
21.5 10.2 2.5
31.6 10.2 3.3
If different output voltages are required, RFB2 should be selected to be between 4.99 kΩto 49.9 kΩand RFB1 can
be calculated using the equation below.
(7)
Adjusting the Operating Frequency (RT)
The operating frequency of the LM20146 can be adjusted by connecting a resistor from the RT pin to ground.
The equation shown below can be used to calculate the value of RTfor a given operating frequency.
where
• fSW is the switching frequency in kHz
• RTis the frequency adjust resistor in kΩ(8)
Please refer to the curve Oscillator Frequency verses RTin the Typical Performance Characteristics section If the
RTresistor is omitted the device will not operate.
Loop Compensation (RC1, CC1)
The purpose of loop compensation is to meet static and dynamic performance requirements while maintaining
adequate stability. Optimal loop compensation depends on the output capacitor, inductor, load, and the device
itself. Table 2 below gives values for the compensation network that will result in a stable system when using a
100 µF, 6.3V ceramic X5R output capacitor and 1 µH inductor.
Table 2. Recommended Compensation for
COUT = 100 µF, L = 1.5 µH & fSW = 500 kHz
VIN VOUT CC1 (nF) RC1 (kΩ)
5.00 3.30 2.2 15.4
5.00 2.50 2.2 13.3
5.00 1.80 2.2 10.7
5.00 1.50 2.2 9.31
5.00 1.20 2.2 7.87
5.00 0.80 2.7 4.42
3.30 2.50 2.7 8.45
3.30 1.80 2.7 7.5
16 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: LM20146
COMP
CC1
RC1
CC2
LM20146
(optional)
AM
fSW/2
0 dB
FREQUENCY (Hz)
GAIN (dB)
Error Amp Zero, fZ(EA)
Complex Double Pole, fP(MOD)
Optional Error Amp
Pole, fP2(EA)
0 dB
0 dB
AEA + AM
Error Amplifier
Transfer Function Modulator and Output Filter
Transfer Function
Compensated Open
Loop Transfer Function
AEA
Error Amp Pole, fP1(EA)
Complex Double Pole, fP(MOD)
Output Filter Zero, fZ(FIL)
Output Filter Pole, fP(FIL)
fC
Error Amp Pole, fP(EA)
LM20146
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SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
Table 2. Recommended Compensation for
COUT = 100 µF, L = 1.5 µH & fSW = 500 kHz (continued)
VIN VOUT CC1 (nF) RC1 (kΩ)
3.30 1.50 2.7 6.81
3.30 1.20 2.7 5.9
3.30 0.80 2.7 4.32
If the desired solution differs from the table above the loop transfer function should be analyzed to optimize the
loop compensation. The overall loop transfer function is the product of the power stage and the feedback network
transfer functions. For stability purposes, the objective is to have a loop gain slope that is -20db/decade from a
very low frequency to beyond the crossover frequency. Figure 31, shown below, shows the transfer functions for
power stage, feedback/compensation network, and the resulting closed loop system for the LM20146.
Figure 31. LM20146 Loop Compensation
The power stage transfer function is dictated by the modulator, output LC filter, and load; while the feedback
transfer function is set by the feedback resistor ratio, error amp gain, and external compensation network.
To achieve a -20dB/decade slope, the error amplifier zero, located at fZ(EA), should positioned to cancel the
output filter pole (fP(FIL)). An additional error amp pole, located at fP2(EA), can be added to cancel the output filter
zero at fZ(FIL). Cancellation of the output filter zero is recommended if larger value, non-ceramic output capacitors
are used.
Compensation of the LM20146 is achieved by adding an RC network as shown in Figure 32 below.
Figure 32. Compensation Network for LM20146
Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LM20146
CC2 = COUT x RESR
RC1
fZ(FIL) = 1
2 x S x COUT x RESR
RC1 =x
CC1
COUT
IOUT
VOUT +D x fSW
48750*VIN
-1
+
1-D
fSW x L 1
2 x fSW x L
-
LM20146
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
www.ti.com
A good starting value for CC1 for most applications is 3.3 nF. Once the value of CC1 is chosen the value of RC
should be calculated using the equation below to cancel the output filter pole (FP(FIL)) as shown in Figure 31.
(9)
A higher crossover frequency can be obtained, usually at the expense of phase margin, by lowering the value of
CC1 and recalculating the value of RC1. Likewise, increasing CC1 and recalculating RC1 will provide additional
phase margin at a lower crossover frequency. As with any attempt to compensate the LM20146 the stability of
the system should be verified for desired transient droop and settling time.
If the output filter zero, FZ(FIL) approaches the crossover frequency (FC), an additional capacitor (CC2) should be
placed at the COMP pin to ground. This capacitor adds a pole to cancel the output filter zero assuring the
crossover frequency will occur before the double pole at fSW/2 degrades the phase margin. The output filter zero
is set by the output capacitor value and ESR as shown in the equation below.
(10)
If needed, the value for CC2 should be calculated using the equation shown below.
where
• RESR is the output capacitor series resistance
• RC1 is the calculated compensation resistance (11)
AVIN Filtering Components (CFand RF)
To prevent high frequency noise spikes from disturbing the sensitive analog circuitry connected to the AVIN and
AGND pins, a high frequency RC filter is required between PVIN and AVIN. These components are shown in
Figure 29. as CFand RF. The required value for RFis 1Ω. CFmust be used. Recommended value of CFis 1.0
µF. The filter capacitor, CFshould be placed as close to the IC as possible with a direct connection from AVIN to
AGND. A good quality X5R or X7R ceramic capacitor should be used for CF.
Sub-Regulator Bypass Capacitor (CVCC)
The capacitor at the VCC pin provides noise filtering and stability for the internal sub-regulator. The
recommended value of CVCC should be no smaller than 1 µF and no greater than 10 µF. The capacitor should be
a good quality ceramic X5R or X7R capacitor. In general, a 1 µF ceramic capacitor is recommended for most
applications.
Setting the Start up Time (CSS)
The addition of a capacitor connected from the SS pin to ground sets the time at which the output voltage will
reach the final regulated value. Larger values for CSS will result in longer start up times. Table 3, shown below
provides a list of soft start capacitors and the corresponding typical start up times.
Table 3. Start Up Times for Different Soft-Start Capacitors
Start Up Time (ms) CSS (nF)
1 none
5 33
10 68
15 100
20 120
18 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: LM20146
SS/TRK
VOUT1
R1
R2
EN LM20146
External
Power Supply
VOUT2
RA = - 1
VTO
VIH_EN x RB
EN
VOUT1
RA
RB
LM20146
External
Power Supply
VOUT2
tSS = 0.8V x CSS
ISS
LM20146
www.ti.com
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
If different start up times are needed the equation shown below can be used to calculate the start up time.
(12)
As shown above, the start up time is influenced by the value of the Soft-Start capacitor CSS(F) and the 5 µA Soft-
Start pin current ISS(A).
While the Soft-Start capacitor can be sized to meet many start up requirements, there are limitations to its size.
The Soft-Start time can never be faster than 1ms due to the internal default 1 ms start up time. When the device
is enabled there is an approximate time interval of 50 µs when the Soft-Start capacitor will be discharged just
prior to the Soft-Start ramp. If the enable pin is rapidly pulsed or the Soft-Start capacitor is large there may not be
enough time for CSS to completely discharge resulting in start up times less than predicted. To aid in the
discharging of the Soft-Start capacitor during long disable periods an external 1 MΩresistor from SS/TRK to
ground can be used without greatly affecting the start-up time.
Using Precision Enable and Power Good
The precision enable (EN) and power good (PGOOD) pins of the LM20146 can be used to address many
sequencing requirements. The turn-on of the LM20146 can be controlled with the precision enable pin by using
two external resistors as shown in Figure 33.
Figure 33. Sequencing LM20146 with Precision Enable
The value for resistor RBcan be selected by the user to control the current through the divider. Typically this
resistor will be selected to be between 10 kΩand 1 MΩ. Once the value for RBis chosen the resistor RAcan be
solved using the equation below to set the desired turn-on voltage.
(13)
When designing for a specific turn-on threshold (VTO) the tolerance on the input supply, enable threshold
(VIH_EN), and external resistors needs to be considered to insure proper turn-on of the device.
The LM20146 features an open drain power good (PGOOD) pin to sequence external supplies or loads and to
provide fault detection. This pin requires an external resistor (RPG) to pull PGOOD high while when the output is
within the PGOOD tolerance window. Typical values for this resistor range from 10 kΩto 100 kΩ.
Tracking an External Supply
By using a properly chosen resistor divider network connected to the SS/TRK pin, as shown in Figure 34, the
output of the LM20146 can be configured to track an external voltage source to obtain a simultaneous or
ratiometric start up.
Figure 34. Tracking an External Supply
Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LM20146
VOUT1
VOUT2
VEN
VOLTAGE
TIME
VOLTAGE
TIME
SIMULTANEOUS START UP
RATIOMETRIC START UP
=1R
VOUT1
VOUT2
VEN
OUT12OUT Vx8.0<V
( ) x
-1
=1R 2R
V1OUT
2R
x
-1
V2OUT
V8.0 ¸
¹
·
¨
©
§
¸
¨
LM20146
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
www.ti.com
Since the Soft-Start charging current ISS is always present on the SS/TRK pin, the size of R2 should be less than
10 kΩto minimize the errors in the tracking output. Once a value for R2 is selected the value for R1 can be
calculated using appropriate equation in Figure 35, to give the desired start up. Figure 35 shows two common
start up sequences; the top waveform shows a simultaneous start up while the waveform at the bottom illustrates
a ratiometric start up.
Figure 35. Common Start Up Sequences
A simultaneous start up is preferred when powering most FPGAs, DSPs, or other microprocessors. In these
systems the higher voltage, VOUT1, usually powers the I/O, and the lower voltage, VOUT2, powers the core. A
simultaneous start up provides a more robust power up for these applications since it avoids turning on any
parasitic conduction paths that may exist between the core and the I/O pins of the processor.
The second most common power on behavior is known as a ratiometric start up. This start up is preferred in
applications where both supplies need to be at the final value at the same time.
Similar to the Soft-Start function, the fastest start up possible is 1ms regardless of the rise time of the tracking
voltage. When using the track feature the final voltage seen by the SS/TRACK pin should exceed 1V to provide
sufficient overdrive and transient immunity.
Thermal Considerations
The thermal characteristics of the LM20146 are specified using the parameter θJA, which relates the junction
temperature to the ambient temperature. Although the value of θJA is dependant on many variables, it still can be
used to approximate the operating junction temperature of the device.
20 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: LM20146
500 LPFM
200 LFPM
No Air Flow
LM20146
www.ti.com
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
To obtain an estimate of the device junction temperature, one may use the following relationship:
TJ= PDθJA + TA(14)
and PD= PIN x (1 - Efficiency) - 1.1 x IOUT2 x DCR
where
• TJis the junction temperature in °C
• PIN is the input power in Watts (PIN = VIN x IIN)
•θJA is the junction to ambient thermal resistance for the LM20146
• TAis the ambient temperature in °C
• IOUT is the output load current
• DCR is the inductor series resistance (15)
It is important to always keep the operating junction temperature (TJ) below 125°C for reliable operation. If the
junction temperature exceeds 160°C the device will cycle in and out of thermal shutdown. If thermal shutdown
occurs it is a sign of inadequate heatsinking or excessive power dissipation in the device.
Figure 36, shown below, provides a better approximation of the θJA for a given PCB copper area on a 4 layer
board. The PCB heatsink area consists of 2oz. copper located on the bottom layer of the PCB directly under the
HTSSOP exposed pad. The bottom copper area is connected to the HTSSOP exposed pad by means of a 4 x 4
array of 12 mil thermal vias.
Figure 36. Thermal Resistance vs PCB Area
PCB Layout Considerations
PC board layout is an important part of DC-DC converter design. Poor board layout can disrupt the performance
of a DC-DC converter and surrounding circuitry by contributing to EMI, ground bounce, and resistive voltage loss
in the traces. These can send erroneous signals to the DC-DC converter resulting in poor regulation or instability.
Good layout can be implemented by following a few simple design rules.
1. Minimize area of switched current loops. In a buck regulator there are two loops where currents are switched
rapidly. The first loop starts from the input capacitor, to the regulator VIN pin, to the regulator SW pin, to the
inductor then out to the output capacitor and load. The second loop starts from the output capacitor ground, to
the regulator PGND pins, to the inductor and then out to the load (see Figure 37). To minimize both loop areas
the input capacitor should be placed as close as possible to the PVIN pin. Grounding for both the input and
output capacitor should consist of a small localized top side plane that connects to PGND and the die attach pad
(DAP). The inductor should be placed as close as possible to the SW pin and output capacitor.
Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 21
Product Folder Links: LM20146
PVIN SW
AGND
FB
PGOOD
EN
SS/TRK
AVIN
COMP
LM20146 L
RF
VCC
PGND
CIN
RC1
CC1
VIN
CSS
CVCC
COUT
VOUT
RFB2
RFB1
CFRT
RT
(optional)
CC2
PVIN SW
PGND
LVOUT
LM20146
CIN COUT
LOOP1 LOOP2
LM20146
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
www.ti.com
2. Minimize the copper area of the switch node. Since the LM20146 has the SW pins on opposite sides of the
package it is recommended to via these pins down to the bottom or internal layer with 2 to 4 vias on each SW
pin. The SW pins should be directly connected with a trace that runs across the bottom of the package. To
minimize IR losses this trace should be no smaller that 50 mils wide, but no larger than 100 mils wide to keep the
copper area to a minimum. In general the SW pins should not be connected on the top layer since it could block
the ground return path for the power ground. The inductor should be placed as close as possible to one of the
SW pins to further minimize the copper area of the switch node.
3. Have a single point ground for all device analog grounds located under the DAP. The ground connections for
the compensation, feedback, and Soft-Start components should be connected together then routed to the AGND
pin of the device. The AGND pin should connect to PGND under the DAP. This prevents any switched or load
currents from flowing in the analog ground plane. If not properly handled poor grounding can result in degraded
load regulation or erratic switching behavior.
4. Minimize trace length to the FB pin. Since the feedback node can be high impedance the trace from the output
resistor divider to FB pin should be as short as possible. This is most important when high value resistors are
used to set the output voltage. The feedback trace should be routed away from the SW pin and inductor to avoid
contaminating the feedback signal with switch noise.
5. Make input and output bus connections as wide as possible. This reduces any voltage drops on the input or
output of the converter and can improve efficiency. If voltage accuracy at the load is important make sure
feedback voltage sense is made at the load. Doing so will correct for voltage drops at the load and provide the
best output accuracy.
6. Provide adequate device heatsinking. Use as many vias as is possible to connect the DAP to the power plane
heatsink. For best results use a 4x4 via array with a minimum via diameter of 12 mils. See the Thermal
Considerations section to insure enough copper heatsinking area is used to keep the junction temperature below
125°C.
Figure 37. Schematic of LM20146 Highlighting Layout Sensitive Nodes
Typical Application Circuits
This section provides several application solutions with a bill of materials. All bill of materials reference the below
figure. The compensation for these solutions were optimized to work over a wide range of input and output
voltages; if a faster transient response is needed reduce the value of CC1 and calculate the new value for RC1 as
outlined in the design guide.
Figure 38.
22 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: LM20146
LM20146
www.ti.com
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
Bill of Materials (VIN = 5V, VOUT = 3.3V, FSW = 500kHz, IOUTMAX = 6A)
Designator Description Part Number Manufacturer Qty
U1 Synchronous Buck Regulator LM20146 Texas Instruments 1
CIN 100µF, 1210, X5R, 6.3V C3225X5R0J107M TDK 1
COUT 100µF, 1210, X5R, 6.3V C3225X5R0J107M TDK 1
L 1µH, 7.8 mΩSPM6530T-1R0M120 TDK 1
RF1Ω, 0603 CRCW06031R0J-e3 Vishay-Dale 1
CF100nF, 0603, X7R, 16V GRM188R71C104KA01 Murata 1
CVCC 1µF, 0603, X5R, 6.3V GRM188R60J105KA01 Murata 1
RC1 14.3 kΩ, 0603 CRCW06031432F-e3 Vishay-Dale 1
CC1 1nF, 0603, COG, 50V GRM1885C1H102JA01 Murata 1
CSS 33nF, 0603, X7R, 25V VJ0603Y333KXXA Vishay-Vitramon 1
RT100kΩ, 0603 CRCW06031003F-e3 Vishay-Dale 1
RFB1 31.6kΩ, 0603 CRCW06033162F-e3 Vishay-Dale 1
RFB2 10.2kΩ, 0603 CRCW06031022F-e3 Vishay-Dale 1
Bill of Materials (VIN = 3.3V to 5V, VOUT = 1.2V, FSW =750kHz, IOUTMAX = 6A)
Designator Description Part Number Manufacturer Qty
U1 Synchronous Buck Regulator LM20146 Texas Instruments 1
CIN 100µF, 1210, X5R, 6.3V C3225X5R0J107M TDK 1
COUT 100µF, 1210, X5R, 6.3V C3225X5R0J107M TDK 1
L 0.68µH, 5.39 mΩSPM6530T-R68M140 TDK 1
RF1Ω, 0603 CRCW06031R0J-e3 Vishay-Dale 1
CF100nF, 0603, X7R, 16V GRM188R71C104KA01 Murata 1
CVCC 1µF, 0603, X5R, 6.3V GRM188R60J105KA01 Murata 1
RC1 4.53kΩ, 0603 CRCW06034532F-e3 Vishay-Dale 1
CC1 1.8nF, 0603, X7R, 25V VJ0603Y182KXXA Vishay-Vitramon 1
CSS 33nF, 0603, X7R, 25V VJ0603Y333KXXA Vishay-Vitramon 1
RT48.7kΩ, 0603 CRCW06034872F-e3 Vishay-Dale 1
RFB1 4.99kΩ, 0603 CRCW06034991F-e3 Vishay-Dale 1
RFB2 10kΩ, 0603 CRCW06031002F-e3 Vishay-Dale 1
Copyright © 2008–2013, Texas Instruments Incorporated Submit Documentation Feedback 23
Product Folder Links: LM20146
LM20146
SNVS563C –FEBRUARY 2008–REVISED APRIL 2013
www.ti.com
REVISION HISTORY
Changes from Revision B (April 2013) to Revision C Page
• Changed layout of National Data Sheet to TI format .......................................................................................................... 23
24 Submit Documentation Feedback Copyright © 2008–2013, Texas Instruments Incorporated
Product Folder Links: LM20146
PACKAGE OPTION ADDENDUM
www.ti.com 10-Dec-2020
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead finish/
Ball material
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM20146MH/NOPB ACTIVE HTSSOP PWP 16 92 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 L20146
MH
LM20146MHE/NOPB ACTIVE HTSSOP PWP 16 250 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 L20146
MH
LM20146MHX/NOPB ACTIVE HTSSOP PWP 16 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 L20146
MH
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
PACKAGE OPTION ADDENDUM
www.ti.com 10-Dec-2020
Addendum-Page 2
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM20146MHE/NOPB HTSSOP PWP 16 250 178.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1
LM20146MHX/NOPB HTSSOP PWP 16 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 20-Jul-2019
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM20146MHE/NOPB HTSSOP PWP 16 250 210.0 185.0 35.0
LM20146MHX/NOPB HTSSOP PWP 16 2500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 20-Jul-2019
Pack Materials-Page 2
www.ti.com
PACKAGE OUTLINE
C
TYP
6.6
6.2
14X 0.65
16X 0.30
0.19
2X
4.55
(0.15) TYP
0 - 8 0.15
0.05
3.3
2.7
3.3
2.7
2X 1.34 MAX
NOTE 5
1.2 MAX
(1)
0.25
GAGE PLANE
0.75
0.50
A
NOTE 3
5.1
4.9
B4.5
4.3
4X 0.166 MAX
NOTE 5
4214868/A 02/2017
PowerPAD HTSSOP - 1.2 mm max heightPWP0016A
PLASTIC SMALL OUTLINE
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm per side.
4. Reference JEDEC registration MO-153.
5. Features may not be present.
PowerPAD is a trademark of Texas Instruments.
TM
116
0.1 C A B
9
8
PIN 1 ID
AREA
SEATING PLANE
0.1 C
SEE DETAIL A
DETAIL A
TYPICAL
SCALE 2.400
THERMAL
PAD
17
www.ti.com
EXAMPLE BOARD LAYOUT
(5.8)
0.05 MAX
ALL AROUND 0.05 MIN
ALL AROUND
16X (1.5)
16X (0.45)
14X (0.65)
(3.4)
NOTE 9
(5)
NOTE 9
(3.3)
(3.3)
( 0.2) TYP
VIA (1.1) TYP
(1.1)
TYP
4214868/A 02/2017
PowerPAD HTSSOP - 1.2 mm max heightPWP0016A
PLASTIC SMALL OUTLINE
SYMM
SYMM
SEE DETAILS
LAND PATTERN EXAMPLE
EXPOSED METAL SHOWN
SCALE:10X
1
89
16
METAL COVERED
BY SOLDER MASK
SOLDER MASK
DEFINED PAD
17
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
9. Size of metal pad may vary due to creepage requirement.
TM
METAL
SOLDER MASK
OPENING
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
PADS 1-16
EXPOSED
METAL
SOLDER MASK
DEFINED
SOLDER MASK
METAL UNDER SOLDER MASK
OPENING
EXPOSED
METAL
www.ti.com
EXAMPLE STENCIL DESIGN
16X (1.5)
16X (0.45)
(3.3)
(3.3)
BASED ON
0.125 THICK
STENCIL
14X (0.65)
(R0.05) TYP
(5.8)
4214868/A 02/2017
PowerPAD HTSSOP - 1.2 mm max heightPWP0016A
PLASTIC SMALL OUTLINE
2.79 X 2.790.175 3.01 X 3.010.15 3.3 X 3.3 (SHOWN)0.125 3.69 X 3.690.1
SOLDER STENCIL
OPENING
STENCIL
THICKNESS
NOTES: (continued)
10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
11. Board assembly site may have different recommendations for stencil design.
TM
SYMM
SYMM
1
89
16
BASED ON
0.125 THICK
STENCIL
BY SOLDER MASK
METAL COVERED SEE TABLE FOR
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
SOLDER PASTE EXAMPLE
EXPOSED PAD
100% PRINTED SOLDER COVERAGE BY AREA
SCALE:10X
17
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