LM2734ZSDX/NOPB - NATIONAL SEMICONDUCTOR

LM2734Z

LM2734Z/LM2734ZQ Thin SOT23 1A Load Step-Down DC-DC Regulator

Literature Number: SNVS334D

June 12, 2008

LM2734Z/LM2734ZQ

Thin SOT23 1A Load Step-Down DC-DC Regulator

General Description

The LM2734Z regulator is a monolithic, high frequency, PWM

step-down DC/DC converter assembled in a 6-pin Thin

SOT23 and LLP non pull back package. It provides all the

active functions to provide local DC/DC conversion with fast

transient response and accurate regulation in the smallest

possible PCB area.

With a minimum of external components and online design

support through WEBENCH®™, the LM2734Z is easy to use.

The ability to drive 1A loads with an internal 300mΩ NMOS

switch using state-of-the-art 0.5µm BiCMOS technology re-

sults in the best power density available. The world class

control circuitry allows for on-times as low as 13ns, thus sup-

porting exceptionally high frequency conversion over the en-

tire 3V to 20V input operating range down to the minimum

output voltage of 0.8V. Switching frequency is internally set

to 3MHz, allowing the use of extremely small surface mount

inductors and chip capacitors. Even though the operating fre-

quency is very high, efficiencies up to 85% are easy to

achieve. External shutdown is included, featuring an ultra-low

stand-by current of 30nA. The LM2734Z utilizes current-mode

control and internal compensation to provide high-perfor-

mance regulation over a wide range of operating conditions.

Additional features include internal soft-start circuitry to re-

duce inrush current, pulse-by-pulse current limit, thermal

shutdown, and output over-voltage protection.

Features

■Thin SOT23-6 package, or 6 lead LLP package

■3.0V to 20V input voltage range

■0.8V to 18V output voltage range

■1A output current

■3MHz switching frequency

■300mΩ NMOS switch

■30nA shutdown current

■0.8V, 2% internal voltage reference

■Internal soft-start

■Current-Mode, PWM operation

■Thermal shutdown

■LM2734ZQ is AEC-Q100 Grade 1 qualified and is

manufactured on an Automotive Grade Flow

Applications

■DSL Modems

■Local Point of Load Regulation

■Battery Powered Devices

■USB Powered Devices

■Automotive

Typical Application Circuit

20130301

Efficiency vs Load Current

VIN = 5V, VOUT = 3.3V

20130345

WEBENCH™ is a trademark of Transim.

LM2734Z/LM2734ZQ Thin SOT23 1A Load Step-Down DC-DC Regulator

Connection Diagrams

20130305

6-Lead TSOT

NS Package Number MK06A

20130360

6-Lead LLP (3mm x 3mm)

NS Package Number SDE06A

Ordering Information

Order Number Package Type NSC Package

Drawing

Package

Marking

Supplied As Features

LM2734ZMK

TSOT-6 MK06A

SFTB 1000 Units on Tape and Reel

LM2734ZMKX SFTB 3000 Units on Tape and Reel

LM2734ZQMKE SVBB 250 Units on Tape and Reel AEC-Q100 Grade 1

Qualified. Automotive-Grade

Production Flow*

LM2734ZQMK SVBB 1000 Units on Tape and Reel

LM2734ZQMKX SVBB 3000 Units on Tape and Reel

LM2734ZSD

6-Lead LLP SDE06A

L163B 1000 Units on Tape and Reel

LM2734ZSDX L163B 4500 Units on Tape and Reel

LM2734ZQSDE L238B 250 Units on Tape and Reel AEC-Q100 Grade 1

Qualified. Automotive-Grade

Production Flow*

LM2734ZQSD L238B 1000 Units on Tape and Reel

LM2734ZQSDX L238B 4500 Units on Tape and Reel

*Automotive Grade (Q) product incorporates enhanced manufacturing and support processes for the automotive market, including defect detection methodologies.

Reliability qualification is compliant with the requirements and temperature grades defined in the AEC-Q100 standard. Automotive grade products are identified

with the letter Q. For more information go to http://www.national.com/automotive.

Pin Descriptions

Pin Name Function

1 BOOST Boost voltage that drives the internal NMOS control switch. A

bootstrap capacitor is connected between the BOOST and SW pins.

2 GND Signal and Power ground pin. Place the bottom resistor of the

feedback network as close as possible to this pin for accurate

regulation.

3 FB Feedback pin. Connect FB to the external resistor divider to set output

voltage.

4 EN Enable control input. Logic high enables operation. Do not allow this

pin to float or be greater than VIN + 0.3V.

5 VIN Input supply voltage. Connect a bypass capacitor to this pin.

6 SW Output switch. Connects to the inductor, catch diode, and bootstrap

capacitor.

DAP GND The Die Attach Pad is internally connected to GND

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LM2734Z/LM2734ZQ

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

VIN -0.5V to 24V

SW Voltage -0.5V to 24V

Boost Voltage -0.5V to 30V

Boost to SW Voltage -0.5V to 6.0V

FB Voltage -0.5V to 3.0V

EN Voltage -0.5V to (VIN + 0.3V)

Junction Temperature 150°C

ESD Susceptibility (Note 2) 2kV

Storage Temp. Range -65°C to 150°C

Soldering Information

Infrared/Convection Reflow (15sec) 220°C

Wave Soldering Lead Temp. (10sec) 260°C

Operating Ratings (Note 1)

VIN 3V to 20V

SW Voltage -0.5V to 20V

Boost Voltage -0.5V to 25V

Boost to SW Voltage 1.6V to 5.5V

Junction Temperature Range −40°C to +125°C

Thermal Resistance θJA (Note 3)

TSOT23–6 118°C/W

Electrical Characteristics

Specifications with standard typeface are for TJ = 25°C, and those in boldface type apply over the full Operating Temperature

Range (TJ = -40°C to 125°C). VIN = 5V, VBOOST - VSW = 5V unless otherwise specified. Datasheet min/max specification limits are

guaranteed by design, test, or statistical analysis.

Symbol Parameter Conditions Min

(Note 4)

Typ

(Note 5)

Max

(Note 4) Units

VFB Feedback Voltage 0.784 0.800 0.816 V

ΔVFB/ΔVIN

Feedback Voltage Line

Regulation

VIN = 3V to 20V 0.01 % / V

IFB Feedback Input Bias Current Sink/Source 10 250 nA

UVLO

Undervoltage Lockout VIN Rising 2.74 2.90

Undervoltage Lockout VIN Falling 2.0 2.3

UVLO Hysteresis 0.30 0.44 0.62

FSW Switching Frequency 2.2 3.0 3.6 MHz

DMAX Maximum Duty Cycle 78 85 %

DMIN Minimum Duty Cycle 8 %

RDS(ON) Switch ON Resistance

VBOOST - VSW = 3V

(TSOT Package)

300 600 mΩ

VBOOST - VSW = 3V

(LLP Package)

340 650 mΩ

ICL Switch Current Limit VBOOST - VSW = 3V 1.2 1.7 2.5 A

IQQuiescent Current Switching 1.5 2.5 mA

Quiescent Current (shutdown) VEN = 0V 30 nA

IBOOST Boost Pin Current (Switching) 4.25 6mA

VEN_TH

Shutdown Threshold Voltage VEN Falling 0.4 V

Enable Threshold Voltage VEN Rising 1.8

IEN Enable Pin Current Sink/Source 10 nA

ISW Switch Leakage 40 nA

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see Electrical Characteristics.

Note 2: Human body model, 1.5kΩ in series with 100pF.

Note 3: Thermal shutdown will occur if the junction temperature exceeds 165°C. The maximum power dissipation is a function of TJ(MAX) , θJA and TA . The

maximum allowable power dissipation at any ambient temperature is PD = (TJ(MAX) – TA)/θJA . All numbers apply for packages soldered directly onto a 3” x 3” PC

board with 2oz. copper on 4 layers in still air. For a 2 layer board using 1 oz. copper in still air, θJA = 204°C/W.

Note 4: Guaranteed to National’s Average Outgoing Quality Level (AOQL).

Note 5: Typicals represent the most likely parametric norm.

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LM2734Z/LM2734ZQ

Typical Performance Characteristics All curves taken at VIN = 5V, VBOOST - VSW = 5V, L1 = 2.2 µH and

TA = 25°C, unless specified otherwise.

Efficiency vs Load Current

VOUT = 5V

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Efficiency vs Load Current

VOUT = 3.3V

20130351

Efficiency vs Load Current

VOUT = 1.5V

20130337

Oscillator Frequency vs Temperature

20130327

Line Regulation

VOUT = 1.5V, IOUT = 500mA

20130354

Line Regulation

VOUT = 3.3V, IOUT = 500mA

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LM2734Z/LM2734ZQ

Block Diagram

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FIGURE 1.

Application Information

THEORY OF OPERATION

The LM2734Z is a constant frequency PWM buck regulator

IC that delivers a 1A load current. The regulator has a preset

switching frequency of 3MHz. This high frequency allows the

LM2734Z to operate with small surface mount capacitors and

inductors, resulting in a DC/DC converter that requires a min-

imum amount of board space. The LM2734Z is internally

compensated, so it is simple to use, and requires few external

components. The LM2734Z uses current-mode control to reg-

ulate the output voltage.

The following operating description of the LM2734Z will refer

to the Simplified Block Diagram (Figure 1) and to the wave-

forms in Figure 2. The LM2734Z supplies a regulated output

voltage by switching the internal NMOS control switch at con-

stant frequency and variable duty cycle. A switching cycle

begins at the falling edge of the reset pulse generated by the

internal oscillator. When this pulse goes low, the output con-

trol logic turns on the internal NMOS control switch. During

this on-time, the SW pin voltage (VSW) swings up to approxi-

mately VIN, and the inductor current (IL) increases with a linear

slope. IL is measured by the current-sense amplifier, which

generates an output proportional to the switch current. The

sense signal is summed with the regulator’s corrective ramp

and compared to the error amplifier’s output, which is propor-

tional to the difference between the feedback voltage and

VREF. When the PWM comparator output goes high, the out-

put switch turns off until the next switching cycle begins.

During the switch off-time, inductor current discharges

through Schottky diode D1, which forces the SW pin to swing

below ground by the forward voltage (VD) of the catch diode.

The regulator loop adjusts the duty cycle (D) to maintain a

constant output voltage.

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FIGURE 2. LM2734Z Waveforms of SW Pin Voltage and

Inductor Current

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LM2734Z/LM2734ZQ

BOOST FUNCTION

Capacitor CBOOST and diode D2 in Figure 3 are used to gen-

erate a voltage VBOOST. VBOOST - VSW is the gate drive voltage

to the internal NMOS control switch. To properly drive the in-

ternal NMOS switch during its on-time, VBOOST needs to be at

least 1.6V greater than VSW. Although the LM2734Z will op-

erate with this minimum voltage, it may not have sufficient

gate drive to supply large values of output current. Therefore,

it is recommended that VBOOST be greater than 2.5V above

VSW for best efficiency. VBOOST – VSW should not exceed the

maximum operating limit of 5.5V.

5.5V > VBOOST – VSW > 2.5V for best performance.

20130308

FIGURE 3. VOUT Charges CBOOST

When the LM2734Z starts up, internal circuitry from the

BOOST pin supplies a maximum of 20mA to CBOOST. This

current charges CBOOST to a voltage sufficient to turn the

switch on. The BOOST pin will continue to source current to

CBOOST until the voltage at the feedback pin is greater than

0.76V.

There are various methods to derive VBOOST:

1. From the input voltage (VIN)

2. From the output voltage (VOUT)

3. From an external distributed voltage rail (VEXT)

4. From a shunt or series zener diode

In the Simplifed Block Diagram of Figure 1, capacitor

CBOOST and diode D2 supply the gate-drive current for the

NMOS switch. Capacitor CBOOST is charged via diode D2 by

VIN. During a normal switching cycle, when the internal NMOS

control switch is off (TOFF) (refer to Figure 2), VBOOST equals

VIN minus the forward voltage of D2 (VFD2), during which the

current in the inductor (L) forward biases the Schottky diode

D1 (VFD1). Therefore the voltage stored across CBOOST is

VBOOST - VSW = VIN - VFD2 + VFD1

When the NMOS switch turns on (TON), the switch pin rises

VSW = VIN – (RDSON x IL),

forcing VBOOST to rise thus reverse biasing D2. The voltage at

VBOOST is then

VBOOST = 2VIN – (RDSON x IL) – VFD2 + VFD1

which is approximately

2VIN - 0.4V

for many applications. Thus the gate-drive voltage of the

NMOS switch is approximately

VIN - 0.2V

An alternate method for charging CBOOST is to connect D2 to

the output as shown in Figure 3. The output voltage should

be between 2.5V and 5.5V, so that proper gate voltage will be

applied to the internal switch. In this circuit, CBOOST provides

a gate drive voltage that is slightly less than VOUT.

In applications where both VIN and VOUT are greater than

5.5V, or less than 3V, CBOOST cannot be charged directly from

these voltages. If VIN and VOUT are greater than 5.5V,

CBOOST can be charged from VIN or VOUT minus a zener volt-

age by placing a zener diode D3 in series with D2, as shown

in Figure 4. When using a series zener diode from the input,

ensure that the regulation of the input supply doesn’t create

a voltage that falls outside the recommended VBOOST voltage.

(VINMAX – VD3) < 5.5V

(VINMIN – VD3) > 1.6V

20130309

FIGURE 4. Zener Reduces Boost Voltage from VIN

An alternative method is to place the zener diode D3 in a

shunt configuration as shown in Figure 5. A small 350mW to

500mW 5.1V zener in a SOT-23 or SOD package can be used

for this purpose. A small ceramic capacitor such as a 6.3V,

0.1µF capacitor (C4) should be placed in parallel with the

zener diode. When the internal NMOS switch turns on, a pulse

of current is drawn to charge the internal NMOS gate capac-

itance. The 0.1 µF parallel shunt capacitor ensures that the

VBOOST voltage is maintained during this time.

Resistor R3 should be chosen to provide enough RMS current

to the zener diode (D3) and to the BOOST pin. A recom-

mended choice for the zener current (IZENER) is 1 mA. The

current IBOOST into the BOOST pin supplies the gate current

of the NMOS control switch and varies typically according to

the following formula:

IBOOST = (D + 0.5) x (VZENER – VD2) mA

where D is the duty cycle, VZENER and VD2 are in volts, and

IBOOST is in milliamps. VZENER is the voltage applied to the

anode of the boost diode (D2), and VD2 is the average forward

voltage across D2. Note that this formula for IBOOST gives typ-

ical current. For the worst case IBOOST, increase the current

by 25%. In that case, the worst case boost current will be

IBOOST-MAX = 1.25 x IBOOST

R3 will then be given by

R3 = (VIN - VZENER) / (1.25 x IBOOST + IZENER)

For example, let VIN = 10V, VZENER = 5V, VD2 = 0.7V, IZENER

= 1mA, and duty cycle D = 50%. Then

IBOOST = (0.5 + 0.5) x (5 - 0.7) mA = 4.3mA

R3 = (10V - 5V) / (1.25 x 4.3mA + 1mA) = 787Ω

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LM2734Z/LM2734ZQ

20130348

FIGURE 5. Boost Voltage Supplied from the Shunt Zener

on VIN

ENABLE PIN / SHUTDOWN MODE

The LM2734Z has a shutdown mode that is controlled by the

enable pin (EN). When a logic low voltage is applied to EN,

the part is in shutdown mode and its quiescent current drops

to typically 30nA. Switch leakage adds another 40nA from the

input supply. The voltage at this pin should never exceed

VIN + 0.3V.

SOFT-START

This function forces VOUT to increase at a controlled rate dur-

ing start up. During soft-start, the error amplifier’s reference

voltage ramps from 0V to its nominal value of 0.8V in approx-

imately 200µs. This forces the regulator output to ramp up in

a more linear and controlled fashion, which helps reduce in-

rush current.

OUTPUT OVERVOLTAGE PROTECTION

The overvoltage comparator compares the FB pin voltage to

a voltage that is 10% higher than the internal reference Vref.

Once the FB pin voltage goes 10% above the internal refer-

ence, the internal NMOS control switch is turned off, which

allows the output voltage to decrease toward regulation.

UNDERVOLTAGE LOCKOUT

Undervoltage lockout (UVLO) prevents the LM2734Z from

operating until the input voltage exceeds 2.74V(typ).

The UVLO threshold has approximately 440mV of hysteresis,

so the part will operate until VIN drops below 2.3V(typ). Hys-

teresis prevents the part from turning off during power up if

VIN is non-monotonic.

CURRENT LIMIT

The LM2734Z uses cycle-by-cycle current limiting to protect

the output switch. During each switching cycle, a current limit

comparator detects if the output switch current exceeds 1.7A

(typ), and turns off the switch until the next switching cycle

begins.

THERMAL SHUTDOWN

Thermal shutdown limits total power dissipation by turning off

the output switch when the IC junction temperature exceeds

165°C. After thermal shutdown occurs, the output switch

doesn’t turn on until the junction temperature drops to ap-

proximately 150°C.

Design Guide

INDUCTOR SELECTION

The Duty Cycle (D) can be approximated quickly using the

ratio of output voltage (VO) to input voltage (VIN):

The catch diode (D1) forward voltage drop and the voltage

drop across the internal NMOS must be included to calculate

a more accurate duty cycle. Calculate D by using the following

formula:

VSW can be approximated by:

VSW = IO x RDS(ON)

The diode forward drop (VD) can range from 0.3V to 0.7V de-

pending on the quality of the diode. The lower VD is, the higher

the operating efficiency of the converter.

The inductor value determines the output ripple current. Low-

er inductor values decrease the size of the inductor, but

increase the output ripple current. An increase in the inductor

value will decrease the output ripple current. The ratio of ripple

current (ΔiL) to output current (IO) is optimized when it is set

between 0.3 and 0.4 at 1A. The ratio r is defined as:

One must also ensure that the minimum current limit (1.2A)

is not exceeded, so the peak current in the inductor must be

calculated. The peak current (ILPK) in the inductor is calculated

by:

ILPK = IO + ΔIL/2

If r = 0.5 at an output of 1A, the peak current in the inductor

will be 1.25A. The minimum guaranteed current limit over all

operating conditions is 1.2A. One can either reduce r to 0.4

resulting in a 1.2A peak current, or make the engineering

judgement that 50mA over will be safe enough with a 1.7A

typical current limit and 6 sigma limits. When the designed

maximum output current is reduced, the ratio r can be in-

creased. At a current of 0.1A, r can be made as high as 0.9.

The ripple ratio can be increased at lighter loads because the

net ripple is actually quite low, and if r remains constant the

inductor value can be made quite large. An equation empiri-

cally developed for the maximum ripple ratio at any current

below 2A is:

r = 0.387 x IOUT-0.3667

Note that this is just a guideline.

The LM2734Z operates at frequencies allowing the use of

ceramic output capacitors without compromising transient re-

sponse. Ceramic capacitors allow higher inductor ripple with-

out significantly increasing output ripple. See the output

capacitor section for more details on calculating output volt-

age ripple.

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LM2734Z/LM2734ZQ

Now that the ripple current or ripple ratio is determined, the

inductance is calculated by:

where fs is the switching frequency and IO is the output cur-

rent. When selecting an inductor, make sure that it is capable

of supporting the peak output current without saturating. In-

ductor saturation will result in a sudden reduction in induc-

tance and prevent the regulator from operating correctly.

Because of the speed of the internal current limit, the peak

current of the inductor need only be specified for the required

maximum output current. For example, if the designed maxi-

mum output current is 0.5A and the peak current is 0.7A, then

the inductor should be specified with a saturation current limit

of >0.7A. There is no need to specify the saturation or peak

current of the inductor at the 1.7A typical switch current limit.

The difference in inductor size is a factor of 5. Because of the

operating frequency of the LM2734Z, ferrite based inductors

are preferred to minimize core losses. This presents little re-

striction since the variety of ferrite based inductors is huge.

Lastly, inductors with lower series resistance (DCR) will pro-

vide better operating efficiency. For recommended inductors

see Example Circuits.

INPUT CAPACITOR

An input capacitor is necessary to ensure that VIN does not

drop excessively during switching transients. The primary

specifications of the input capacitor are capacitance, voltage,

RMS current rating, and ESL (Equivalent Series Inductance).

The recommended input capacitance is 10µF, although 4.7µF

works well for input voltages below 6V. The input voltage rat-

ing is specifically stated by the capacitor manufacturer. Make

sure to check any recommended deratings and also verify if

there is any significant change in capacitance at the operating

input voltage and the operating temperature. The input ca-

pacitor maximum RMS input current rating (IRMS-IN) must be

greater than:

It can be shown from the above equation that maximum RMS

capacitor current occurs when D = 0.5. Always calculate the

RMS at the point where the duty cycle, D, is closest to 0.5.

The ESL of an input capacitor is usually determined by the

effective cross sectional area of the current path. A large

leaded capacitor will have high ESL and a 0805 ceramic chip

capacitor will have very low ESL. At the operating frequencies

of the LM2734Z, certain capacitors may have an ESL so large

that the resulting impedance (2πfL) will be higher than that

required to provide stable operation. As a result, surface

mount capacitors are strongly recommended. Sanyo

POSCAP, Tantalum or Niobium, Panasonic SP or Cornell

Dubilier ESR, and multilayer ceramic capacitors (MLCC) are

all good choices for both input and output capacitors and have

very low ESL. For MLCCs it is recommended to use X7R or

X5R dielectrics. Consult capacitor manufacturer datasheet to

see how rated capacitance varies over operating conditions.

OUTPUT CAPACITOR

The output capacitor is selected based upon the desired out-

put ripple and transient response. The initial current of a load

transient is provided mainly by the output capacitor. The out-

put ripple of the converter is:

When using MLCCs, the ESR is typically so low that the ca-

pacitive ripple may dominate. When this occurs, the output

ripple will be approximately sinusoidal and 90° phase shifted

from the switching action. Given the availability and quality of

MLCCs and the expected output voltage of designs using the

LM2734Z, there is really no need to review any other capac-

itor technologies. Another benefit of ceramic capacitors is

their ability to bypass high frequency noise. A certain amount

of switching edge noise will couple through parasitic capaci-

tances in the inductor to the output. A ceramic capacitor will

bypass this noise while a tantalum will not. Since the output

capacitor is one of the two external components that control

the stability of the regulator control loop, most applications will

require a minimum at 10 µF of output capacitance. Capaci-

tance can be increased significantly with little detriment to the

regulator stability. Like the input capacitor, recommended

multilayer ceramic capacitors are X7R or X5R. Again, verify

actual capacitance at the desired operating voltage and tem-

perature.

Check the RMS current rating of the capacitor. The RMS cur-

rent rating of the capacitor chosen must also meet the follow-

ing condition:

CATCH DIODE

The catch diode (D1) conducts during the switch off-time. A

Schottky diode is recommended for its fast switching times

and low forward voltage drop. The catch diode should be

chosen so that its current rating is greater than:

ID1 = IO x (1-D)

The reverse breakdown rating of the diode must be at least

the maximum input voltage plus appropriate margin. To im-

prove efficiency choose a Schottky diode with a low forward

voltage drop.

BOOST DIODE

A standard diode such as the 1N4148 type is recommended.

For VBOOST circuits derived from voltages less than 3.3V, a

small-signal Schottky diode is recommended for greater effi-

ciency. A good choice is the BAT54 small signal diode.

BOOST CAPACITOR

A ceramic 0.01µF capacitor with a voltage rating of at least

6.3V is sufficient. The X7R and X5R MLCCs provide the best

performance.

OUTPUT VOLTAGE

The output voltage is set using the following equation where

R2 is connected between the FB pin and GND, and R1 is

connected between VO and the FB pin. A good value for R2

is 10kΩ.

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LM2734Z/LM2734ZQ

PCB Layout Considerations

When planning layout there are a few things to consider when

trying to achieve a clean, regulated output. The most impor-

tant consideration when completing the layout is the close

coupling of the GND connections of the CIN capacitor and the

catch diode D1. These ground ends should be close to one

another and be connected to the GND plane with at least two

through-holes. Place these components as close to the IC as

possible. Next in importance is the location of the GND con-

nection of the COUT capacitor, which should be near the GND

connections of CIN and D1.

There should be a continuous ground plane on the bottom

layer of a two-layer board except under the switching node

island.

The FB pin is a high impedance node and care should be

taken to make the FB trace short to avoid noise pickup and

inaccurate regulation. The feedback resistors should be

placed as close as possible to the IC, with the GND of R2

placed as close as possible to the GND of the IC. The VOUT

trace to R1 should be routed away from the inductor and any

other traces that are switching.

High AC currents flow through the VIN, SW and VOUT traces,

so they should be as short and wide as possible. However,

making the traces wide increases radiated noise, so the de-

signer must make this trade-off. Radiated noise can be de-

creased by choosing a shielded inductor.

The remaining components should also be placed as close

as possible to the IC. Please see Application Note AN-1229

for further considerations and the LM2734Z demo board as

an example of a four-layer layout.

Calculating Efficiency, and Junction

Temperature

The complete LM2734Z DC/DC converter efficiency can be

calculated in the following manner.

Calculations for determining the most significant power loss-

es are shown below. Other losses totaling less than 2% are

not discussed.

Power loss (PLOSS) is the sum of two basic types of losses in

the converter, switching and conduction. Conduction losses

usually dominate at higher output loads, where as switching

losses remain relatively fixed and dominate at lower output

loads. The first step in determining the losses is to calculate

the duty cycle (D).

VSW is the voltage drop across the internal NFET when it is

on, and is equal to:

VSW = IOUT x RDSON

VD is the forward voltage drop across the Schottky diode. It

can be obtained from the Electrical Characteristics section. If

the voltage drop across the inductor (VDCR) is accounted for,

the equation becomes:

This usually gives only a minor duty cycle change, and has

been omitted in the examples for simplicity.

The conduction losses in the free-wheeling Schottky diode

are calculated as follows:

PDIODE = VD x IOUT(1-D)

Often this is the single most significant power loss in the cir-

cuit. Care should be taken to choose a Schottky diode that

has a low forward voltage drop.

Another significant external power loss is the conduction loss

in the output inductor. The equation can be simplified to:

PIND = IOUT2 x RDCR

The LM2734Z conduction loss is mainly associated with the

internal NFET:

PCOND = IOUT2 x RDSON x D

Switching losses are also associated with the internal NFET.

They occur during the switch on and off transition periods,

where voltages and currents overlap resulting in power loss.

The simplest means to determine this loss is to empirically

measuring the rise and fall times (10% to 90%) of the switch

at the switch node:

PSWF = 1/2(VIN x IOUT x freq x TFALL)

PSWR = 1/2(VIN x IOUT x freq x TRISE)

PSW = PSWF + PSWR

Typical Rise and Fall Times vs Input Voltage

VIN TRISE TFALL

5V 8ns 4ns

10V 9ns 6ns

15V 10ns 7ns

Another loss is the power required for operation of the internal

circuitry:

PQ = IQ x VIN

IQ is the quiescent operating current, and is typically around

1.5mA. The other operating power that needs to be calculated

is that required to drive the internal NFET:

PBOOST = IBOOST x VBOOST

VBOOST is normally between 3VDC and 5VDC. The IBOOST rms

current is approximately 4.25mA. Total power losses are:

9 www.national.com

LM2734Z/LM2734ZQ

Design Example 1:

Operating Conditions

VIN 5.0V POUT 2.5W

VOUT 2.5V PDIODE 151mW

IOUT 1.0A PIND 75mW

VD0.35V PSWF 53mW

Freq 3MHz PSWR 53mW

IQ1.5mA PCOND 187mW

TRISE 8ns PQ7.5mW

TFALL 8ns PBOOST 21mW

RDSON 330mΩPLOSS 548mW

INDDCR 75mΩ

D0.568

η = 82%

Calculating the LM2734Z Junction Temperature

Thermal Definitions:

TJ = Chip junction temperature

TA = Ambient temperature

RθJC = Thermal resistance from chip junction to device case

RθJA = Thermal resistance from chip junction to ambient air

20130373

FIGURE 6. Cross-Sectional View of Integrated Circuit Mounted on a Printed Circuit Board.

Heat in the LM2734Z due to internal power dissipation is re-

moved through conduction and/or convection.

Conduction: Heat transfer occurs through cross sectional ar-

eas of material. Depending on the material, the transfer of

heat can be considered to have poor to good thermal con-

ductivity properties (insulator vs conductor).

Heat Transfer goes as:

silicon→package→lead frame→PCB.

Convection: Heat transfer is by means of airflow. This could

be from a fan or natural convection. Natural convection occurs

when air currents rise from the hot device to cooler air.

Thermal impedance is defined as:

Thermal impedance from the silicon junction to the ambient

air is defined as:

This impedance can vary depending on the thermal proper-

ties of the PCB. This includes PCB size, weight of copper

used to route traces and ground plane, and number of layers

within the PCB. The type and number of thermal vias can also

make a large difference in the thermal impedance. Thermal

vias are necessary in most applications. They conduct heat

from the surface of the PCB to the ground plane. Four to six

thermal vias should be placed under the exposed pad to the

ground plane if the LLP package is used. If the Thin SOT23-6

package is used, place two to four thermal vias close to the

ground pin of the device.

The datasheet specifies two different RθJA numbers for the

Thin SOT23–6 package. The two numbers show the differ-

ence in thermal impedance for a four-layer board with 2oz.

copper traces, vs. a four-layer board with 1oz. copper. RθJA

equals 120°C/W for 2oz. copper traces and GND plane, and

235°C/W for 1oz. copper traces and GND plane.

www.national.com 10

LM2734Z/LM2734ZQ

Method 1:

To accurately measure the silicon temperature for a given

application, two methods can be used. The first method re-

quires the user to know the thermal impedance of the silicon

junction to case. (RθJC) is approximately 80°C/W for the Thin

SOT23-6 package. Knowing the internal dissipation from the

efficiency calculation given previously, and the case temper-

ature, which can be empirically measured on the bench we

have:

Therefore:

TJ = (RθJC x PLOSS) + TC

Design Example 2:

Operating Conditions

VIN 5.0V POUT 2.5W

VOUT 2.5V PDIODE 151mW

IOUT 1.0A PIND 75mW

VD0.35V PSWF 53mW

Freq 3MHz PSWR 53mW

IQ1.5mA PCOND 187mW

TRISE 8ns PQ7.5mW

TFALL 8ns PBOOST 21mW

RDSON 330mΩPLOSS 548mW

INDDCR 75mΩ

D0.568

The second method can give a very accurate silicon junction

temperature. The first step is to determine RθJA of the appli-

cation. The LM2734Z has over-temperature protection cir-

cuitry. When the silicon temperature reaches 165°C, the

device stops switching. The protection circuitry has a hys-

teresis of 15°C. Once the silicon temperature has decreased

to approximately 150°C, the device will start to switch again.

Knowing this, the RθJA for any PCB can be characterized dur-

ing the early stages of the design by raising the ambient

temperature in the given application until the circuit enters

thermal shutdown. If the SW-pin is monitored, it will be obvi-

ous when the internal NFET stops switching indicating a

junction temperature of 165°C. Knowing the internal power

dissipation from the above methods, the junction temperature

and the ambient temperature, RθJA can be determined.

Once this is determined, the maximum ambient temperature

allowed for a desired junction temperature can be found.

Design Example 3:

Operating Conditions

Package SOT23-6

VIN 12.0V POUT 2.475W

VOUT 3.30V PDIODE 523mW

IOUT 750mA PIND 56.25mW

VD0.35V PSWF 108mW

Freq 3MHz PSWR 108mW

IQ1.5mA PCOND 68.2mW

IBOOST 4mA PQ18mW

VBOOST 5V PBOOST 20mW

TRISE 8ns PLOSS 902mW

TFALL 8ns

RDSON 400mΩ

INDDCR 75mΩ

D30.3%

Using a standard National Semiconductor Thin SOT23-6

demonstration board to determine the RθJA of the board. The

four layer PCB is constructed using FR4 with 1/2oz copper

traces. The copper ground plane is on the bottom layer. The

ground plane is accessed by two vias. The board measures

2.5cm x 3cm. It was placed in an oven with no forced airflow.

The ambient temperature was raised to 94°C, and at that

temperature, the device went into thermal shutdown.

If the junction temperature was to be kept below 125°C, then

the ambient temperature cannot go above 54.2°C.

TJ - (RθJA x PLOSS) = TA

The method described above to find the junction temperature

in the Thin SOT23-6 package can also be used to calculate

the junction temperature in the LLP package. The 6 pin LLP

package has a RθJC = 20°C/W, and RθJA can vary depending

on the application. RθJA can be calculated in the same manner

as described in method #2 (see example 3).

11 www.national.com

LM2734Z/LM2734ZQ

LLP Package

The LM2734Z is packaged in a Thin SOT23-6 package and

the 6–pin LLP. The LLP package has the same footprint as

the Thin SOT23-6, but is thermally superior due to the ex-

posed ground paddle on the bottom of the package.

20130374

No Pullback LLP Configuration

RθJA of the LLP package is normally two to three times better

than that of the Thin SOT23-6 package for a similar PCB con-

figuration (area, copper weight, thermal vias).

20130370

FIGURE 7. Dog Bone

For certain high power applications, the PCB land may be

modified to a "dog bone" shape (see Figure 7). By increasing

the size of ground plane, and adding thermal vias, the RθJA

for the application can be reduced.

Design Example 4:

Operating Conditions

Package LLP-6

VIN 12.0V POUT 2.475W

VOUT 3.3V PDIODE 523mW

IOUT 750mA PIND 56.25mW

VD0.35V PSWF 108mW

Freq 3MHz PSWR 108mW

IQ1.5mA PCOND 68.2mW

IBOOST 4mA PQ18mW

VBOOST 5V PBOOST 20mW

TRISE 8ns PLOSS 902mW

TFALL 8ns

RDSON 400mΩ

INDDCR 75mΩ

D30.3%

This example follows example 2, but uses the LLP package.

Using a standard National Semiconductor LLP-6 demonstra-

tion board, use Method 2 to determine RθJA of the board. The

four layer PCB is constructed using FR4 with 1/2oz copper

traces. The copper ground plane is on the bottom layer. The

ground plane is accessed by four vias. The board measures

2.5cm x 3cm. It was placed in an oven with no forced airflow.

The ambient temperature was raised to 113°C, and at that

temperature, the device went into thermal shutdown.

If the junction temperature is to be kept below 125°C, then the

ambient temperature cannot go above 73.2°C.

TJ - (RθJA x PLOSS) = TA

www.national.com 12

LM2734Z/LM2734ZQ

Package Selection

To determine which package you should use for your specific

application, variables need to be known before you can de-

termine the appropriate package to use.

1. Maximum ambient system temperature

2. Internal LM2734Z power losses

3. Maximum junction temperature desired

4. RθJA of the specific application, or RθJC (LLP or Thin

SOT23-6)

The junction temperature must be less than 125°C for the

worst-case scenario.

13 www.national.com

LM2734Z/LM2734ZQ

LM2734Z Design Examples

20130342

FIGURE 8. VBOOST Derived from VIN

Operating Conditions: 5V to 1.5V/1A

Bill of Materials for Figure 8

Part ID Part Value Part Number Manufacturer

U1 1A Buck Regulator LM2734ZX National Semiconductor

C1, Input Cap 10µF, 6.3V, X5R C3216X5ROJ106M TDK

C2, Output Cap 10µF, 6.3V, X5R C3216X5ROJ106M TDK

C3, Boost Cap 0.01uF, 16V, X7R C1005X7R1C103K TDK

D1, Catch Diode 0.3VF Schottky 1A, 10VR MBRM110L ON Semi

D2, Boost Diode 1VF @ 50mA Diode 1N4148W Diodes, Inc.

L1 2.2µH, 1.8A ME3220–222MX Coilcraft

R1 8.87kΩ, 1% CRCW06038871F Vishay

R2 10.2kΩ, 1% CRCW06031022F Vishay

R3 100kΩ, 1% CRCW06031003F Vishay

www.national.com 14

LM2734Z/LM2734ZQ

20130343

FIGURE 9. VBOOST Derived from VOUT

12V to 3.3V/1A

Bill of Materials for Figure 9

Part ID Part Value Part Number Manufacturer

U1 1A Buck Regulator LM2734ZX National Semiconductor

C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK

C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK

C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK

D1, Catch Diode 0.34VF Schottky 1A, 30VR SS1P3L Vishay

D2, Boost Diode 0.6VF @ 30mA Diode BAT17 Vishay

L1 3.3µH, 1.3A ME3220–332MX Coilcraft

R1 31.6kΩ, 1% CRCW06033162F Vishay

R2 10.0 kΩ, 1% CRCW06031002F Vishay

R3 100kΩ, 1% CRCW06031003F Vishay

15 www.national.com

LM2734Z/LM2734ZQ

20130344

FIGURE 10. VBOOST Derived from VSHUNT

18V to 1.5V/1A

Bill of Materials for Figure 10

Part ID Part Value Part Number Manufacturer

U1 1A Buck Regulator LM2734ZX National Semiconductor

C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK

C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK

C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK

C4, Shunt Cap 0.1µF, 6.3V, X5R C1005X5R0J104K TDK

D1, Catch Diode 0.4VF Schottky 1A, 30VR SS1P3L Vishay

D2, Boost Diode 1VF @ 50mA Diode 1N4148W Diodes, Inc.

D3, Zener Diode 5.1V 250Mw SOT-23 BZX84C5V1 Vishay

L1 3.3µH, 1.3A ME3220–332MX Coilcraft

R1 8.87kΩ, 1% CRCW06038871F Vishay

R2 10.2kΩ, 1% CRCW06031022F Vishay

R3 100kΩ, 1% CRCW06031003F Vishay

R4 4.12kΩ, 1% CRCW06034121F Vishay

www.national.com 16

LM2734Z/LM2734ZQ

20130349

FIGURE 11. VBOOST Derived from Series Zener Diode (VIN)

15V to 1.5V/1A

Bill of Materials for Figure 11

Part ID Part Value Part Number Manufacturer

U1 1A Buck Regulator LM2734ZX National Semiconductor

C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK

C2, Output Cap 22µF, 6.3V, X5R C3216X5ROJ226M TDK

C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK

D1, Catch Diode 0.4VF Schottky 1A, 30VR SS1P3L Vishay

D2, Boost Diode 1VF @ 50mA Diode 1N4148W Diodes, Inc.

D3, Zener Diode 11V 350Mw SOT-23 BZX84C11T Diodes, Inc.

L1 3.3µH, 1.3A ME3220–332MX Coilcraft

R1 8.87kΩ, 1% CRCW06038871F Vishay

R2 10.2kΩ, 1% CRCW06031022F Vishay

R3 100kΩ, 1% CRCW06031003F Vishay

17 www.national.com

LM2734Z/LM2734ZQ

20130350

FIGURE 12. VBOOST Derived from Series Zener Diode (VOUT)

15V to 9V/1A

Bill of Materials for Figure 12

Part ID Part Value Part Number Manufacturer

U1 1A Buck Regulator LM2734ZX National Semiconductor

C1, Input Cap 10µF, 25V, X7R C3225X7R1E106M TDK

C2, Output Cap 22µF, 16V, X5R C3216X5R1C226M TDK

C3, Boost Cap 0.01µF, 16V, X7R C1005X7R1C103K TDK

D1, Catch Diode 0.4VF Schottky 1A, 30VR SS1P3L Vishay

D2, Boost Diode 1VF @ 50mA Diode 1N4148W Diodes, Inc.

D3, Zener Diode 4.3V 350mw SOT-23 BZX84C4V3 Diodes, Inc.

L1 2.2µH, 1.8A ME3220–222MX Coilcraft

R1 102kΩ, 1% CRCW06031023F Vishay

R2 10.2kΩ, 1% CRCW06031022F Vishay

R3 100kΩ, 1% CRCW06031003F Vishay

www.national.com 18

LM2734Z/LM2734ZQ

Physical Dimensions inches (millimeters) unless otherwise noted

6-Lead SOT23 Package

NS Package Number MK06A

6-Lead LLP Package

NS Package Number SDE06A

19 www.national.com

LM2734Z/LM2734ZQ

Notes

LM2734Z/LM2734ZQ Thin SOT23 1A Load Step-Down DC-DC Regulator

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