Vout
T1
T1
Vin
ISOLATED
FEEDBACK
SSOFF
VCC
REF AGNDPGNDSSRESRT
UVLO
VIN
HO1 BST1 HS1 LO1 CS LO2 HS2 BST2 HO2
COMP
LM5046 PHASE-SHIFTED
FULL-BRIDGE CONTROLLER
WITH INTEGRATED GATE DRIVERS
RD1 RD2
GATE
DRIVE
ISOLATION
SR1
SR2
VCC VCC
ISOLATION
BOUNDARY
SLOPE RAMP
SSSR
OVP
Q1
Q2
Q3
Q4
Product
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LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
LM5046 Phase-Shifted Full-Bridge PWM Controller With Integrated MOSFET Drivers
1 Features 3 Description
The LM5046 PWM controller contains all of the
1• Highest Integration Controller for Small Form features necessary to implement a phase-shifted full-
Factor, High-Density Power Converters bridge topology power converter using either current
• High-Voltage Start-Up Regulator mode or voltage mode control. This device is
• Intelligent Sync Rectifier Start-Up Allows Linear intended to operate on the primary side of an isolated
DC-DC converter with input voltage up to 100 V. This
Turn-on into Prebiased Loads highly integrated controller-driver provides dual 2-A
• Synchronous Rectifiers Disabled in UVLO Mode high- and low-side gate drivers for the four external
and Hiccup Mode bridge MOSFETs, plus control signals for the
• Two Independent, Programmable Dead-Time secondary-side synchronous rectifier MOSFETs.
Adjustments to Enable Zero-Volt Switching External resistors program the dead-time to enable
zero-volt switching of the primary FETs. Intelligent
• Four High-Current 2-A Bridge Gate Drivers startup of the synchronous rectifiers allows monotonic
• Wide-Bandwidth Opto-Coupler Interface turnon of the power converter even with prebias load
• Configurable for Either Current Mode or Voltage conditions. Additional features include cycle-by-cycle
Mode Control current limiting, hiccup mode restart, programmable
soft-start, synchronous rectifier soft-start, and a 2-
• Dual-Mode Overcurrent Protection MHz capable oscillator with synchronization capability
• Resistor Programmed 2-MHz Oscillator and thermal shutdown.
• Programmable Line UVLO and OVP Device Information(1)
2 Applications PART NUMBER PACKAGE BODY SIZE (NOM)
• E-Bike HTSSOP (28) 9.70 mm x 4.40 mm
LM5046
• Military: Radar/Electronic Warfare WSON (28) 5.00 mm x 5.00 mm
• Power: Telecom DC/DC Module: Analog (1) For all available packages, see the orderable addendum at
the end of the datasheet.
• Private Branch Exchange (PBX)
• Solar Power Inverters Simplified Phase-Shifted Full-Bridge Power
• Vector Signal Generator Converter
• Microwave Oven
• Point-to-Point Microwave Backhaul
• Power: Telecom/Server AC/DC Supply: Dual
Controller: Analog
• Solar Micro-Inverter
• TETRA Base Station
• Washing Machine: Low-End
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
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Table of Contents
7.3 Feature Description................................................. 13
1 Features.................................................................. 17.4 Device Functional Modes........................................ 22
2 Applications ........................................................... 18 Application and Implementation ........................ 25
3 Description............................................................. 18.1 Application Information............................................ 25
4 Revision History..................................................... 28.2 Typical Application.................................................. 25
5 Pin Configuration and Functions......................... 39 Power Supply Recommendations...................... 37
6 Specifications......................................................... 610 Layout................................................................... 37
6.1 Absolute Maximum Ratings ...................................... 610.1 Layout Guidelines ................................................. 37
6.2 Handling Ratings....................................................... 610.2 Layout Example .................................................... 37
6.3 Recommended Operating Conditions....................... 611 Device and Documentation Support................. 39
6.4 Thermal Information.................................................. 711.1 Trademarks........................................................... 39
6.5 Electrical Characteristics........................................... 711.2 Electrostatic Discharge Caution............................ 39
6.6 Typical Characteristics............................................ 10 11.3 Glossary................................................................ 39
7 Detailed Description............................................ 12 12 Mechanical, Packaging, and Orderable
7.1 Overview................................................................. 12 Information........................................................... 39
7.2 Functional Block Diagram....................................... 12
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision G (March 2013) to Revision H Page
• Added Pin Configuration and Functions section, Handling Rating table, Feature Description section, Device
Functional Modes,Application and Implementation section, Power Supply Recommendations section, Layout
section, Device and Documentation Support section, and Mechanical, Packaging, and Orderable Information
section ................................................................................................................................................................................... 1
Changes from Revision F (March 2013) to Revision G Page
• Changed layout of National Data Sheet to TI format ........................................................................................................... 25
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5 mm x 5 mm
WQFN 28
SS
RES
HS2
HO2
BST2
SSSR
REF
RT
SLOPE
RD2
COMP
AGND
RD1
PGND
LO1
VCC
LO2
BST1
SS OFF
SR1
OVP
RAMP
UVLO
CS
HS1
HO1
VIN
SR2
TSSOP28
VIN
PGND
LO1
VCC
LO2
SR2
SR1
HS2
HO2
BST2
HS1
BST1
HO1
COMP
RD2
UVLO
OVP
CS
RAMP
SSSR
AGND
SLOPE
RT
REF
RD1
SS
SS OFF
RES
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5 Pin Configuration and Functions
28-Pin
HTSSOP
Top View
28-Pin
WQFN
Top View
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Pin Functions
PIN
38 PIN 28 PIN I/O DESCRIPTION APPLICATION INFORMATION
WQFN
NAME TSSOP TSSOP NO.
NO. NO.
An external voltage divider from the power source sets the
shutdown and standby comparator levels. When UVLO
Line reaches the 0.4 V threshold the VCC and REF regulators
UVLO 1 1 25 I Undervoltage are enabled. At the 1.25 V threshold, the SS pin is released
Lockout and the controller enters the active mode. Hysteresis is set
by an internal current sink that pulls 20 µA from the external
resistor divider.
An external voltage divider from the input power supply sets
the shutdown level during an over-voltage condition.
OVP/O Overvoltage Alternatively, an external NTC thermistor voltage divider can
2 2 26 I
TP Protection be used to set the shutdown temperature. The threshold is
1.25 V. Hysteresis is set by an internal current that sources
20 µA of current into the external resistor divider.
Modulation ramp for the PWM comparator. This ramp can
be a signal representative of the primary current (current
Input to PWM
RAMP 4 3 27 I mode) or proportional to the input voltage (feed-forward
Comparator voltage mode). This pin is reset to GND at the end of every
cycle.
If CS exceeds 750 mV the PWM output pulse will be
Current Sense terminated, entering cycle-by-cycle current limit. An internal
CS 6 4 28 I Input switch holds CS low for 40 nS after either output switches
high to blank leading edge transients.
A ramping current source from 0 to 100 µA is provided for
Slope slope compensation in current mode control. This pin can be
SLOPE 7 5 1 O Compensation connected through an appropriate resistor to the CS pin to
Current provide slope compensation. If slope compensation is not
required, SLOPE must be tied to ground.
An external opto-coupler connected to the COMP pin
sources current into an internal NPN current mirror. The
Input to the Pulse PWM duty cycle is at maximum with zero input current,
COMP 8 6 2 I Width Modulator while 1 mA reduces the duty cycle to zero. The current
mirror improves the frequency response by reducing the AC
voltage across the opto-coupler.
Output of a 5V Maximum output current is 15 mA. Locally decouple with a
REF 9 7 3 O reference 0.1µF capacitor.
Oscillator The resistance connected between RT and AGND sets the
Frequency oscillator frequency. Synchronization is achieved by AC
RT/SY 10 8 4 I Control and coupling a pulse to the RT/SYNC pin that raises the voltage
NC Frequency at least 1.5 V above the 2 V nominal bias level.
Synchronization
AGND 11 9 5 I Analog Ground Connect directly to the Power Ground.
The resistance connected between RD1 and AGND sets the
Passive to Active
RD1 12 10 6 I delay from the falling edge of HO1/SR1 or LO1/SR2 and the
Delay rising edge of LO1 or HO1 respectively.
The resistance connected between RD2 and AGND sets the
Active to Passive
RD2 13 11 7 I delay from the falling edge of LO2 or HO2 and the rising
Delay edge of HO2 or LO2 respectively.
Whenever the CS pin exceeds the 750 mV cycle-by-cycle
current limit threshold, 30 µA current is sourced into the
RES capacitor for the remainder of the PWM cycle. If the
RES capacitor voltage reaches 1.0 V, the SS capacitor is
discharged to disable the HO1, HO2, LO1, LO2 and SR1,
RES 16 12 8 I Restart Timer SR2 outputs. The SS pin is held low until the voltage on the
RES capacitor has been ramped between 2 V and 4 V eight
times by 10 µA charge and 5 µA discharge currents. After
the delay sequence, the SS capacitor is released to initiate
a normal start-up sequence.
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Pin Functions (continued)
PIN
38 PIN 28 PIN I/O DESCRIPTION APPLICATION INFORMATION
WQFN
NAME TSSOP TSSOP NO.
NO. NO.
An internal 20 µA current source charges the SS pin during
start-up. The input to the PWM comparator gradually rises
as the SS capacitor charges to steadily increase the PWM
SS 17 13 9 I Soft-Start Input duty cycle. Pulling the SS pin to a voltage below 200 mV
stops PWM pulses at HO1,2 and LO1,2 and turns off the
synchronous rectifier FETs to a low state.
An external capacitor and an internal 20 µA current source
Secondary Side set the soft-start ramp for the synchronous rectifiers. The
SSSR 18 14 10 I Soft-Start SSSR capacitor charge-up is enabled after the first output
pulse and SS > 2 V and Icomp < 800 µA
When SS OFF pin is connected to the AGND, the LM5046
soft-stops in the event of a VIN UVLO and Hiccup mode
SSOFF 19 15 11 I Soft-Stop Disable current limit condition. If the SSOFF pin is connected to REF
pin, the controller hard-stops on any fault condition. Refer to
Table 1 for more details.
Synchronous Control output for synchronous rectifier gate. Capable of
SR2 25 19 15 O Rectifier Driver peak sourcing 100 mA and sinking 400 mA.
The output voltage of the start-up regulator is initially
regulated to 9.5V. Once the secondary side soft-start (SSSR
Output of Start- pin) reaches 1 V, the VCC output is reduced to 7.7 V. If an
VCC 27 21 17 I Up Regulator auxiliary winding raises the voltage on this pin above the
regulation set-point, the internal start-up regulator will
shutdown, thus reducing the IC power dissipation.
PGND 28 22 18 I Power Ground Connect directly to Analog Ground
LO1, Low Side Output Alternating output of the PWM gate driver. Capable of 1.5A
29, 26 23, 20 19, 16 O
LO2 Driver peak source and 2A peak sink current.
Synchronous Control output for synchronous rectifier gate. Capable of
SR1 30 24 20 O Rectifier Driver peak sourcing 100 mA and sinking 400 mA.
Bootstrap capacitors connected between BST1, 2 and SW1,
Gate Drive 2 provide bias supply for the high side HO1,2 gate drivers.
BST1,2 33, 22 25, 18 21, 14 I Bootstrap External diodes are required between VCC and BST1,2 to
charge the bootstrap capacitors when SW1,2 are low.
High side PWM outputs capable of driving the upper
High Side Output
HO1,2 34, 21 26, 17 22, 13 O MOSFET of the bridge with 1.5A peak source and 2A peak
Driver sink current.
Common connection of the high side FET source, low side
HS1,2 35, 20 27, 16 23, 12 O Switch Node FET drain and transformer primary winding.
Input to the Start-up Regulator. Operating input range is 14
Input Power V to 100 V. For power sources outside of this range, the
VIN 38 28 24 I Source LM5046 can be biased directly at VCC by an external
regulator.
3, 5, 14,
15, 23,
NC 24, 31, - - - No Connect
32, 36,
37
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6 Specifications
6.1 Absolute Maximum Ratings(1)
MIN MAX UNIT
VIN to GND –0.3 105 V
HS to GND(2) –5 105 V
BST1/BST2 to GND –0.3 116 V
BST1/BST2 to HS1/HS2 –0.3 16 V
HO1/HO2 to HS1/HS2(3) –0.3 BST1/BST2+0.3 V
LO1/LO2/SR1/SR2(3) –0.3 VCC+0.3 V
VCC to GND –0.3 16 V
REF,SSOFF,RT,OVP,UVLO to GND –0.3 7 V
RAMP –0.3 7 V
COMP –0.3 V
COMP Input Current 10 mA
All other inputs to GND(3) –0.3 REF+0.3 V
Junction Temperature 150 °C
(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which
operation of the device is intended to be functional. For specifications and test conditions, see Electrical Characteristics.
(2) The negative HS voltage must never be more negative than VCC-16V. For example, if VCC = 12 V, the negative transients at HS must
not exceed –4 V.
(3) These pins are output pins and as such should not be connected to an external voltage source. The voltage range listed is the limits the
internal circuitry is designed to reliably tolerate in the application circuit.
6.2 Handling Ratings MIN MAX UNIT
Tstg Storage temperature range –55 150 °C
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all 2 kV
pins(1)
V(ESD) Electrostatic discharge Charged device model (CDM), per JEDEC specification 750 V
JESD22-C101, all pins(2)
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
6.3 Recommended Operating Conditions
over operating free-air temperature range (unless otherwise noted) MIN MAX UNIT
VIN Voltage 14 100 V
External Voltage Applied to VCC 10 14 V
Junction Temperature –40 125 °C
SLOPE –0.3 2 V
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6.4 Thermal Information LM5046
THERMAL METRIC(1) PWP RSG UNIT
28 PINS
RθJA Junction-to-ambient thermal resistance 33.9 37.4
RθJC(top) Junction-to-case (top) thermal resistance 18.1 21.8
RθJB Junction-to-board thermal resistance 15.7 10.1 °C/W
ψJT Junction-to-top characterization parameter 0.4 0.2
ψJB Junction-to-board characterization parameter 15.6 10
RθJC(bot) Junction-to-case (bottom) thermal resistance 2.0 2.6
(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
6.5 Electrical Characteristics
Limits in standard typeface are for TJ= 25°C only; for the MIN and MAX apply the junction temperature range of –40°C to
125°C. Unless otherwise specified, the following conditions apply: VIN = 48 V, RT = 25 kΩ, RD1 = RD2 = 20 kΩ. No load on
HO1, HO2, LO1, LO2, SR1, SR2, COMP=0 V, UVLO = 2.5 V, OVP = 0 V, SSOFF = 0 V.
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
STARTUP REGULATOR (VCC PIN)
VCC1 VCC voltage ICC= 10 mA (SSSR < 1 V) 9.3 9.6 9.9 V
VCC2 VCC voltage ICC= 10 mA (SSSR > 1 V) 7.5 7.8 8.1 V
ICC(Lim) VCC current limit VCC= 6 V 60 80 mA
ICC(ext) VCC supply current Supply current into VCC from an externally 4.6 mA
applied source. VCC = 10 V
VCC load regulation ICC from 0 to 50 mA 35 mV
VCC(UV) VCC under-voltage threshold Positive going VCC VCC1–0.2 VCC1–0.1 V
VCC under-voltage threshold Negative going VCC 5.9 6.3 6.7 V
IIN VIN operating current 4 mA
VIN shutdown current VIN = 20 V, VUVLO = 0 V 300 520 µA
VVIN = 100 V, VUVLO = 0 V 350 550 µA
VIN start-up regulator leakage VCC = 10 V 160 µA
VOLTAGE REFERENCE REGULATOR (REF PIN)
VREF REF Voltage IREF = 0 mA 4.85 5 5.15 V
REF voltage regulation IREF = 0 to 10 mA 25 50 mV
IREF(Lim) REF current limit VREF = 4.5 V 15 20 mA
VREFUV VREF under-voltage threshold Positive going VREF 4.3 4.5 4.7 V
Hysteresis 0.25 V
UNDERVOLTAGE LOCK OUT AND SHUTDOWN (UVLO PIN)
VUVLO Under-voltage threshold 1.18 1.25 1.32 V
IUVLO Hysteresis current UVLO pin sinking current when VUVLO < 16 20 24 µA
1.25 V
Under-voltage standby enable UVLO voltage rising 0.32 0.4 0.48 V
threshold
Hysteresis 0.05 V
VOVP OVP shutdown threshold OVP rising 1.18 1.25 1.32 V
OVP hysteresis current OVP sources current when OVP > 1.25 V 16 20 24 µA
SOFT-START (SS PIN)
ISS SS charge current VSS = 0 V 16 20 24 µA
SS threshold for SSSR charge ICOMP < 800 µA 1.93 2.0 2.20 V
current enable
SS output low voltage Sinking 100 µA 40 mV
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Electrical Characteristics (continued)
Limits in standard typeface are for TJ= 25°C only; for the MIN and MAX apply the junction temperature range of –40°C to
125°C. Unless otherwise specified, the following conditions apply: VIN = 48 V, RT = 25 kΩ, RD1 = RD2 = 20 kΩ. No load on
HO1, HO2, LO1, LO2, SR1, SR2, COMP=0 V, UVLO = 2.5 V, OVP = 0 V, SSOFF = 0 V.
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
SS threshold to disable switching 200 mV
ISSSR SSSR charge current VSS > 2 V, ICOMP < 800 µA 16 20 24 µA
ISSSR-DIS1 SSSR discharge current 1 VUVLO < 1.25 V 54 65 75 µA
ISSSR-DIS2 SSSR discharge current 2 VRES > 1 V 109 125 147 µA
SSSR output low voltage Sinking 100 µA 50 mV
SSSR threshold to enable SR1/SR2 1.2 V
CURRENT SENSE INPUT (CS PIN)
VCS Current limit threshold 0.710 0.750 0.785 V
CS delay to output 65 ns
CS leading edge blanking 50 ns
RCS CS sink impedance (clocked) Internal FET sink impedance 18 45 Ω
SOFT-STOP DISABLE (SS OFF PIN)
VIH(min) SSOFF Input Threshold 2.8 V
SSOFF pull down resistance 200 kΩ
CURRENT LIMIT RESTART (RES PIN)
RRES RES pull-down resistance Termination of hiccup timer 37 Ω
VRES RES hiccup threshold 1 V
RES upper counter threshold 4 V
RES lower counter threshold 2 V
IRES-SRC1 Charge current source 1 VRES < 1 V, VCS> 750 mV 30 µA
IRES-SRC2 Charge current source 2 1 V < VRES < 4 V 10 µA
IRES-DIS2 Discharge current source 1 VCS < 750 mV 5 µA
IRES-DIS2 Discharge current source 2 2 V < VRES < 4 V 5 µA
Ratio of time in hiccup mode to time VRES > 1 V, Hiccup counter 147
in current limit
VOLTAGE FEED-FORWARD (RAMP PIN)
RAMP sink impedance (Clocked) 5.5 20 Ω
OSCILLATOR (RT PIN)
FSW1 Frequency (LO1, half oscillator RT= 25 kΩ185 200 215 kHz
frequency)
FSW2 Frequency (LO1, half oscillator RT= 10 kΩ420 480 540 kHz
frequency)
DC level 2.0 V
RT sync threshold 2.8 3 3.3 V
ZVS TIMING CONTROL (RD1 & RD2 PINS)
TPA HO1/SR1 turn-off to LO1 turn-on RD1 = 20 kΩ39 65 89 ns
LO1/SR2 turn-off to HO1 turn-on RD1 = 100 kΩ230 300 391 ns
TAP LO2 turn-off to HO2 turn-on RD2 = 20 kΩ27 55 78 ns
HO2 turn-off to LO2 turn-on RD2 = 100 kΩ214 300 378 ns
COMP PIN
VPWM-OS COMP current to RAMP offset VRAMP = 0 V 680 800 940 µA
VSS-OS SS to RAMP offset VRAMP = 0 V 0.78 1.0 1.22 V
COMP current to RAMP gain ΔRAMP/ΔICOMP 2400 Ω
SS to RAMP gain ΔSS/ΔRAMP 0.5
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Electrical Characteristics (continued)
Limits in standard typeface are for TJ= 25°C only; for the MIN and MAX apply the junction temperature range of –40°C to
125°C. Unless otherwise specified, the following conditions apply: VIN = 48 V, RT = 25 kΩ, RD1 = RD2 = 20 kΩ. No load on
HO1, HO2, LO1, LO2, SR1, SR2, COMP=0 V, UVLO = 2.5 V, OVP = 0 V, SSOFF = 0 V.
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
COMP current for SSSR charge VSS > 2 V 690 800 915 µA
current enable
COMP to output delay 120 ns
Minimum duty cycle ICOMP = 1 mA 0 %
SLOPE COMPENSATION (SLOPE PIN)
ISLOPE Slope compensation current ramp Peak of RAMP current 100 µA
BOOST (BST PIN)
VBst uv BST under-voltage threshold VBST – VHS rising 3.8 4.7 5.6 V
Hysteresis 0.5 V
HO1, HO2, LO1, LO2 GATE DRIVERS
VOL Low-state output voltage IHO/LO = 100 mA 0.16 0.32 V
VOH High-state output voltage IHO/LO = 100 mA 0.27 0.495 V
VOHL = VCC – VLO
VOHH = VBST – VHO
Rise Time C-load = 1000 pF 16 ns
Fall Time C-load = 1000 pF 11 ns
IOHL Peak Source Current VHO/LO = 0 V 1.5 - A
IOLL Peak Sink Current VHO/LO = VCC 2 - A
SR1, SR2 GATE DRIVERS
VOL Low-state output voltage ISR1/SR2 = 10 mA 0.05 0.10 V
VOH High-state output voltage ISR1/SR2 = 10 mA, 0.17 0.28 V
VOH = VREF – VSR
Rise Time C-load = 1000 pF 60 ns
Fall Time C-load = 1000 pF 20 ns
IOHL Peak Source Current VSR = 0 V 0.1 - A
IOLL Peak Sink Current VSR = VREF 0.4 - A
THERMAL
TSD Thermal Shutdown Temp 160 °C
Thermal Shutdown Hysteresis 25 °C
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0 20 40 60 80 100
0
1
2
3
4
5
6
IIN(V)
VIN(V)
VUVLO=3V
VUVLO=1V
VUVLO=0V
5 10 15 20 25 30
50
60
70
80
90
100
EFFICIENCY (%)
LOAD CURRENT (A)
36V
48V
72V
VOUT= 3.3V
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6.6 Typical Characteristics
Figure 2. VCC vs ICC
Figure 1. Application Board Efficiency
Figure 3. VVCC and VREF vs. VVIN Figure 4. IIN vs. VIN
Figure 5. VREF vs. IREF Figure 6. Oscillator Frequency vs. RT
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Typical Characteristics (continued)
Figure 7. Dead-Time TPA, TAP vs. Temperature Figure 8. Dead-Time TPA, TAP vs. RD1, RD2
Figure 9. CS Threshold vs. Temperature
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7 Detailed Description
7.1 Overview
The LM5046 PWM controller contains all of the features necessary to implement a Phase-Shifted Full-Bridge
(PSFB) topology power converter using either current mode or voltage mode control. This device is intended to
operate on the primary side of an isolated dc-dc converter with input voltage up to 100 V. This highly integrated
controller-driver provides dual 2A high and low side gate drivers for the four external bridge MOSFETs plus
control signals for secondary side synchronous rectifiers. External resistors program the dead-time to enable
Zero-Volt Switching (ZVS) of the primary FETs. Please refer to the Application and Implementation section for
details on the operation of the PSFB topology. Intelligent startup of synchronous rectifier allows turn-on of the
power converter into the pre-bias loads. Cycle-by-cycle current limit protects the power components from load
transients while hiccup mode protection limits average power dissipation during extended overload conditions.
Additional features include programmable soft-start, soft-start of the synchronous rectifiers, and a 2 MHz capable
oscillator with synchronization capability and thermal shutdown.
7.2 Functional Block Diagram
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7.3 Feature Description
7.3.1 High-Voltage Start-Up Regulator
The LM5046 contains an internal high voltage start-up regulator that allows the input pin (VIN) to be connected
directly to the supply voltage over a wide range from 14 V to 100 V. The input can withstand transients up to 105
V. When the UVLO pin potential is greater than 0.4 V, the VCC regulator is enabled to charge an external
capacitor connected to the VCC pin. The VCC regulator provides power to the voltage reference (REF) and the
gate drivers (HO1/HO2 and LO1/LO2). When the voltage on the VCC pin exceeds its Under Voltage (UV)
threshold, the internal voltage reference (REF) reaches its regulation set point of 5V and the UVLO voltage is
greater than 1.25 V, the soft-start capacitor is released and normal operation begins. The regulator output at
VCC is internally current limited. The value of the VCC capacitor depends on the total system design, and its
start-up characteristics. The recommended range of values for the VCC capacitor is 0.47 μF to 10 µF.
The internal power dissipation of the LM5046 can be reduced by powering VCC from an external supply. The
output voltage of the VCC regulator is initially regulated to 9.5 V. After the synchronous rectifiers are engaged
(which is approximately when the output voltage in within regulation), the VCC voltage is reduced to 7.7 V. In
typical applications, an auxiliary transformer winding is connected through a diode to the VCC pin. This winding
must raise the VCC voltage above 8 V to shut off the internal start-up regulator. Powering VCC from an auxiliary
winding improves efficiency while reducing the controller’s power dissipation. The VCC UV circuit will still function
in this mode, requiring that VCC never falls below its UV threshold during the start-up sequence. The VCC
regulator series pass transistor includes a diode between VCC and VIN that should not be forward biased in
normal operation. Therefore, the auxiliary VCC voltage should never exceed the VIN voltage.
An external DC bias voltage can be used instead of the internal regulator by connecting the external bias voltage
to both the VCC and the VIN pins. This implementation is shown in the Application and Implementation section.
The external bias must be greater than 10 V and less than the VCC maximum voltage rating of 14 V.
7.3.2 Line Undervoltage Detector
The LM5046 contains a dual level Under-Voltage Lockout (UVLO) circuit. When the UVLO pin voltage is below
0.4 V, the controller is in a low current shutdown mode. When the UVLO pin voltage is greater than 0.4 V but
less than 1.25 V, the controller is in standby mode. In standby mode the VCC and REF bias regulators are active
while the controller outputs are disabled. When the VCC and REF outputs exceed their respective under-voltage
thresholds and the UVLO pin voltage is greater than 1.25 V, the soft-start capacitor is released and the normal
operation begins. An external set-point voltage divider from VIN to GND can be used to set the minimum
operating voltage of the converter. The divider must be designed such that the voltage at the UVLO pin will be
greater than 1.25 V when VIN enters the desired operating range. UVLO hysteresis is accomplished with an
internal 20μA current sink that is switched on or off into the impedance of the set-point divider. When the UVLO
threshold is exceeded, the current sink is deactivated to quickly raise the voltage at the UVLO pin. When the
UVLO pin voltage falls below the 1.25 V threshold, the current sink is enabled causing the voltage at the UVLO
pin to quickly fall. The hysteresis of the 0.4V shutdown comparator is internally fixed at 50 mV.
The UVLO pin can also be used to implement various remote enable / disable functions. Turning off the
converter by forcing the UVLO pin to standby condition (0.4 V < UVLO < 1.25 V) provides a controlled soft-stop.
Refer to the Soft-Stop section for more details.
7.3.3 Overvoltage Protection
An external voltage divider can be used to set either an over voltage or an over temperature protection. During
an OVP condition, the SS and SSSR capacitors are discharged and all the outputs are disabled. The divider
must be designed such that the voltage at the OVP pin is greater than 1.25 V when over voltage/temperature
condition exists. Hysteresis is accomplished with an internal 20μA current source. When the OVP pin voltage
exceeds 1.25 V, the 20 μA current source is activated to quickly raise the voltage at the OVP pin. When the OVP
pin voltage falls below the 1.25 V threshold, the current source is deactivated causing the voltage at the OVP to
quickly fall. Refer to the Application and Implementation section for more details.
7.3.4 Reference
The REF pin is the output of a 5 V linear regulator that can be used to bias an opto-coupler transistor and
external housekeeping circuits. The regulator output is internally current limited to 15mA. The REF pin needs to
be locally decoupled with a ceramic capacitor, the recommended range of values are from 0.1 μF to 10 μF
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 13
Product Folder Links: LM5046
RT =1
FOSC x 1 x 10-10
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Feature Description (continued)
7.3.5 Oscillator, Sync Input
The LM5046 oscillator frequency is set by a resistor connected between the RT pin and AGND. The RT resistor
should be located very close to the device. To set a desired oscillator frequency (FOSC), the necessary value of
RT resistor can be calculated from Equation 1:
(1)
For example, if the desired oscillator frequency is 400 kHz i.e. each phase (LO1 or LO2) at 200 kHz, the value of
RTwill be 25 kΩ. If the LM5046 is to be synchronized to an external clock, that signal must be coupled into the
RT pin through a 100 pF capacitor. The RT pin voltage is nominally regulated at 2.0 V and the external pulse
amplitude should lift the pin to between 3.5 V and 5.0 V on the low-to-high transition. The synchronization pulse
width should be between 15 and 200 ns. The RT resistor is always required, whether the oscillator is free running
or externally synchronized and the SYNC frequency must be equal to, or greater than the frequency set by the
RT resistor. When syncing to an external clock, it is recommended to add slope compensation by connecting an
appropriate resistor from the VCC pin to the CS pin. Also disable the SLOPE pin by grounding it.
7.3.6 Cycle-by-Cycle Current Limit
The CS pin is to be driven by a signal representative of the transformer’s primary current. If the voltage on the
CS pin exceeds 0.75 V, the current sense comparator immediately terminates the PWM cycle. A small RC filter
connected to the CS pin and located near the controller is recommended to suppress noise. An internal 18 Ω
MOSFET discharges the external current sense filter capacitor at the conclusion of every cycle. The discharge
MOSFET remains on for an additional 40 ns after the start of a new PWM cycle to blank leading edge spikes.
The current sense comparator is very fast and may respond to short duration noise pulses. Layout is critical for
the current sense filter and the sense resistor. The capacitor associated with CS filter must be placed very close
to the device and connected directly to the CS and AGND pins. If a current sense transformer is used, both the
leads of the transformer secondary should be routed to the filter network, which should be located close to the
IC. When designing with a current sense resistor, all of the noise sensitive low power ground connections should
be connected together near the AGND pin, and a single connection should be made to the power ground (sense
resistor ground point).
7.3.7 Hiccup Mode
The LM5046 provides a current limit restart timer to disable the controller outputs and force a delayed restart (i.e.
Hiccup mode) if a current limit condition is repeatedly sensed. The number of cycle-by-cycle current limit events
required to trigger the restart is programmed by the external capacitor at the RES pin. During each PWM cycle,
the LM5046 either sources or sinks current from the RES capacitor. If current limit is detected, the 5 μA current
sink is disabled and a 30 μA current source is enabled. If the RES voltage reaches the 1.0 V threshold, the
following restart sequence occurs, as shown in Figure 10:
• The SS and SSSR capacitors are fully discharged
• The 30 μA current source is turned-off and the 10 μA current source is turned-on.
• Once the voltage at the RES pin reaches 4.0V the 10 μA current source is turned-off and a 5 μA current sink
is turned-on, ramping the voltage on the RES capacitor down to 2.0 V.
• Once RES capacitor reaches 2.0V, threshold, the 10μA current source is turned-on again. The RES capacitor
voltage is ramped between 4.0V and 2.0V eight times.
• When the counter reaches eight, the RES pin voltage is pulled low and the soft-start capacitor is released to
begin a soft-start sequence. The SS capacitor voltage slowly increases. When the SS voltage reaches 1.0 V,
the PWM comparator will produce the first narrow pulse.
• If the overload condition persists after restart, cycle-by-cycle current limiting will begin to increase the voltage
on the RES capacitor again, repeating the hiccup mode sequence.
• If the overload condition no longer exists after restart, the RES pin will be held at ground by the 5 μA current
sink and the normal operation resumes.
The hiccup mode function can be completely disabled by connecting the RES pin to the AGND pin. In this
configuration the cycle-by-cycle protection will limit the maximum output current indefinitely, no hiccup restart
sequences will occur.
14 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
Hiccup Mode off-time
Soft-Start
Restart delay
1V
2V
4V
Count to Eight
1V
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Feature Description (continued)
Figure 10. Hiccup Mode Delay and Soft-Start Timing Diagram
7.3.8 PWM Comparator
The LM5046 pulse width modulator (PWM) comparator is a three input device, it compares the signal at the
RAMP pin to the loop error signal or the soft-start, whichever is lower, to control the duty cycle. This comparator
is optimized for speed in order to achieve minimum controllable duty cycles. The loop error signal is received
from the external feedback and isolation circuit in the form of a control current into the COMP pin. The COMP pin
current is internally mirrored by a matching pair of NPN transistors which sink current through a 5 kΩresistor
connected to the 5 V reference. The resulting control voltage passes through a 1 V offset, followed by a 2:1
resistor divider before being applied to the PWM comparator.
An opto-coupler detector can be connected between the REF pin and the COMP pin. Because the COMP pin is
controlled by a current input, the potential difference across the opto-coupler detector is nearly constant. The
bandwidth limiting phase delay which is normally introduced by the significant capacitance of the opto-coupler is
thereby greatly reduced. Higher loop bandwidths can be realized since the bandwidth limiting pole associated
with the opto-coupler is now at a much higher frequency. The PWM comparator polarity is configured such that
with no current flowing into the COMP pin, the controller produces maximum duty cycle.
7.3.9 RAMP Pin
The voltage at the RAMP pin provides the modulation ramp for the PWM comparator. The PWM comparator
compares the modulation ramp signal at the RAMP pin to the loop error signal to control the duty cycle. The
modulation ramp signal can be implemented either as a ramp proportional to the input voltage, known as feed-
forward voltage mode control, or as a ramp proportional to the primary current, known as current mode control.
The RAMP pin is reset by an internal MOSFET with an RDS(ON) of 5.5 Ωat the conclusion of each PWM cycle.
The ability to configure the RAMP pin for either voltage mode or current mode allows the controller to be
implemented for the optimum control method depending upon the design constraints. Refer to the Application
and Implementation section for more details on configuring the RAMP pin for feed-forward voltage mode control
and peak current mode control.
7.3.10 Slope Pin
For duty cycles greater than 50% (25% for each phase), peak current mode control is subject to sub-harmonic
oscillation. Sub-harmonic oscillation is normally characterized by observing alternating wide and narrow duty
cycles. This can be eliminated by adding an artificial ramp, known as slope compensation, to the modulating
signal at the RAMP pin. The SLOPE pin provides a current source ramping from 0 to 100μA, at the frequency set
by the RT resistor, for slope compensation. The ramping current source at the SLOPE pin can be utilized in a
couple of different ways to add slope compensation to the RAMP signal:
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Product Folder Links: LM5046
CS
RFILTER
RCS
LM5046
RAMP
CLK + LEB
CLK
Current
Sense
CFILTER
SLOPE
100 PA
0
CS
RFILTER
RCS
LM5046
RAMP
CLK + LEB
CLK
Current
Sense
CFILTER
SLOPE
100 PA
0
RSLOPE
(a) (b)
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Feature Description (continued)
1) As shown in Figure 11(a), the SLOPE and RAMP pins can be connected together through an appropriate
resistor to the CS pin. This configuration will inject current sense signal plus slope compensation to the RAMP
pin but CS pin will not see any slope compensation. Therefore, in this scheme slope compensation will not affect
the current limit.
2) In a second configuration, as shown in Figure 11(b), the SLOPE, RAMP and CS pins can be tied together. In
this configuration the ramping current source from the SLOPE pin will flow through the filter resistor and filter
capacitor, therefore both the CS pin and the RAMP pin will see the current sense signal plus the slope
compensation ramp. In this scheme, the current limit is compensated by the slope compensation and the current
limit onset point will vary.
If slope compensation is not required, for example in feed-forward voltage mode control, the SLOPE pin must be
connected to the AGND pin. When the RT pin is synched to an external clock, it is recommended to disable the
SLOPE pin and add slope compensation externally by connecting an appropriate resistor from the VCC pin to the
CS pin. Please refer to the Application and Implementation section for more details.
(a) Slope Compensation Configured for PWM Only (No Current Limit Slope)
(b) Slope Compensation Configured for PWM and Current Limit
Figure 11. Slope Compensation Configuration
7.3.11 Soft-Start
The soft-start circuit allows the power converter to gradually reach a steady state operating point, thereby
reducing the start-up stresses and current surges. When bias is supplied to the LM5046, the SS capacitor is
discharged by an internal MOSFET. When the UVLO, VCC and REF pins reach their operating thresholds, the
SS capacitor is released and is charged with a 20µA current source. Once the SS pin voltage crosses the 1 V
offset, SS controls the duty cycle. The PWM comparator is a three input device; it compares the RAMP signal
against the lower of the signals between the soft-start and the loop error signal. In a typical isolated application,
as the secondary bias is established, the error amplifier on the secondary side soft-starts and establishes closed-
loop control, steering the control away from the SS pin.
One method to shutdown the regulator is to ground the SS pin. This forces the internal PWM control signal to
ground, reducing the output duty cycle quickly to zero. Releasing the SS pin begins a soft-start cycle and normal
operation resumes. A second shutdown method is presented in the UVLO section.
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Product Folder Links: LM5046
1
FOSC
DMAX = - (TPA)
( )
2
FOSC)(
LM5046
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SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Feature Description (continued)
7.3.12 Gate Driver Outputs
The LM5046 provides four gate drivers: two floating high side gate drivers HO1 and HO2 and two ground
referenced low side gate drivers LO1 and LO2. Each internal driver is capable of sourcing 1.5A peak and sinking
2A peak. The low-side gate drivers are powered directly by the VCC regulator. The HO1 and HO2 gate drivers
are powered from a bootstrap capacitor connected between BST1/BST2 and HS1/HS2 respectively. An external
diode connected between VCC (anode pin) and BST (cathode pin) provides the high side gate driver power by
charging the bootstrap capacitor from VCC when the corresponding switch node (HS1/HS2 pin) is low. When the
high side MOSFET is turned on, BST1 rises to a peak voltage equal to VCC + VHS1 where VHS1 is the switch
node voltage.
The BST and VCC capacitors should be placed close to the pins of the LM5046 to minimize voltage transients
due to parasitic inductances since the peak current sourced to the MOSFET gates can exceed 1.5A. The
recommended value of the BST capacitor is 0.1 μF or greater. A low ESR / ESL capacitor, such as a surface
mount ceramic, should be used to prevent voltage droop during the HO transitions.
Figure 12 illustrates the sequence of the LM5046 gate-drive outputs. Initially, the diagonal HO1 and LO2 are
turned-on together during the power transfer cycle, followed by the freewheel cycle, where HO1 and HO2 are
kept on. In the subsequent phase, the diagonal HO2 and LO1 are turned-on together during the power transfer
cycle, followed by a freewheel cycle, where LO1 and LO2 are kept on. The power transfer mode is often called
the active mode and the freewheel mode is often called as the passive mode. The dead-time between the
passive mode and the active mode, TPA, is set by the RD1 resistor and the dead-time between the active mode
and the passive mode, TAP, is set by the RD2 resistor. Refer to the Application and Implementation section for
more details on the operation of the phase-shifted full-bridge topology.
If the COMP pin is open circuit, the outputs will operate at maximum duty cycle. The maximum duty cycle for
each phase is limited by the dead-time set by the RD1 resistor. If the RD1 resistor is set to zero then the
maximum duty cycle is slightly less than 50% due to the internally fixed dead-time. The internally fixed dead-time
is 30 ns which does not vary with the operating frequency. The maximum duty cycle for each output can be
calculated from Equation 2:
(2)
Where, TPA is the time set by the RD1 resistor and FOSC is the frequency of the oscillator. For example, if the
oscillator frequency is set at 400 kHz and the TPA time set by the RD1 resistor is 60 ns, the resulting DMAX will be
equal to 0.488.
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 17
Product Folder Links: LM5046
HO1
LO2
HO2
CLK
IL
LO1
HO1,LO2 HO1,HO2 HO2,LO1 LO1,LO2 HO1,LO2 HO1,HO2
Power
Transfer Freewheel Power
Transfer Freewheel Power
Transfer Freewheel
TAP
TAP
TAP
TPA TPA
FOSC
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Feature Description (continued)
Figure 12. Timing Diagram Illustrating the Sequence of Gate-Driver Outputs in the PSFB Topology
7.3.13 Synchronous Rectifier Control Outputs (SR1 & SR2)
Synchronous rectification (SR) of the transformer secondary provides higher efficiency, especially for low output
voltage converters, compared to the diode rectification. The reduction of rectifier forward voltage drop (0.5 V –
1.5 V) to 10 mV – 200 mV VDS voltage for a MOSFET significantly reduces rectification losses. In a typical
application, the transformer secondary winding is center tapped, with the output power inductor in series with the
center tap. The SR MOSFETs provide the ground path for the energized secondary winding and the inductor
current. From Figure 13 it can be seen that when the HO1/LO2 diagonal is turned ON, power transfer is enabled
from the primary. During this period, the SR1 MOSFET is enabled and the SR2 MOSFET is turned-off. The
secondary winding connected to the SR2 MOSFET drain is twice the voltage of the center tap at this time. At the
conclusion of the HO1/LO2 pulse, the inductor current continues to flow through the SR2 MOSFET body diode.
Since the body diode causes more loss than the SR MOSFET, efficiency can be improved by minimizing the
TSRON period. In the LM5046, the time TSRON is internally fixed to be 30ns. The 30ns internally fixed dead-time,
along with inherent system delays due to galvanic isolation, plus the gate drive ICs, will provide sufficient margin
to prevent the shoot-through current.
During the freewheeling period, the inductor current flows in both the SR1 and SR2 MOSFETs, which effectively
shorts the transformer secondary. The SR MOSFETs are disabled at the rising edge of the CLK, which also
disables HO1 or LO1. As shown in Figure 13, SR1 is disabled at the same instant as HO1 is disabled, and SR2
is disabled at the same instant as LO1 is disabled. The dead-times, TSROFF and TPA achieve two different things
but are set by single resistor, RD1. Therefore, RD1 value should be selected such that the SR1/SR2 turns-off
before the next power transfer cycle is initiated by TPA.
The SR drivers are powered by the REF regulator and each SR output is capable of sourcing 0.1A and sinking
0.4A peak. The amplitude of the SR drivers is limited to 5 V. The 5 V SR signals enable the LM5046 to transfer
SR control across the isolation barrier either through a solid-state isolator or a pulse transformer. The actual gate
sourcing and sinking currents for the synchronous MOSFETs are provided by the secondary-side bias and gate
drivers.
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Product Folder Links: LM5046
HO1
LO2
HO2
CLK
IL
LO1
HO1,LO2 HO1,HO2 HO2,LO1 LO1,LO2 HO1,LO2 HO1,HO2
Power
Transfer Freewheel Power
Transfer Freewheel Power
Transfer Freewheel
TAP
TAP
TAP
SR1
SR2
TSRON
TSRON
TPA = TSROFF
TPA = TSROFF
RD(1,2) = TPA, TAP
3 pF ; For 20k < (1,2) < 100k
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Feature Description (continued)
TPA and TAP can be programmed by connecting a resistor between RD1 and RD2 pins and AGND. It should be
noted that while RD1 effects the maximum duty cycle, RD2 does not. The RD1 and RD2 resistors should be
located very close to the device. The formula for RD1 and RD2 resistors are given in Equation 3:
(3)
If the desired dead-time for TPA is 60 ns, then the RD1 will be 20 kΩ.
Figure 13. Synchronous Rectifier Timing Diagram
7.3.14 Soft-Start of the Synchronous Rectifiers
In addition to the basic soft-start already described, the LM5046 contains a second soft-start function that
gradually turns on the synchronous rectifiers to their steady-state duty cycle. This function keeps the
synchronous rectifiers off during the basic soft-start allowing a linear start-up of the output voltage even into pre-
biased loads. Then the SR output duty cycle is gradually increased to prevent output voltage disturbances due to
the difference in the voltage drop between the body diode and the channel resistance of the synchronous
MOSFETs. Initially, when bias is supplied to the LM5046, the SSSR capacitor is discharged by an internal
MOSFET. When the SS capacitor reaches a 2 V threshold and once it is established that COMP is in control of
the duty cycle i.e. ICOMP < 800 µA, the SSSR discharge is released and SSSR capacitor begins charging with a
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 19
Product Folder Links: LM5046
Power
Transfer
(HO2, LO1)
SR1
SR2
CLK
TSROFF TSRON
TSROFFTSRON
Power
Transfer
(HO1, LO2)
(a) (b)
TSRON is internally fixed to 30 ns
TSROFF is set by a resistor on the RD1 pin
Freewheel
Power
Transfer
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Feature Description (continued)
20 µA current source. Once the SSSR cap crosses the internal 1 V threshold, the LM5046 begins the soft-start of
the synchronous FETs. The SR soft-start follows a leading edge modulation technique, that is, the leading edge
of the SR pulse is soft-started as opposed trailing edge modulation of the primary FETs. As shown in
Figure 14(a), SR1 and SR2 are turned-on simultaneously with a narrow pulse-width during the freewheeling
cycle. At the end of the freewheel cycle i.e. at the rising edge of the internal CLK, the SR FET in-phase with the
next power transfer cycle is kept on while the SR FET out of phase with it is turned-off. The in-phase SR FET is
kept on throughout the power transfer cycle and at the end of it, both the primary FETs and the in-phase SR
FETs are turned-off together. The synchronous rectifier outputs can be disabled by grounding the SSSR pin.
Figure 14. (a) Waveforms during Soft-Start (b) Waveforms after Soft-Start
7.3.15 Pre-Bias Startup
A common requirement for power converters is to have a monotonic output voltage start-up into a pre-biased
load i.e. a pre-charged output capacitor. In a pre-biased load condition, if the synchronous rectifiers are engaged
prematurely they will sink current from the pre-charged output capacitors resulting in an undesired output voltage
dip. This condition is undesirable and could potentially damage the power converter. The LM5046 utilizes unique
control circuitry to ensure intelligent turn-on of the synchronous rectifiers such that the output has a monotonic
startup. Initially, the SSSR capacitor is held at ground to disable the synchronous MOSFETs allowing the body
diode to conduct. The synchronous rectifier soft-start is initiated once it is established the duty cycle is controlled
by the COMP instead of the soft-start capacitor i.e. ICOMP < 800 µA and the voltage at the SS pin > 2 V. The
SSSR capacitor is then released and is charged by a 20 µA current source. Further, as shown in Figure 15,a1
V offset on the SSSR pin is used to provide additional delay. This delay ensures the output voltage is in
regulation avoiding any reverse current when the synchronous MOSFETs are engaged.
7.3.16 Soft-Stop
As shown in Figure 16, if the UVLO pin voltage falls below the 1.25 V standby threshold, but above the 0.4 V
shutdown threshold, the SSSR capacitor is soft-stopped with a 60 µA current source (3 times the charging
current). Once the SSSR pin reaches the 1.0 V threshold, both the SS and SSSR pins are immediately
discharged to GND. Soft-stopping the power converter gradually winds down the energy in the output capacitors
and results in a monotonic decay of the output voltage. During the hiccup mode, the same sequence is executed
except that the SSSR is discharged with a 120 µA current source (6 times the charging current). In case of an
OVP, VCC UV, thermal limit or a VREF UV condition, the power converter hard-stops, whereby all of the control
outputs are driven to a low state immediately.
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Product Folder Links: LM5046
VIN UVLO
SS
SSSR
1.25V 1.25V
0.45V
1.0V
SS
Primary
SR1, SR2
SSSR
1.0V
VOUT
1.0V
Prebiased Load
COMP
Secondary
Bias
2.0V
LM5046
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SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Feature Description (continued)
Figure 15. Pre-Bias Voltage Startup Waveforms
Figure 16. Stop-Stop Waveforms during a UVLO Event
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Feature Description (continued)
7.3.17 Soft-Stop Off
The Soft-Start Off (SSOFF) pin gives additional flexibility by allowing the power converter to be configured for
hard-stop during line UVLO and hiccup mode condition. If the SS OFF pin is pulled up to the 5 V REF pin, the
power converter hard-stops in any fault condition. Hard-stop drives each control output to a low state
immediately. Refer to Table 1 for more details.
Table 1. Soft-Stop in Fault Conditions
FAULT CONDITION SSSR
UVLO Soft-Stop
(UVLO<1.25V) 3x the charging rate
OVP Hard-Stop
(OVP>1.25V)
Hiccup Soft-Stop
(CS>0.75 and RES>1V) 6x the charging rate
VCC/VREF UV Hard-Stop
Internal Thermal Limit Hard-Stop
Note: All the above conditions are valid with SSOFF pin tied to GND. If SSOFF = 5 V, the LM5046 hard-stops in
all the conditions. The SS pin remains high in all the conditions until the SSSR pin reaches 1 V.
7.3.18 Thermal Protection
Internal thermal shutdown circuitry is provided to protect the integrated circuit in the event the maximum rated
junction temperature is exceeded. When activated, typically at 160°C, the controller is forced into a shutdown
state with the output drivers, the bias regulators (VCC and REF) disabled. This helps to prevent catastrophic
failures from accidental device overheating. During thermal shutdown, the SS and SSSR capacitors are fully
discharged and the controller follows a normal start-up sequence after the junction temperature falls to the
operating level (140°C).
7.4 Device Functional Modes
7.4.1 Control Method Selection
The LM5045 is a versatile PWM control IC that can be configured for either current mode control or voltage
mode control. The choice of the control method usually depends upon the designer preference. The following
must be taken into consideration while selecting the control method. Current mode control can inherently balance
flux in both phases of the full-bridge topology. The full-bridge topology, like other double ended topologies, is
susceptible to the transformer core saturation. Any asymmetry in the volt-second product applied between the
two alternating phases results in flux imbalance that causes a dc buildup in the transformer. This continual dc
buildup may eventually push the transformer into saturation. The volt-second asymmetry can be corrected by
employing current mode control. In current mode control, a signal representative of the primary current is
compared against an error signal to control the duty cycle. In steady-state, this results in each phase being
terminated at the same peak current by adjusting the pulse-width and thus applying equal volt-seconds to both
the phases.
Current mode control can be susceptible to noise and sub-harmonic oscillation, while voltage mode control
employs a larger ramp for PWM and is usually less susceptible. Voltage-mode control with input line feed-
forward also has excellent line transient response. When configuring for voltage mode control, a dc blocking
capacitor can be placed in series with the primary winding of the power transformer to avoid any flux imbalance
that may cause transformer core saturation.
7.4.2 Voltage Mode Control Using the LM5045
To configure the LM5045 for voltage mode control, an external resistor (RFF) and capacitor (CFF) connected to
VIN, AGND, and the RAMP pins is required to create a saw-tooth modulation ramp signal shown in Figure 17.
The slope of the signal at RAMP will vary in proportion to the input line voltage. The varying slope provides line
feed-forward information necessary to improve line transient response with voltage mode control. With a constant
error signal, the on-time (TON) varies inversely with the input voltage (VIN) to stabilize the Volt- Second product
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RFF =-1 VRAMP
VINMIN
(1-
FOSC x CFF x In )
VIN
RFF
CFF
LM5045
SLOPE
PROPORTIONAL
TO VIN
COMP
Gate Drive
1V
CLK
VIN R
R
5V
5k
1:1
RAMP
LM5046
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SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Device Functional Modes (continued)
of the transformer primary. Using a line feed-forward ramp for PWM control requires very little change in the
voltage regulation loop to compensate for changes in input voltage, as compared to a fixed slope oscillator ramp.
Furthermore, voltage mode control is less susceptible to noise and does not require leading edge filtering.
Therefore, it is a good choice for wide input range power converters. Voltage mode control requires a Type-III
compensation network, due to the complex-conjugate poles of the L-C output filter.
Figure 17. Feed-Forward Voltage Mode Configuration
The recommended capacitor value range for CFF is from 100 pF to 1800 pF. Referring to Figure 17, it can be
seen that CFF value must be small enough to be discharged with in the clock pulse-width which is typically within
50ns. The RDS(ON) of the internal discharge FET is 5.5 Ω.
The value of RFF required can be calculated from
(4)
For example, assuming a VRAMP of 1.5 V (a good compromise of signal range and noise immunity), at VINMIN of
36 V (oscillator frequency of 400 kHz and CFF = 470 pF results in a value for RFF of 125 kΩ.
7.4.3 Current Mode Control Using the LM5045
The LM5045 can be configured for current mode control by applying a signal proportional to the primary current
to the RAMP pin. One way to achieve this is shown in Figure 18. The primary current can be sensed using a
current transformer or sense resistor, the resulting signal is filtered and applied to the RAMP pin through a
resistor used for slope compensation. It can be seen that the signal applied to the RAMP pin consists of the
primary current information from the CS pin plus an additional ramp for slope compensation, added by the
resistor RSLOPE.
The current sense resistor is selected such that during over current condition, the voltage across the current
sense resistor is above the minimum CS threshold of 728 mV.
In general, the amount of slope compensation required to avoid sub-harmonic oscillation is equal to at least one-
half the down-slope of the output inductor current, transformed to the primary. To mitigate sub-harmonic
oscillation after one switching period, the slope compensation has to be equal to one times the down slope of the
filter inductor current transposed to primary. This is known as deadbeat control. The slope compensation resistor
required to implement dead-beat control can be calculated as follows:
where
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 23
Product Folder Links: LM5046
CS
RFILTER
RCS
LM5045
RAMP
CLK + LEB
CLK
Current
Sense
CFILTER
SLOPE
100 PA
0
RSLOPE
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Device Functional Modes (continued)
• NTR is the turns-ratio with respect to the secondary (5)
For example, for a 3.3 V output converter with a turns-ratio between primary and secondary of 9:1, an output
filter inductance (LFILTER) of 800 nH and a current sense resistor (RSENSE) of 150 mΩ, RSLOPE of 1.67 kΩwill
suffice.
Figure 18. Current Mode Configuration
24 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
LM5046
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SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM5046 is a highly integrated PWM controller that contains all of the features necessary for implementing
Phase Shifted Full Bridge topology power converters using either current mode or voltage mode control. The
device targets DC to DC converter applications with input voltages of up to 100 Vdc and output power in the
range 100W to 1kW.
8.2 Typical Application
The following schematic shows an example of a 100W phase-shifted full-bridge converter controlled by LM5046.
The operating input voltage range is 36 V to 75 V, and the output voltage is 3.3 V. The output current capability
is 30 Amps. The converter is configured for current mode control with external slope compensation. An auxiliary
winding is used to raise the VCC voltage to reduce the controller power dissipation.
Figure 19. Evaluation Board Schematic
8.2.1 Design Requirements
PARAMETERS VALUE
Input operating range 36 V to 75 V
Output voltage 3.3 V
Measured efficiency at 48 V 92% @ 30A
Frequency of operation 420 kHz
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 25
Product Folder Links: LM5046
LLeak
SW1
LMag
LO2
HO2
IO + Imag
HO1
LO1
SW1
VIN
LMag
LO2
HO2
HO1
LO1
SW2 SR1 SR2
VOUT
VIN
SW2
SW1
VIN
LMag
LO2
HO2
HO1
LO1
SW2 SR1 SR2
VOUT
LLeakage
GND CParasitic
LMag
HO1 HO2
LO2
LO1
VIN
SW1 SW2
Power Transfer/Active Mode Active to Passive
Transition
Freewheel/Passive Mode Passive to Active
Transition
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Typical Application (continued)
PARAMETERS VALUE
Board size 2.28 x 1.45 x 0.5 inches
Load Regulation 0.2%
Line Regulation 0.1%
Line UVLO 34V/32V on/off
Hiccup Mode Current Limit
8.2.2 Detailed Design Procedure
8.2.2.1 Phase-Shifted Full-Bridge Operation
Figure 20. Operating States of the PSFB Topology
The phase shifted full-bridge topology is a derivative of the conventional full-bridge topology. When tuned
appropriately the PSFB topology achieves zero voltage switching (ZVS) of the primary FETs while maintaining
constant switching frequency. The ZVS feature is highly desirable as it reduces both the switching losses and the
EMI emissions. The realization of the PSFB topology using the LM5046 is explained as follows:
8.2.2.1.1 Operating State 1 (Power Transfer/Active Mode)
The power transfer mode of the PSFB topology is similar to the hard switching full-bridge i.e. When the FETs in
the diagonal of the bridge are turned-on (HO1 & LO2 or HO2 & LO1), a power transfer cycle from the primary to
the secondary is initiated. Figure 20 depicts the case where the diagonal switches HO1 and LO2 are activated. In
this state, full VIN is applied to the primary of the power transformer, which is typically stepped down on the
secondary winding.
8.2.2.1.2 Operating State 2 (Active to Passive Transition)
At the end of the power transfer cycle, PWM turns off switch LO2. In the primary side, the reflected load current
plus the magnetizing current propels the SW2 node towards VIN. The active to passive transition is finished
when either the body diode of HO2 is forward-biased or HO2 is turned-on, whichever happens earlier. A delay
can be introduced by setting RD2 to an appropriate value, such that HO2 is turned-on only after the body-diode
is forward biased. In this mode, the Imag+ILpeak act as a current source charging the parasitic capacitor located at
the node SW2. At light load conditions, it takes a longer time to propel SW node towards VIN.
The active to passive transition time can be approximated by using Equation 6:
26 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
Vout
T1
T1
Vin
HO1 HO2
LO2LO1
SR1
SR2
Turn-off
controlled by
CLK
Turn-off
controlled by
PWM
Passive to Active
Transition at SW1 Active to Passive
Transition at SW2
SW1 SW2
TPA =(Lleakage + Lcommutation) x Cparasitic
'
2
TAP = Cparasitic x VIN
ILpeak
NTR )(Im +
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
(6)
Where, Imis the magnetizing current, NTR is the power transformer’s turns ratio, ILpeak is the peak output filter
inductor current and Cparasitic is the parasitic capacitance at the node SW2.
8.2.2.1.3 Operating State 3 (Freewheel/Passive Mode)
In the freewheel mode, unlike the conventional full-bridge topology where all the four primary FETs are off, in the
PSFB topology the primary of the power transformer is shorted by activating either both the top FETs (HO1 and
HO2) or both of the bottom FETs (LO1 and LO2) alternatively. In the current CLK cycle, the top FETs HO1 and
HO2 are kept on together. Further in this mode, on the secondary side, similar to the classic full-bridge topology
the synchronous FETs are both activated. During this state there is no energy transfer from the primary and the
filter inductor current in the secondary freewheels through both the synchronous FETs.
8.2.2.1.4 Operating State 4 (Passive to Active Transition)
At the end of the switching cycle i.e. after the oscillator times out the current CLK cycle, the primary switch HO1
and the secondary FET SR1 are turned-off simultaneously. The voltage at the node SW1 begins to fall towards
the GND. This is due to the resonance between leakage inductance of the power transformer plus any additional
commutation inductor and the parasitic capacitances at SW1. The magnetizing inductor is shorted in the
freewheel mode and therefore it does not play any role in this transition. The LC resonance results in a half-wave
sinusoid whose period is determined by the leakage inductor and parasitic capacitor. The peak of the half-wave
sinusoid is a function of the load current. The passive to active transition time can be approximated by using
Equation 7:
(7)
When tuned appropriately either by deliberately increasing the leakage inductance or by adding an extra
commutating inductor, the sinusoidal resonant waveform peaks such that it is clamped by the body-diode of the
LO1 switch. At this instant, ZVS can be realized by turning on the LO1 switch.
The switching sequence in this CLK cycle is as follows: activation of the switch LO1 turns the diagonal LO1 and
HO2 on, resulting in power transfer. The power transfer cycle ends when PWM turns off HO2, which is followed
by an active to passive transition where LO2 is turned on. In the freewheel mode, LO1 and LO2 are both
activated. From this sequence, it can be inferred that the FETs on the right side of the bridge (HO2 and LO2) are
always terminated by the PWM ending a power transfer cycle and the SW2 node always sees an active to
passive transition. Further, the FETs on the left side of the bridge (HO1 and LO1) are always turned-off by the
CLK ending a freewheel cycle and the SW1 node always sees a passive to active transition.
Figure 21. Simplified PSFB Topology Showing the Turn-Off Mechanism
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 27
Product Folder Links: LM5046
RFF =-1 VRAMP
VINMIN
(1-
FOSC x CFF x In )
VIN
RFF
CFF
LM5046
SLOPE
PROPORTIONAL
TO VIN
COMP
Gate Drive
1V
CLK
VIN R
R
5V
5k
1:1
RAMP
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
8.2.2.2 Control Method Selection
The LM5046 is a versatile PWM control IC that can be configured for either current mode control or voltage
mode control. The choice of the control method usually depends upon the designer preference. The following
must be taken into consideration while selecting the control method. Current mode control can inherently balance
flux in both phases of the PSFB topology. The PSFB topology, like other double ended topologies, is susceptible
to the transformer core saturation. Any asymmetry in the volt-second product applied between the two alternating
phases results in flux imbalance that causes a dc buildup in the transformer. This continual dc buildup may
eventually push the transformer into saturation. The volt-second asymmetry can be corrected by employing
current mode control. In current mode control, a signal representative of the primary current is compared against
an error signal to control the duty cycle. In steady-state, this results in each phase being terminated at the same
peak current by adjusting the pulse-width and thus applying equal volt-seconds to both the phases.
Current mode control can be susceptible to noise and sub-harmonic oscillation, while voltage mode control
employs a larger ramp for PWM and is usually less susceptible. Voltage-mode control with input line feed-
forward also has excellent line transient response. When configuring for voltage mode control, a dc blocking
capacitor can be placed in series with the primary winding of the power transformer to avoid any flux imbalance
that may cause transformer core saturation.
8.2.2.3 Voltage Mode Control Using the LM5046
To configure the LM5046 for voltage mode control, an external resistor (RFF) and capacitor (CFF) connected to
VIN, AGND, and the RAMP pins is required to create a saw-tooth modulation ramp signal shown in Figure 22.
The slope of the signal at RAMP will vary in proportion to the input line voltage. The varying slope provides line
feed-forward information necessary to improve line transient response with voltage mode control. With a constant
error signal, the on-time (TON) varies inversely with the input voltage (VIN) to stabilize the Volt- Second product
of the transformer primary. Using a line feed-forward ramp for PWM control requires very little change in the
voltage regulation loop to compensate for changes in input voltage, as compared to a fixed slope oscillator ramp.
Furthermore, voltage mode control is less susceptible to noise and does not require leading edge filtering.
Therefore, it is a good choice for wide input range power converters. Voltage mode control requires a Type-III
compensation network, due to the complex-conjugate poles of the L-C output filter.
Figure 22. Feed-Forward Voltage Mode Configuration
The recommended capacitor value range for CFF is from 100 pF to 1800 pF. Referring to Figure 22, it can be
seen that CFF value must be small enough to be discharged with in the clock pulse-width which is typically within
50 ns. The RDS(ON) of the internal discharge FET is 5.5 Ω.
The value of RFF required can be calculated from Equation 8.
(8)
For example, assuming a VRAMP of 1.5 V (a good compromise of signal range and noise immunity), at VINMIN of
36 V (oscillator frequency of 400 kHz and CFF = 470 pF results in a value for RFF of 125 kΩ.
28 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
CS
RFILTER
RCS
LM5046
RAMP
CLK + LEB
CLK
Current
Sense
CFILTER
SLOPE
100 PA
0
RSLOPE
OUT CS
SLOPE
FILTER OSC SLOPE TR
V R
R
L F I N
´
=
´ ´ ´
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
8.2.2.4 Current Mode Control Using the LM5046
The LM5046 can be configured for current mode control by applying a signal proportional to the primary current
to the RAMP pin. One way to achieve this is shown in Figure 23. The primary current can be sensed using a
current transformer or sense resistor, the resulting signal is filtered and applied to the RAMP pin through a
resistor used for slope compensation. It can be seen that the signal applied to the RAMP pin consists of the
primary current information from the CS pin plus an additional ramp for slope compensation, added by the
resistor RSLOPE.
The current sense resistor is selected such that during over current condition, the voltage across the current
sense resistor is above the minimum CS threshold of 728 mV.
In general, the amount of slope compensation required to avoid sub-harmonic oscillation is equal to at least one-
half the down-slope of the output inductor current, transformed to the primary. To mitigate sub-harmonic
oscillation after one switching period, the slope compensation has to be equal to one times the down slope of the
filter inductor current transposed to primary. This is known as deadbeat control. The slope compensation resistor
required to implement dead-beat control can be calculated using Equation 9:
(9)
Where NTR is the turns-ratio with respect to the secondary. For example, for a 3.3 V output converter with a
turns-ratio between primary and secondary of 9:1, an output filter inductance (LFILTER) of 800 nH and a current
sense resistor (RSENSE) of 150 mΩ, RSLOPE of 1.67 kΩwill suffice.
Figure 23. Current Mode Configuration
8.2.2.5 VIN and VCC
The voltage applied to the VIN pin, which may be the same as the system voltage applied to the power
transformer’s primary (VPWR), can vary in the range of the 14 to 100 V. It is recommended that the filter shown in
Figure 24 be used to suppress the transients that may occur at the input supply. This is particularly important
when VIN is operated close to the maximum operating rating of the LM5046. The current into VIN depends
primarily on the LM5046’s operating current, the switching frequency, and any external loads on the VCC pin,
that typically include the gate capacitances of the power MOSFETs. In typical applications, an auxiliary
transformer winding is connected through a diode to the VCC pin. This pin must raise VCC voltage above 8 V to
shut off the internal start-up regulator.
After the outputs are enabled and the external VCC supply voltage has begun supplying power to the IC, the
current into the VIN pin drops below 1 mA. VIN should remain at a voltage equal to or above the VCC voltage to
avoid reverse current through the internal body diode of the internal VCC regulator.
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 29
Product Folder Links: LM5046
R2 = 1.25V x R1
VPWR-OFF -1.25V - (20 PA x R1)
R1 = VHYS
20 PA
VPWR VIN VCC
10V - 16V
(from aux winding)
LM5046
11V
0.1 PF
50
LM5046
VIN
VPWR
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Figure 24. Input Transient Protection
8.2.2.6 For Applications With > 100 V Input
For applications where the system input voltage exceeds 100 V, VIN can be powered from an external start-up
regulator as shown in Figure 25. In this configuration, the VIN and VCC pins should be connected together. The
voltage at the VCC and VIN pins must be greater than 10 V (> Max VCC reference voltage) yet not exceed 16 V.
To enable operation the VCC voltage must be raised above 10 V. The voltage at the VCC pin must not exceed
16 V. The voltage source at the right side of Figure 25 is typically derived from the power stage, and becomes
active once the LM5046’s outputs are active.
Figure 25. Start-Up Regulator for VPWR > 100 V
8.2.2.7 UVLO and OVP Voltage Divider Selection
Two dedicated comparators connected to the UVLO and OVP pins are used to detect under voltage and over
voltage conditions. The threshold values of both these comparators are set at 1.25 V. The two functions can be
programmed independently with two separate voltage dividers from VIN to AGND as shown in Figure 26 and
Figure 27, or with a three-resistor divider as shown in Figure 28. Independent UVLO and OVP pins provide
greater flexibility for the user to select the operational voltage range of the system. When the UVLO pin voltage is
below 0.4 V, the controller is in a low current shutdown mode. For a UVLO pin voltage greater than 0.4 V but
less than 1.25 V the controller is in standby mode. Once the UVLO pin voltage is greater than 1.25 V, the
controller is fully enabled. Two external resistors can be used to program the minimum operational voltage for the
power converter as shown in Figure 26. When the UVLO pin voltage falls below the 1.25 V threshold, an internal
20 µA current sink is enabled to lower the voltage at the UVLO pin, thus providing threshold hysteresis.
Resistance values for R1and R2can be determined from Equation 10:
(10)
Where VPWR is the desired turn-on voltage and VHYS is the desired UVLO hysteresis at VPWR.
For example, if the LM5046 is to be enabled when VPWR reaches 33 V, and disabled when VPWR is decreased
to 31 V, R1should be 100 kΩ, and R2should be 4.2 kΩ. The voltage at the UVLO pin should not exceed 7 V at
any time.
30 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
OVP
LM5046
1.25V
20 PA
STANDBY
VPWR
5V
R1
R2
UVLO
LM5046
1.25V
20 PA
SHUTDOWN
VPWR
R1
R2
STANDBY
0.4V
R2 = 1.25V x R1
VPWR -1.25V + (20 PA x R1)
VHYS
20 PA
R1 =
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Two external resistors can be used to program the maximum operational voltage for the power converter as
shown in Figure 27. When the OVP pin voltage rises above the 1.25 V threshold, an internal 20 µA current
source is enabled to raise the voltage at the OVP pin, thus providing threshold hysteresis. Resistance values for
R1and R2can be determined from Equation 11:
(11)
If the LM5046 is to be disabled when VPWR-OFF reaches 80 V and enabled when it is decreased to 78 V. R1
should be 100 kΩ, and R2should be 1.5 kΩ. The voltage at the OVP pin should not exceed 7 V at any time.
Figure 26. Basic UVLO Configuration
Figure 27. Basic OVP Configuration
The UVLO and OVP can also be set together using a 3 resistor divider ladder as shown in Figure 28. R1is
calculated as explained in the basic UVLO divider selection. Using the same values, as in the above two
examples, for the UVLO and OVP set points, R1and R3remain the same at 100 kΩand 1.5 kΩ. The R2is 2.7
kΩobtained by subtracting R3from 4.2 kΩ.
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 31
Product Folder Links: LM5046
UVLO
LM5046
1.25V
20 PA
SHUTDOWN
VPWR
R1
STANDBY
0.4V
STANDBY SHUTDOWN R2
1.25V
20 PA
0.4V SHUTDOWN
OVP
1.25V STANDBY
VPWR
5V
R1
R2
R3
LM5046
STANDBY
UVLO
20 PA
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Figure 28. UVLO/OVP Divider
Remote configuration of the controller’s operational modes can be accomplished with open drain device(s)
connected to the UVLO pin as shown in Figure 29.
Figure 30 shows an application of the OVP comparator for Remote Thermal Protection using a thermistor (or
multiple thermistors) which may be located near the main heat sources of the power converter. The negative
temperature coefficient (NTC) thermistor is nearly logarithmic, and in this example a 100 kΩthermistor with the β
material constant of 4500 Kelvin changes to approximately 2 kΩat 130ºC. Setting R1to one-third of this
resistance (665 Ω) establishes 130ºC as the desired trip point (for VREF = 5 V). In a temperature band from
20ºC below to 20ºC above the OVP threshold, the voltage divider is nearly linear with 25mV per ºC sensitivity.
R2provides temperature hysteresis by raising the OVP comparator input by R2x 20 µA. For example, if a 22 kΩ
resistor is selected for R2, then the OVP pin voltage will increase by 22k x 20 µA = 506 mV. The NTC
temperature must therefore fall by 506 mV / 25 mV per ºC = 20ºC before the LM5046 switches from standby
mode to the normal mode.
Figure 29. Remote Standby and Disable Control
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Product Folder Links: LM5046
Q1
Q2
Q3
Q4
NP
CS
LM5046 RF
CFRCS
VIN
AGND
Q1
Q2
Q3
Q4
VPWR
NP
NS1
NS2
RF
CFRCS
CS
LM5046
VIN
AGND
NTC
THERMISTOR
OVP
LM5046
1.25V
20 PA
STANDBY
VPWR 5V
R1
R2
T
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Figure 30. Remote Thermal Protection
8.2.2.8 Current Sense
The CS pin receives an input signal representative of its transformer’s primary current, either from a current
sense transformer or from a resistor located at the junction of source pin of the primary switches, as shown in
Figure 31 and Figure 32, respectively. In both the cases, the filter components RFand CFshould be located as
close to the IC as possible, and the ground connection from the current sense transformer, or RSENSE should be
a dedicated trace to the appropriate GND pin. Please refer to the layout section for more layout tips.
The current sense components must provide a signal > 710 mV at the CS pin during an over-load event. Once
the voltage on the CS pin crosses the current limit threshold, the current sense comparator terminates the PWM
pulse and starts to charge the RES pin. Depending on the configuration of the RES pin, the LM5046 will
eventually initiate a hiccup mode restart or be in continuous current limit.
Figure 31. Transformer Current Sense
Figure 32. Resistor Current Sense
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 33
Product Folder Links: LM5046
Hiccup Mode off-time
Soft-Start
Restart delay
1V
2V
4V
Count to Eight
1V
THICCUP = 5 µA
CRES x (2.0Vx8) 10 µA
CRES x ((2.0Vx8) + 1.0V)
+
TCS = CRES x 1.0V
30 PA
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
8.2.2.9 Hiccup Mode Current Limit Restart
The operation of the hiccup mode restart circuit is explained in the Overview section. During a continuous current
limit condition, the RES pin is charged with 30 µA current source. The restart delay time required to reach the 1.0
V threshold is given by Equation 12:
(12)
This establishes the number of current limit events allowed before the IC initiates a hiccup restart sequence. For
example, if the CRES = 0.01 µF, the time TCS as noted in Figure 33 is 334 µs. Once the RES pin reaches 1.0 V,
the 30 µA current source is turned-off and a 10 µA current source is turned-on during the ramp up to 4 V and a 5
µA is turned on during the ramp down to 2 V. The hiccup mode off-time is given by Equation 13:
(13)
With a CRES= 0.01 µF, the hiccup time is 49 ms. Once the hiccup time is finished, the RES pin is pulled-low and
the SS pin is released allowing a soft-start sequence to commence. Once the SS pin reaches 1 V, the PWM
pulses will commence. The hiccup mode provides a cool-down period for the power converter in the event of a
sustained overload condition thereby lowering the average input current and temperature of the power
components during such an event.
Figure 33. Hiccup Mode Delay and Soft-Start Timing Diagram
8.2.2.10 Augmenting the Gate Drive Strength
The LM5046 includes powerful 2A integrated gate drivers. However, in certain high power applications (> 500W),
it might be necessary to augment the strength of the internal gate driver to achieve higher efficiency and better
thermal performance. In high power applications, typically, the I2xR loss in the primary MOSFETs is significantly
higher than the switching loss. In order to minimize the I2xR loss, either the primary MOSFETs are paralleled or
MOSFETs with low RDS (on) are employed. Both these scenarios increase the total gate charge to be driven by
the controller IC. An increase in the gate charge increases the FET transition time and hence increases the
switching losses. Therefore, to keep the total losses within a manageable limit the transition time needs to be
reduced.
Generally, during the miller capacitance charge/discharge the total available driver current is lower during the
turn-off process than during the turn-on process and often it is enough to speed-up the turn-off time to achieve
the efficiency and thermal goals. This can be achieved simply by employing a PNP device, as shown in
Figure 34, from gate to source of the power FET. During the turn-on process, when the LO1 goes high, the
current is sourced through the diode D1 and the BJT Q1 provides the path for the turn-off current. Q1 should be
located as close to the power FET as possible so that the turn-off current has the shortest possible path to the
ground and does not have to pass through the controller.
34 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
LO1
PGND
VCC
BST1
HO1
HS1
VIN
LM5046
Q1
Q2
LO1
PGND
VCC
BST1
HO1
HS1
VIN
LM5046
Q1
D1
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
Figure 34. Circuit to Speed-up the Turn-off Process
Depending on the gate charge characteristics of the primary FET, if it is required to speed up both the turn-on
and the turn-off time, a bipolar totem pole structure as shown in Figure 35 can be used. When LO1 goes high,
the gate to source current is sourced through the NPN transistor Q1 and similar to the circuit shown in Figure 34
when LO1 goes low the PNP transistor Q2 expedites the turn-off process.
Figure 35. Bipolar Totem Pole Arrangement
Alternatively, a low side gate driver such as LM5112 can be utilized instead of the discrete totem pole. The
LM5112 comes in a small package with a 3A source and a 7A sink capability. While driving the high-side FET,
the HS1 acts as a local ground and the boot capacitor between the BST and HS pins acts as VCC.
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 35
Product Folder Links: LM5046
LO1
PGND
LM5112
LM5112
VCC
BST1
HO1
HS1
VIN
LM5046
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Figure 36. Using a Low Side Gate Driver to Augment Gate Drive Strength
8.2.3 Application Curve
Figure 37. Application Board Efficiency
36 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
RD1
R_FILTER
RAMP
RD2
RT
RES SS SSSR
To Current Sensing
Network
To Voltage Feedback
Network
C_FILTER
HTSSOP28
VIN
PGND
LO1
VCC
LO2
SR2
SR1
HS2
HO2
BST2
HS1
BST1
HO1
COMP
RD2
UVLO
OVP
CS
RAMP
SSSR
AGND
SLOPE
RT
REF
RD1
SS
SS OFF
RES
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
9 Power Supply Recommendations
The LM5046 can be used to control power levels up to 1kW. Therefore the current levels can be considerable.
Care should be taken that components with the correct current rating are chosen. This would include magnetic
components, power MOSFETS and diodes, connectors and wire sizes. Input and output capacitors should have
the correct ripple current rating. The use of a multilayer PCB is recommended with a copper area chosen to
ensure the LM5046 is operating below its maximum junction temperature. Full power loading should never be
attempted without providing adequate cooling.
10 Layout
10.1 Layout Guidelines
The LM5046 current sense and PWM comparators are very fast and respond to short duration noise pulses. The
components at the CS, COMP, SLOPE, RAMP, SS, SSSR, RES, UVLO, OVP, RD1, RD2, and RT pins should
be physically close as possible to the IC, thereby minimizing noise pickup on the PC board trace inductance.
Eliminating or minimizing via’s in these critical connections are essential. Layout consideration is critical for the
current sense filter. If a current sense transformer is used, both leads of the transformer secondary should be
routed to the sense filter components and to the IC pins. The ground side of the transformer should be
connected via a dedicated PC board trace to the AGND pin, rather than through the ground plane. If the current
sense circuit employs a sense resistor in the drive transistor source, low inductance resistors should be used. In
this case, all the noise sensitive, low-current ground trace should be connected in common near the IC, and then
a single connection made to the power ground (sense resistor ground point).
The gate drive outputs of the LM5046 should have short, direct paths to the power MOSFETs in order to
minimize inductance in the PC board. The boot-strap capacitors required for the high side gate drivers should be
located very close to the IC and connected directly to the BST and HS pins. The VCC and REF capacitors
should also be placed close to their respective pins with short trace inductance. Low ESR and ESL ceramic
capacitors are recommended for the boot-strap, VCC and the REF capacitors. The two ground pins (AGND,
PGND) must be connected together directly underneath the IC with a short, direct connection, to avoid jitter due
to relative ground bounce.
10.2 Layout Example
Figure 38. Layout of Components Around RAMP, CS, SLOPE, COMP, RT, RD1, RD2, RES, SS, and SSR
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 37
Product Folder Links: LM5046
R1
R2
C1
INPUT DC VOLTAGE
R3
C2
C3
C4
C4
VIN
PGND
LO1
VCC
LO2
SR2
SR1
HS2
HO2
BST2
HS1
BST1
HO1
COMP
RD2
UVLO
OVP
CS
RAMP
SSSR
AGND
SLOPE
RT
REF
RD1
SS
SS OFF
RES
LM5046
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
www.ti.com
Layout Example (continued)
Figure 39. Layout of Components Around VIN, VCC, AGND, PGND UVLO, OVP, REF, BST1, BST2, HS1,
and HS2
38 Submit Documentation Feedback Copyright © 2011–2014, Texas Instruments Incorporated
Product Folder Links: LM5046
LM5046
www.ti.com
SNVS703H –FEBRUARY 2011–REVISED NOVEMBER 2014
11 Device and Documentation Support
11.1 Trademarks
All trademarks are the property of their respective owners.
11.2 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.3 Glossary
SLYZ022 —TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
Copyright © 2011–2014, Texas Instruments Incorporated Submit Documentation Feedback 39
Product Folder Links: LM5046
PACKAGE OPTION ADDENDUM
www.ti.com 10-Dec-2020
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead finish/
Ball material
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM5046MH/NOPB ACTIVE HTSSOP PWP 28 48 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM5046
MH
LM5046MHX/NOPB ACTIVE HTSSOP PWP 28 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM5046
MH
LM5046SQ/NOPB ACTIVE WQFN RSG 28 1000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 L5046
LM5046SQX/NOPB ACTIVE WQFN RSG 28 4500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 L5046
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two
lines if the finish value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
PACKAGE OPTION ADDENDUM
www.ti.com 10-Dec-2020
Addendum-Page 2
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM5046MHX/NOPB HTSSOP PWP 28 2500 330.0 16.4 6.8 10.2 1.6 8.0 16.0 Q1
LM5046SQ/NOPB WQFN RSG 28 1000 178.0 12.4 5.3 5.3 1.3 8.0 12.0 Q1
LM5046SQX/NOPB WQFN RSG 28 4500 330.0 12.4 5.3 5.3 1.3 8.0 12.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM5046MHX/NOPB HTSSOP PWP 28 2500 367.0 367.0 35.0
LM5046SQ/NOPB WQFN RSG 28 1000 210.0 185.0 35.0
LM5046SQX/NOPB WQFN RSG 28 4500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 11-Oct-2013
Pack Materials-Page 2
www.ti.com
PACKAGE OUTLINE
C
SEE TERMINAL
DETAIL
28X 0.3
0.2
3.6 0.1
28X 0.5
0.3
0.8 MAX
(A) TYP
0.05
0.00
24X 0.5
2X
3
2X 3
3.6 0.1
A5.1
4.9 B
5.1
4.9
0.3
0.2
0.5
0.3
WQFN - 0.8 mm max heightRSG0028A
PLASTIC QUAD FLATPACK - NO LEAD
4214982/A 08/2016
DIM A
OPT 1 OPT 1
(0.1) (0.2)
PIN 1 INDEX AREA
0.08
SEATING PLANE
1
715
21
814
28 22
(OPTIONAL)
PIN 1 ID 0.1 C A B
0.05
EXPOSED
THERMAL PAD
29 SYMM
SYMM
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. The package thermal pad must be soldered to the printed circuit board for thermal and mechanical performance.
SCALE 2.600
DETAIL
OPTIONAL TERMINAL
TYPICAL
www.ti.com
EXAMPLE BOARD LAYOUT
0.07 MIN
ALL AROUND
0.07 MAX
ALL AROUND
28X (0.25)
28X (0.6)
( 0.2) TYP
VIA
24X (0.5)
(4.8)
(4.8)
4X
(1.55)
(3.6)
(R0.05)
TYP
4X (1.55)
(1.23)
TYP
(1.23) TYP
WQFN - 0.8 mm max heightRSG0028A
PLASTIC QUAD FLATPACK - NO LEAD
4214982/A 08/2016
SYMM
1
7
814
15
21
22
28
SYMM
LAND PATTERN EXAMPLE
SCALE:15X
29
NOTES: (continued)
4. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
number SLUA271 (www.ti.com/lit/slua271).
5. Vias are optional depending on application, refer to device data sheet. If any vias are implemented, refer to their locations shown
on this view. It is recommended that vias under paste be filled, plugged or tented.
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
SOLDER MASK
DEFINED
METAL
SOLDER MASK
OPENING
SOLDER MASK DETAILS
NON SOLDER MASK
DEFINED
(PREFERRED)
www.ti.com
EXAMPLE STENCIL DESIGN
28X (0.6)
28X (0.25)
24X (0.5)
(4.8)
(4.8)
9X (1.03)
(1.23)
TYP
(1.23) TYP
(R0.05) TYP
WQFN - 0.8 mm max heightRSG0028A
PLASTIC QUAD FLATPACK - NO LEAD
4214982/A 08/2016
NOTES: (continued)
6. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
SYMM
METAL
TYP
SOLDER PASTE EXAMPLE
BASED ON 0.125 mm THICK STENCIL
EXPOSED PAD 29
73.7% PRINTED SOLDER COVERAGE BY AREA UNDER PACKAGE
SCALE:20X
SYMM
1
7
814
15
21
22
28
29
www.ti.com
PACKAGE OUTLINE
C
TYP
6.6
6.2
1.1 MAX
26X 0.65
28X 0.30
0.19
2X
8.45
TYP
0.20
0.09
0 - 8
0.10
0.02
5.65
5.25
3.15
2.75
(1)
0.25
GAGE PLANE
0.7
0.5
A
NOTE 3
9.8
9.6
B
NOTE 4
4.5
4.3
4214870/A 10/2014
PowerPAD - 1.1 mm max heightPWP0028A
PLASTIC SMALL OUTLINE
NOTES:
1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing
per ASME Y14.5M.
2. This drawing is subject to change without notice.
3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not
exceed 0.15 mm, per side.
4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm, per side.
5. Reference JEDEC registration MO-153, variation AET.
PowerPAD is a trademark of Texas Instruments.
TM
128
0.1 C A B
15
14
PIN 1 ID
AREA
SEATING PLANE
0.1 C
SEE DETAIL A
DETAIL A
TYPICAL
SCALE 1.800
THERMAL
PAD
www.ti.com
EXAMPLE BOARD LAYOUT
0.05 MAX
ALL AROUND
0.05 MIN
ALL AROUND
28X (1.3)
(6.1)
28X (0.45)
(0.9) TYP
28X (1.5)
28X (0.45)
26X
(0.65)
(3)
(3.4)
NOTE 9
(5.5)
SOLDER
MASK
OPENING
(9.7)
(1.3)
(1.3) TYP
(5.8)
( ) TYP
VIA
0.2
(0.65) TYP
4214870/A 10/2014
PowerPAD - 1.1 mm max heightPWP0028A
PLASTIC SMALL OUTLINE
SOLDER MASK
DEFINED PAD
LAND PATTERN EXAMPLE
SCALE:6X
HV / ISOLATION OPTION
0.9 CLEARANCE CREEPAGE
OTHER DIMENSIONS IDENTICAL TO IPC-7351
TM
NOTES: (continued)
6. Publication IPC-7351 may have alternate designs.
7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.
8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature
numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).
9. Size of metal pad may vary due to creepage requirement.
METAL
SOLDER MASK
OPENING
NON SOLDER MASK
DEFINED
SOLDER MASK DETAILS
SOLDER MASK
OPENING
METAL UNDER
SOLDER MASK
SOLDER MASK
DEFINED
SYMM
SYMM
SEE DETAILS
1
14 15
28
METAL COVERED
BY SOLDER MASK
SOLDER
MASK
OPENING
IPC-7351 NOMINAL
0.65 CLEARANCE CREEPAGE
www.ti.com
EXAMPLE STENCIL DESIGN
28X (1.3)
28X (0.45)
(6.1)
28X (1.5)
28X (0.45)
26X (0.65)
(3)
(5.5)
BASED ON
0.127 THICK
STENCIL
(5.8)
4214870/A 10/2014
PowerPAD - 1.1 mm max heightPWP0028A
PLASTIC SMALL OUTLINE
2.66 X 4.770.178
2.88 X 5.160.152
3.0 X 5.5 (SHOWN)0.127
3.55 X 6.370.1
SOLDER STENCIL
OPENING
STENCIL
THICKNESS
SEE TABLE FOR
DIFFERENT OPENINGS
FOR OTHER STENCIL
THICKNESSES
NOTES: (continued)
10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate
design recommendations.
11. Board assembly site may have different recommendations for stencil design.
TM
SOLDER PASTE EXAMPLE
EXPOSED PAD
100% PRINTED SOLDER COVERAGE AREA
SCALE:6X
HV / ISOLATION OPTION
0.9 CLEARANCE CREEPAGE
OTHER DIMENSIONS IDENTICAL TO IPC-7351
SYMM
SYMM
1
14 15
28
BASED ON
0.127 THICK
STENCIL BY SOLDER MASK
METAL COVERED
IPC-7351 NOMINAL
0.65 CLEARANCE CREEPAGE
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