LM26003
SW
SW
BOOT
FB
COMP
VDD
PGND
FPWM
FREQ
SS
SYNC
EN
PGOOD
AVIN
4
5
16
12
17
6
13
8
14
7
11
9
18
19
VIN
VOUT
VIN
SW
3
20
C9
C3
R5
C5
R3
C6
R2
R1 C7
C1
R4
VDD
21EP
R6
EN
SYNC
+
VIN
VIN
2
1
AGND 10
VBIAS 15
PGOOD D1
L1
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LM26003-xx 3-A Switching Regulator With High Efficiency Sleep Mode
1 Features 3 Description
The LM26003 is a switching regulator designed for
1• LM26003-Q1 is an Automotive-Grade Product the high-efficiency requirements of applications with
That is AEC-Q100 Grade 1 Qualified (–40°C to standby modes. The device features a low-current
+125°C Operating Junction Temperature) sleep mode to maintain efficiency under light-load
• High-Efficiency Sleep Mode conditions and current-mode control for accurate
regulation over a wide input voltage range. Quiescent
• 40-µA Typical Iq in Sleep Mode current is reduced to 10.8 µA typically in shutdown
• 10.8-µA Typical Iq in Shutdown Mode mode and less than 40 µA in sleep mode. Forced
• 3.0-V Minimum Input Voltage PWM mode is also available to disable sleep mode.
• 4.0-V to 38-V Continuous Input Range The LM26003 device can deliver up to 3 A of
• 1.5% Reference Accuracy continuous load current with a fixed current limit,
• Cycle-by-Cycle Current Limit through the internal N-channel switch. The part has a
wide input voltage range of 4.0 V to 38 V and can
• Adjustable Frequency (150 kHz to 500 kHz) operate with input voltages as low as 3 V during line
• Synchronizable to an External Clock transients.
• Power Good Flag Operating frequency is adjustable from 150 kHz to
• Forced PWM Function 500 kHz with a single resistor and can be
• Adjustable Soft-Start synchronized to an external clock.
• 20-Pin HTSSOP Package Other features include Power Good, adjustable soft-
• Thermal Shut Down start, enable pin, input undervoltage protection, and
an internal bootstrap diode for reduced component
count.
2 Applications
• Automotive Telematics Device Information(1)
• Navigation Systems PART NUMBER PACKAGE BODY SIZE (NOM)
• In-Dash Instrumentation LM26003 HTSSOP (20) 6.50 mm x 4.40 mm
• Battery-Powered Applications LM26003-Q1
• Standby Power for Home Gateways and Set-Top (1) For all available packages, see the orderable addendum at
Boxes the end of the datasheet.
4 Typical Application Circuit
1
An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM26003
,
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SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
www.ti.com
Table of Contents
8.3 Feature Description................................................. 11
1 Features.................................................................. 18.4 Device Functional Modes........................................ 14
2 Applications ........................................................... 19 Application and Implementation ........................ 17
3 Description............................................................. 19.1 Application Information............................................ 17
4 Typical Application Circuit ................................... 19.2 Typical Application.................................................. 17
5 Revision History..................................................... 210 Power Supply Recommendations ..................... 24
6 Pin Configuration and Functions......................... 311 Layout................................................................... 24
7 Specifications......................................................... 411.1 Layout Guidelines ................................................. 24
7.1 Absolute Maximum Ratings ...................................... 411.2 Layout Example .................................................... 25
7.2 ESD Ratings: LM26003 ............................................ 411.3 Thermal Considerations and TSD......................... 25
7.3 ESD Ratings: LM26003-Q1 ...................................... 412 Device and Documentation Support................. 26
7.4 Recommended Operating Conditions....................... 412.1 Device Support .................................................... 26
7.5 Thermal Information.................................................. 512.2 Documentation Support ........................................ 26
7.6 Electrical Characteristics........................................... 512.3 Related Links ........................................................ 26
7.7 Switching Characteristics.......................................... 712.4 Trademarks........................................................... 26
7.8 Typical Characteristics.............................................. 812.5 Electrostatic Discharge Caution............................ 26
8 Detailed Description............................................ 10 12.6 Glossary................................................................ 26
8.1 Overview................................................................. 10 13 Mechanical, Packaging, and Orderable
8.2 Functional Block Diagram....................................... 10 Information........................................................... 26
5 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
Changes from Revision E (December 2014) to Revision F Page
• Changed from 0.575V ........................................................................................................................................................... 6
• Changed 1.365V .................................................................................................................................................................... 6
Changes from Revision D (March 2013) to Revision E Page
• Added Pin Configuration and Functions section, ESD Ratings table, Feature Description section, Device Functional
Modes,Application and Implementation section, Power Supply Recommendations section, Layout section, Device
and Documentation Support section, and Mechanical, Packaging, and Orderable Information section .............................. 1
Changes from Revision C (March 2013) to Revision D Page
• Changed layout of National Data Sheet to TI format ........................................................................................................... 25
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Exposed Pad
Connect to GND
COMP 8
VIN 1
VIN 2
VIN 3
AVIN 4
PGOOD 5
EN 6
SS 7 14 SYNC
15 VBIAS
16 VDD
17 BOOT
18 SW
19 SW
20 SW
AGND 10
FB 9
11 PGND
13 FPWM
12 FREQ
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SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
6 Pin Configuration and Functions
20-Pin
TSSOP Package
Top View
Pin Functions
PIN I/O DESCRIPTION
NAME NO.
VIN 1 I Power supply input for high side FET
VIN 2 I Power supply input for high side FET
VIN 3 I Power supply input for high side FET
AVIN 4 I Power supply input for IC supply
PGOOD 5 O Power Good pin. An open-drain output which goes high when the output voltage is greater than 92% of
nominal.
EN 6 I Enable is an analog level input pin. When pulled below 0.8 V, the device enters shutdown mode.
SS 7 I Soft-start pin. Connect a capacitor from this pin to GND to set the soft-start time.
COMP 8 I Compensation pin. Connect to a resistor capacitor pair to compensate the control loop.
FB 9 I Feedback pin. Connect to a resistor divider between VOUT and GND to set output voltage.
AGND 10 GND Analog GND as IC reference
PGND 11 GND Power GND is GND for the switching stage of the regulator
FREQ 12 O Frequency adjust pin. Connect a resistor from this pin to GND to set the operating frequency.
FPWM 13 I FPWM is a logic level input pin. For normal operation, connect to GND. When pulled high, sleep mode
operation is disabled.
SYNC 14 I Frequency synchronization pin. Connect to an external clock signal for synchronized operation. SYNC must be
pulled low for non-synchronized operation.
VBIAS 15 I Connect to an external 3-V or greater supply to bypass the internal regulator for improved efficiency. If not
used, VBIAS should be tied to GND.
VDD 16 O The output of the internal regulator. Bypass with a minimum 1.0-µF capacitor.
BOOT 17 I Bootstrap capacitor pin. Connect a 0.1-µF minimum ceramic capacitor from this pin to SW to generate the gate
drive bootstrap voltage.
SW 18 O Switch pin. The source of the internal N-channel switch.
SW 19 O Switch pin. The source of the internal N-channel switch.
SW 20 O Switch pin. The source of the internal N-channel switch.
EP EP GND Exposed Pad thermal connection. Connect to GND.
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7 Specifications
7.1 Absolute Maximum Ratings
over operating free-air temperature range (unless otherwise noted) (1)(2)
MIN MAX UNIT
VIN –0.3 40 V
SW –1 40 V
VDD –0.3 7 V
VBIAS –0.3 10 V
FB -0.3 7 V
Voltages from the
indicated BOOT VSW-0.3 VSW+7 V
pins to GND PGOOD –0.3 7 V
FREQ –0.3 7 V
SYNC –0.3 7 V
EN –0.3 40 V
FPWM –0.3 7 V
Power Dissipation 3.1 W
Recommended Vapor Phase (70s) 215 °C
Lead Temperature Infrared (15s) 220 °C
Storage temperature, Tstg –65 150 °C
(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/Distributors for availability and
specifications.
7.2 ESD Ratings: LM26003 VALUE UNIT
Human body model (HBM), per ANSI/ESDA/JEDEC JS-001, all pins(1) ±2000
Charged device model (CDM), per JEDEC specification JESD22-C101, all ±1000
V(ESD) Electrostatic discharge V
pins(2)
Charged machine model ±200000
(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.
(2) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.
7.3 ESD Ratings: LM26003-Q1 VALUE UNIT
Human body model (HBM), per AEC Q100-002(1) ±2000
Corner pins (1, 10, 11, and 20) ±1000
Charged device model (CDM), per
V(ESD) Electrostatic discharge V
AEC Q100-011 Other pins ±1000
Charged machine model ±200
(1) AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
7.4 Recommended Operating Conditions MIN NOM MAX UNIT
Operating Junction Temperature −40 125 °C
Supply Voltage 3.0 38 V
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7.5 Thermal Information LM26003,
LM26003-Q1
THERMAL METRIC(1) UNIT
PWP
20 PINS
RθJA Junction-to-ambient thermal resistance 25.4
RθJC(top) Junction-to-case (top) thermal resistance 19.6
RθJB Junction-to-board thermal resistance 16.5 °C/W
ψJT Junction-to-top characterization parameter 0.5
ψJB Junction-to-board characterization parameter 16.3
RθJC(bot) Junction-to-case (bottom) thermal resistance 0.8
(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.
7.6 Electrical Characteristics
Unless otherwise stated, Vin = 12 V, TJ= 25°C. Minimum and Maximum limits are ensured through test, design, or statistical
correlation. PARAMETER TEST CONDITIONS MIN TYP(1) MAX UNIT
SYSTEM
ISD(2) Shutdown Current EN = 0 V 10.8 µA
EN = 0 V, –40°C ≤TJ≤125°C 20
IqSleep_VB(2) Quiescent Current Sleep mode, VBIAS = 5 V 40 µA
Sleep mode, VBIAS = 5 V, 70
–40°C ≤TJ≤125°C
IqSleep_VDD Quiescent Current Sleep mode, VBIAS = GND 76 µA
Sleep mode, VBIAS = GND, 125
–40°C ≤TJ≤125°C
IqPWM_VB Quiescent Current PWM mode, VBIAS = 5 V 0.16 0.23 mA
FPWM = 2 V
IqPWM_VDD Quiescent Current PWM mode, VBIAS = GND 0.65 0.85 mA
FPWM = 2 V
IBIAS_Sleep(2) Bias Current Sleep mode, VBIAS = 5 V 33 µA
Sleep mode, VBIAS = 5 V, 60
–40°C ≤TJ≤125°C
IBIAS_PWM Bias Current PWM mode, VBIAS = 5 V 0.5 0.7 mA
VFB Feedback Voltage 5 V < Vin < 38 V 1.236 V
5 V < Vin < 38 V, –40°C ≤TJ≤1.217 1.255
125°C
IFB FB Bias Current VFB = 1.20 V ±200 nA
ΔVOUT/ΔVIN Output Voltage Line Regulation 5 V < Vin < 38 V 0.00025 %/V
ΔVOUT/ΔIOUT Output Voltage Load Regulation 0.8 V < VCOMP < 1.15 V 0.08 %/A
VDD VDD Pin Output Voltage 7 V < Vin < 35 V, IVDD= 0 mA to 5.99 V
5 mA
7 V < Vin < 35 V, IVDD= 0 mA to 5.50 6.50
5 mA, –40°C ≤TJ≤125°C
ISS_Source Soft-start Source Current 2.5 µA
–40°C ≤TJ≤125°C 1.5 4.6
Vbias_th VBIAS On Voltage Specified at IBIAS = 92.5% of full 2.64 2.9 3.07 V
value
(1) Min and Max limits are 100% production tested at 25°C. Limits over the operating temperature range are ensured through correlation
using Statistical Quality Control (SQC) methods. Limits are used to calculate Average Outgoing Quality Level (AOQL).
(2) Iq and ISD specify the current into the VIN and AVIN pins. IBIAS is the current into the VBIAS pin when the VBIAS voltage is greater
than 3 V. All quiescent current specifications apply to non-switching operation.
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Electrical Characteristics (continued)
Unless otherwise stated, Vin = 12 V, TJ= 25°C. Minimum and Maximum limits are ensured through test, design, or statistical
correlation. PARAMETER TEST CONDITIONS MIN TYP(1) MAX UNIT
PROTECTION
ILIMPK Peak Current Limit 4.7 A
–40°C ≤TJ≤125°C 3.15 6.05
VFB_SC Short Circuit Frequency Foldback Measured at FB falling 0.87 V
Threshold
F_min_sc Min Frequency in Foldback VFB < 0.3 V 45 kHz
VTH_PGOOD Power Good Threshold Measured at FB, PGOOD rising 92%
Measured at FB, PGOOD rising, 89% 95%
–40°C ≤TJ≤125°C
PGOOD Hysteresis 2% 6% 8%
IPGOOD_HI PGOOD Leakage Current PGOOD = 5 V 1.25 nA
RDS_PGOOD PGOOD On Resistance PGOOD sink current = 500 µA 150 Ω
VUVLO Under-voltage Lock-Out Threshold Vin falling , shutdown, VDD = 2.96 V
VIN
Vin falling , shutdown, VDD = 2.70 3.30
VIN, –40°C ≤TJ≤125°C
Vin rising, soft-start, VDD = VIN 3.99
Vin rising, soft-start, VDD = VIN, 3.70 4.30
–40°C ≤TJ≤125°C
TSD Thermal Shutdown Threshold 160 °C
θJA Thermal Resistance Power dissipation = 1W, 0 lfpm 32 °C/W
air flow
LOGIC
VthEN Enable Threshold Voltage Enable rising 1.18 V
Enable rising, –40°C ≤TJ≤0.8 1.4
125°C
Enable Hysteresis 180 mV
IEN_Source EN Source Current EN = 0 V 4.85 µA
VTH_FPWM FPWM Threshold 1.24 V
–40°C ≤TJ≤125°C 0.8 1.6
IFPWM FPWM Leakage Current FPWM = 5 V 3 nA
EA
gm Error Amp Trans-conductance 675 µmho
–40°C ≤TJ≤125°C 400 1000
ICOMP COMP Source Current VCOMP = 0.9 V 57 µA
COMP Sink Current VCOMP = 0.9 V 57 µA
VCOMP COMP Pin Voltage Range 0.64 1.27 V
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7.7 Switching Characteristics
over operating free-air temperature range (unless otherwise noted)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
Isw = 2A 0.095
RDS(ON) Switch On Resistance Ω
Isw = 2A, –40°C ≤TJ≤125°C 0.040 0.200
Vin = 38 V, VSW = 0 V 0.002
Isw_off Switch Off State Leakage Current μA
Vin = 38 V, VSW = 0 V, –40°C ≤TJ5.0
≤125°C
RFREQ = 62k, 124k, 240k, –40°C ≤±10%
fsw Switching Frequency TJ≤125°C
VFREQ FREQ Voltage 1.0 V
fSW range Switching Frequency Range –40°C ≤TJ≤125°C 150 500 kHz
SYNC rising 1.23
SYNC rising, –40°C ≤TJ≤125°C 1.6
Sync Pin Threshold V
VSYNC SYNC falling 1.10
SYNC falling, –40°C ≤TJ≤125°C 0.8
Sync Pin Hysteresis 135 mV
ISYNC SYNC Leakage Current 2 nA
Upper Frequency Synchronization As compared to nominal fSW, –40°C +30%
FSYNC_UP Range ≤TJ≤125°C
Lower Frequency Synchronization As compared to nominal fSW, –40°C –20%
FSYNC_DN Range ≤TJ≤125°C
TOFFMIN Minimum Off-time 300 ns
TONMIN Minimum On-time 190 ns
THSLEEP_HYS Sleep Mode Threshold Hysteresis VFB rising, % of THWAKE 101.3%
Measured at falling FB, COMP = 0.6 1.236
THWAKE Wake Up Threshold V
V
BOOT = 6 V, SW = GND 0.001
IBOOT BOOT Pin Leakage Current μA
BOOT = 6 V, SW = GND, –40°C ≤5.0
TJ≤125°C
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7.8 Typical Characteristics
Unless otherwise specified the following conditions apply: Vin = 12 V, TJ= 25°C.
Figure 1. Efficiency vs Load Current (300 kHz) Figure 2. Efficiency vs Load Current (500 kHz)
Figure 3. VFB vs Temperature Figure 4. VFB vs Vin (IDC = 300 mA)
Figure 5. IQ and IVBIAS vs Temperature (Sleep Mode) Figure 6. IQ and IVBIAS vs Temperature (PWM Mode)
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Typical Characteristics (continued)
Unless otherwise specified the following conditions apply: Vin = 12 V, TJ= 25°C.
Figure 0. UNDEFINED
Figure 7. Peak Current Limit vs Temperature
Figure 0. UNDEFINED
Figure 8. Normalized Switching Frequency vs Temperature
(300 kHz)
Figure 9. UVLO Threshold vs Temperature (VDD = VIN)
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sleep
wake
0.6V
2.5 PA
+
-
0.9V
V clamp
+
-
+
-
SS
logic
+
-
TSD
VDD_low
ss end
blanking
qn
qn
fpwm
Sync and
bootstrap
control
fpwm
Sync
Sync
5 PA
VDD_low
EN
FB
SW
SW
FPWM
COMP
PGOOD
SS
AGND
SYNC
BOOT
VIN
VBIAS
VDD
AVIN
SD
+
-
+
-
+
-
Clock / Sync
BG
BG
+
I Sense
PWM
Comp
BG
IREF
TSD
FPWM / Sleep
Peak Current
Control
Sleep
Reset
EA
PG
0.92BG
+
-
Corrective
Ramp
PWM Control
Logic
+
-
+
-Sleep
Set
LDO
UVLO
on
Switchover
control
VREG
sleep
EP
frequency
foldback
FREQ
soft start on
SD
ff
ff
PGND
1 Ps/DIV
Vout
10 mV/Div
IL
500 mA/Div
ID
1A/Div
VSW
5V/Div
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8 Detailed Description
8.1 Overview
The LM26003 device is a current mode PWM buck regulator. At the beginning of each clock cycle, the internal
high-side switch turns on, allowing current to ramp-up in the inductor. The inductor current is internally monitored
during each switching cycle. A control signal derived from the inductor current is compared to the voltage control
signal at the COMP pin, derived from the feedback voltage. When the inductor current reaches its threshold, the
high-side switch is turned off and inductor current ramps-down. While the switch is off, inductor current is
supplied through the catch diode. This cycle repeats at the next clock cycle. In this way, duty-cycle and output
voltage are controlled by regulating inductor current. Current mode control provides superior line and load
regulation. Other benefits include cycle-by-cycle current limiting and a simplified compensation scheme. Typical
PWM waveforms are shown in Figure 10.
Figure 10. PWM Waveforms 1A-Load, Vin = 12 V
8.2 Functional Block Diagram
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Iss x tss
1.236V
Css =
Vout
10 mV/Div
VSW
5V/Div
IL
200 mA/Div
1 Ps/DIV
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8.3 Feature Description
8.3.1 FPWM
Pulling the FPWM pin high disables sleep mode and forces the LM26003 device to always operate in PWM
mode. Light load efficiency is reduced in PWM mode, but switching frequency remains stable. The FPWM pin
can be connected to the VDD pin to pull it high. In FPWM mode, under light load conditions, the regulator
operates in discontinuous conduction mode (DCM). In discontinuous conduction mode, current through the
inductor starts at zero and ramps-up to its peak, then ramps-down to zero again. Until the next cycle, the inductor
current remains at zero. At nominal load currents, in FPWM mode, the device operates in continuous conduction
mode, where positive current always flows in the inductor. Typical discontinuous operation waveforms are shown
in Figure 11.
Figure 11. Discontinuous Mode Waveforms 75-mA Load, Vin = 12 V
At very light load, in FPWM mode, the LM26003 device may enter sleep mode. This is to prevent an overvoltage
condition from occurring. However, the FPWM sleep threshold is much lower than in normal operation.
8.3.2 Soft-Start
The soft-start feature provides a controlled output voltage ramp-up at startup. This reduces inrush current and
eliminates output overshoot at turn-on. The soft-start pin, SS, must be connected to GND through a capacitor. At
power-on, enable, or UVLO recovery, an internal 2.5-µA (typical) current charges the soft-start capacitor. During
soft-start, the error amplifier output voltage is controlled by both the soft-start voltage and the feedback loop. As
the SS pin voltage ramps-up, the duty-cycle increases proportional to the soft-start ramp, causing the output
voltage to ramp-up. The rate at which the duty-cycle increases depends on the capacitance of the soft-start
capacitor. The higher the capacitance, the slower the output voltage ramps-up. The soft-start capacitor value can
be calculated with the following equation:
where
• tss is the desired soft-start time
• Iss is the soft-start source current. (1)
During soft-start, current limit and synchronization remain in effect, while sleep mode and frequency foldback are
disabled. Soft-start mode ends when the SS pin voltage reaches 1.23 V typical. At this point, output voltage
control is transferred to the FB pin and the SS pin is discharged.
8.3.3 Current Limit
The peak current limit is set internally by directly measuring peak inductor current through the internal switch. To
ensure accurate current sensing, AVIN should be bypassed with a minimum 100-nF ceramic capacitor as close
as possible to AVIN and GND pins. Also the PVIN pin should be bypassed with at least 2.2 µF to ensure low
jitter operation.
When the inductor current reaches the current limit threshold, the internal FET turns off immediately allowing
inductor current to ramp-down until the next cycle. This reduction in duty-cycle corresponds to a reduction in
output voltage.
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050 100 150 200 250 300
RFREQ (k:)
100
200
300
400
500
600
SWITCHING FREQUENCY (kHz)
(Vin ± Vout) x Vout
fsw x L x Vin
Iripple =
Iripple
2
Iloadmax = 3.15A -
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Feature Description (continued)
The current limit comparator is disabled for less than 150 ns at the leading edge for increased immunity to
switching noise.
Because the current limit monitors peak inductor current, the DC load current limit threshold varies with
inductance and frequency. Assuming a minimum current limit of 3.15 A, maximum load current can be calculated
as follows:
(2)
Where Iripple is the peak-to-peak inductor ripple current, calculated as shown below:
(3)
To find the worst case (lowest) current limit threshold, use the maximum input voltage and minimum current limit
specification.
During high overcurrent conditions, such as output short circuit, the LM26003 device employs frequency foldback
as a second level of protection. If the feedback voltage falls below the short circuit threshold of 0.9 V, operating
frequency is reduced, thereby reducing average switch current. This is especially helpful in short circuit
conditions, when inductor current can rise very high during the minimum on-time. Frequency reduction begins at
20% below the nominal frequency setting. The minimum operating frequency in foldback mode is 45 kHz
(typical).
If the FB voltage falls below the frequency foldback threshold during frequency synchronized operation, the
SYNC function is disabled. Operating frequency versus FB voltage in short circuit conditions is shown in the
Typical Characteristics section.
Under conditions where the on-time is close to minimum (less than 200 ns typically), such as high input voltage
and high switching frequency, the current limit may not function properly. This is because the current limit circuit
cannot reduce the on-time below minimum which prevents entry into frequency foldback mode. There are two
ways to ensure proper current limit and foldback operation under high input voltage conditions. First, the
operating frequency can be reduced to increase the nominal on-time. Second, the inductor value can be
increased to slow the current ramp and reduce the peak overcurrent.
8.3.4 Frequency Adjustment and Synchronization
The switching frequency of the LM26003 device can be adjusted between 150 kHz and 500 kHz using a single
external resistor. This resistor is connected from the FREQ pin to ground as shown in the typical application. The
resistor value can be calculated with the following empirically derived equation:
RFREQ = (6.25 x 1010) x fSW-1.042 (4)
Figure 12. Switching Frequency vs RFREQ
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Feature Description (continued)
The switching frequency can also be synchronized to an external clock signal using the SYNC pin. The SYNC
pin allows the operating frequency to be varied above and below the nominal frequency setting. The adjustment
range is from 30% above nominal to 20% below nominal. External synchronization requires a 1.23-V minimum
(typical) peak signal level at the SYNC pin. The FREQ resistor must always be connected to initialize the nominal
operating frequency. The operating frequency is synchronized to the falling edge of the SYNC input. When
SYNC goes low, the high-side switch turns on. This allows any duty-cycle to be used for the sync signal when
synchronizing to a frequency higher than nominal. When synchronizing to a lower frequency, however, there is a
minimum duty-cycle requirement for the SYNC signal, given in the equation below:
where
• fnom is the nominal switching frequency set by the FREQ resistor
• fsync is a square wave. (5)
If the SYNC pin is not used, it must be pulled low for normal operation. A 10 kΩpulldown resistor is
recommended to protect against a missing sync signal. Although the LM26003 device is designed to operate at
up to 500 kHz, maximum load current may be limited at higher frequencies due to increased temperature rise.
See the Thermal Considerations and TSD section.
8.3.5 VBIAS
The VBIAS pin is used to bypass the internal regulator which provides the bias voltage to the LM26003 device.
When the VBIAS pin is connected to a voltage greater than 3 V, the internal regulator automatically switches
over to the VBIAS input. This reduces the current into VIN (Iq) and increases system efficiency. Using the VBIAS
pin has the added benefit of reducing power dissipation within the device.
For most applications where 3 V < Vout < 10 V, VBIAS can be connected to VOUT. If not used, VBIAS should be
tied to GND.
If VBIAS drops below 2.9 V (typical), the device automatically switches over to supply the internal bias voltage
from Vin.
When the LM26003 device is powered with the circuit's output voltage through VBIAS, especially at low output
voltages such as 3.3 V, output ripple noise can couple in through the Vbias pin causing some falling edge jitter
on the switch node. To avoid this, additional bypassing close to the VBIAS pin with a low ESR capacitor can be
implemented. The circuit diagram in Figure 16 shows this bypass capacitor C8.
8.3.6 Low VIN Operation and UVLO
The LM26003 device is designed to remain operational during short line transients when the input voltage may
drop as low as 3.0 V. Minimum nominal operating input voltage is 4.0 V. Below this voltage, switch RDS(ON)
increases, due to the lower gate drive voltage from VDD. The minimum voltage required at VDD is approximately
3.5 V for normal operation within specification.
VDD can also be used as a pullup voltage for functions such as PGOOD and FPWM. Note that if VDD is used
externally, the pin is not recommended for loads greater than 1 mA.
If the input voltage approaches the nominal output voltage, the duty-cycle is maximized to hold up the output
voltage. In this mode of operation, once the duty-cycle reaches its maximum, the LM26003 device can skip a
maximum of seven off pulses, effectively increasing the duty-cycle and thus minimizing the dropout from input to
output. Typical off-pulse skipping waveforms are shown in Figure 13.
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Vout
20 mV/Div
VSW
2V/Div
IL
100 mA/Div
4 Ps/DIV
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,
LM26003-Q1
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Feature Description (continued)
Figure 13. Off-Pulse Skipping Waveforms Vin = 3.5 V, Vnom = 3.3 V, fnom = 305 kHz
UVLO is sensed at both VIN and VDD, and is activated when either voltage falls below 2.96 V (typical). Although
VDD is typically less than 200 mV below VIN, it will not discharge through VIN. Therefore when the VIN voltage
drops rapidly, VDD may remain high, especially in sleep mode. For fast line voltage transients, using a larger
capacitor at the VDD pin can help to hold off a UVLO shutdown by extending the VDD discharge time. By
holding up VDD, a larger cap can also reduce the RDS(ON) (and dropout voltage) in low VIN conditions.
Alternately, under heavy loading the VDD voltage can fall several hundred mV below VIN. In this case, UVLO
may be triggered by VDD even though the VIN voltage is above the UVLO threshold.
When UVLO is activated the LM26003 device enters a standby state in which VDD remains charged. As input
voltage and VDD voltage rise above 3.99 V (typical) the device will restart from soft-start mode.
8.3.7 PGOOD
A Power Good pin, PGOOD, is available to monitor the output voltage status. The pin is internally connected to
an open-drain MOSFET, which remains open while the output voltage is within operating range. PGOOD goes
low (low impedance to ground) when the output falls below 89% of nominal or EN is pulled low. When the output
voltage returns to within 95% of nominal, as measured at the FB pin, PGOOD returns to a high state. For
improved noise immunity, there is a 5-µs delay between the PGOOD threshold and the PGOOD pin going low.
8.4 Device Functional Modes
The LM26003 device has three basic operation modes: shutdown, sleep or light load operation and full operation.
The part enters shutdown mode when the EN pin is pulled low. In this mode, the converter is disabled and the
quiescent current is minimized. See the Enable section for more details.
The part enters sleep mode when the converter is active (EN high) and the output current is low. Sleep mode is
activated as the COMP voltage naturally falls below a typical 0.6 V threshold in light load operation. When
operating in sleep mode, the switching events of the converter are reduced in order to lower the current
consumption of the system. Forcing the FPWM pin high will prevent sleep mode operation. Refer to Sleep Mode
for details about operating in Sleep mode as well as entering and exiting sleep mode.
When the part in enabled and the output load is higher, the part will be in full PWM operation. In addition to these
normal functioning modes, the LM26003 device has a frequency foldback operating mode which reduces the
operating frequency to protect from short circuits. See Current Limit for more details.
8.4.1 Enable
The LM26003 device provides a shutdown function via the EN pin to disable the device when the output voltage
does not need to be maintained. EN is an analog level input with typically 180 mV of hysteresis. The device is
active when the EN pin is above 1.18 V (typical) and in shutdown mode when EN is below this threshold. When
EN goes high, the internal VDD regulator turns on and charges the VDD capacitor. When VDD reaches 3.9 V
(typical), the soft-start pin begins to source current. In shutdown mode, the VDD regulator shuts down and total
quiescent current is reduced to 10.8 µA (typical). Because the EN pin sources 4.85 µA (typical) of pullup current,
this pin can be left open for always-on operation. When open, EN will be pulled up to VIN.
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Product Folder Links: LM26003 LM26003-Q1
ISleep = Imin + 0.23 PVin - Vout
L
2
xfsw x L
D x 2 x (Vin ± Vout)
Vout
50 mV/Div
VSW
5V/Div
IL
400 mA/Div
100 Ps/DIV
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,
LM26003-Q1
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SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
Device Functional Modes (continued)
If EN is connected to VIN, it must be connected through a 10 kΩresistor to limit noise spikes. EN can also be
driven externally with a maximum voltage of 38 V or VIN + 15 V, whichever is lower.
8.4.2 Sleep Mode
In light load conditions, the LM26003 device automatically switches into sleep mode for improved efficiency. As
loading decreases, the voltage at FB increases and the COMP voltage decreases. When the COMP voltage
reaches the 0.6-V (typical) clamp threshold and the FB voltage rises 1% above nominal, sleep mode is enabled
and switching stops. The regulator remains in sleep mode until the FB voltage falls to the reset threshold, at
which point switching resumes. This 1% FB window limits the corresponding output ripple requirement to
approximately 1% of nominal output voltage. The sleep cycle will repeat until load current is increased. Figure 14
shows typical switching and output voltage waveforms in sleep mode.
Figure 14. Sleep Mode Waveforms 25-mA Load, Vin = 12 V
In sleep mode, quiescent current is reduced to less than 40 µA (typical) when not switching. The DC sleep mode
threshold can roughly be calculated according to the equation below:
where
• Imin = Ilim/16 (4.7A/16 typically)
• D = duty-cycle, defined as (Vout + Vdiode)/Vin. (6)
When load current increases above this limit, the LM26003 device is forced back into PWM operation. The sleep
mode threshold varies with frequency, inductance, and duty-cycle as shown in Figure 15.
Figure 15. Sleep Mode Threshold vs Vin Vout = 3.3 V
Below the sleep threshold, decreasing load current results in longer sleep cycles, which can be quantified as
shown below:
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Device Functional Modes (continued)
Dwake = Iload/Isleep
where
• Dwake is the percentage of time awake when the load current is below the sleep threshold. (7)
Sleep mode combined with low IQ operation minimizes the input supply current. Input supply current in sleep
mode can be calculated based on the wake duty cycle, as shown below:
Iin = Iq + (IQG x Dwake) + (Io x D) (8)
Where IQG is the gate drive current, calculated as:
IQG = (9.2 x 10-9) x fSW
And Io is the sum of Iload, Ibias, and current through the feedback resistors.
Because this calculation applies only to sleep mode, use the Iq_Sleep_VB and IBIAS_SLEEP values from the Electrical
Characteristics. If VBIAS is connected to ground, use the same equation with Ibias equal to zero and Iq_Sleep_VDD.
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Product Folder Links: LM26003 LM26003-Q1
+
+
FB
COMP
SS
SYNC
EN
PGOOD
AVIN
4
5
8
6
9
14
7
VIN
3
VIN
VIN SW
SW
BOOT
VDD
VBIAS
FPWM
FREQ
13
17
15
12
16
18
19
SW 20
PGND 10
11
AGND
EP
21
VIN: 4V ± 38V
C10
100 PFC9
6.8 PFR7
0:
VDD C2
100 nF
R4
200 k:
PGOOD
EN
SYNC R6
10 k:
C3
22 nF C4
220 pF
C5
4.7 nF
R2
33.2
k:
R1
56.2 k:C11
**
R5
124 k:
C7
120 PF
C1
10 PF
VDD
C6
0.1 PF
D1
5A
L1 VOUT: 3.3V/3A
15 PH
R3
12.1 k:
GND
C8
10 PF
1
2
** optional component
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,
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SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
9 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
9.1 Application Information
The LM26003 is a switching regulator designed for the high-efficiency requirements of applications with standby
modes.
9.2 Typical Application
The following sections detail the design of a typical buck converter with sleep mode operation enabled (FPWM
low). Figure 16 shows a complete typical application schematic. The components have been selected based on
the design criteria given in the following sections.
Figure 16. Example Circuit 3A, 300 kHz
9.2.1 Design Requirements
The following parameters are needed to properly design the application and size the components:
Table 1. Design Parameters
PARAMETERS VALUES
Vout Output voltage
Vin min Maximum input voltage
Vin max Minimum input voltage
Iout max Maximum output current
Fsw Switching Frequency
Fbw Bandwidth of the converter
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(Vin ± Vout) x Vout x Re
Vin x fsw x Vrip
LMIN =
Iripple
2
Ipeak = Iload +
(Vin ± Vout) x Vout
fsw x L x Vin
Iripple =
R2 = Vout
Vfb
©
§¹
·
R1
-1
LM26003
,
LM26003-Q1
SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
www.ti.com
9.2.2 Detailed Design Procedure
Table 2. Bill of Materials
REFERENCE NUMBER MANUFACTURER PART NUMBER
C7 Nippon Chemi-Con APXE6R3ARA121ME61G
C9 TDK C4532X7R1H685M
C10 Panasonic EEE-FK1J101P
D1 Central Semiconductor CMSH 5-40
L1 Wurth 744770115
9.2.2.1 Setting Output Voltage
The output voltage is set by the ratio of a voltage divider at the FB pin as shown in the typical application. The
resistor values can be determined by the following equation:
where
• Vfb = 1.236 V typically. (9)
A maximum value of 150 kΩis recommended for the sum of R1 and R2.
As input voltage decreases towards the nominal output voltage, the LM26003 device can skip up to seven off-
pulses as described in the Low VIN Operation and UVLO section. In low output voltage applications, if the on-
time reaches TonMIN, the device will skip on-pulses to maintain regulation. There is no limit to the number of
pulses that are skipped. In this mode of operation, however, output ripple voltage may increase slightly.
9.2.2.2 Inductor
The output inductor should be selected based on inductor ripple current. The amount of inductor ripple current
compared to load current, or ripple content, is defined as Iripple/Iload. Ripple content should be less than 40%.
Inductor ripple current, Iripple, can be calculated as shown below:
(10)
Larger ripple content increases losses in the inductor and reduces the effective current limit.
Larger inductance values result in lower output ripple voltage and higher efficiency, but a slightly degraded
transient response. Lower inductance values allow for smaller case size, but the increased ripple lowers the
effective current limit threshold.
Remember that inductor value also affects the sleep mode threshold as shown in Figure 15.
When choosing the inductor, the saturation current rating must be higher than the maximum peak inductor
current and the RMS current rating should be higher than the maximum load current. Peak inductor current,
Ipeak, is calculated as:
(11)
For example, at a maximum load of 3 A and a ripple content of 10%, peak inductor current is equal to 3.15 A
which is safely at the minimum current limit of 3.15 A. By increasing the inductor size, ripple content and peak
inductor current are lowered, which increases the current limit margin.
The size of the output inductor can also be determined using the desired output ripple voltage, Vrip. The
equation to determine the minimum inductance value based on Vrip is as follows:
(12)
Where Re is the ESR of the output capacitors, and Vrip is a peak-to-peak value. This equation assumes that the
output capacitors have some amount of ESR. It does not apply to ceramic output capacitors.
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Irms = Vin
Vout x (Vin ± Vout)
Iload x
CMIN = ('Vt)2 - ('It x Re)2
'Vt -
L x ©
§
©
§
Vout x Re
2
'Vt
'It
ReMAX =
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SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
If this method is used, ripple content should still be verified to be less than 40% and that the peak currents do not
exceed the minimum current threshold.
9.2.2.3 Output Capacitor
The primary criterion for selecting an output capacitor is equivalent series resistance, or ESR.
ESR (Re) can be selected based on the requirements for output ripple voltage and transient response. Once an
inductor value has been selected, ripple voltage can be calculated for a given Re using the equation above for
LMIN. Lower ESR values result in lower output ripple.
Re can also be calculated from the following equation:
where
•ΔVt is the allowed voltage excursion during a load transient
•ΔIt is the maximum expected load transient. (13)
If the total ESR is too high, the load transient requirement cannot be met, no matter how large the output
capacitance.
If the ESR criteria for ripple voltage and transient excursion cannot be met, more capacitors should be used in
parallel.
For non-ceramic capacitors, the minimum output capacitance is of secondary importance, and is determined only
by the load transient requirement.
If there is not enough capacitance, the output voltage excursion will exceed the maximum allowed value even if
the maximum ESR requirement is met. The minimum capacitance is calculated as follows:
(14)
It is assumed the total ESR, Re, is no greater than ReMAX. Also, it is assumed that L has already been selected.
Generally speaking, the output capacitance requirement decreases with Re, ΔIt, and L. A typical value greater
than 120 µF works well for most applications.
9.2.2.4 Input Capacitor
In a switching converter, very fast switching pulse currents are drawn from the input rail. Therefore, input
capacitors are required to reduce noise, EMI, and ripple at the input to the LM26003 device. Capacitors must be
selected that can handle both the maximum ripple RMS current at highest ambient temperature as well as the
maximum input voltage. The equation for calculating the RMS input ripple current is shown below:
(15)
For noise suppression, a ceramic capacitor in the range of 1.0 µF to 10 µF should be placed as close as possible
to the PVIN pin. For the AVIN pin also some decoupling is necessary. It is very important that the pin is
decoupled with such a capacitor close to the AGND pin and the GND pin of the IC to avoid switching noise to
couple into the IC. Also some RC input filtering can be implemented using a small resistor between PVIN and
AVIN. In Figure 16 the resistor value of R7 is selected to be 0Ωbut can be increased to filter with different time
constants depending on the capacitor value used. When using a R7 resistor, keep in mind that the resistance will
increase the minimum input voltage threshold due to the voltage drop across the resistor.
The PVIN decoupling should be implemented in a way to minimize the trace length between the Cin capacitor
gnd and the Schottky diode gnd. A larger, high ESR input capacitor should also be used. This capacitor is
recommended for damping input voltage spikes during power-on and for holding up the input voltage during
transients. In low input voltage applications, line transients may fall below the UVLO threshold if there is not
enough input capacitance. Both tantalum and electrolytic type capacitors are suitable for the bulk capacitor.
However, large tantalums may not be available for high input voltages and their working voltage must be derated
by at least 2X.
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9.2.2.5 Bootstrap
The drive voltage for the internal switch is supplied via the BOOT pin. This pin must be connected to a ceramic
capacitor, Cboot, from the switch node, shown as C6 in the typical application. The LM26003 device provides the
VDD voltage internally, so no external diode is needed. A maximum value of 0.1 µF is recommended for Cboot.
Values smaller than 0.022 µF may result in insufficient hold up time for the drive voltage and increased power
dissipation.
During low Vin operation, when the on-time is extended, the bootstrap capacitor is at risk of discharging. If the
Cboot capacitor is discharged below approximately 2.5 V, the LM26003 device enters a high frequency re-charge
mode. The Cboot cap is re-charged via the synchronous FET shown in the block diagram. Switching returns to
normal when the Cboot cap has been recharged.
9.2.2.6 Catch Diode
When the internal switch is off, output current flows through the catch diode. Alternately, when the switch is on,
the diode sees a reverse voltage equal to Vin. Therefore, the important parameters for selecting the catch diode
are peak current and peak inverse voltage. The average current through the diode is given by:
IDAVE = Iload x (1-D)
where
• D is the duty-cycle, defined as Vout/Vin. (16)
The catch diode conducts the largest currents during the lowest duty-cycle. Therefore IDAVE should be calculated
assuming maximum input voltage. The diode should be rated to handle this current continuously. For overcurrent
or short-circuit conditions, the catch diode should be rated to handle peak currents equal to the peak current
limit.
The peak inverse voltage rating of the diode must be greater than maximum input voltage.
A Schottky diode must be used. It's low forward voltage maximizes efficiency and BOOT voltage, while also
protecting the SW pin against large negative voltage spikes.
When selecting the catch diode for high efficiency low output load applications, select a Schottky diode with low
reverse leakage current. Also keep in mind that the reverse leakage current of a Schottky diode increases with
temperature and with reverse voltage. Reverse voltage equals roughly the input voltage in a buck converter. At
hot, the diode reverse leakage current may be larger than the current consumption of the LM26003 device.
9.2.2.7 Compensation
The purpose of loop compensation is to ensure stable operation while maximizing dynamic performance. Stability
can be analyzed with loop gain measurements, while dynamic performance is analyzed with both loop gain and
load transient response. Loop gain is equal to the product of control-output transfer function (power stage) and
the feedback transfer function (the compensation network).
For stability purposes, our target is to have a loop gain slope that is –20dB/decade from a very low frequency to
beyond the crossover frequency. Also, the crossover frequency should not exceed one-fifth of the switching
frequency, that is, 60 kHz in the case of 300 kHz switching frequency.
For dynamic purposes, the higher the bandwidth, the faster the load transient response. The downside to high
bandwidth is that it increases the regulators susceptibility to board noise which ultimately leads to excessive
falling edge jitter of the switch node voltage.
A large DC gain means high DC regulation accuracy (that is, DC voltage changes little with load or line
variations).
To achieve this loop gain, the compensation components should be set according to the shape of the control-
output bode plot. A typical plot is shown in Figure 17.
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1
20 x S x Ro x Co
fp = +0.5
2 x S x L x fsw x Co
1
2S x Re x Co
fz =
2
fsw
fn =
B
fzc fpc1
FREQUENCY
GAIN (dB)
fpc
(0Hz)
-20dB/dec
-20dB/dec
0dB/dec
-60
-40
-20
0
20
0.01 0.1 1 10 100
FREQUENCY (kHz)
GAIN (dB)
1000
0
-45
-90
-135
-180
PHASE (º)
fp fz fn
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Figure 17. Control-Output Transfer Function
The control-output transfer function consists of one pole (fp), one zero (fz), and a double pole at fn (half the
switching frequency).
Referring to Figure 17, the following should be done to create a –20dB/decade roll-off of the loop gain:
1. Place a pole at 0Hz (fpc)
2. Place a zero at fp (fzc)
3. Place a second pole at fz (fpc1)
The resulting feedback (compensation) bode plot is shown below in Figure 18. Adding the control-output
response to the feedback response will then result in a nearly continuous -20db/decade slope.
Figure 18. Feedback Transfer Function
The control-output corner frequencies can be determined approximately by the following equations:
where
• Co is the output capacitance
• Ro is the load resistance
• Re is the output capacitor ESR
• fsw is the switching frequency. (17)
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1
2 x S x R1 x C11
fzff =
Vfb
fpff = fzff x Vout
1
2 x S x fz x R3
C4 =
1
2 x S x fpMAX x R3
C5 =
R3 = R1 + R2
R2
©
§¹
·
B
gm x
FB
COMP
To Vout
R2
R3
C4
C5
R1 C11
8
9
LM26003
,
LM26003-Q1
SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
www.ti.com
The effects of slope compensation and current sense gain are included in this equation. However, the equation is
an approximation intended to simplify loop compensation calculations.
Since fp is determined by the output network, it shifts with loading. Determine the range of frequencies
(fpmin/max) across the expected load range. Then determine the compensation values as described below and
shown in Figure 19.
Figure 19. Compensation Network
1. The compensation network automatically introduces a low frequency pole (fpc), which is close to 0 Hz.
2. Once the fp range is determined, R5 should be calculated using:
(18)
Where B is the desired feedback gain in v/v between fp and fz, and gm is the transconductance of the error
amplifier. A gain value around 10 dB (3.3 V/V) is generally a good starting point. Bandwidth increases with
increasing values of R3.
3. Next, place a zero (fzc) near fp using C5. C5 can be determined with the following equation:
(19)
The selected value of C5 should place fzc within a decade above or below fpmax and not less than fpmin. A
higher C5 value (closer to fpmin) generally provides a more stable loop, but too high a value will slow the
transient response time. Conversely, a smaller C5 value will result in a faster transient response, but lower
phase margin.
4. A second pole (fpc1) can also be placed at fz. This pole can be created with a single capacitor, C4. The
minimum value for this capacitor can be calculated by:
(20)
C4 may not be necessary in all applications. However if the operating frequency is being synchronized below
the nominal frequency, C4 is recommended. Although it is not required for stability, C4 is very helpful in
suppressing noise.
A phase lead capacitor can also be added to increase the phase and gain margins. The phase lead capacitor
is most helpful for high input voltage applications or when synchronizing to a frequency greater than nominal.
This capacitor, shown as C11 in Figure 19, should be placed in parallel with the top feedback resistor, R1.
C11 introduces an additional zero and pole to the compensation network. These frequencies can be
calculated as shown below:
(21)
A phase lead capacitor will boost loop phase around the region of the zero frequency, fzff. fzff should be
placed somewhat below the fpz1 frequency set by C4. However, if C11 is too large, it will have no effect.
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1 Ps/DIV
Vout
1V/Div
PGOOD
5V/Div
SS
1V/Div
EN
10V/Div
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9.2.3 Application Curves
Refer to Typical Characteristics.
Figure 20. Startup Waveforms Figure 21. Load Transient Response
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10 Power Supply Recommendations
The LM26003 device is designed to operate from various DC power supplies including a car battery. If so, VIN
input should be protected from reversal voltage and voltage dump over 48 V. The impedance of the input supply
rail should be low enough that the input current transient does not cause a drop below VIN UVLO level. If the
input supply is connected by using long wires, additional bulk capacitance may be required in addition to normal
input capacitor.
11 Layout
11.1 Layout Guidelines
Good board layout is critical for switching regulators such as the LM26003 device. First, the ground plane area
must be sufficient for thermal dissipation purposes, and second, appropriate guidelines must be followed to
reduce the effects of switching noise.
Switch mode converters are very fast switching devices. In such devices, the rapid increase of input current
combined with parasitic trace inductance generates unwanted Ldi/dt noise spikes at the SW node and also at the
VIN node. The magnitude of this noise tends to increase as the output current increases. This parasitic spike
noise may turn into electromagnetic interference (EMI) and can also cause problems in device performance.
Therefore, care must be taken in layout to minimize the effect of this switching noise.
The current sensing circuit in current mode devices can be easily affected by switching noise. This noise can
cause duty-cycle jitter which leads to increased spectrum noise. Although the LM26003 device has 150 ns
blanking time at the beginning of every cycle to ignore this noise, some noise may remain after the blanking time.
Following the important guidelines below will help minimize switching noise and its effect on current sensing.
The switch node area should be as small as possible. The catch diode, input capacitors, and output capacitors
should be grounded to the same local ground, with the bulk input capacitor grounded as close as possible to the
catch diode anode. Additionally, the ground area between the catch diode and bulk input capacitor is very noisy
and should be somewhat isolated from the rest of the ground plane.
A ceramic input capacitor must be connected as close as possible to the AVIN pin as well as PVIN pin. The
capacitor between AVIN and ground should be grounded close to the GND pins of the LM26003 device and the
PVIN capacitor should be grounded close to the Schottky diode ground. Often, the AVIN bypass capacitor is
most easily located on the bottom side of the PCB. It increases trace inductance due to the vias, it reduces trace
length however.
The above layout recommendations are illustrated in Figure 22.
It is a good practice to connect the EP, GND pin, and small signal components (COMP, FB, FREQ) to a separate
ground plane, shown in Figure 22 as EP GND, and in the schematics as a signal ground symbol. Both the
exposed pad and the GND pin must be connected to ground. This quieter plane should be connected to the high
current ground plane at a quiet location, preferably near the Vout ground as shown by the dashed line in
Figure 22.
The EP GND plane should be made as large as possible, since it is also used for thermal dissipation. Several
vias can be placed directly below the EP to increase heat flow to other layers when they are available. The
recommended via hole diameter is 0.3mm.
The trace from the FB pin to the resistor divider should be short and the entire feedback trace must be kept away
from the inductor and switch node. See AN-1229 SIMPLE SWITCHER ® PCB Layout Guidelines,SNVA054, for
more information regarding PCB layout for switching regulators.
24 Submit Documentation Feedback Copyright © 2008–2015, Texas Instruments Incorporated
Product Folder Links: LM26003 LM26003-Q1
PswAC = Vin x Iload x fsw x Vin x 10-9
1.33
©
§¹
·
Vin Vout
GND
SW
EP
GND
+
LM26003
,
LM26003-Q1
www.ti.com
SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
11.2 Layout Example
Figure 22. Example PCB Layout
11.3 Thermal Considerations and TSD
Although the LM26003 device has a built in current limit, at ambient temperatures above 80°C, device
temperature rise may limit the actual maximum load current. Therefore, temperature rise must be taken into
consideration to determine the maximum allowable load current.
Temperature rise is a function of the power dissipation within the device. The following equations can be used to
calculate power dissipation (PD) and temperature rise, where total PD is the sum of FET switching losses, FET
DC losses, drive losses, Iq, and VBIAS losses:
PDTOTAL = PswAC + PswDC + PQG + PIq + PVBIAS (22)
(23)
PswDC = D x Iload2x (0.095 + 0.00065 x (Tj- 25)) (24)
PQG = Vin x 9.2 x 10-9 x fsw (25)
PIq = Vin x Iq (26)
PVBIAS = Vbias x IVBIAS (27)
Given this total power dissipation, junction temperature can be calculated as follows:
Tj = Ta + (PDTOTAL xθJA) (28)
Where θJA= 32°C/W (typically) when using a multi-layer board with a large copper plane area. θJA varies with
board type and metallization area.
To calculate the maximum allowable power dissipation, assume Tj = 125°C. To ensure that junction temperature
does not exceed the maximum operating rating of 125°C, power dissipation should be verified at the maximum
expected operating frequency, maximum ambient temperature, and minimum and maximum input voltage. The
calculated maximum load current is based on continuous operation and may be exceeded during transient
conditions.
If the power dissipation remains above the maximum allowable level, device temperature will continue to rise.
When the junction temperature exceeds its maximum, the LM26003 device engages Thermal Shut Down (TSD).
In TSD, the part remains in a shutdown state until the junction temperature falls to within normal operating limits.
At this point, the device restarts in soft-start mode.
Copyright © 2008–2015, Texas Instruments Incorporated Submit Documentation Feedback 25
Product Folder Links: LM26003 LM26003-Q1
LM26003
,
LM26003-Q1
SNVS576F –AUGUST 2008–REVISED FEBRUARY 2015
www.ti.com
12 Device and Documentation Support
12.1 Device Support
12.1.1 Third-Party Products Disclaimer
TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT
CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES
OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER
ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.
12.2 Documentation Support
12.2.1 Related Documentation
AN-1229 SIMPLE SWITCHER®PCB Layout Guidelines,SNVA054
12.3 Related Links
The table below lists quick access links. Categories include technical documents, support and community
resources, tools and software, and quick access to sample or buy.
Table 3. Related Links
TECHNICAL TOOLS & SUPPORT &
PARTS PRODUCT FOLDER SAMPLE & BUY DOCUMENTS SOFTWARE COMMUNITY
LM26003 Click here Click here Click here Click here Click here
LM26003-Q1 Click here Click here Click here Click here Click here
12.4 Trademarks
SIMPLE SWITCHER is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
12.5 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
12.6 Glossary
SLYZ022 —TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
13 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
26 Submit Documentation Feedback Copyright © 2008–2015, Texas Instruments Incorporated
Product Folder Links: LM26003 LM26003-Q1
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM26003MH/NOPB ACTIVE HTSSOP PWP 20 73 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM26003
MH
LM26003MHX/NOPB ACTIVE HTSSOP PWP 20 2500 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM26003
MH
LM26003QMH/NOPB ACTIVE HTSSOP PWP 20 73 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM26003
QMH
LM26003QMHX/NOPB ACTIVE HTSSOP PWP 20 2500 Green (RoHS
& no Sb/Br) SN Level-1-260C-UNLIM -40 to 125 LM26003
QMH
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
PACKAGE OPTION ADDENDUM
www.ti.com 6-Feb-2020
Addendum-Page 2
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
OTHER QUALIFIED VERSIONS OF LM26003, LM26003-Q1 :
•Catalog: LM26003
•Automotive: LM26003-Q1
NOTE: Qualified Version Definitions:
•Catalog - TI's standard catalog product
•Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM26003MHX/NOPB HTSSOP PWP 20 2500 330.0 16.4 6.95 7.1 1.6 8.0 16.0 Q1
LM26003QMHX/NOPB HTSSOP PWP 20 2500 330.0 16.4 6.95 7.1 1.6 8.0 16.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 14-Sep-2017
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM26003MHX/NOPB HTSSOP PWP 20 2500 367.0 367.0 35.0
LM26003QMHX/NOPB HTSSOP PWP 20 2500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 14-Sep-2017
Pack Materials-Page 2
MECHANICAL DATA
PWP0020A
www.ti.com
MXA20A (Rev C)
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