LM3421MH/NOPB - NATIONAL SEMICONDUCTOR

VIN

COMP

CSH

HSN

HSP

RPD

RCT

AGND

OVP

nDIM

VCC

GATE

PGND

DDRV

9PWM

ILED

VIN

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Folder

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An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

LM342x N -Channel Controllers for Constant-Current LED Drivers

1 Features

1• VIN Range From 4.5 V to 75 V

• High-Side Adjustable Current Sense

• 2-Ω, 1-A Peak MOSFET Gate Driver

• Input Undervoltage and Output Overvoltage

Protection

• PWM and Analog Dimming

• Cycle-by-Cycle Current Limit

• Programmable Switching Frequency

• Zero Current Shutdown and Thermal Shutdown

• LED Output Status Flag (LM3423 and LM3423-Q0

Only)

• Fault Status Flag and Timer(LM3423 and

LM3423-Q0 Only)

2 Applications

• LED Drivers: Buck, Boost, Buck-Boost, and

SEPIC

• Indoor and Outdoor Area SSL

• Automotive

• General Illumination

• Constant-Current Regulators

3 Description

The LM3421 and LM3423 family of devices are

versatile high voltage N-channel MOSFET controllers

for LED drivers. They can be easily configured in

buck, boost, buck-boost and SEPIC topologies. This

flexibility, along with an input voltage rating of 75 V,

makes the these controllers ideal for illuminating

LEDs in a large family of applications.

Adjustable high-side current sense voltage allows for

tight regulation of the LED current with the highest

efficiency possible. The LM3421 and LM3423 devices

use predictive off-time (PRO) control, which is a

combination of peak current-mode control and a

predictive off-timer. This method of control eases the

design of loop compensation while providing inherent

input voltage feed-forward compensation.

The LM3421 and LM3423 devices include a high-

voltage start-up regulator that operates over a wide

input range of 4.5 V to 75 V. The internal PWM

controller is designed for adjustable switching

frequencies of up to 2 MHz, thus enabling compact

solutions.

Device Information(1)

PART NUMBER PACKAGE BODY SIZE (NOM)

LM3421 HTSSOP (16) 5.00 mm × 4.40 mm

LM3423 HTSSOP (20) 6.50 mm × 4.40 mm

(1) For all available packages, see the orderable addendum at

the end of the data sheet.

Typical Boost Application

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

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Product Folder Links: LM3421 LM3423

Table of Contents

1 Features.................................................................. 1

2 Applications ........................................................... 1

3 Description............................................................. 1

4 Revision History..................................................... 2

5 Device Comparison ............................................... 3

6 Pin Configuration and Functions......................... 4

7 Specifications......................................................... 5

7.1 Absolute Maximum Ratings ..................................... 5

7.2 ESD Ratings ............................................................ 6

7.3 Recommended Operating Conditions....................... 6

7.4 Thermal Information.................................................. 6

7.5 Electrical Characteristics .......................................... 7

7.6 Typical Characteristics ........................................... 11

8 Detailed Description............................................ 13

8.1 Overview................................................................. 13

8.2 Functional Block Diagram....................................... 13

8.3 Feature Description................................................. 14

9 Application and Implementation ........................ 28

9.1 Application Information............................................ 28

9.2 Typical Applications ................................................ 32

10 Power Supply Recommendations ..................... 65

10.1 General Recommendations .................................. 65

10.2 Input Supply Current Limit .................................... 65

11 Layout................................................................... 65

11.1 Layout Guidelines ................................................. 65

11.2 Layout Example .................................................... 66

12 Device and Documentation Support................. 67

12.1 Device Support...................................................... 67

12.2 Related Links ........................................................ 67

12.3 Community Resources.......................................... 67

12.4 Trademarks........................................................... 67

12.5 Electrostatic Discharge Caution............................ 67

12.6 Glossary................................................................ 67

13 Mechanical, Packaging, and Orderable

Information........................................................... 67

4 Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Revision F (July 2015) to Revision G Page

• Deleted references to automotive grade (LM342x-Q1 and LM342x-Q0) devices, now available in data sheet SNVSB95... 1

• Corrected typographic error in Table 1................................................................................................................................... 3

• Changed EN pulldown resistance specification minimum value from: 0.45 MΩto: 0.245 MΩElectrical

Characteristics table. ............................................................................................................................................................. 7

• Changed EN pulldown resistance specification maximum value from: 1.3 MΩto: 2.85 MΩin Electrical

Characteristics table. ............................................................................................................................................................. 7

Changes from Revision E (April 2013) to Revision F Page

• Added ESD Ratings table, Feature Description section, Device Functional Modes,Application and Implementation

section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and

Mechanical, Packaging, and Orderable Information section ................................................................................................. 1

Changes from Revision D (May 2013) to Revision E Page

• Changed layout of National Data Sheet to TI format ........................................................................................................... 64

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

5 Device Comparison

Table 1. Device Comparison

DEVICE FLAG FEATURES QUALIFACTION TEMPERATURE

RANGE, TA

LED OUTPUT FAULT

STATUS

LM3421-Q0 No No AEC-Q100 Grade 0 –40°C to +150°C

LM3421-Q1 No No AEC-Q100 Grade 1 –40°C to +125°C

LM3423-Q0 Yes Yes AEC-Q100 Grade 0 –40°C to +150°C

LM3423-Q1 Yes Yes AEC-Q100 Grade 1 –40°C to +125°C

LM3421 No No Commercial Grade –40°C to +125°C

LM3423 Yes Yes Commercial Grade –40°C to +125°C

120

IADJ

VREF

COMP

CSH

OVP

AGND

RCT

nDIM

VIN

FLT

TIMR

VCC

RPD

PGND

DDRV

HSP

HSN

GATE

21 Thermal

Pad

VCC

COMP

CSH

RPD

PGND

DDRV

HSP

HSN

OVP

GATE

AGND

RCT

nDIM

VIN 116

17 Thermal

Pad

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

6 Pin Configuration and Functions

PWP Package

16-Pin HTSSOP

Top View

PWP Package

20-Pin HTSSOP

Top View

(1) G = Ground, I = Input, O = Output

Pin Functions

PIN I/O(1) FUNCTION

NAME LM3423 LM3421

AGND 6 6 G Analog ground. Connect to PGND through the DAP copper pad to provide ground return

for CSH, COMP, RCT, and TIMR.

COMP 3 3 I Compensation. Connect a capacitor to AGND to set the compensation.

CSH 4 4 I Current sense high. Connect a resistor to AGND to set the signal current. For analog

dimming, connect a controlled current source or a potentiometer to AGND as detailed in

the Analog Dimming section.

DDRV 13 9 O Dim gate drive output. Connect to the gate of the dimming MOSFET.

DPOL 12 — I Dim polarity. Connect to AGND if dimming with a series P-channel MOSFET or leave

open when dimming with series N-channel MOSFET.

EN 2 2 I Enable. Connect to AGND for zero current shutdown or apply more than 2.4 V to enable

device.

FLT 9 — I Fault flag. Connect to pullup resistor from VIN and N-channel MOSFET open-drain

output is high when a fault condition is latched by the timer.

GATE 15 11 O Main gate drive output. Connect to the gate of the main switching MOSFET.

HSN 20 16 I LED current sense negative. Connect through a series resistor to the negative side of

the LED current sense resistor.

HSP 19 15 I LED current sense positive. Connect through a series resistor to the positive side of the

LED current sense resistor.

IS 17 13 I Main switch current sense. Connect to the drain of the main N-channel MOSFET switch

for RDS-ON sensing or to a sense resistor installed in the source of the same device.

LRDY 11 — O LED ready flag. Connect to pullup resistor from VIN and N-channel MOSFET open-drain

output pulls down when the LED current is not in regulation.

nDIM 8 8 I Dimming input and undervoltage protection. Connect a PWM signal for dimming as

detailed in the PWM Dimming section and/or a resistor divider from VIN to program input

undervoltage lockout (UVLO). Turnon threshold is 1.24 V and hysteresis for turnoff is

provided by a 23-µA current source.

OVP 7 7 I Overvoltage protection. Connect to a resistor divider from VOto program output

overvoltage lockout (OVLO). Turnoff threshold is 1.24 V and hysteresis for turnon is

provided by 23-µA current source.

PGND 14 10 G Power ground. Connect to AGND through the DAP copper pad to provide ground return

for GATE and DDRV.

RCT 5 5 I Resistor capacitor timing. External RC network sets the predictive off-time and thus the

switching frequency.

LM3421

LM3423

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Product Folder Links: LM3421 LM3423

Pin Functions (continued)

PIN I/O(1) FUNCTION

NAME LM3423 LM3421

RPD 18 14 I Resistor pulldown. Connect the low side of all external resistor dividers (VIN UVLO,

OVP) to implement zero-current shutdown.

TIMR 10 — I Fault timer. Connect a capacitor to AGND to set the time delay before a sensed fault

condition is latched.

VIN 1 1 I Input voltage. Bypass with 100-nF capacitor to AGND as close to the device as possible

in the printed-circuit-board layout.

VCC 16 12 I Internal regulator output. Bypass with 2.2-µF to 3.3-µF ceramic capacitor to PGND.

Thermal PAD G Thermal PAD on bottom of IC. Star ground, connecting AGND and PGND.

DAP DAP (21) DAP (17) G Star ground, connecting AGND and PGND.

(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(2) If Military/Aerospace specified devices are required, contact the Texas Instruments Sales Office/Distributors for availability and

specifications.

(3) Refer to http://www.ti.com/packaging for more detailed information and mounting techniques.

7 Specifications

7.1 Absolute Maximum Ratings

over operating free-air temperature range (unless otherwise noted)(1)(2)

MIN MAX UNIT

VIN, EN, RPD, nDIM –0.3 76 V

–1 continuous mA

OVP, HSP, HSN, LRDY, FLT, DPOL –0.3 76 V

–100 continuous µA

RCT –0.3 76 V

–1 continuous 5 continuous mA

IS –0.3 76 V

–2 for 100 ns

–1 continuous mA

VCC –0.3 8 V

TIMR –0.3 7 V

–100 continuous 100 continuous µA

COMP, CSH –0.3 6 V

–200 continuous 200 continuous µA

GATE, DDRV –0.3 VCC V

–2.5 for 100 ns VCC+ 2.5 for 100 ns

–1 continuous 1 continuous mA

PGND –0.3 0.3 V

–2.5 for 100 ns 2.5 for 100 ns

Continuous power dissipation Internally Limited

Maximum junction temperature Internally Limited

Maximum lead temperature (solder and reflow) (3) 260 °C

Storage temperature –65 150 °C

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.

7.2 ESD Ratings VALUE UNIT

V(ESD) Electrostatic discharge

Human-body model (HBM), per ANSI/ESDA/JEDEC JS-

001(1) ±2000 V

Charged-device model (CDM), per JEDEC specification

JESD22-C101 ±500

7.3 Recommended Operating Conditions MIN MAX UNIT

Operating junction temperature, TJLM3421 , LM3423 −40 125 °C

Input voltage, VIN 4.5 75 V

(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application

report, SPRA953.

7.4 Thermal Information

THERMAL METRIC(1) LM3421 LM3423

UNITPWP (HTSSOP) PWP (HTSSOP)

16 PINS 20 PINS

RθJA Junction-to-ambient thermal resistance 38.9 36.7 °C/W

RθJC(top) Junction-to-case (top) thermal resistance 23.1 21.5 °C/W

RθJB Junction-to-board thermal resistance 16.8 18 °C/W

ψJT Junction-to-top characterization parameter 0.6 0.5 °C/W

ψJB Junction-to-board characterization parameter 16.6 17.8 °C/W

RθJC(bot) Junction-to-case (bottom) thermal resistance 1.7 1.9 °C/W

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

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7.5 Electrical Characteristics

VIN = 14, −40°C ≤TJ≤125°C unless otherwise specified. Minimum and maximum limits are specified through test, design, or

statistical correlation. Typical values represent the most likely parametric norm at TJ= 25°C, and are provided for reference

purposes only.PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

START-UP REGULATOR

VCCREG VCC regulation ICC = 0 mA 6.3 7.35 V

ICC = 0 mA, TA= 25°C 6.9

ICCLIM VCC current limit VCC = 0 V 20 mA

VCC = 0 V, TA= 25°C 25

IQQuiescent Current VEN = 3 V, Static 3 mA

VEN = 3 V, Static, TA= 25°C 2

ISD Shutdown current VEN = 0 V 1 µA

VEN = 0 V, TA= 25°C 0.1

VCC SUPPLY

VCCUV VCC UVLO Threshold

VCC Increasing 4.5

VCC Increasing, TA= 25°C 4.17

VCC Decreasing 3.7

VCC Decreasing, TA= 25°C 4.08

VCCHYS VCC UVLO Hysteresis TA= 25°C 0.1 V

ENABLE THRESHOLDS

ENST EN start-up threshold VEN Increasing 2.4 V

VEN Increasing, TA= 25°C 1.75

ENST EN start-up threshold VEN Decreasing 0.8 V

VEN Decreasing, TA= 25°C 1.63

ENSTHYS EN start-up hysteresis TA= 25°C 0.1 V

REN EN pulldown resistance VEN = 1 V 0.245 2.85 MΩ

VEN = 1 V, TA= 25°C 0.82

CSH THRESHOLDS

CSH high fault CSH Increasing, TA= 25°C 1.6 V

CSH low condition on LRDY

Pin CSH increasing, TA= 25°C LM3423 1 V

OV THRESHOLDS

OVPCB OVP OVLO threshold OVP Increasing 1.185 1.285 V

OVP Increasing, TA= 25°C 1.24

OVPHYS OVP hysteresis source current OVP Active (high) 20 25 µA

OVP Active (high), TA= 25°C 23

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

Electrical Characteristics (continued)

VIN = 14, −40°C ≤TJ≤125°C unless otherwise specified. Minimum and maximum limits are specified through test, design, or

statistical correlation. Typical values represent the most likely parametric norm at TJ= 25°C, and are provided for reference

purposes only.PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

(1) Specified by design. Not production tested.

(2) f = 25/(CT× RT

DPOL THRESHOLDS

DPOLTHRSH DPOL logic threshold DPOL Increasing 2 2.6 V

DPOL Increasing, TA= 25°C 2.3

RDPOL DPOL pullup resistance 1200 kΩ

TA= 25°C 500

FAULT TIMER

VFLTTH Fault threshold 1.185 1.29 V

TA= 25°C 1.24

IFLT FAULT pin source current 10 13 µA

TA= 25°C 11.5

ERROR AMPLIFIER

VREF CSH reference voltage w/r/t to AGND 1.21 1.26 V

w/r/t to AGND, TJ= 25°C 1.235

Error amplifier input bias

current TJ= 25°C –0.6 0 0.6 µA

COMP sink or source current 22 35 µA

TJ= 25°C 30

Transconductance TJ= 25°C 100 µA/V

Linear input range (1), TJ= 25°C ±125 mV

Transconductance bandwidth –6dB Unloaded Response (1),

MIN = TJ= 25°C 0.5 1 MHz

OFF TIMER

tOFF(min) Minimum OFF-time RCT = 1 V through 1 kΩ75 ns

RCT = 1 V through 1 kΩ, TJ= 25°C 35

RRCT RCT reset pulldown resistance 120 Ω

TJ= 25°C 36

VRCT VIN/25 reference voltage VIN = 14 V 540 585 mV

VIN = 14 V, TJ= 25°C 565

fContinuous conduction

switching frequency 2.2 nF > CT> 470 pF, TJ= 25°C (See (2)) Hz

PWM COMPARATOR

COMP-to-PWM offset voltage 700 900 mV

TJ= 25°C 800

CURRENT LIMIT (IS)

ILIM Current limit threshold 215 275 mV

TJ= 25°C 245

Current limit delay-to-output 75 ns

TJ= 25°C 35

tLEB Leading edge blanking (LEB)

time 115 325 ns

TJ= 25°C 210

HIGH SIDE TRANSCONDUCTANCE AMPLIFIER

Input bias current TJ= 25°C 11.5 µA

gMTransconductance 20 mA/V

TJ= 25°C 119

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Electrical Characteristics (continued)

VIN = 14, −40°C ≤TJ≤125°C unless otherwise specified. Minimum and maximum limits are specified through test, design, or

statistical correlation. Typical values represent the most likely parametric norm at TJ= 25°C, and are provided for reference

purposes only.PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

Input offset current –1.5 1.5 µA

TJ= 25°C 0

Input offset voltage –7 7 mV

TJ= 25°C 0

gM(BW) Transconductance bandwidth ICSH = 100 µA(1), TJ= 25°C 250 500 kHz

GATE DRIVER (GATE)

RSRC(GATE) GATE sourcing resistance GATE = High 6 Ω

GATE = High, TJ= 25°C 2

RSNK(GATE) GATE sinking resistance GATE = Low 4.5 Ω

GATE = Low, TJ= 25°C 1.3

DIM DRIVER (DIM, DDRV)

nDIMVTH nDIM / UVLO threshold 1.185 1.285 V

TJ= 25°C 1.24

nDIMHYS nDIM hysteresis current 20 25 µA

TJ= 25°C 23

RSRC(DDRV) DDRV sourcing resistance DDRV = High 30 Ω

DDRV = High, TJ= 25°C 13.5

RSNK(DDRV) DDRV sinking resistance DDRV = Low 10 Ω

DDRV = Low, TJ= 25°C 3.5

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

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Product Folder Links: LM3421 LM3423

Electrical Characteristics (continued)

VIN = 14, −40°C ≤TJ≤125°C unless otherwise specified. Minimum and maximum limits are specified through test, design, or

statistical correlation. Typical values represent the most likely parametric norm at TJ= 25°C, and are provided for reference

purposes only.PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

PULLDOWN N-CHANNEL MOSFETS

RRPD RPD pulldown resistance 300 Ω

TJ= 25°C 145

RFLT FLT pulldown resistance 300 Ω

TJ= 25°C 145

RLRDY LRDY pulldown resistance 300 Ω

TJ= 25°C 135

THERMAL SHUTDOWN

TSD Thermal shutdown threshold(1) TJ= 25°C 165 °C

THYS Thermal shutdown

hysteresis(1) TJ= 25°C 25 °C

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

7.6 Typical Characteristics

TA= 25°C, VIN = 14 V unless otherwise specified

VO= 32 V (9 LEDs)

Figure 1. Boost Efficiency vs. Input Voltage VO= 21 V (6 LEDs)

Figure 2. Buck-Boost Efficiency vs. Input Voltage

VO= 32V (9 LEDs)

Figure 3. Boost LED Current vs. Input Voltage VO= 21 V (6 LEDs)

Figure 4. Buck-Boost LED Current vs. Input Voltage

VO= 21 V (6 LEDs), VIN = 24 V

Figure 5. Analog Dimming VO= 32 V (9 LEDs), VIN = 24 V

Figure 6. PWM Dimming

TEMPERATURE (°C)

tON-MIN (ns)

225

220

215

210

205

200

195

-50 -14 22 58 94 130

TEMPERATURE (°C)

VRCT (mV)

567

566

565

564

563

562

-50 -14 22 58 94 130

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

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Product Folder Links: LM3421 LM3423

Typical Characteristics (continued)

TA= 25°C, VIN = 14 V unless otherwise specified

Figure 7. VCSH vs Junction Temperature Figure 8. VCC vs Junction Temperature

Figure 9. VRCT vs Junction Temperature Figure 10. VLIM vs Junction Temperature

Figure 11. tON(min) vs Junction Temperature

AGND

PWM

Start new on time

nDIM

COMP

REFERENCE

CURRENT

LIMIT

CSH

HSN

LOGIC

HSP

OVP

HYSTERESIS

LED CURRENT LOW

UVLO

DDRV

GATE

PGND

OVP

LRDY

RPD

LatchOff TIMR

OVLO

FLT

LED CURRENT HIGH

Regulator

Thermal

Limit

TLIM

OVLO

PGND

LatchOff

Standby

Dimming

Reset

Dominant

RCT

800 mV

STOP

820k

500k

LEB

In the LM3421, TIMR is internally shorted to AGND.

1.235V

6.9V LDO

(4.1V)

DPOL

Grey pins are available in the LM3423 only.

LEB

W = 150 ns

VIN/25

VIN UVLO

HYSTERESIS

1.24V

VCC

1.235V

VCC

1.24V

0.245V

1.0V

1.6V

VCC UVLO

VCC

VCC UVLO

VIN

23 PA

11.5 PA

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

8 Detailed Description

8.1 Overview

The LM3421 and LM3423 are N-channel MOSFET ( N-channel FET ) controllers for buck, boost and buck-boost

current regulators which are ideal for driving LED loads. The controller has wide input voltage range allowing for

regulation of a variety of LED loads. The high-side differential current sense, with low adjustable threshold

voltage, provides an excellent method for regulating output current while maintaining high system efficiency.

The devices use a Predictive Off-time (PRO) control architecture that allows the regulator to be operated using

minimal external control loop compensation, while providing an inherent cycle-by-cycle current limit. The

adjustable current sense threshold provides the capability to amplitude (analog) dim the LED current and the

output enable and disable function with external dimming FET driver allows for fast PWM dimming of the LED

load. The maximum attainable LED current is not internally limited because the device is a controller. Instead,

current is a function of the system operating point, component choices, and switching frequency that allows the

device to easily provide constant currents up to 5 A. This controller contains all the features necessary to

implement a high-efficiency versatile LED driver.

8.2 Functional Block Diagram

D=INO VV +

D= INO VV -

D= O

iL (t)

ÂiL-PP

IL-MAX

IL-MIN

tON = DTStOFF = (1-D)TS

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

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Product Folder Links: LM3421 LM3423

8.3 Feature Description

8.3.1 Current Regulators

Figure 12. Ideal CCM Regulator Inductor Current iL(t)

Current regulators can create three basic topologies: buck, boost, or buck-boost. All three topologies in their

most basic form contain a main switching MOSFET, a recirculating diode, an inductor and capacitors. The

controller is designed to drive a ground referenced N-channel FET which is perfect for a standard boost

regulator. However, buck and buck-boost regulators usually have a high-side switch. When driving an LED load,

a ground referenced load is often not necessary, therefore a ground referenced switch drives a floating load

instead. The controller can then be used to drive all three basic topologies as shown in the Basic Topology

Schematics section. Other topologies such as the SEPIC and flyback converter (both derivatives of the buck-

boost) can be implemented as well.

Looking at the buck-boost design, the basic operation of a current regulator can be analyzed. During the time

that the N-channel FET (Q1) is turned on (tON), the input voltage source stores energy in the inductor (L1) while

the output capacitor (CO) provides energy to the LED load. When Q1 is turned off (tOFF), the re-circulating diode

(D1) becomes forward biased and L1 provides energy to both COand the LED load. Figure 12 shows the

inductor current (iL(t)) waveform for a regulator operating in CCM.

The average output LED current (ILED) is proportional to the average inductor current (IL) , therefore if ILis tightly

controlled, ILED is well regulated. As the system changes input voltage or output voltage, the ideal duty cycle (D)

is varied to regulate ILand ultimately ILED. For any current regulator, D is a function of the conversion ratio:

Buck

(1)

Boost

(2)

Buck-boost

(3)

8.3.2 Predictive Off-Time (PRO) Control

PRO control is used by the device to control ILED. It is a combination of average peak current control and a one-

shot off-timer that varies with input voltage. The LM3421 and LM3423 use peak current control to regulate the

average LED current through an array of HBLEDs. This method of control uses a series resistor in the LED path

to sense LED current and can use either a series resistor in the MOSFET path or the MOSFET RDS-ON for both

cycle-by-cycle current limit and input voltage feed forward. D is indirectly controlled by changes in both tOFF and

tON, which vary depending on the operating point.

VSNS = 1.24V x RCSH

RHSP

ICSH = RHSP

VSNS

RSNS

ILED RHSP

RHSN HSN

HSP High-Side

Sense Amplifier

CSH 1.24V

CCMP

RCSH

COMP

Error Amplifier

VSNS

To PWM

Comparator

LM3421/23

ICSH

LM3421

LM3423

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Feature Description (continued)

Even though the off-time control is quasi-hysteretic, the input voltage proportionality in the off-timer creates an

essentially constant switching frequency over the entire operating range for boost and buck-boost topologies.

The buck topology can be designed to give constant ripple over either input voltage or output voltage, however

switching frequency is only constant at a specific operating point .

This type of control minimizes the control loop compensation necessary in many switching regulators, simplifying

the design process. The averaging mechanism in the peak detection control loop provides extremely accurate

LED current regulation over the entire operating range.

PRO control was designed to mitigate current mode instability (also called sub-harmonic oscillation) found in

standard peak current mode control when operating near or above 50% duty cycles. When using standard peak

current mode control with a fixed switching frequency, this condition is present, regardless of the topology.

However, using a constant off-time approach, current mode instability cannot occur, enabling easier design and

control.

Predictive off-time advantages:

• There is no current mode instability at any duty cycle.

• Higher duty cycles or voltage transformation ratios are possible, especially in the boost regulator.

The only disadvantage is that synchronization to an external reference frequency is generally not available.

8.3.3 Average LED Current

Figure 13. LED Current Sense Circuitry

The LM3421 and LM3423 use an external current sense resistor (RSNS) placed in series with the LED load to

convert the LED current (ILED) into a voltage (VSNS) as shown in Figure 13. The HSP and HSN pins are the inputs

to the high-side sense amplifier which are forced to be equal potential (VHSP=VHSN) through negative feedback.

Because of this, the VSNS voltage is forced across RHSP to generate the signal current (ICSH) which flows out of

the CSH pin and through the RCSH resistor. The error amplifier regulates the CSH pin to 1.24 V, therefore ICSH

can be calculated using Equation 4.

(4)

This application regulates VSNS as described in Equation 5.

(5)

Calculate ILED using Equation 6.

CSH

RCSH

LM3421/23

VCC

RBIAS

RMAX

RADJ

Variable Current Source

Variable

Resistance

VREF

VO = 76V - VIN

VHSP < 76V

VHSN > 3.5V

ILED = RSNS

1.24V

RSNS

VSNS RCSH

RHSP

= x

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

Feature Description (continued)

(6)

The selection of the three resistors (RSNS, RCSH, and RHSP) is not arbitrary. For matching and noise performance,

the suggested signal current ICSH is approximately 100 µA. This current does not flow in the LEDs and does not

affect either the off-state LED current or the regulated LED current. ICSH can be above or below this value, but

the high-side amplifier offset characteristics may be affected slightly. In addition, to minimize the effect of the

high-side amplifier voltage offset on LED current accuracy, the minimum VSNS is suggested to be 50 mV. Place a

resistor (RHSN = RHSP) in series with the HSN pin to cancel out the effects of the input bias current (approximately

10 µA) of both inputs of the high-side sense amplifier.

The sense resistor (RSNS) can be placed anywhere in the series string of LEDs as long as the voltage at the HSN

and HSP pins (VHSP and VHSN) satisfies the following conditions.

(7)

Typically, for a buck-boost configuration, RSNS is placed at the bottom of the string (LED-) which allows for

greater flexibility of input and output voltage. However, if there is substantial input voltage ripple allowed, it can

help to place RSNS at the top of the string (LED+) which limits the output voltage of the string to:

(8)

The CSH pin can also be used as a low-side current sense input regulated to 1.24 V. The high-side sense

amplifier is disabled if HSP and HSN are tied to AGND (or VHSN > VHSP) .

8.3.4 Analog Dimming

The CSH pin can be used to analog dim the LED current by adjusting the current sense voltage (VSNS). There

are several different methods to adjust VSNS using the CSH pin:

1. External variable resistance: Adjust a potentiometer placed in series with RCSH to vary VSNS.

2. External variable current source: Source current (0 µA to ICSH) into the CSH pin to adjust VSNS.

Figure 14. Analog Dimming Circuitry

In general, analog dimming applications require a lower switching frequency to minimize the effect of the leading

edge blanking circuit. As the LED current is reduced, the output voltage and the duty cycle decreases.

Eventually, the minimum on-time is reached. The lower the switching frequency, the wider the linear dimming

range. Figure 14 shows how both CSH methods are physically implemented.

Method 1 uses an external potentiometer in the CSH path which is a simple addition to the existing circuitry.

However, the LEDs cannot dim completely because there is always some resistance causing signal current to

flow. This method is also susceptible to noise coupling at the CSH pin because the potentiometer increases the

size of the signal current loop.

ILIM = 245 mV

RLIM

LM3421/23

PWM

COMP

RLIM

Q1 GATE

LEB

PGND

0.245V

0.8V

RDS-ON

Sensing

RLIM

Sensing

ILED = (ICSH - IADD) x RHSP

RSNS¸

RADJ x VREF

IADD = RADJ + RMAX - VBE-Q6

RBIAS

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Feature Description (continued)

Method 2 provides a complete dimming range and better noise performance, though it is more complex. It

consists of a PNP current mirror and a bias network consisting of an NPN, 2 resistors and a potentiometer

(RADJ), where RADJ controls the amount of current sourced into the CSH pin. A higher resistance value sources

more current into the CSH pin, causing less regulated signal current through RHSP, effectively dimming the LEDs.

VREF should be a precise external voltage reference, while Q7 and Q8 should be a dual pair PNP for best

matching and performance. The additional current (IADD) sourced into the CSH pin can be calculated using

Equation 9.

(9)

The corresponding LED current ( ILED) for a specific IADD is:

(10)

8.3.5 Current Sense and Current Limit

The LM3421 and LM3423 achieve peak current mode control using a comparator that monitors the main

MOSFET (Q1) transistor current, comparing it with the COMP pin voltage as shown in Figure 15. The controller

incorporates a cycle-by-cycle overcurrent protection function. Aredundant internal current sense comparator

provides the current limit functionality . If the voltage at the current sense comparator input (IS pin) exceeds 245

mV (typical), the on cycle is immediately terminated. The IS input pin has an internal N-channel MOSFET which

pulls it down at the conclusion of every cycle. The discharge device remains on for an additional 210 ns (typical)

after the beginning of a new cycle to blank the leading edge spike on the current sense signal. The leading edge

blanking (LEB) determines the minimum achievable on-time (tON-MIN).

Figure 15. Current Sense / Current Limit Circuitry

There are two possible methods to sense the transistor current. The RDS-ON of the main power MOSFET can be

used as the current sense resistance because the IS pin was designed to withstand the high voltages present on

the drain when the MOSFET is in the off state. Alternatively, a sense resistor located in the source of the

MOSFET may be used for current sensing; however, TI suggests a low inductance (ESL) type. The cycle-by-

cycle current limit (ILIM) can be calculated using either method as the limiting resistance (RLIM):

(11)

1P =Z1+D

OD Cr x

T=SNSCSH RR500VD xxx

c620VD x

( ) LIM

LED RID1 xx+

( ) LIMHSP RRD1 xx+

§+s

1Z1P ¸

T¨

§-s

1Z1Z ¸

LM3421/23

VIN

nDIM

VIN

RPD

OVP

Enable

RUV2

RUV1

ROV2

ROV1

L1 D1

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

Feature Description (continued)

8.3.6 Overcurrent Protection

The LM3421 and LM3423 controllers have a secondary method of overcurrent protection. Switching action is

disabled whenever the current in the LEDs is more than 30% above the regulation set point. The dimming

MOSFET switch driver (DDRV) is not disabled however as this would immediately remove the fault condition and

cause oscillatory behavior.

8.3.7 Zero Current Shutdown

The LM3421 and LM3423 controllers implement zero current shutdown through the EN and RPD pins. When

pulled low, the EN pin places the devices into near-zero current state, where only the leakage currents occurs at

the pins (typical 0.1 µA). The applications circuits frequently have resistor dividers to set UVLO, OVLO, or other

similar functions. The RPD pin is an open-drain N-channel MOSFET that is enabled only when the device is

enabled. Tying the bottom of all resistor dividers to the RPD pin as shown in Figure 16 allows them to float

during shutdown, thus removing their current paths and providing true application-wide zero current shutdown.

Figure 16. Zero Current Shutdown Circuit

8.3.8 Control Loop Compensation

The control loop is modeled as most typical current mode controllers. Using a first order approximation, the

uncompensated loop can be modeled as a single pole created by the output capacitor and, in the boost and

buck-boost topologies, a right half plane zero created by the inductor, where both have a dependence on the

LED string dynamic resistance. There is also a high-frequency pole in the model; however, it is near the

switching frequency and plays no part in the compensation design process. Therefore, it is neglected. Because

ceramic capacitance is recommended for use with LED drivers, due to long lifetimes and high ripple current

rating, the ESR of the output capacitor can also be neglected in the loop analysis. The DC gain of the

uncompensated loop depends on internal controller gains and the external sensing network.

This section describes a buck-boost regulator as an example case.

Use Equation 12 to calculate the uncompensated loop gain for a buck-boost regulator.

(12)

Where the uncompensated DC loop gain of the system is calculated using Equation 13.

(13)

And the output pole (ωP1) is approximated using Equation 14.

(14)

RSNS

ILED RHSP

RHSN HSN

HSP High-Side

Sense Amplifier

CSH 1.24V

CCMP

RCSH

COMP

Error Amplifier

VSNS

To PWM

Comparator

LM3421/23

CFS

RFS

sets öP3

sets öP2

PHASE (°)

FREQUENCY (Hz)

GAIN (dB)

100

-20

-40

-60

135

-45

-90

-135

-180

-225

1e-1 1e1 1e3 1e5 1e7

0° Phase Margin

öP1

PHASE

GAIN

öZ1

=Dr 2

ZL1Dx

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Feature Description (continued)

And the right half plane zero (ωZ1) is:

(15)

Figure 17. Uncompensated Loop Gain Frequency Response

Figure 17 shows the uncompensated loop gain in a worst-case scenario when the RHP zero is below the output

pole. This occurs at high duty cycles when the regulator is trying to boost the output voltage significantly. The

RHP zero adds 20dB/decade of gain while losing 45°/decade of phase, which places the crossover frequency

(when the gain is zero dB) extremely high because the gain only starts falling again due to the high-frequency

pole (not shown in Figure 17). The phase is below –180° at the crossover frequency, which means there is no

phase margin (180° + phase at crossover frequency) causing system instability. Even if the output pole is below

the RHP zero, the phase reaches –180° before the crossover frequency in most cases yielding instability.

Figure 18. Compensation Circuitry

PHASE (°)

FREQUENCY (Hz)

GAIN (dB)

-20

-40

-60

-80

-45

-90

-135

-180

-225

-270

1e-1 1e1 1e3 1e5 1e7

GAIN

60° Phase Margin

PHASE

öP2

öP3

öP1

öZ1

x= 0U

TT -1 ¸

§s

xx +1 ¸

§s

Z3P

+1 ¸

§s

Z2P

+1 ¸

§s

Z1P

3P =Z

FSFS CR x

P2 6CMP

5 10 C

u u

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

Feature Description (continued)

To mitigate this problem, a compensator should be designed to give adequate phase margin (above 45°) at the

crossover frequency. A simple compensator using a single capacitor at the COMP pin (CCMP) adds a dominant

pole to the system, which ensures adequate phase margin if placed low enough. At high duty cycles (as shown

in Figure 17), the RHP zero places extreme limits on the achievable bandwidth with this type of compensation.

However, because an LED driver is essentially free of output transients (except catastrophic failures open or

short), the dominant pole approach, even with reduced bandwidth, is usually the best approach. The dominant

compensation pole (ωP2) is determined by CCMP and the output resistance (RO) of the error amplifier (typically 5

MΩ) as demonstrated in Equation 16.

(16)

It may also be necessary to add one final pole at least one decade above the crossover frequency to attenuate

switching noise and, in some cases, provide better gain margin. This pole can be placed across RSNS to filter the

ESL of the sense resistor at the same time. Figure 18 shows how the compensation is physically implemented in

the system.

The high-frequency pole (ωP3) can be calculated using Equation 17.

(17)

The total system transfer function becomes:

(18)

The resulting compensated loop gain frequency response shown in Figure 19 indicates that the system has

adequate phase margin (above 45°) if the dominant compensation pole is placed low enough, ensuring stability.

Figure 19. Compensated Loop Gain Frequency Response

8.3.9 Start-Up Regulator

The controller includes a high voltage, low dropout bias regulator. When power is applied, the regulator is

enabled and sources current into an external capacitor (CBYP) connected to the VCC pin. The recommended

bypass capacitance for the VCC regulator is 2.2 µF to 3.3 µF. The output of the VCC regulator is monitored by an

internal UVLO circuit that protects the device from attempting to operate with insufficient supply voltage and the

supply is also internally current limited. Figure 20 shows the typical start-up waveforms.

-OFFTURN 24V.1V x+

OV1 OV2

R5.0 R

1OV

1¸

-OFFTURN 24V.1V 1OV

R+2OVOV RR

COCMPVCCSU tttt ++=

OCO Ct x

LED

CMPCMP

CMP C36Ct x

=PA

0.9V

BYPBYPVCC C168Ct x:

=mA25 V2.4

VCMP

0tVCC tCMP tCO t

0.9V

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Feature Description (continued)

Figure 20. Start-Up Waveforms

First, CBYP is charged to be above VCC UVLO threshold (approximately 4.2 V). The CVCC charging time (tVCC) can

be estimated using Equation 19.

(19)

CCMP is then charged to 0.9 V over the charging time (tCMP), which can be estimated using Equation 20.

(20)

Once CCMP = 0.9 V, the part starts switching to charge COuntil the LED current is in regulation. The COcharging

time (tCO) can be roughly estimated using Equation 21.

(21)

The system start-up time (tSU) is defined using Equation 22.

(22)

In some configurations, the start-up waveform overshoots the steady state COMP pin voltage. In this case, the

LED current and output voltage overshoots also, which can trip the overvoltage or protection, causing a race

condition. The easiest way to prevent this is to use a larger compensation capacitor (CCMP), thereby slowing

down the control loop.

8.3.10 Overvoltage Lockout (OVLO)

The LM3421 and LM3423 can be configured to detect an output (or input) overvoltage condition through the OVP

pin. The pin features a precision 1.24-V threshold with 23 µA (typical) of hysteresis current as shown in

Figure 21. When the OVLO threshold is exceeded, the GATE pin is immediately pulled low and a 23-µA current

source provides hysteresis to the lower threshold of the OVLO hysteretic band.

If the LEDs are referenced to a potential other than ground (floating), as in the buck-boost and buck

configuration, the output voltage (VO) should be sensed and translated to ground by using a single PNP as

shown in Figure 22.

The overvoltage turnoff threshold (VTURN-OFF) is defined:

Ground Referenced

(23)

Floating

(24)

LM3421/23

OVP

ROV2

ROV1

LED+

LED-

1.24V

23 PA

LM3421/23

ROV2

ROV1

OVLO

OVP

2OVHYSO RA23V xP=

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

Feature Description (continued)

In the ground referenced configuration, the voltage across ROV2 is VO– 1.24 V whereas in the floating

configuration it is VO– 620 mV where 620 mV approximates VBE of the PNP.

The overvoltage hysteresis (VHYSO) is defined using Equation 25.

(25)

Figure 21. Overvoltage Protection Circuitry

Figure 22. Floating Output OVP Circuitry

The OVLO feature can cause some interesting results if the OVLO trip-point is set too close to VO. At turnon, the

converter has a modest amount of voltage overshoot before the control loop gains control of ILED. If the overshoot

exceeds the OVLO threshold, the controller shuts down, opening the dimming MOSFET. This isolates the LED

load from the converter and the output capacitance. The voltage then discharges very slowly through the HSP

and HSN pins until VOdrops below the lower threshold, where the process repeats. This looks like the LEDs are

blinking at around 2 Hz. This mode can be escaped if the input voltage is reduced.

8.3.11 Input Undervoltage Lockout (UVLO)

The nDIM pin is a dual-function input that features an accurate 1.24-V threshold with programmable hysteresis

as shown in Figure 23. This pin functions as both the PWM dimming input for the LEDs and as a VIN UVLO.

When the pin voltage rises and exceeds the 1.24-V threshold, 23 µA (typical) of current is driven out of the nDIM

pin into the resistor divider providing programmable hysteresis.

PA23

HYS

V¨

§+

2UV

R¸

1UV

x1UV

2UV

UVH

2UV

RA23 x

HYS

1¸

-ONTURN 24V.1V 1UV

R+2UVUV RR

1.24V

23 PA

LM3421/23

RUV2

RUV1

VIN

UVLO

nDIM

RUVH

(optional)

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Feature Description (continued)

Figure 23. UVLO Circuit

When using the nDIM pin for UVLO and PWM dimming concurrently, the UVLO circuit can have an extra series

resistor to set the hysteresis. This allows the standard resistor divider to have smaller resistor values minimizing

PWM delays due to a pulldown MOSFET at the nDIM pin (see PWM Dimming section). In general, at least 3 V of

hysteresis is preferable when PWM dimming, if operating near the UVLO threshold.

The turnon threshold (VTURN-ON) is defined using Equation 26.

(26)

The hysteresis (VHYS) is defined as follows:

8.3.11.1 UVLO Only

(27)

8.3.11.2 PWM Dimming and UVLO

(28)

When zero current shutdown and UVLO are implemented together, the EN pin can be used to escape UVLO.

The nDIM pin pulls up to VIN when EN is pulled low. Therefore, if VIN is within the UVLO hysteretic window when

EN is pulled high again, the controller starts-up even though VTURN-ON is not exceeded.

8.3.12 PWM Dimming

The active low nDIM pin can be driven with a PWM signal which controls the main N-channel FET and the

dimming FET (dimFET). The brightness of the LEDs can be varied by modulating the duty cycle of this signal.

LED brightness is approximately proportional to the PWM signal duty cycle, (that is, 30% duty cycle equals

approximately 30% LED brightness). This function can be ignored if PWM dimming is not required by using nDIM

solely as a VIN UVLO input as described in Input Undervoltage Lockout (UVLO) or by tying it directly to VCC or

VIN.

tPULSE = 2 x ILED x VO X L1

VIN2

LM3421/23

RUV2

RUV1

VIN

nDIM

RUVH

QDIM

DDIM

Inverted

PWM

Standard

PWM

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

Feature Description (continued)

Figure 24. PWM Dimming Circuit

STOPPED DD EDITING HERELM3421 and LM3423

Figure 24 shows how the PWM signal is applied to nDIM:

1. Connect the dimming MOSFET (QDIM) with the drain to the nDIM pin and the source to AGND. Apply an

external logic-level PWM signal to the gate of QDIM.

2. Connect the anode of a Schottky diode (DDIM) to the nDIM pin. Apply an inverted external logic-level PWM

signal to the cathode of the same diode.

The DDRV pin is a PWM output that follows the nDIM PWM input signal. When the nDIM pin rises, the DDRV pin

rises and the PWM latch reset signal is removed allowing the main MOSFET Q1 to turn on at the beginning of

the next clock set pulse. In boost and buck-boost topologies, the DDRV pin is used to control a N-channel

MOSFET placed in series with the LED load, while it would control a P-channel MOSFET in parallel with the load

for a buck topology.

The series dimFET opens the LED load, when nDIM is low, effectively speeding up the rise and fall times of the

LED current. Without any dimFET, the rise and fall times are limited by the inductor slew rate and dimming

frequencies above 1 kHz are impractical. Using the series dimFET, dimming frequencies up to 30 kHz are

achievable. With a parallel dimFET (buck topology), even higher dimming frequencies are achievable.

When using the PWM functionality in a boost regulator, the PWM signal can drive a ground referenced FET.

However, with buck-boost and buck topologies, level shifting circuitry is necessary to translate the PWM dim

signal to the floating dimFET as shown in Figure 25 and Figure 26. If high side dimming is necessary in a boost

regulator using the LM3423, level shifting can be added providing the polarity inverting DPOL pin is pulled low

(see LM3423 Only: DPOL, FLT, TIMR, and LRDY section) as shown in Figure 27.

When using a series dimFET to PWM dim the LED current, more output capacitance is always better. Typical

applications use a minimum of 40 µF for PWM dimming. For most applications, a capacitance of 40 µF provides

adequate energy storage at the output when the dimFET turns off and opens the LED load. Then when the

dimFET is turned back on, the capacitance helps source current into the load, improving the LED current rise

time.

A minimum on-time must be maintained in order for PWM dimming to operate in the linear region of its transfer

function. Because the controller is disabled during dimming, the PWM pulse must be long enough such that the

energy intercepted from the input is greater than or equal to the energy being put into the LEDs. For boost and

buck-boost regulators, the minimum dimming pulse length in seconds (tPULSE) is:

(29)

Even maintaining a dimming pulse greater than tPULSE, preserving linearity at low dimming duty cycles is difficult.

RSNS

10V

10 k:

LM3421/23

DDRV

VCC

100 pF

100

DPOL

RSNS

LM3421/23

DDRV

100 nF

100

10V

RSNS

LED+

Q6 Q4

10V VIN

5 k:

LM3421/23

500:

DDRV

VCC

100 pF

100 nF

10:

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Feature Description (continued)

The second helpful modification is to remove the CFS capacitor and RFS resistor, eliminating the high-frequency

compensation pole. Typically, this does not affect stability, but it speeds up the response of the CSH pin,

specifically at the rising edge of the LED current when PWM dimming, thus improving the achievable linearity at

low dimming duty cycles.

Figure 25. Buck-boost Level-Shifted PWM Circuit

Figure 26. Buck Level-Shifted PWM Circuit

Figure 27. Boost Level-Shifted PWM Circuit

TMR

C=11.5

FLT

txPA

V1.24

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

Feature Description (continued)

8.3.13 LM3423 Only: DPOL, FLT, TIMR, and LRDY

The LM3423 has four additional pins: DPOL, FLT, TIMR, and LRDY. The DPOL pin is simply used to invert the

DDRV polarity . If DPOL is left open, then it is internally pulled high and the polarity is correct for driving a series

N-channel dimFET. If DPOL is pulled low then the polarity is correct for using a series P-channel dimFET in high-

side dimming applications. For a parallel P-channel dimFET, as used in the buck topology, leave DPOL open for

proper polarity.

The additional TIMR and FLT pins can be used in conjunction with an input disconnect MOSFET switch as

shown in Figure 28 to protect the module from various fault conditions.

A fault is detected and an 11.5 µA (typical) current is sourced from the TIMR pin whenever any one of the

following conditions exists.

• LED current is above regulation by more than 30%.

• OVLO has engaged.

• Thermal shutdown has engaged.

An external capacitor (CTMR) from TIMR to AGND programs the fault filter time as follows:

(30)

When the voltage on the TIMR pin reaches 1.24 V, the device is latched off and the N-channel MOSFET open-

drain FLT pin transitions to a high impedance state. The controller immediatly pulls the TIMR pin to ground

(resets) if the fault condition is removed at any point during the filter period. Otherwise, if the timer expires, the

fault remains latched until one of these situations occurs:

• The EN pin is pulled low long enough for the VCC pin to drop below 4.1 V (approximately 200 ms) or

• the TIMR pin is pulled to ground or

• a complete power cycle occurs

When using the EN and OVP pins in conjunction with the RPD pulldown pin, a race condition exists when exiting

the disabled (EN low) state. When disabled, controller pulls up the OVP pin to the output voltage because the

RPD pulldown is disabled, and this appears as if it is a real OVLO condition. The timer pin immediately rises and

latches the controller to the fault state. To protect against this behavior, a minimum timer capacitor (CTMR = 220

pF) should be used. If fault latching is not required, short the TMR pin to AGND, which disables the FLT flag

function.

The LM3423 also includes an LED Ready (LRDY) flag to notify the system that the LEDs are in proper

regulation. The N-channel MOSFET open-drain LRDY pin is pulled low whenever any of the following conditions

are met:

1. VCC UVLO has engaged.

2. LED current is below regulation by more than 20%.

3. LED current is above regulation by more than 30%.

4. Overvoltage protection has engaged

5. Thermal shutdown has engaged.

6. A fault has latched the device off.

The LRDY pin is pulled low during start-up of the device and remains low until the LED current is in regulation.

LM3421/23

VIN

FLT

TIMR

LRDY

VIN

VSW

High = LED in regulation

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Feature Description (continued)

Figure 28. Fault Detection and LED Status Circuit

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component

specification, and TI does not warrant its accuracy or completeness. TI’s customers are

responsible for determining suitability of components for their purposes. Customers should

validate and test their design implementation to confirm system functionality.

9.1 Application Information

9.1.1 Inductor

The inductor (L1) is the main energy storage device in a switching regulator. Depending on the topology, energy

is stored in the inductor and transfered to the load in different ways (as an example, buck-boost operation is

detailed in the Current Regulators section). The size of the inductor, the voltage across it, and the length of the

switching subinterval (tON or tOFF) determines the inductor current ripple (ΔiL-PP). In the design process, L1 is

chosen to provide a desired ΔiL-PP. For a buck regulator the inductor has a direct connection to the load, which is

good for a current regulator. This requires little to no output capacitance therefore ΔiL-PP is basically equal to the

LED ripple current ΔiLED-PP. However, for boost and buck-boost regulators, there is always an output capacitor

which reduces ΔiLED-PP; therefore, the inductor ripple can be larger than in the buck regulator case where output

capacitance is minimal or completely absent.

In general, ΔiLED-PP is recommended by manufacturers to be less than 40% of the average LED current (ILED).

Therefore, for the buck regulator with no output capacitance, ΔiL-PP should also be less than 40% of ILED. For the

boost and buck-boost topologies, ΔiL-PP can be much higher depending on the output capacitance value.

However, ΔiL-PP is suggested to be less than 100% of the average inductor current (IL) to limit the RMS inductor

current.

L1 is also suggested to have an RMS current rating at least 25% higher than the calculated minimum allowable

RMS inductor current (IL-RMS).

9.1.2 LED Dynamic Resistance

When the load is a string of LEDs, the output load resistance is the LED string dynamic resistance plus RSNS.

LEDs are PN junction diodes, and their dynamic resistance shifts as their forward current changes. Dividing the

forward voltage of a single LED (VLED) by the forward current (ILED) leads to an incorrect calculation of the

dynamic resistance of a single LED (rLED). The result can be 5 to 10 times higher than the true rLED value.

Figure 29. Dynamic Resistance

Obtaining rLED is accomplished by referring to the manufacturer's LED I-V characteristic. It can be calculated as

the slope at the nominal operating point as shown in Figure 29. For any application with more than 2 series

LEDs, RSNS can be neglected allowing rDto be approximated as the number of LEDs multiplied by rLED.

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9.1.3 Output Capacitor

For boost and buck-boost regulators, the output capacitor (CO) provides energy to the load when the recirculating

diode (D1) is reverse biased during the first switching subinterval. An output capacitor in a buck topology simplys

reduce the LED current ripple (ΔiLED-PP) below the inductor current ripple (ΔiL-PP). In all cases, COis sized to

provide a desired ΔiLED-PP. As mentioned in the Inductor section, ΔiLED-PP is recommended by manufacturers to be

less than 40% of the average LED current (ILED).

COshould be carefully chosen to account for derating due to temperature and operating voltage. It must also

have the necessary RMS current rating. Ceramic capacitors are the best choice due to their high ripple current

rating, long lifetime, and good temperature performance. An X7R dieletric rating is suggested.

9.1.4 Input Capacitors

The input capacitance (CIN) provides energy during the discontinuous portions of the switching period. For buck

and buck-boost regulators, CIN provides energy during tON and during tOFF, the input voltage source charges up

CIN with the average input current (IIN). For boost regulators, CIN only needs to provide the ripple current due to

the direct connection to the inductor. CIN is selected given the maximum input voltage ripple (ΔvIN-PP) which can

be tolerated. ΔvIN-PP is suggested to be less than 10% of the input voltage (VIN).

An input capacitance at least 100% greater than the calculated CIN value is recommended to account for derating

due to temperature and operating voltage. When PWM dimming, even more capacitance can be helpful to

minimize the large current draw from the input voltage source during the rising transition of the LED current

waveform.

The chosen input capacitors must also have the necessary RMS current rating. Ceramic capacitors are again the

best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R

dielectric rating is suggested.

For most applications, TI recommends bypassing the VIN pin with an 0.1 µF ceramic capacitor placed as close as

possible to the pin. In situations where the bulk input capacitance may be far from the controller, a 10-Ωseries

resistor can be placed between the bulk input capacitance and the bypass capacitor, creating a 150-kHz filter to

eliminate undesired high-frequency noise.

9.1.5 Main MOSFET / Dimming MOSFET

The controller requires an external N-channel FET (Q1) as the main power MOSFET for the switching regulator.

TI recommends Q1 have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe

operation during the ringing of the switch node. In practice, all switching regulators have some ringing at the

switch node due to the diode parasitic capacitance and the lead inductance. TI recommends the current rating be

at least 10% higher than the average transistor current. The power rating is then verified by calculating the power

loss given the RMS transistor current and the N-channel FET on-resistance (RDS-ON).

When PWM dimming, the controller requires another MOSFET (Q2) placed in series (or parallel for a buck

regulator) with the LED load. This MOSFET should have a voltage rating greater than the output voltage (VO)

and a current rating at least 10% higher than the nominal LED current (ILED) . The power rating is simply RDS-ON

multiplied by ILED, assuming 100% dimming duty cycle (continuous operation) occurs.

For most applications, choose an N-channel FET that minimizes total gate charge (Qg) when fSW is high. It that is

not possible. minimize the on-resistance RDS(on) to minimize the dominant power losses in the system.

Frequently, higher current N-channel FETs in larger packages yield better thermal performance.

9.1.6 Re-Circulating Diode

The controller requires a recirculating diode (D1) to carry the inductor current during the off time (tOFF). The most

efficient choice for D1 is a Schottky diode due to low forward voltage drop and near-zero reverse recovery time.

Similar to Q1, TI recommends D1 have a voltage rating at least 15% higher than the maximum transistor voltage

to ensure safe operation during the ringing of the switch node and a current rating at least 10% higher than the

average diode current. The power rating is verified by calculating the power loss through the diode. This is

accomplished by checking the typical diode forward voltage from the I-V curve on the product data sheet and

multiplying by the average diode current. In general, higher current diodes have a lower forward voltage and

come in better performing packages minimizing both power losses and temperature rise.

( )

OOIN VVV25 -

f=2

INTT VCR xx

fSW = 25 x VIN - VO

RT x CT X VIN

( )

VIN VO

L1 D1

Boost Inrush Diode

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Application Information (continued)

9.1.7 Boost Inrush Current

When configured as a boost converter, there is a phantom power path comprised of the inductor, the output

diode, and the output capacitor. This path causes two things to happen when power is applied:

1. a very large inrush of current to charge the output capacitor

2. the energy stored in the inductor during this inrush collects in the output capacitor, charging it to a higher

potential than the input voltage

Depending on the state of the EN pin, the output capacitor discharges by:

1. EN < 1.3 V: no discharge path (leakage only).

2. EN > 1.3 V, the OVP divider resistor path, if present, and 10 µA into each of the HSP & HSN pins.

In applications using the OVP divider and with EN > 1.3 V, the output capacitor voltage can charge higher than

VTURN-OFF. In this situation, the FLT pin (LM3423 only) is open and the PWM dimming MOSFET is turned off.

This condition (the system appearing disabled) can persist for an undesirably long time. Possible solutions to this

condition include:

• Add an inrush diode from VIN to the output as shown in Figure 30.

• Add an NTC thermistor in series with the input to prevent the inrush from overcharging the output capacitor

too high.

• Use a current limited source supply.

• Raise the OVP threshold.

Figure 30. Boost Topology with Inrush Diode

9.1.8 Switching Frequency

An external resistor (RT) connected between the RCT pin and the switch node (where D1, Q1, and L1 connect),

in combination with a capacitor (CT) between the RCT and AGND pins, sets the off-time (tOFF) as shown in

Figure 31. For boost and buck-boost topologies, the VIN proportionality ensures a virtually constant switching

frequency (fSW).

For a buck topology, RTand CTare also used to set tOFF, however the iinput voltage (VIN) proportionality does

not ensure a constant switching frequency. Instead, constant ripple operation can be achieved. Changing the

connection of RTin Figure 31 from VSW to VIN provides a constant ripple over varying VIN. Adding a PNP

transistor as shown in Figure 32 provides constant ripple over varying VO.

The switching frequency is defined:

Buck (Constant Ripple vs. VIN)

(31)

Buck (Constant Ripple vs. VO)

(32)

Boost and Buck-boost

LM3421/23

RCT

Start tON

VIN/25

Reset timer

RSNS

VIN

LED-

VSW

LM3421/23

RCT

Start tON

VIN/25

Reset timer

SW CR 25

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Application Information (continued)

(33)

For all topologies, the CTcapacitor is recommended to be 1 nF and should be located very close to the LM34xx-

Q1.

Figure 31. Off-timer Circuitry for Boost and Buck-

boost Regulators Figure 32. Off-timer Circuitry for Buck Regulators

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CIN

CBYP

RLIM

CCMP

RCSH

ROV2

ROV1

RUV2

RUV1

RHSN

RHSP

ILED

RSNS

RFS

CFS

COV

Q3 PWM

RUVH

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9.2 Typical Applications

9.2.1 Basic Topology Schematics

Figure 33. Boost Regulator (VIN < VO)

RLIM

ROV2

ROV1

ILED

RSNS

RFS

CFS

COV

L1 Q5

DIM

RPU

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CIN

CBYP

CCMP

RCSH

RUV2

RUV1

RHSN

RHSP

Q3 PWM

RUVH

CDIM

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Typical Applications (continued)

Figure 34. Buck Regulator (VIN > VO)

ROV2

ROV1

ILED

RSNS

RFS

CFS

COV

VIN

DIM

RPU

RSER

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

RLIM

CCMP

RCSH

RUV2

RUV1

RHSN

RHSP

Q3 PWM

RUVH

CIN

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Typical Applications (continued)

Figure 35. Buck-Boost Regulator

D=INO VV +

D= INO VV -

D= O

rD = N x rLED

VO = N x VLED

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Typical Applications (continued)

9.2.1.1 Design Requirements

Number of series LEDs: N

Single LED forward voltage: VLED

Single LED dynamic resistance: rLED

Nominal input voltage: VIN

Input voltage range: VIN-MAX, VIN-MIN

Switching frequency: fSW

Current sense voltage: VSNS

Average LED current: ILED

Inductor current ripple: ΔiL-PP

LED current ripple: ΔiLED-PP

Peak current limit: ILIM

Input voltage ripple: ΔvIN-PP

Output OVLO characteristics: VTURN-OFF, VHYSO

Input UVLO characteristics: VTURN-ON, VHYS

9.2.1.2 Detailed Design Procedure

9.2.1.2.1 Operating Point

Given the number of series LEDs (N), the forward voltage (VLED) and dynamic resistance (rLED) for a single LED,

solve for the nominal output voltage (VO) and the nominal LED string dynamic resistance (rD):

(34)

(35)

Solve for the ideal nominal duty cycle (D):

Buck:

(36)

Boost:

(37)

Buck-Boost:

(38)

Using the same equations, find the minimum duty cycle (DMIN) using maximum input voltage (VIN-MAX) and the

maximum duty cycle (DMAX) using the minimum input voltage (VIN-MIN). Also, remember that D' = 1 - D.

9.2.1.2.2 Switching Frequency

Set the switching frequency (fSW) by assuming a CTvalue of 1 nF and solving for RT:

Buck (Constant Ripple vs. VIN)

IL-RMS = 1 + 1

12 xILED

'IL-PP x D' ¸

§2

ILED x

IL-RMS = ILED x 1 + 1

12 xILED

'IL-PP

§2

L PP SW

V D

L1 i f



' u

IN O

L PP SW

(V V ) D

L1 i f



 u

' u

1.24V

RHSP =RRI SNSCSHLED xx

SNS

R = SNS

LED

T25

R = TSW Cf x

( )

T25

R=2

OOIN VVV -

xx 2

INTSW VCf xx

RT = 25 x VIN - VO

fSW x CT X VIN

( )

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Typical Applications (continued)

(39)

Buck (Constant Ripple vs. VO)

(40)

Boost and Buck-Boost

(41)

9.2.1.2.3 Average LED Current

For all topologies, set the average LED current (ILED) knowing the desired current sense voltage (VSNS) and

solving for RSNS:

(42)

If the calculated RSNS is too far from a desired standard value, then VSNS requires adjustment to obtain a

standard value.

Setup the suggested signal current of 100 µA by assuming RCSH = 12.4 kΩand solving for RHSP:

(43)

If the calculated RHSP is too far from a desired standard value, then RCSH can be adjusted to obtain a standard

value.

9.2.1.2.4 Inductor Ripple Current

Set the nominal inductor ripple current (ΔiL-PP) by solving for the appropriate inductor (L1):

Buck

(44)

Boost and Buck-Boost

(45)

To set the worst case inductor ripple current, use VIN-MAX and DMIN when solving for L1.

The minimum allowable inductor RMS current rating (IL-RMS) can be calculated as:

Buck

(46)

Boost and Buck-Boost

(47)

1P =Z1

OD Cr x

§+s

1Z1P ¸

T¨

§-s

1Z1Z ¸

§+s

1Z1P ¸

LIM

R=LIM

245 mV

1-DMAX

DMAX

ICO-RMS = ILED x

IRMSCO =

-12PP-LED

iü

LED

OD LED PP SW

I D

Cr i f



u ' u

C = PPL

PPLEDDSW irf8 -

'xxx

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Typical Applications (continued)

9.2.1.2.5 LED Ripple Current

Set the nominal LED ripple current (ΔiLED-PP), by solving for the output capacitance (CO):

Buck

(48)

Boost and Buck-boost

(49)

To set the worst case LED ripple current, use DMAX when solving for CO. Remember, when PWM dimming, TI

recommends using a minimum of 40 µF of output capacitance to improve performance.

The minimum allowable RMS output capacitor current rating (ICO-RMS) can be approximated:

Buck

(50)

Boost and Buck-boost

(51)

9.2.1.2.6 Peak Current Limit

Set the peak current limit (ILIM) by solving for the transistor path sense resistor (RLIM):

(52)

9.2.1.2.7 Loop Compensation

Using a simple first order peak current mode control model, neglecting any output capacitor ESR dynamics, the

necessary loop compensation can be determined.

First, the uncompensated loop gain (TU) of the regulator can be approximated:

Buck

(53)

Boost and Buck-Boost

(54)

Where the pole (ωP1) is approximated:

Buck

(55)

FS P3

C10

u Z

max

3P =

Z( ) 10, 1Z1P x

CMP 6

C5 10

Z u u

1Z1P )

,min( Z

5x0U

2P =

T=SNSCSH RR500VD xxx

c620VD x

( ) LIM

LED RID1 xx+

( ) LIMHSP RRD1 xx+

T=SNSCSH RR500VD xxx

c310VD x

LIMLED RI x

LIMHSP RR2 xx

SNS 620V

RR500V =

xx CSH

LIMLED RI x

T=LIMHSP RR x

=Dr 2

ZL1Dx

=Dr 2

ZL1

1P =Z1+D

OD Cr x

1P =Z2

OD Cr x

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Typical Applications (continued)

Boost

(56)

Buck-Boost

(57)

And the RHP zero (ωZ1) is approximated:

Boost

(58)

Buck-Boost

(59)

And the uncompensated DC loop gain (TU0) is approximated:

Buck

(60)

Boost

(61)

Buck-Boost

(62)

For all topologies, the primary method of compensation is to place a low frequency dominant pole (ωP2), which

ensures that there is ample phase margin at the crossover frequency. This is accomplished by placing a

capacitor (CCMP) from the COMP pin to AGND, which is calculated according to the lower value of the pole and

the RHP zero of the system (shown as a minimizing function):

(63)

(64)

If analog dimming is used, CCMP should be approximately 4× larger to maintain stability as the LEDs are dimmed

to zero.

A high-frequency compensation pole (ωP3) can be used to attenuate switching noise and provide better gain

margin. Assuming RFS = 10 Ω, CFS is calculated according to the higher value of the pole and the RHP zero of

the system (shown as a maximizing function):

(65)

(66)

The total system loop gain (T) can then be written as:

MAXINMAXT VV -- =

1-DMAX

DMAX

ICIN-RMS = ILED x

ICIN-RMS = 'iL-PP

(1-DMID)DII LEDRMSCIN xx=

- MID

CIN = 'VIN-PP x fSW

ILED x D

CIN = 8 x 'VIN-PP x fSW

'iL-PP

CIN = ILED x (1 - D) x D

'VIN-PP x fSW

x= 0U

TT -1 ¸

§s

xx +1 ¸

§s

Z3P

+1 ¸

§s

Z2P

+1 ¸

§s

Z1P

x= 0U

TT 1

xx +1 ¸

§s

Z3P

+1 ¸

§s

Z2P

+1 ¸

§s

Z1P

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Typical Applications (continued)

Buck

(67)

Boost and Buck-Boost

(68)

9.2.1.2.8 Input Capacitance

Set the nominal input voltage ripple (ΔvIN-PP) by solving for the required capacitance (CIN):

Buck

(69)

Boost

(70)

Buck-Boost

(71)

Use DMAX to set the worst case input voltage ripple, when solving for CIN in a buck-boost regulator and DMID = 0.5

when solving for CIN in a buck regulator.

The minimum allowable RMS input current rating (ICIN-RMS) can be approximated:

Buck

(72)

Boost

(73)

Buck-Boost

(74)

9.2.1.2.9 N-channel FET

The N-channel FET voltage rating should be at least 15% higher than the maximum N-channel FET drain-to-

source voltage (VT-MAX):

Buck

(75)

Boost

FDDD VIP x=

ID-MAX = ILED

ID-MAX = (1 - DMIN) x ILED

VRD-MAX = VIN-MAX + VO

VRD-MAX = VO

VRD-MAX = VIN-MAX

DSON

RMSTT RIP x= -

IRMST =

-D

ILED

DIT- ILEDRMS x=

IT-MAX = x ILED

1 - DMAX

DMAX

IT-MAX = DMAX x ILED

OMAXINMAXT VVV +

=--

MAXT

V-

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Typical Applications (continued)

(76)

Buck-Boost

(77)

The current rating should be at least 10% higher than the maximum average N-channel FET current (IT-MAX):

Buck

(78)

Boost and Buck-Boost

(79)

Approximate the nominal RMS transistor current (IT-RMS) :

Buck

(80)

9.2.1.2.9.1 Boost and Buck-Boost

(81)

Given an N-channel FET with on-resistance (RDS-ON), solve for the nominal power dissipation (PT):

(82)

9.2.1.2.10 Diode

The Schottky diode voltage rating should be at least 15% higher than the maximum blocking voltage (VRD-MAX):

Buck

(83)

Boost

(84)

Buck-Boost

(85)

The current rating should be at least 10% higher than the maximum average diode current (ID-MAX):

Buck

(86)

Boost and Buck-Boost

(87)

Replace DMAX with D in the ID-MAX equation to solve for the average diode current (ID). Given a diode with forward

voltage (VFD), solve for the nominal power dissipation (PD):

(88)

( )

RR +x 2UV1UV

A23P

( )

xHYS A23V xP- R 2UV

R1UV

UVH

R =

RUV1 =1.24VV ONTURN -

R1.24V UV2

RUV2 =A23P

VHYS

ROV1=R1.24V OV2

xVm620V OFFTURN -

ROV1 =1.24VV OFFTURN -

R1.24V OV2

ROV2 =VHYSO

A23P

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Typical Applications (continued)

9.2.1.2.11 Output OVLO

For boost and buck-boost regulators, output OVLO is programmed with the turn-off threshold voltage (VTURN-OFF)

and the desired hysteresis (VHYSO). To set VHYSO, solve for ROV2:

(89)

To set VTURN-OFF, solve for ROV1:

Boost

(90)

Buck-Boost

(91)

A small filter capacitor (COVP = 47 pF) should be added from the OVP pin to ground to reduce coupled switching

noise.

9.2.1.2.12 Input UVLO

For all topologies, input UVLO is programmed with the turnon threshold voltage (VTURN-ON) and the desired

hysteresis (VHYS).

Method 1: If no PWM dimming is required, a two resistor network can be used. To set VHYS, solve for RUV2:

(92)

To set VTURN-ON, solve for RUV1:

(93)

Method 2: If PWM dimming is required, a three resistor network is suggested. To set VTURN-ON, assume RUV2 =

10 kΩand solve for RUV1 as in Method 1. To set VHYS, solve for RUVH:

(94)

9.2.1.2.13 PWM Dimming Method

PWM dimming can be performed several ways:

Method 1: Connect the dimming MOSFET (Q3) with the drain to the nDIM pin and the source to AGND. Apply an

external PWM signal to the gate of QDIM. A pulldown resistor may be necessary to properly turn off Q3.

Method 2: Connect the anode of a Schottky diode to the nDIM pin. Apply an external inverted PWM signal to the

cathode of the same diode.

The DDRV pin should be connected to the gate of the dimFET with or without level-shifting circuitry as described

in the PWM Dimming section. The dimFET should be rated to handle the average LED current and the nominal

output voltage.

9.2.1.2.14 Analog Dimming Method

Analog dimming can be performed several ways:

Method 1: Place a potentiometer in series with the RCSH resistor to dim the LED current from the nominal ILED to

near zero.

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Typical Applications (continued)

Method 2: Connect a controlled current source as detailed in the Analog Dimming section to the CSH pin.

Increasing the current sourced into the CSH node decreases the LEDs from the nominal ILED to zero current in

the same manner as the thermal foldback circuit.

ROV2

ROV1

ILED

RSNS

RFS

CFS

COV

VIN

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

RLIM

CCMP

RCSH

RUV2

RUV1

RHSN

RHSP

CIN

10V ± 70V

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Typical Applications (continued)

9.2.2 LM3421 Buck-Boost Application

Figure 36. LM3421 Buck-Boost Application

9.2.2.1 Design Requirements

N=6

VLED = 3.5 V

rLED = 325 mΩ

VIN = 24 V

VIN-MIN = 10 V

VIN-MAX = 70 V

fSW = 500 kHz

VSNS = 100 mV

ILED =1A

ΔiL-PP = 700 mA

ΔiLED-PP = 12 mA

ΔvIN-PP = 100 mV

=1.24V1.24V

RHSP :

=k0.1

x0.1k12.4A1

xx RRI SNSCSHLED

=== 1.0

RSNS ILED

mV100

VSNS

CT = 1 nF

RT = 49.9 k:

2525 =49.9 k: x 1 nF

fSW = RT x CT= 501 kHz

2525 =500 kHz x 1 nF = 50 k:

RT =fSW x CT

DMAX == 677.0

V21 V10V21 +

VV IN-MINO +

21V

=21V + 70V = 0.231

DMIN = VO + VIN-MAX

533.0467.01D1'D =

D== 467.0

V21 V24V21 +

VOVV INO +

=95.1m3256rNr LEDD

V21V5.36VNV LEDO =

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Typical Applications (continued)

ILIM =6A

VTURN-ON = 10 V

VHYS =3V

VTURN-OFF = 40 V

VHYSO = 10 V

9.2.2.2 Detailed Design Procedure

9.2.2.2.1 Operating Point

Solve for VOand rD:

(95)

(96)

Solve for D, D', DMAX, and DMIN:

(97)

(98)

(99)

(100)

9.2.2.2.2 Switching Frequency

Assume CT= 1 nF and solve for RT:

(101)

The closest standard resistor is 49.9 kΩ; therefore, fSW is:

(102)

The chosen component from step 2 is:

(103)

9.2.2.2.3 Average LED Current

Solve for RSNS:

(104)

Assume RCSH = 12.4 kΩand solve for RHSP:

(105)

DILED x

'iPP-LED SW

fxrDxCO

= =

kHz01595.1 xx:1 mA

467.0A1 x F40P

'iPP-LED

rSWPP-LEDD xx DILED x

CO=

= F39.8P=

kHz015mA195.1 xx:

467.0A1 x

H331L P

A89.1

IRMSL =

ILED

RMSL x

-12

+¸

§Di PPL c

ILED

533.0mA678 2

x¸

§A1

533.0 A1

PP- ==

LDVIN xf1L SW

xkHz015H33 xP

467.0V42 x mA678

== DVIN xfSW

x467.0V42 x PH

1L PP-

'iLkHz015700 mA x

k4.21RCSH =

k1RR HSN ==

HSP

0.1R NSS =:

ILED = = k0.11.24V :xA0.1=

k4.121.0 :x:

RR CSHSNS xR1.24V HSP

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Typical Applications (continued)

The closest standard resistor for RSNS is actually 0.1 Ωand for RHSP is actually 1 kΩ; therefore, ILED is:

(106)

The chosen components from step 3 are:

(107)

9.2.2.2.4 Inductor Ripple Current

Solve for L1:

(108)

The closest standard inductor is 33 µH; therefore, ΔiL-PP is:

(109)

Determine minimum allowable RMS current rating:

(110)

The chosen component from step 4 is:

(111)

9.2.2.2.5 Output Capacitance

Solve for CO:

(112)

The closest capacitance totals 40 µF; therefore, ΔiLED-PP is:

(113)

Determine minimum allowable RMS current rating:

ZP3 = (max ZP1, ZZ1) x 10 = ZZ1 x 10

ZP3 = 36k sec

rad x 10 = 360ksec

rad

CMP 66

1 1

C 0.3 F

rad

5 10 0.675 5 10

sec

Z u u : u u :

= = sec

rad

675.0=

sec

rad

k19

min(ZP1, ZZ1)

ZP2 = 5 x TU0

ZP1

5 x 5630 5 x 5630

=T 0U = 5630=

04.0A1467.1 :

xx V620533.0 x

( )

D1+RI LIMLED xx V620D x

sec

rad

k36=== 533.095.1 2

x:H33467.0 Px

Dr 2

xL1Dx

sec

rad

k19=== 1.467 F40

1.95:Px

rDxD1+

0.04:RLIM =

:04.0 === 6.13A

mV245mV245

ILIM RLIM

:=== 041.0

RLIM mV245mV245

ILIM

CO = 4 x 10 PF

=ILED

IRMSCO- =

1- 0.677

677.0 1.45A

x1- DMAX

DMAX =

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

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Product Folder Links: LM3421 LM3423

Typical Applications (continued)

(114)

The chosen components from step 5 are:

(115)

9.2.2.2.6 Peak Current Limit

Solve for RLIM:

(116)

The closest standard resistor is 0.04 Ω; therefore, ILIM is:

(117)

The chosen component from step 6 is:

(118)

9.2.2.2.7 Loop Compensation

ωP1 is approximated:

(119)

ωZ1 is approximated:

(120)

TU0 is approximated:

(121)

To ensure stability, calculate ωP2:

(122)

Solve for CCMP:

(123)

To attenuate switching noise, calculate ωP3:

(124)

Assume RFS = 10 Ωand solve for CFS:

Q1 o 32A, 100V, DPAK

mW82m50A28.1RIP 2

DSON

RMSTT =

IRMST =

-ILED

Dc=xA28.1

0.467

533.0

=A2.1A1 =

677.01- 677.0

IMAXT-

V91V21V70VVV OMAXINMAXT =+=+= --

CIN = 4 x 4.7 PF

=ILED

IRMSIN- =

1- 0.677

677.0 1.45A

x1- DMAX

DMAX =

CIN == kHz504mV100 x 467.0A1 x F27.9 P=

'vSWPPIN- x

DILED x

F27.0CFS P=

F0.33CCMP P=

:10RFS =

= F28.0

1P==CFS 10:xsec

rad

k360

10:3P

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Typical Applications (continued)

(125)

The chosen components from step 7 are:

(126)

9.2.2.2.8 Input Capacitance

Solve for the minimum CIN:

(127)

To minimize power supply interaction a 200% larger capacitance of approximately 20 µF is used, therefore the

actual ΔvIN-PP is much lower. Because high voltage ceramic capacitor selection is limited, four 4.7-µF X7R

capacitors are chosen.

Determine minimum allowable RMS current rating:

(128)

The chosen components from step 8 are:

(129)

9.2.2.2.9 N-channel FET

Determine minimum Q1 voltage rating and current rating:

(130)

(131)

A 100-V N-channel FET is chosen with a current rating of 32 A due to the low RDS-ON = 50 mΩ. Determine IT-RMS

and PT:

(132)

(133)

The chosen component from step 9 is:

(134)

9.2.2.2.10 Diode

Determine minimum D1 voltage rating and current rating:

:== k13.6

-0.62VV OFFTURN

xR1.24V OV2

=ROV1

:x k4321.24V- 0.62V40V

29.94VA23k432A23RV OVHYSO =

=== A

10V

ROV2 P:k435

VHYSO

A23P

k:301R 2UV =k:18.2R 1UV =

RUV1

()

RR1.24V UV2UV1+x

VONTURN =

= V10.1=

( )

k130k18.21.24V :+:xk18.2 :

VONTURN-

:== k4.18

-1.24VV ONTURN

xR1.24V UV2

=RUV1

:x k1301.24V-1.24V10V

2.99VA23k130A23RV 2UVHYS =

=PP

=== A

RUV2 P:k130

VHYS

A23P

D1 o 12A, 100V, DPAK

mW600mV600A1VIP FDDD =

A1II LEDMAXD ==

V91V21V70VVV OMAXINMAXRD =

=--

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

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Product Folder Links: LM3421 LM3423

Typical Applications (continued)

(135)

(136)

A 100-V diode is chosen with a current rating of 12 A and VDF = 600 mV. Determine PD:

(137)

The chosen component from step 10 is:

(138)

9.2.2.2.11 Input UVLO

Solve for RUV2:

(139)

The closest standard resistor is 130 kΩ; therefore, VHYS is:

(140)

Solve for RUV1:

(141)

The closest standard resistor is 18.2 kΩ, making VTURN-ON:

(142)

The chosen components from step 11 are:

(143)

9.2.2.2.12 Output OVLO

Solve for ROV2:

(144)

The closest standard resistor is 432 kΩ; therefore, VHYSO is:

(145)

Solve for ROV1:

(146)

The closest standard resistor is 13.7 kΩ, making VTURN-OFF:

ROV1 = 13.7 k:

ROV2 = 432 k:

VOFFTURN =

V= 39.7

ROV1

k7.13 :

( )

k432k7.131.24V :+:x

()

RR1.24V OV2OV1+x

VOFFTURN-

0.5x

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LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

Typical Applications (continued)

(147)

The chosen components from step 12 are:

(148)

Table 2. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3421 Buck-boost controller TI LM3421MH

1 CBYP 2.2-µF X7R 10% 16V MURATA GRM21BR71C225KA12L

1 CCMP 0.33-µF X7R 10% 25V MURATA GRM21BR71E334KA01L

1 CFS 0.27-µF X7R 10% 25V MURATA GRM21BR71E274KA01L

4 CIN 4.7-µF X7R 10% 100V TDK C5750X7R2A475K

4 CO10-µF X7R 10% 50V TDK C4532X7R1H106K

1 COV 47-pF COG/NPO 5% 50V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50V MURATA GRM2165C1H102JA01D

1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF

1 L1 33 µH 20% 6.3 A COILCRAFT MSS1278-333MLB

1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682

1 Q2 PNP 150 V 600 mA FAIRCHILD MMBT5401

1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA

2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA

1 RLIM 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA

1 ROV1 13.7 kΩ1% VISHAY CRCW080513K7FKEA

1 ROV2 432 kΩ1% VISHAY CRCW0805432KFKEA

1 RSNS 0.1 Ω1% 1 W VISHAY WSL2512R1000FEA

1 RT49.9 kΩ1% VISHAY CRCW080549K9FKEA

1 RUV1 18.2 kΩ1% VISHAY CRCW080518K2FKEA

1 RUV2 130 kΩ1% VISHAY CRCW0805130KFKEA

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

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Product Folder Links: LM3421 LM3423

9.2.2.3 Application Curve

VOUT = 21 V

Figure 37. Sample Buck-Boost Efficiency vs Input Voltage.

ROV2

ROV1

RSNS

RFS

CFS

COV

8V ± 28V

ILED

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

RLIM

CCMP

RCSH

RUV2

RUV1

RHSN

RHSP

Q3 PWM

RUVH

CIN

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

9.2.3 LM3421 BOOST Application

Figure 38. LM3421 BOOST Application

9.2.3.1 Design Requirements

• Input: 8 V to 28 V

• Output: 9 LEDs at 1 A

• PWM Dimming up to 30kHz

• Switching Frequency: 700-kHz

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9.2.3.2 Detailed Design Procedure

Table 3. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3421 Boost controller TI LM3421MH

1 CBYP 2.2-µF X7R 10% 16 V MURATA GRM21BR71C225KA12L

1 CCMP 0.1-µF X7R 10% 25 V MURATA GRM21BR71E104KA01L

0 CFS DNP

4 CIN 4.7-µF X7R 10% 100 V TDK C5750X7R2A475K

4 CO10-µF X7R 10% 50 V TDK C4532X7R1H106K

1 COV 47-pF COG/NPO 5% 50 V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D

2 D1, D2 Schottky 60 V 5 A COMCHIP CDBC560-G

1 L1 33-µH 20% 6.3 A COILCRAFT MSS1278-333MLB

2 Q1, Q2 NMOS 60 V 8 A VISHAY SI4436DY

1 Q3 NMOS 60 V 115 mA ON-SEMI 2N7002ET1G

2 RCSH, ROV1 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RFS 0Ω1% VISHAY CRCW08050000Z0EA

2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA

1 RLIM 0.06 Ω1% 1 W VISHAY WSL2512R0600FEA

1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA

1 RSNS 0.1 Ω1% 1 W VISHAY WSL2512R1000FEA

1 RUV2 10 kΩ1% VISHAY CRCW080510K0FKEA

1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA

1 RUV1 1.82 kΩ1% VISHAY CRCW08051K82FKEA

1 RUVH 17.8 kΩ1% VISHAY CRCW080517K8FKEA

ROV2

ROV1

ILED

RSNS

RFS

CFS

COV

VIN

10V ± 30V

Q6 Q4

VIN

DIM

RPU

RSER

DIM

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

RLIM

CCMP

RCSH

RUV2

RUV1

RHSN

RHSP

Q3 PWM

RUVH

CIN

RPOT

LM3421

LM3423

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Product Folder Links: LM3421 LM3423

9.2.4 LM3421 Buck-Boost Application

Figure 39. LM3421 Buck-Boost Application

9.2.4.1 Design Requirements

• Input: 10 V to 30 V

• Output: 4 LEDs at 2 A

• PWM Dimming: up to 10 kHz

• Analog Dimming

• Switching Frequency: 600-kHz

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9.2.4.2 Detailed Design Procedure

Table 4. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3421 Buck-boost controller TI LM3421MH

1 CB100-pF COG/NPO 5% 50 V MURATA GRM2165C1H101JA01D

1 CBYP 2.2-µF X7R 10% 16 V MURATA GRM21BR71C225KA12L

3 CCMP, CREF, CSS 1-µF X7R 10% 25 V MURATA GRM21BR71E105KA01L

1 CF0.1-µF X7R 10% 25 V MURATA GRM21BR71E104KA01L

0 CFS DNP

4 CIN 6.8-µF X7R 10% 50 V TDK C5750X7R1H685K

4 CO10-µF X7R 10% 50 V TDK C4532X7R1H106K

1 COV 47-pF COG/NPO 5% 50 V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D

1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF

1 D2 Zener 10 V 500 mA ON-SEMI BZX84C10LT1G

1 L1 22 µH 20% 7.2 A COILCRAFT MSS1278-223MLB

2 Q1, Q2 NMOS 60 V 8 A VISHAY SI4436DY

1 Q3 NMOS 60 V 260 mA ON-SEMI 2N7002ET1G

1 Q4 PNP 40 V 200 mA FAIRCHILD MMBT5087

1 Q5 PNP 150 V 600 mA FAIRCHILD MMBT5401

1 Q6 NPN 300 V 600 mA FAIRCHILD MMBTA42

1 Q7 NPN 40 V 200 mA FAIRCHILD MMBT6428

1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RF10 Ω1% VISHAY CRCW080510R0FKEA

1 RFS 0Ω1% VISHAY CRCW08050000Z0EA

1 RUV2 10 kΩ1% VISHAY CRCW080510K0FKEA

2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA

1 RLIM 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA

1 ROV1 18.2 kΩ1% VISHAY CRCW080518K2FKEA

1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA

1 RPOT 1-MΩpotentiometer BOURNS 3352P-1-105

1 RPU 4.99 kΩ1% VISHAY CRCW08054K99FKEA

1 RSER 499 Ω1% VISHAY CRCW0805499RFKEA

1 RSNS 0.05 Ω1% 1 W VISHAY WSL2512R0500FEA

1 RT41.2 kΩ1% VISHAY CRCW080541K2FKEA

1 RUV1 1.43 kΩ1% VISHAY CRCW08051K43FKEA

1 RUVH 17.4 kΩ1% VISHAY CRCW080517K4FKEA

ROV2

ROV1

RSNS

RFS

CFS

COV

700 mA

ILED

RBIAS2

RMAX

RADJ

RCSH

VCC

18V ± 38V

OVP

nDIM

LM3423

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

RLIM

CCMP

RUV2

RUV1

RHSN

RHSP

CIN

FLT DPOL

9 12

TIMR LRDY

10 11

VREF

VCC

D3 Q2

RPU

RSER

CDIM

RPD

External

Enable

Q3 PWM

RUVH

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

9.2.5 LM3423 Boost Application

Figure 40. LM3423 Boost Application

9.2.5.1 Design Requirements

• Input: 18 V to 38 V

• Output: 12 LEDs at 700 mA

• High-Side PWM Dimming: up to 30 kHz

• Dimming: Analog

• Zero Current Shutdown

• Switching Frequency: 700-kHz

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9.2.5.2 Detailed Design Procedure

Table 5. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3423 Boost controller TI LM3423MH

1 CBYP 2.2-µF X7R 10% 16 V MURATA GRM21BR71C225KA12L

1 CCMP 1-µF X7R 10% 25 V MURATA GRM21BR71E105KA01L

1 CFS 0.1-µF X7R 10% 25 V MURATA GRM21BR71E104KA01L

4 CIN 4.7-µF X7R 10% 100 V TDK C5750X7R2A475K

4 CO10-µF X7R 10% 50 V TDK C4532X7R1H106K

1 COV 47-pF COG/NPO 5% 50 V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D

2 D1, D2 Schottky 60 V 5 A COMCHIP CDBC560-G

1 D3 Zener 10 V 500 mA ON-SEMI BZX84C10LT1G

1 L1 47 µH 20% 5.3 A COILCRAFT MSS1278-473MLB

1 Q1 NMOS 60 V 8 A VISHAY SI4436DY

1 Q2 PMOS 70 V 5.7 A ZETEX ZXMP7A17K

1 Q3 NMOS 60 V 260 mA ON-SEMI 2N7002ET1G

1 Q4, Q5 (dual pack) Dual PNP 40 V 200 mA FAIRCHILD FFB3906

1 Q6 NPN 300 V 600 mA FAIRCHILD MMBTA42

1 Q7 NPN 40 V 200 mA FAIRCHILD MMBT3904

1 RADJ 100-kΩpotentiometer BOURNS 3352P-1-104

1 RBIAS2 17.4 kΩ1% VISHAY CRCW080517K4FKEA

2 RCSH, ROV1 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA

3 RHSP, RHSN, RMAX 1 kΩ1% VISHAY CRCW08051K00FKEA

1 RLIM 0.06 Ω1% 1W VISHAY WSL2512R0600FEA

1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA

1 RSNS 0.15 Ω1% 1W VISHAY WSL2512R1500FEA

1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA

1 RUV1 1.43 kΩ1% VISHAY CRCW08051K43FKEA

1 RUV2 10 kΩ1% VISHAY CRCW080510K0FKEA

1 RUVH 16.9 kΩ1% VISHAY CRCW080516K9FKEA

ROV2

ROV1

500 mA

ILED

RSNS

RFS

CFS

COV

VIN

10V ± 70V

Q6 Q4

VIN

DIM

RPU

RSER

DIM

PWM

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

CCMP

RUV2

RUV1

RHSN

RHSP

CIN

RCSH

RUVH

CCSH

External

Enable

CEN

RCT

LM3421

LM3423

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SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

9.2.6 LM3421 Buck-Boost Application

Figure 41. LM3421 Buck-Boost Application

9.2.6.1 Design Requirements

• Input: 10 V to 70 V

• Output: 6 LEDs at 500 mA

• PWM Dimming up to 10 kHz

• Slow Fade Out

• MOSFET RDS-ON Sensing

• 700-kHz Switching Frequency

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9.2.6.2 Detailed Design Procedure

Table 6. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3421 Buck-boost controller TI LM3421MH

1 CB100-pF COG/NPO 5% 50 V MURATA GRM2165C1H101JA01D

1 CBYP 2.2-µF X7R 10% 16 V MURATA GRM21BR71C225KA12L

1 CCMP 1-µF X7R 10% 25 V MURATA GRM21BR71E105KA01L

1 CF0.1-µF X7R 10% 25 V MURATA GRM21BR71E104KA01L

0 CFS DNP

4 CIN 4.7-µF X7R 10% 100 V TDK C5750X7R2A475K

4 CO10-µF X7R 10% 50 V TDK C4532X7R1H106K

1 COV 47-pF COG/NPO 5% 50 V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D

1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF

1 D2 Zener 10 V 500 mA ON-SEMI BZX84C10LT1G

1 L1 68 µH 20% 4.3 A COILCRAFT MSS1278-683MLB

2 Q1, Q2 NMOS 100 V 32 A FAIRCHILD FDD3682

1 Q3 NMOS 60 V 260 mA ON-SEMI 2N7002ET1G

2 Q4, Q8 PNP 40 V 200 mA FAIRCHILD MMBT5087

1 Q5 PNP 150 V 600 mA FAIRCHILD MMBT5401

1 Q6 NPN 300 V 600 mA FAIRCHILD MMBTA42

2 Q7, Q9 NPN 40 V 200 mA FAIRCHILD MMBT6428

1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RFS 0Ω1% VISHAY CRCW08050000Z0EA

1 RUV2 10 kΩ1% VISHAY CRCW080510K0FKEA

2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA

1 ROV1 15.8 kΩ1% VISHAY CRCW080515K8FKEA

1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA

1 RPU 4.99 kΩ1% VISHAY CRCW08054K99FKEA

1 RSER 499 Ω1% VISHAY CRCW0805499RFKEA

1 RSNS 0.2 Ω1% 1 W VISHAY WSL2512R2000FEA

1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA

1 RUV1 1.43 kΩ1% VISHAY CRCW08051K43FKEA

1 RUVH 17.4 kΩ1% VISHAY CRCW080517K4FKEA

RLIM

ROV1

COV

15V ± 50V

PWM

ROV2

1.25A

ILED

RSNS

RFS

CFS

L1 Q4

RPU

CDIM

OVP

nDIM

LM3423

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

CCMP

RUV2

RUV1

RHSN

RHSP

CIN

FLT DPOL

9 12

TIMR LRDY

10 11

RCSH

RUVH

VIN

LED

STATUS

LIGHT

RPU2

RPD

External

Enable

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

9.2.7 LM3423 Buck Application

Figure 42. LM3423 Buck Application

9.2.7.1 Design Requirements

• Input: 15 V to 50 V

• Output: 3 LEDs at 1.25 A

• PWM Dimming up to 50 kHz

• LED Status Indicator

• Zero Current Shutdown

• 700-kHz Switching Frequency

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9.2.7.2 Detailed Design Procedure

Table 7. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3423 Buck controller TI LM3423MH

1 CBYP 2.2-µF X7R 10% 16 V MURATA GRM21BR71C225KA12L

2 CCMP, CDIM 0.1 µF X7R 10% 25 V MURATA GRM21BR71E104KA01L

0 CFS DNP

4 CIN 4.7-µF X7R 10% 100 V TDK C5750X7R2A475K

0 CODNP

1 COV 47-pF COG/NPO 5% 50 V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D

1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF

1 D2 Zener 10 V 500 mA ON-SEMI BZX84C10LT1G

1 L1 22 µH 20% 7.3 A COILCRAFT MSS1278-223MLB

1 Q1 NMOS 60 V 8 A VISHAY SI4436DY

1 Q2 PMOS 30 V 6.2 A VISHAY SI3483DV

1 Q3 NMOS 60 V 115 mA ON-SEMI 2N7002ET1G

1 Q4 PNP 150 V 600 mA FAIRCHILD MMBT5401

1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RFS 0Ω1% VISHAY CRCW08050000OZEA

2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA

1 RLIM 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA

1 ROV1 21.5 kΩ1% VISHAY CRCW080521K5FKEA

1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA

3 RPU, RPU2, RUV2 100 kΩ1% VISHAY CRCW0805100KFKEA

1 RT35.7 kΩ1% VISHAY CRCW080535K7FKEA

1 RSNS 0.08 Ω1% 1 W VISHAY WSL2512R0800FEA

1 RUV1 11.5 kΩ1% VISHAY CRCW080511K5FKEA

ROV2

ROV1

2.5A

ILED

RSNS

RFS

CFS

COV

VIN

15V ± 60V

OVP

nDIM

LM3423

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

CCMP

RUV2

RUV1

RHSN

RHSP

CIN

FLT DPOL

9 12

TIMR LRDY

10 11

RCSH

VIN

CTMR

RFLT

RPU D2

RPD

External

Enable

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

9.2.8 LM3423 Buck-Boost Application

Figure 43. LM3423 Buck-Boost Application

9.2.8.1 Design Requirements

• Input: 15 V to 60 V

• Output: 8 LEDs at 2.5 A

• Fault Input Disconnect

• Zero Current Shutdown

• 500-kHz Switching Frequency

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9.2.8.2 Detailed Design Procedure

Table 8. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3423 Buck-boost controller TI LM3423MH

1 CBYP 2.2-µF X7R 10% 16 V MURATA GRM21BR71C225KA12L

1 CCMP 0.33-µF X7R 10% 25 V MURATA GRM21BR71E334KA01L

1 CFS 0.1-µF X7R 10% 25 V MURATA GRM21BR71E104KA01L

4 CIN 4.7-µF X7R 10% 100 V TDK C5750X7R2A475K

4 CO10-µF X7R 10% 50 V TDK C4532X7R1H106K

1 COV 47-pF COG/NPO 5% 50 V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D

1 CTMR 220-pF COG/NPO 5% 50 V MURATA GRM2165C1H221JA01D

1 D1 Schottky 100 V 12 A VISHAY 12CWQ10FNPBF

1 D2 Zener 10 V 500 mA ON-SEMI BZX84C10LT1G

1 L1 22 µH 20% 7.2 A COILCRAFT MSS1278-223MLB

1 Q1 NMOS 100 V 32 A FAIRCHILD FDD3682

1 Q2 PMOS 70 V 5.7 A ZETEX ZXMP7A17K

1 Q5 PNP 150 V 600 mA FAIRCHILD MMBT5401

2 RCSH, ROV1 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RFS 10 Ω1% VISHAY CRCW080510R0FKEA

2 RFLT, RPU2 100 kΩ1% VISHAY CRCW0805100KFKEA

2 RHSP, RHSN 1 kΩ1% VISHAY CRCW08051K00FKEA

2 RLIM, RSNS 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA

1 ROV1 15.8 kΩ1% VISHAY CRCW080515K8FKEA

1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA

1 RT49.9 kΩ1% VISHAY CRCW080549K9FKEA

1 RUV1 13.7 kΩ1% VISHAY CRCW080513K7FKEA

1 RUV2 150 kΩ1% VISHAY CRCW0805150KFKEA

ROV2

ROV1

RSNS

RFS

CFS

COV

9V ± 36V

750 mA

ILED

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

CBYP

RLIM

CCMP

RCSH

RUV2

RUV1

RHSN

RHSP

Q3 PWM

RUVH

CIN

CSEP

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

9.2.9 LM3421 SEPIC Application

Figure 44. LM3421 SEPIC Application

9.2.9.1 Design Procedure

• Input: 9 V to 36 V

• Output: 5 LEDs at 750 mA

• PWM Dimming up to 30 kHz

• 500-kHz Switching Frequency

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

9.2.9.2 Detailed Design Procedure

Table 9. Bill of Materials

QTY PART ID PART VALUE MANUFACTURER PART NUMBER

1 LM3421 SEPIC controller TI LM3421MH

1 CBYP 2.2-µF X7R 10% 16 V MURATA GRM21BR71C225KA12L

1 CCMP 0.47-µF X7R 10% 25 V MURATA GRM21BR71E474KA01L

0 CFS DNP

4 CIN 4.7-µF X7R 10% 100 V TDK C5750X7R2A475K

4 CO10-µF X7R 10% 50 V TDK C4532X7R1H106K

1 CSEP 1-µF X7R 10% 100 V TDK C4532X7R2A105K

1 COV 47-pF COG/NPO 5% 50 V AVX 08055A470JAT2A

1 CT1000-pF COG/NPO 5% 50 V MURATA GRM2165C1H102JA01D

1 D1 Schottky 60 V 5 A COMCHIP CDBC560-G

2 L1, L2 68 µH 20% 4.3 A COILCRAFT DO3340P-683

2 Q1, Q2 NMOS 60 V 8 A VISHAY SI4436DY

1 Q3 NMOS 60 V 115 mA ON-SEMI 2N7002ET1G

1 RCSH 12.4 kΩ1% VISHAY CRCW080512K4FKEA

1 RFS 0Ω1% VISHAY CRCW08050000OZEA

2 RHSP, RHSN 750 Ω1% VISHAY CRCW0805750RFKEA

1 RLIM 0.04 Ω1% 1 W VISHAY WSL2512R0400FEA

1 ROV1 15.8 kΩ1% VISHAY CRCW080515K8FKEA

1 ROV2 499 kΩ1% VISHAY CRCW0805499KFKEA

2 RREF1, RREF2 49.9 kΩ1% VISHAY CRCW080549K9FKEA

1 RSNS 0.1 Ω1% 1 W VISHAY WSL2512R1000FEA

1 RT49.9 kΩ1% VISHAY CRCW080549K9FKEA

1 RUV1 1.62 kΩ1% VISHAY CRCW08051K62FKEA

1 RUV2 10 kΩ1% VISHAY CRCW080510K0FKEA

1 RUVH 16.9 kΩ1% VISHAY CRCW080516K9FKEA

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

10 Power Supply Recommendations

10.1 General Recommendations

The device is designed to operate from an input voltage supply range from 4.5 V to 75 V. This input supply

should be well regulated. If the input supply is located more than a few inches from the EVM or PCB, additional

bulk capacitance may be required in addition to the ceramic bypass capacitors.

10.2 Input Supply Current Limit

It is important to set the output current limit of your input supply to an appropriate value to avoid delays in your

converter analysis and optimization. If not set high enough, current limit can be tripped during start-up or when

your converter output power is increased, causing a foldback or shutdown condition. It is a common oversight

when powering up a converter for the first time.

11 Layout

11.1 Layout Guidelines

• The performance of any switching regulator depends as much upon the layout of the PCB as the component

selection. Following a few simple guidelines allows maximum noise rejection and minimal generation of EMI

within the circuit.

• Discontinuous currents are the most likely to generate EMI, therefore care should be taken when routing

these paths. The main path for discontinuous current in the LM34xx-Q1 buck regulator contains the input

capacitor (CIN), the recirculating diode (D1), the N-channel MOSFET (Q1), and the sense resistor (RLIM). In

the LM34xx-Q1 boost regulator, the discontinuous current flows through the output capacitor (CO), D1, Q1,

and RLIM. In the buck-boost regulator, both loops are discontinuous and should be carefully layed out. These

loops should be kept as small as possible and the connections between all the components should be short

and thick to minimize parasitic inductance. In particular, the switch node (where L1, D1 and Q1 connect)

should be just large enough to connect the components. To minimize excessive heating, large copper pours

can be placed adjacent to the short current path of the switch node.

• The RT, COMP, CSH, IS, HSP and HSN pins are all high-impedance inputs which couple external noise

easily; therefore, the loops containing these nodes should be minimized whenever possible.

• In some applications the LED or LED array can be far away (several inches or more) from the controller or on

a separate PCB connected by a wiring harness. When an output capacitor is used and the LED array is large

or separated from the rest of the regulator, the output capacitor should be placed close to the LEDs to reduce

the effects of parasitic inductance on the AC impedance of the capacitor.

OVP

nDIM

LM3421

AGND

VIN

PGND

DDRV

DAP

GATE

COMP

VIN

CSH

RCT

HSP

RPD

VCC

HSN

ILED

PWM

Minimize power loop containing discontinuous currents

Note critical paths and component placement:

x Ground plane under IC for signal routing helps minimize noise coupling

Power Ground

GND

Input

Power

Minimize signal current loops (components close to IC)

discontinuous switching

frequency currents

STAR GROUND

LM3421

LM3423

SNVS574G –JULY 2008–REVISED JULY 2019

www.ti.com

Product Folder Links: LM3421 LM3423

11.2 Layout Example

Figure 45. LM3421 Boost Layout Guideline

LM3421

LM3423

www.ti.com

SNVS574G –JULY 2008–REVISED JULY 2019

Product Folder Links: LM3421 LM3423

12 Device and Documentation Support

12.1 Device Support

12.1.1 Third-Party Products Disclaimer

TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT

CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES

OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER

ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.

12.2 Related Links

The table below lists quick access links. Categories include technical documents, support and community

resources, tools and software, and quick access to sample or buy.

Table 10. Related Links

PARTS PRODUCT FOLDER SAMPLE & BUY TECHNICAL

DOCUMENTS TOOLS &

SOFTWARE SUPPORT &

COMMUNITY

LM3421 Click here Click here Click here Click here Click here

LM3421-Q1 Click here Click here Click here Click here Click here

LM3423 Click here Click here Click here Click here Click here

LM3423-Q1 Click here Click here Click here Click here Click here

12.3 Community Resources

The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective

contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of

Use.

TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration

among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help

solve problems with fellow engineers.

Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and

contact information for technical support.

12.4 Trademarks

E2E is a trademark of Texas Instruments.

All other trademarks are the property of their respective owners.

12.5 Electrostatic Discharge Caution

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

12.6 Glossary

SLYZ022 —TI Glossary.

This glossary lists and explains terms, acronyms, and definitions.

13 Mechanical, Packaging, and Orderable Information

The following pages include mechanical, packaging, and orderable information. This information is the most

current data available for the designated devices. This data is subject to change without notice and revision of

this document. For browser-based versions of this data sheet, refer to the left-hand navigation.

PACKAGE OPTION ADDENDUM

www.ti.com 10-Dec-2020

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead finish/

Ball material

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM3421MH/NOPB ACTIVE HTSSOP PWP 16 92 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3421

LM3421MHX/NOPB ACTIVE HTSSOP PWP 16 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3421

LM3423MH/NOPB ACTIVE HTSSOP PWP 20 73 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3423

LM3423MHX/NOPB ACTIVE HTSSOP PWP 20 2500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 LM3423

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance

do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may

reference these types of products as "Pb-Free".

RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.

Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based

flame retardants must also meet the <=1000ppm threshold requirement.

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two

lines if the finish value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

PACKAGE OPTION ADDENDUM

www.ti.com 10-Dec-2020

Addendum-Page 2

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

OTHER QUALIFIED VERSIONS OF LM3421, LM3423 :

•Automotive: LM3421-Q1, LM3423-Q1

NOTE: Qualified Version Definitions:

•Automotive - Q100 devices qualified for high-reliability automotive applications targeting zero defects

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

LM3421MHX/NOPB HTSSOP PWP 16 2500 330.0 12.4 6.95 5.6 1.6 8.0 12.0 Q1

LM3423MHX/NOPB HTSSOP PWP 20 2500 330.0 16.4 6.95 7.1 1.6 8.0 16.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 9-Nov-2018

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

LM3421MHX/NOPB HTSSOP PWP 16 2500 367.0 367.0 35.0

LM3423MHX/NOPB HTSSOP PWP 20 2500 367.0 367.0 35.0

PACKAGE MATERIALS INFORMATION

www.ti.com 9-Nov-2018

Pack Materials-Page 2

www.ti.com

PACKAGE OUTLINE

TYP

6.6

6.2

14X 0.65

16X 0.30

0.19

4.55

(0.15) TYP

0 - 8 0.15

0.05

3.3

2.7

3.3

2.7

2X 1.34 MAX

NOTE 5

1.2 MAX

(1)

0.25

GAGE PLANE

0.75

0.50

NOTE 3

5.1

4.9

B4.5

4.3

4X 0.166 MAX

NOTE 5

4214868/A 02/2017

PowerPAD HTSSOP - 1.2 mm max heightPWP0016A

PLASTIC SMALL OUTLINE

NOTES:

1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing

per ASME Y14.5M.

2. This drawing is subject to change without notice.

3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not

exceed 0.15 mm per side.

4. Reference JEDEC registration MO-153.

5. Features may not be present.

PowerPAD is a trademark of Texas Instruments.

116

0.1 C A B

PIN 1 ID

AREA

SEATING PLANE

0.1 C

SEE DETAIL A

DETAIL A

TYPICAL

SCALE 2.400

THERMAL

PAD

www.ti.com

EXAMPLE BOARD LAYOUT

(5.8)

0.05 MAX

ALL AROUND 0.05 MIN

ALL AROUND

16X (1.5)

16X (0.45)

14X (0.65)

(3.4)

NOTE 9

(5)

NOTE 9

(3.3)

( 0.2) TYP

VIA (1.1) TYP

(1.1)

TYP

4214868/A 02/2017

PowerPAD HTSSOP - 1.2 mm max heightPWP0016A

PLASTIC SMALL OUTLINE

SYMM

SEE DETAILS

LAND PATTERN EXAMPLE

EXPOSED METAL SHOWN

SCALE:10X

METAL COVERED

BY SOLDER MASK

SOLDER MASK

DEFINED PAD

NOTES: (continued)

6. Publication IPC-7351 may have alternate designs.

7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.

8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature

numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).

9. Size of metal pad may vary due to creepage requirement.

METAL

SOLDER MASK

OPENING

NON SOLDER MASK

DEFINED

SOLDER MASK DETAILS

PADS 1-16

EXPOSED

METAL

SOLDER MASK

DEFINED

SOLDER MASK

METAL UNDER SOLDER MASK

OPENING

EXPOSED

METAL

www.ti.com

EXAMPLE STENCIL DESIGN

16X (1.5)

16X (0.45)

(3.3)

BASED ON

0.125 THICK

STENCIL

14X (0.65)

(R0.05) TYP

(5.8)

4214868/A 02/2017

PowerPAD HTSSOP - 1.2 mm max heightPWP0016A

PLASTIC SMALL OUTLINE

2.79 X 2.790.175 3.01 X 3.010.15 3.3 X 3.3 (SHOWN)0.125 3.69 X 3.690.1

SOLDER STENCIL

OPENING

STENCIL

THICKNESS

NOTES: (continued)

10. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate

design recommendations.

11. Board assembly site may have different recommendations for stencil design.

SYMM

BASED ON

0.125 THICK

STENCIL

BY SOLDER MASK

METAL COVERED SEE TABLE FOR

DIFFERENT OPENINGS

FOR OTHER STENCIL

THICKNESSES

SOLDER PASTE EXAMPLE

EXPOSED PAD

100% PRINTED SOLDER COVERAGE BY AREA

SCALE:10X

MECHANICAL DATA

PWP0020A

www.ti.com

MXA20A (Rev C)

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