FB
VIN VCC
RT
HB
VOUT
GND
CSG
RAMP
SS
CS
LO
VOUT
HO
HS
COMP
LM5118-Q1
010 20 30 40 50 60
VIN (V)
75
80
85
90
95
100
EFFICIENCY (%)
70
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An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,
intellectual property matters and other important disclaimers. PRODUCTION DATA.
LM5118-Q1
SNVSAX9 –JUNE 2017
LM5118-Q1 Wide Voltage Range Buck-Boost Controller
1
1 Features
1• AEC-Q100 Qualified for Automotive Applications
– Device Temperature Grade 1: –40°C to
+125°C Ambient Operating Temperature
– Device HBM ESD Classification Level 2
– Device CDM ESD Classification Level C6
• Ultra-Wide Input Voltage Range From 3 V to 75 V
• Emulated Peak Current Mode Control
• Smooth Transition Between Step-Down and Step-
Up Modes
• Switching Frequency Programmable to 500 KHz
• Oscillator Synchronization Capability
• Internal High Voltage Bias Regulator
• Integrated High and Low-Side Gate Drivers
• Programmable Soft-Start Time
• Ultra-Low Shutdown Current
• Enable Input Wide Bandwidth Error Amplifier
• 1.5% Feedback Reference Accuracy
• Thermal Shutdown
• Package: 20-Pin HTSSOP (Exposed Pad)
• Create a Custom Design Using the LM5118-Q1
With the WEBENCH®Power Designer
2 Applications
• Automotive Infotainment
• Automotive Start and Stop Systems
• Industrial Buck-Boost Supplies
3 Description
The LM5118-Q1 wide voltage range Buck-Boost
switching regulator controller features all of the
functions necessary to implement a high-
performance, cost-efficient Buck-Boost regulator
using a minimum of external components. The Buck-
Boost topology maintains output voltage regulation
when the input voltage is either less than or greater
than the output voltage, making it especially suitable
for automotive applications. The LM5118-Q1
operates as a buck regulator while the input voltage
is sufficiently greater than the regulated output
voltage and gradually transitions to the buck-boost
mode as the input voltage approaches the output.
This dual-mode approach maintains regulation over a
wide range of input voltages with optimal conversion
efficiency in the buck mode and a glitch-free output
during mode transitions. This easy-to-use controller
includes drivers for the high-side buck MOSFET and
the low-side boost MOSFET. The regulators control
method is based upon current mode control using an
emulated current ramp. Emulated current mode
control reduces noise sensitivity of the pulse-width
modulation circuit, allowing reliable control of the very
small duty cycles necessary in high input voltage
applications. Additional protection features include
current limit, thermal shutdown and an enable input.
The device is available in a power-enhanced, 20-pin
HTSSOP package featuring an exposed die attach
pad to aid thermal dissipation.
Device Information(1)
PART NUMBER PACKAGE BODY SIZE (NOM)
LM5118-Q1 HTSSOP (20) 6.50 mm × 4.40 mm
(1) For all available packages, see the orderable addendum at
the end of the data sheet.
Simplified Schematic Efficiency vs VIN and IOUT, VOUT = 12 V
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Table of Contents
1 Features.................................................................. 1
2 Applications ........................................................... 1
3 Description............................................................. 1
4 Revision History..................................................... 2
5 Pin Configuration and Functions......................... 3
6 Specifications......................................................... 5
6.1 Absolute Maximum Ratings ...................................... 5
6.2 ESD Ratings.............................................................. 5
6.3 Recommended Operating Conditions....................... 5
6.4 Thermal Information.................................................. 5
6.5 Electrical Characteristics........................................... 6
6.6 Typical Characteristics.............................................. 9
7 Detailed Description............................................ 11
7.1 Overview................................................................. 11
7.2 Functional Block Diagram....................................... 11
7.3 Feature Description................................................. 12
7.4 Device Functional Modes........................................ 19
8 Application and Implementation ........................ 22
8.1 Application Information............................................ 22
8.2 Typical Application ................................................. 22
9 Power Supply Recommendations...................... 34
9.1 Thermal Considerations.......................................... 34
9.2 Bias Power Dissipation Reduction.......................... 35
10 Layout................................................................... 37
10.1 Layout Guidelines ................................................. 37
10.2 Layout Example .................................................... 37
11 Device and Documentation Support................. 38
11.1 Device Support...................................................... 38
11.2 Trademarks........................................................... 38
11.3 Electrostatic Discharge Caution............................ 38
11.4 Glossary................................................................ 38
12 Mechanical, Packaging, and Orderable
Information........................................................... 38
4 Revision History
NOTE: Page numbers for previous revisions may differ from page numbers in the current version.
DATE REVISION NOTES
June 2017 * Initial release. Moved the automotive device from
the SNVS566 to a standalone data sheet
UVLO
RT
EN
RAMP
AGND
SS
FB
VIN
COMP
VOUT
PGND
CSG
CS
SYNC
LO
VCC
VCCX
HB
HO
HS
1
2
3
4
5
7
6
14
13
12
11
8
15
16
9
17
18
10
19
20
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(1) G = Ground, I = Input, O = Output, P = Power
5 Pin Configuration and Functions
PWP Package
20-Pin HTSSOP
(Top View)
Pin Descriptions
PIN TYPE(1) DESCRIPTION
NO. NAME
1 VIN P/I Input supply voltage.
2 UVLO I If the UVLO pin is below 1.23 V, the regulator will be in standby mode (VCC regulator running, switching
regulator disabled). When the UVLO pin exceeds 1.23 V, the regulator enters the normal operating mode.
An external voltage divider can be used to set an undervoltage shutdown threshold. A fixed 5-µA current is
sourced out of the UVLO pin. If a current limit condition exists for 256 consecutive switching cycles, an
internal switch pulls the UVLO pin to ground and then releases.
3 RT I The internal oscillator frequency is set with a single resistor between this pin and the AGND pin. The
recommended frequency range is 50 kHz to 500 kHz.
4 EN I If the EN pin is below 0.5 V, the regulator will be in a low power state drawing less than 10 µA from VIN. EN
must be raised above 3 V for normal operation.
5 RAMP I Ramp control signal. An external capacitor connected between this pin and the AGND pin sets the ramp
slope used for emulated current mode control.
6 AGND G Analog ground.
7 SS I Soft Start. An external capacitor and an internal 10-µA current source set the rise time of the error amp
reference. The SS pin is held low when VCC is less than the VCC undervoltage threshold (< 3.7 V), when
the UVLO pin is low (< 1.23 V), when EN is low (< 0.5 V) or when thermal shutdown is active.
8 FB I Feedback signal from the regulated output. Connect to the inverting input of the internal error amplifier.
9 COMP O Output of the internal error amplifier. The loop compensation network should be connected between COMP
and the FB pin.
10 VOUT I Output voltage monitor for emulated current mode control. Connect this pin directly to the regulated output.
11 SYNC I Sync input for switching regulator synchronization to an external clock.
12 CS I Current sense input. Connect to the diode side of the current sense resistor.
13 CSG I Current sense ground input. Connect to the ground side of the current sense resistor.
14 PGND G Power Ground.
15 LO O Boost MOSFET gate drive output. Connect to the gate of the external boost MOSFET.
16 VCC P/I/O Output of the bias regulator. Locally decouple to PGND using a low ESR/ESL capacitor located as close to
the controller as possible.
17 VCCX P/I Optional input for an externally supplied bias supply. If the voltage at the VCCX pin is greater than 3.9 V, the
internal VCC regulator is disabled and the VCC pin is internally connected to VCCX pin supply. If VCCX is
not used, connect to AGND.
4
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Pin Descriptions (continued)
PIN TYPE(1) DESCRIPTION
NO. NAME
18 HB I High-side gate driver supply used in bootstrap operation. The bootstrap capacitor supplies current to charge
the high-side MOSFET gate. This capacitor should be placed as close to the controller as possible and
connected between HB and HS.
19 HO O Buck MOSFET gate drive output. Connect to the gate of the high-side buck MOSFET through a short, low
inductance path.
20 HS I Buck MOSFET source pin. Connect to the source terminal of the high-side buck MOSFET and the bootstrap
capacitor.
— EP — Solder to the ground plane under the IC to aid in heat dissipation.
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(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings
only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended
Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.
6 Specifications
6.1 Absolute Maximum Ratings
over operating free air temperature range (unless otherwise noted)(1)
MIN MAX UNIT
VIN, EN, VOUT to GND –0.3 76 V
VCC, LO, VCCX, UVLO to GND –0.3 15 V
HB to HS –0.3 15 V
HO to HS –0.3 HB + 0.3 V
HS to GND –4 76 V
CSG, CS to GND –0.3 0.3 V
RAMP, SS, COMP, FB, SYNC, RT to GND –0.3 7 V
Junction temperature –40 150 °C
Storage temperature, Tstg –55 150 °C
(1) AEC Q100-002 indicates that HBM stressing shall be in accordance with the ANSI/ESDA/JEDEC JS-001 specification.
6.2 ESD Ratings VALUE UNIT
V(ESD) Electrostatic
discharge Human-body model (HBM), per AEC Q100-002(1) ±2000 V
Charged-device model (CDM), per AEC Q100-011 ±1000
(1) The Recommended Operating Conditions indicate conditions for which the device is intended to be functional, but does not ensure
specific performance limits. For specifications and test conditions see Electrical Characteristics.
(2) 5-V VIN is required to initially start the controller.
6.3 Recommended Operating Conditions
over operating free air temperature range (unless otherwise noted)(1)
MIN MAX UNIT
VIN (2) 3 75 V
VCC, VCCX 4.75 14 V
Junction temperature –40 +125 °C
(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report.
6.4 Thermal Information
THERMAL METRIC(1) LM5118-Q1
UNITPWP (HTSSOP)
20 PINS
RθJA Junction-to-ambient thermal resistance 40 °C/W
RθJC(bot) Junction-to-case (bottom) thermal resistance 4 °C/W
6
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(1) Minimum and maximum limits are 100% production tested at 25°C. Limits over the operating temperature range are ensured through
correlation using Statistical Quality Control (SQC) methods. Limits are used to calculate Average Outgoing Quality Level (AOQL).
6.5 Electrical Characteristics
Unless otherwise specified, the following conditions apply: VIN = 48 V, VCCX = 0 V, EN = 5 V, RT = 29.11 kΩ, No load on
LO and HO. Typical values apply for TJ= 25°C; minimum and maximum values apply over the full junction temperature range
for operation, −40°C to +125°C.(1)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
VIN SUPPLY
IBIAS VIN operating current VCCX = 0 V 4.5 5.5 mA
IBIASX VIN operating current VCCX = 5 V 1 1.85 mA
ISTDBY VIN shutdown current EN = 0 V 1 10 µA
VCC REGULATOR
VCC(REG) VCC regulation VCCX = 0 V 6.8 7 7.2 V
VCC(REG) VCC regulation VCCX = 0 V, VIN = 6 V 5 5.25 5.5 V
VCC sourcing current limit VCC = 0 21 35 mA
VCCX switch threshold VCCX rising 3.68 3.85 4.02 V
VCCX switch hysteresis 0.2 V
VCCX switch RDS(ON) ICCX = 10 mA 5 12 Ω
VCCX switch leakage VCCX = 0 V 0.5 1 µA
VCCCX pulldown resistance VCCX = 3 V 70 kΩ
VCC undervoltage lockout voltage VCC rising 3.52 3.7 3.86 V
VCC undervoltage hysteresis 0.21 V
HB DC bias current HB-HS = 15 V 205 260 µA
VC LDO mode turnoff 10 V
EN INPUT
VIL max EN input low threshold 0.5 V
VIH min EN input high threshold 3 V
EN input bias current VEN = 3 V –1 1 µA
EN input bias current VEN = 0.5 V –1 1 µA
EN input bias current VEN = 75 V 50 µA
UVLO THRESHOLDS
UVLO standby threshold UVLO rising 1.191 1.231 1.271 V
UVLO threshold hysteresis 0.105 V
UVLO pullup current source UVLO = 0 V 5 µA
UVLO pulldown RDS(ON) 100 200 Ω
SOFT START
SS current source SS = 0V 7.5 10.5 13.5 µA
SS to FB offset FB = 1.23 V 150 mV
SS output low voltage Sinking 100 µA, UVLO = 0 V 7 mV
ERROR AMPLIFIER
VREF FB reference voltage Measured at FB pin,
FB = COMP 1.212 1.23 1.248 V
FB input bias current FB = 2 V 20 200 nA
COMP sink/source current 3 mA
AOL DC gain 80 dB
fBW Unity bain bandwidth 3 MHz
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Electrical Characteristics (continued)
Unless otherwise specified, the following conditions apply: VIN = 48 V, VCCX = 0 V, EN = 5 V, RT = 29.11 kΩ, No load on
LO and HO. Typical values apply for TJ= 25°C; minimum and maximum values apply over the full junction temperature range
for operation, −40°C to +125°C.(1)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
PWM COMPARATORS
tHO(OFF) Forced HO off-time 305 400 495 ns
TON(MIN) Minimum HO on-time 70 ns
COMP to comparator offset 200 mV
OSCILLATOR (RT PIN)
fSW1 Frequency 1 RT = 29.11 kΩ178 200 224 kHz
fSW2 Frequency 2 RT = 9.525 kΩ450 515 575 kHz
SYNC
Sync threshold falling 1.3 V
CURRENT LIMIT
VCS(TH) Cycle-by-cycle sense voltage
threshold (CS-CSG) RAMP = 0 buck mode –103 –125 –147 mV
VCS(THX) Cycle-by-cycle sense voltage
threshold (CS-CSG) RAMP = 0 buck-boost mode –218 –255 –300 mV
CS bias current CS = 0 V 45 60 µA
CSG bias current CSG = 0 V 45 60 µA
Current limit fault timer 256 cycles
RAMP GENERATOR
IR1 RAMP current 1 VIN = 60 V, VOUT = 10 V 245 305 365 µA
IR2 RAMP current 2 VIN = 12 V, VOUT = 12 V 95 115 135 µA
IR3 RAMP current 3 VIN = 5 V, VOUT = 12 V 65 80 95 µA
VOUT bias current VOUT = 48 V 245 µA
LOW-SIDE (LO) GATE DRIVER
VOLL LO low-state output voltage ILO = 100 mA 0.14 0.23 V
VOHL LO high-state output voltage ILO = -100 mA
VOHL = VCC-VLO 0.25 V
LO rise time C-load = 1 nF, VCC = 8 V 16 ns
LO fall time C-load = 1 nF, VCC = 8 V 14 ns
IOHL Peak LO source current VLO = 0 V, VCC = 8 V 2.2 A
IOLL Peak LO sink current VLO = VCC = 8 V 2.7 A
HIGH-SIDE (HO) GATE DRIVER
VOLH HO low-state output voltage IHO = 100 mA 0.135 0.21 V
VOHH HO high-state output voltage IHO = -100 mA,
VOHH = VHB-VOH 0.25 V
HO rise time C-load = 1 nF, VCC = 8 V 14 ns
HO fall time C-load = 1 nF, VCC = 8 V 12 ns
IOHH Peak HO source current VHO = 0V, VCC = 8 V 2.2 A
IOLH Peak HO sink current VHO = VCC = 8 V 3.5 A
HB-HS undervoltage lockout 3 V
8
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Electrical Characteristics (continued)
Unless otherwise specified, the following conditions apply: VIN = 48 V, VCCX = 0 V, EN = 5 V, RT = 29.11 kΩ, No load on
LO and HO. Typical values apply for TJ= 25°C; minimum and maximum values apply over the full junction temperature range
for operation, −40°C to +125°C.(1)
PARAMETER TEST CONDITIONS MIN TYP MAX UNIT
(2) When the duty cycle exceeds 75%, the LM5118-Q1 controller gradually phases into the Buck-Boost mode.
BUCK-BOOST CHARACTERISTICS
Buck-boost mode Buck duty cycle (2) 69% 75% 80%
THERMAL
TSD Thermal shutdown temperature 165 °C
Thermal shutdown hysteresis 25 °C
0 1 2 3 4 5 6 7 8
OUTPUT VOLTAGE (V)
0
0.5
1
1.5
2
2.5
3
3.5
4
CURRENT (A)
LO Source
LO Sink
HO Sink
HO Source
1E+04 1E+05 1E+06 1E+07
FREQUENCY (Hz)
-10
50
GAIN (dB)
0
10
20
30
40
-30
150
0
30
60
90
120
PHASE (°)
0 10 20 30 40 50
0
2
4
6
8
10
VCC (V)
IVCC (mA)
02 4 6 8 10 12
VIN (V)
0
2
4
6
8
10
VCC (V)
65 110
-275
-100
CURRENT LIMIT THRESHOLD (mV)
VOUT/VIN DC (%)
70 75 80 85 90 95 100 105
-250
-225
-200
-175
-150
-125
010 20 30 40 50 60
VIN (V)
75
80
85
90
95
100
EFFICIENCY (%)
70
1 AMP
2 AMP
3 AMP
x
x
x
x
x
x
x
x
x
x
x
x
x
x
x
x
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xx
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x
x
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9
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6.6 Typical Characteristics
Figure 1. Efficiency vs VIN and IOUT, VOUT = 12 V Figure 2. Current Limit Threshold vs VOUT/VIN
VOUT = 12 V
Figure 3. VCC vs VIN Figure 4. VCC vs IVCC
Figure 5. Error Amplifier Gain/Phase Figure 6. LO and HO Peak Gate Current vs Output Voltage
VCC = 8 V
VOUT
FB
VIN VCC
RT
7V
REGULATOR
Vin
HB
LEVEL
SHIFT
EN
5V
THERMAL
SHUTDOWN
L1
C8
LM5118-Q1
VCC UVLO
HB UVLO
ERROR
AMP
COMP
TRACK
and
HOLD
CLK
S
R
Q
Q
OSCILLATOR
AGND
CLK
CSG
CLK
RAMP GENERATOR
= 5µA x Vin +
+
SHUTDOWN
AND
STANDBY
MODE
CONTROL
PWM
I-LIMIT
RAMP
SS
Vin
10 µA
C2
C1
Vin
C9
IRAMP BUCK-BOOST
MODE CONTROL
CS
PGND
LO
VOUT
HO
HS
1
2
3
4
514
6
15
10
13
12
20
19
18
3.9V
17
HICCUP MODE
FAULT TIMER
UVLO
VIN
7
8
9
16
VCCX
SYNC
11
D1
D2
R5
R6
Rs
Q1
Q2
C4
C5
R4
C6
R3
A=10
R8
C12
R1
R2
C1
1
R7
1.23V
C13
DISABLE
1.23V
IRAMP (buck-boost)
C3
C1
0
50 µA
3V
5 µA
IRAMP (buck) = (5µA x (Vin -Vout)) + 50 µA
Vth(buck) = 1.25V
(buck-boost) = 2.50V
Vth
10 Rs V/A
11
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7 Detailed Description
7.1 Overview
The LM5118-Q1 high voltage switching regulator features all of the functions necessary to implement an efficient
high voltage buck or buck-boost regulator using a minimum of external components. The regulator switches
smoothly from buck to buck-boost operation as the input voltage approaches the output voltage, allowing
operation with the input greater than or less than the output voltage. This easy to use regulator integrates high-
side and low-side MOSFET drivers capable of supplying peak currents of 2 A. The regulator control method is
based on current mode control using an emulated current ramp. Peak current mode control provides inherent line
feed-forward, cycle-by-cycle current limiting and ease of loop compensation. The use of an emulated control
ramp reduces noise sensitivity of the pulse-width modulation circuit, allowing reliable processing of very small
duty cycles necessary in high input voltage applications. The operating frequency is user programmable from 50
kHz to 500 kHz. An oscillator synchronization pin allows multiple LM5118-Q1 regulators to self synchronize or be
synchronized to an external clock. Fault protection features include current limiting, thermal shutdown, and
remote shutdown capability. An undervoltage lockout input allows regulator shutdown when the input voltage is
below a user selected threshold, and a low state at the enable pin will put the regulator into an extremely low
current shutdown state. The device is available in the HTSSOP-20EP package featuring an exposed pad to aid
in thermal dissipation.
7.2 Functional Block Diagram
HB
VOUT
CSG
CS
LO
HO
HS
LM5118-Q1
VIN
D1
Q2 (OFF)
D2
Q1
Buck Switch
Current
Diode Current
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7.3 Feature Description
A buck-boost regulator can maintain regulation for input voltages either higher or lower than the output voltage.
The challenge is that buck-boost power converters are not as efficient as buck regulators. The LM5118-Q1 has
been designed as a dual-mode controller whereby the power converter acts as a buck regulator while the input
voltage is above the output. As the input voltage approaches the output voltage, a gradual transition to the buck-
boost mode occurs. The dual-mode approach maintains regulation over a wide range of input voltages, while
maintaining the optimal conversion efficiency in the normal buck mode. The gradual transition between modes
eliminates disturbances at the output during transitions. Figure 8 shows the basic operation of the LM5118-Q1
regulator in the buck mode. In buck mode, transistor Q1 is active and Q2 is disabled. The inductor current ramps
in proportion to the VIN – VOUT voltage difference when Q1 is active and ramps down through the recirculating
diode D1 when Q1 is off. The first order buck mode transfer function is VOUT/VIN = D, where D is the duty cycle
of the buck switch, Q1.
Figure 8. Buck Mode Operation
18 17 16 15 14 13 12 0
10
20
30
40
50
60
70
80
90
100
DUTY CYCLE (%)
VIN (V)
HB
VOUT
CSG
CS
LO
HO
HS
LM5118-Q1 D1
Q2
D2
Q1
Buck Switch
Current
Diode Current
VIN
13
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Feature Description (continued)
Figure 9 shows the basic operation of buck-boost mode. In buck-boost mode both Q1 and Q2 are active for the
same time interval each cycle. The inductor current ramps up (proportional to VIN) when Q1 and Q2 are active
and ramps down through the recirculating diode during the off time. The first order buck-boost transfer function is
VOUT/VIN = D/(1-D), where D is the duty cycle of Q1 and Q2.
Figure 9. Buck-Boost Mode Operation
Figure 10. Mode Dependence on Duty Cycle (VOUT = 12 V)
7.3.1 UVLO
An undervoltage lockout pin is provided to disable the regulator when the input is below the desired operating
range. If the UVLO pin is below 1.13 V, the regulator enters a standby mode with the outputs disabled, but with
VCC regulator operating. If the UVLO input exceeds 1.23 V, the regulator will resume normal operation. A
voltage divider from the input to ground can be used to set a VIN threshold to disable the regulator in brownout
conditions or for low input faults.
SYNC
LM5118-Q1
SYNC
LM5118-Q1
UP TO FIVE
LM5118-Q1 DEVICES
RT = 6.4 x 109
f- 3.02 x 103
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Feature Description (continued)
If a current limit fault exists for more than 256 clock cycles, the regulator will enter a hiccup mode of current
limiting and the UVLO pin will be pulled low by an internal switch. This switch turns off when the UVLO pin
approaches ground potential allowing the UVLO pin to rise. A capacitor connected to the UVLO pin will delay the
return to a normal operating level and thereby set the off-time of the hiccup mode fault protection. An internal 5-
µA pullup current pulls the UVLO pin to a high state to ensure normal operation when the VIN UVLO function is
not required and the pin is left floating.
7.3.2 Oscillator and Sync Capability
The LM5118-Q1 oscillator frequency is set by a single external resistor connected between the RT pin and the
AGND pin. The RTresistor should be located very close to the device and connected directly to the pins of the
IC. To set a desired oscillator frequency (f), the necessary value for the RTresistor can be calculated from
Equation 1:
(1)
The SYNC pin can be used to synchronize the internal oscillator to an external clock. The external clock must be
of higher frequency than the free-running frequency set by the RTresistor. A clock circuit with an open-drain
output is the recommended interface from the external clock to the SYNC pin. The clock pulse duration should
be greater than 15 ns.
Multiple LM5118-Q1 devices can be synchronized together simply by connecting the SYNC pins together as in
Figure 11. In this configuration, all of the devices are synchronized to the highest frequency device. Figure 12
shows the SYNC input and output features of the LM5118-Q1. The internal oscillator circuit drives the SYNC pin
with a strong pull down or weak pullup inverter. When the SYNC pin is pulled low, either by the internal oscillator
or an external clock, the ramp cycle of the oscillator is terminated and forced 400 ns off-time is initiated before a
new oscillator cycle begins. If the SYNC pins of several LM5118-Q1 IC’s are connected together, the IC with the
highest internal clock frequency will pull all the connected SYNC pins low and terminate the oscillator ramp
cycles of the other ICs. The LM5118-Q1 with the highest programmed clock frequency will serve as the master
and control the switching frequency of all the devices with lower oscillator frequencies.
Figure 11. Sync From Multiple Devices
Emulated
Ramp
Ton
Buck:
Buck - Boost:
t
v
Pedestal Level = (volts/amp)
Rsx10
Cramp
ton
x)
PA
50+)- VoutVin(x
PA
5(
Cramp
ton
x)
PA
50+Vinx
PA
5(
SYNC
100 PA
S
R
Q
Q
DEADTIME
ONE - SHOT
1.23V
I 1/RT
15
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Feature Description (continued)
Figure 12. Simplified Oscillator and Block Diagram With Sync I/O Circuit
7.3.3 Error Amplifier and PWM Comparator
The internal high gain error amplifier generates an error signal proportional to the difference between the
regulated output voltage and an internal precision reference (1.23 V). The output of the error amplifier is
connected to the COMP pin. Loop compensation components, typically a type II network shown in are connected
between the COMP and FB pins. This network creates a low frequency pole, a zero, and a noise reducing high
frequency pole. The PWM comparator compares the emulated current sense signal from the RAMP generator to
the error amplifier output voltage at the COMP pin. The same error amplifier is used for operation in buck and
buck-boost mode.
Figure 13. Composition of Emulated Current Signal
gm x L
CRAMP
RS x A =
gm x L
CRAMP =A x RS
5 PA
Vx VIN + 50 PA
IRAMP (buck - boost) =
5 PA
Vx (VIN - VOUT) + 50 PA
IRAMP (buck) =
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Feature Description (continued)
7.3.4 Ramp Generator
The ramp signal of a pulse-width modulator with current mode control is typically derived directly from the buck
switch drain current. This switch current corresponds to the positive slope portion of the inductor current signal.
Using this signal for the PWM ramp simplifies the control loop transfer function to a single pole response and
provides inherent input voltage feed-forward compensation. The disadvantage of using the buck switch current
signal for PWM control is the large leading edge spike due to circuit parasitics. The leading edge spike must be
filtered or blanked to avoid early termination of the PWM pulse. Also, the current measurement may introduce
significant propagation delays. The filtering, blanking time and propagation delay limit the minimal achievable
pulse width. In applications where the input voltage may be relatively large in comparison to the output voltage,
controlling a small pulse width is necessary for regulation. The LM5118-Q1 uses a unique ramp generator which
does not actually measure the buck switch current but instead creates a signal representing or emulating the
inductor current. The emulated ramp provides signal to the PWM comparator that is free of leading edge spikes
and measurement or filtering delays. The current reconstruction is comprised of two elements, a sample-and-
hold pedestal level and a ramp capacitor which is charged by a controlled current source. Refer to Figure 13 for
details.
The sample-and-hold pedestal level is derived from a measurement of the recirculating current through a current
sense resistor in series with the recirculating diode of the buck regulator stage. A small value current sensing
resistor is required between the recirculating diode anode and ground. The CS and CSG pins should be Kelvin
connected directly to the sense resistor. The voltage level across the sense resistor is sampled and held just
prior to the onset of the next conduction interval of the buck switch. The current sensing and sample-and-hold
provide the DC level of the reconstructed current signal. The sample and hold of the recirculating diode current is
valid for both buck and buck-boost modes. The positive slope inductor current ramp is emulated by an external
capacitor connected from the RAMP pin to the AGND and an internal voltage controlled current source. In buck
mode, the ramp current source that emulates the inductor current is a function of the VIN and VOUT voltages per
Equation 2:
(2)
In buck-boost mode, the ramp current source is a function of the input voltage VIN, per Equation 3:
(3)
Proper selection of the RAMP capacitor (CRAMP) depends upon the value of the output inductor (L) and the
current sense resistor (RS). For proper current emulation, the sample and hold pedestal value and the ramp
amplitude must have the same relative relationship to the actual inductor current. That is:
where
• gmis the ramp generator transconductance (5 µA/V)
• A is the current sense amplifier gain (10V/V) (4)
The ramp capacitor should be located very close to the device and connected directly to the RAMP and AGND
pins.
1 +VOUT
Iout x VIN
A = 10k
1k + RG
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Feature Description (continued)
The relationship between the average inductor current and the pedestal value of the sampled inductor current
can cause instability in certain operating conditions. This instability is known as sub-harmonic oscillation, which
occurs when the inductor ripple current does not return to its initial value by the start of the next switching cycle.
Sub-harmonic oscillation is normally characterized by observing alternating wide and narrow pulses at the switch
node. Adding a fixed slope voltage ramp (slope compensation) to the current sense signal prevents this
oscillation. The 50 µA of offset current provided from the emulated current source adds enough slope
compensation to the ramp signal for output voltages less than or equal to 12 V. For higher output voltages,
additional slope compensation may be required. In such applications, the ramp capacitor can be decreased from
the nominal calculated value to increase the ramp slope compensation.
The pedestal current sample is obtained from the current sense resistor (Rs) connected to the CS and CSG pins.
It is sometimes helpful to adjust the internal current sense amplifier gain (A) to a lower value in order to obtain
the higher current limit threshold. Adding a pair of external resistors RG in a series with CS and CSG as in
Figure 14 reduces the current sense amplifier gain A according to Equation 5:
(5)
7.3.5 Current Limit
In the buck mode the average inductor current is equal to the output current (IOUT). In buck-boost mode the
average inductor current is approximately equal to:
(6)
Consequently, the inductor current in buck-boost mode is much larger especially when VOUT is large relative to
VIN. The LM5118-Q1 provides a current monitoring scheme to protect the circuit from possible over-current
conditions. When set correctly, the emulated current sense signal is proportional to the buck switch current with a
scale factor determined by the current sense resistor. The emulated ramp signal is applied to the current limit
comparator. If the peak of the emulated ramp signal exceeds 1.25 V when operating in the buck mode, the PWM
cycle is immediately terminated (cycle-by-cycle current limiting). In buck-boost mode the current limit threshold is
increased to 2.50 V to allow higher peak inductor current. To further protect the external switches during
prolonged overload conditions, an internal counter detects consecutive cycles of current limiting. If the counter
detects 256 consecutive current limited PWM cycles, the LM5118-Q1 enters a low power dissipation hiccup
mode. In the hiccup mode, the output drivers are disabled, the UVLO pin is momentarily pulled low, and the soft-
start capacitor is discharged. The regulator is restarted with a normal soft-start sequence once the UVLO pin
charges back to 1.23 V. The hiccup mode off-time can be programmed by an external capacitor connected from
UVLO pin to ground. This hiccup cycle will repeat until the output overload condition is removed.
In applications with low output inductance and high input voltage, the switch current may overshoot due to the
propagation delay of the current limit comparator and control circuitry. If an overshoot should occur, the sample-
and-hold circuit will detect the excess recirculating diode current. If the sample-and-hold pedestal level exceeds
the internal current limit threshold, the buck switch will be disabled and will skip PWM cycles until the inductor
current has decayed below the current limit threshold. This approach prevents current runaway conditions due to
propagation delays or inductor saturation since the inductor current is forced to decay before the buck switch is
turned on again.
Vout
D = Vin + Vout
0100 200 300 400 500 600
FREQUENCY (kHz)
0.75
0.8
0.85
0.9
0.95
1
MAX DUTY CYCLE
CSG
CS
IL
TRACK
and HOLD
RG
RAMP
1k RG
1k
RS
+
-
10k
10k
0.2V
+
-
RAMP
RESET
Vth
CRAMP
CURRENT LIMIT
COMPARATOR
IRAMP
CURRENT SENSE
AMPLIFIER
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Feature Description (continued)
Figure 14. Current Limit and Ramp Circuit
7.3.6 Maximum Duty Cycle
Each conduction cycle of the buck switch is followed by a forced minimum off-time of 400 ns to allow sufficient
time for the recirculating diode current to be sampled. This forced off-time limits the maximum duty cycle of the
controller. The actual maximum duty cycle will vary with the operating frequency as follows:
DMAX = 1 - f × 400 × 10–9
where
• f is the oscillator frequency in Hz (7)
Figure 15. Maximum Duty Cycle vs Frequency
Limiting the maximum duty cycle will limit the maximum boost ratio (VOUT/VIN) while operating in buck-boost
mode. For example, from Figure 15, at an operating frequency of 500 kHz, DMAX is 80%. Using the buck-boost
transfer function.
(8)
With D = 80%, solving for VOUT results in:
VOUT = 4 × VIN (9)
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Feature Description (continued)
With a minimum input voltage of 5 V, the maximum possible output voltage is 20 V at f = 500 kHz. The buck-
boost step-up ratio can be increased by reducing the operating frequency which increases the maximum duty
cycle.
7.3.7 Soft Start
The soft-start feature allows the regulator to gradually reach the initial steady-state operating point, thus reducing
start-up stresses and surges. The internal 10-µA soft-start current source gradually charges an external soft-start
capacitor connected to the SS pin. The SS pin is connected to the positive input of the internal error amplifier.
The error amplifier controls the pulse-width modulator such that the FB pin approximately equals the SS pin as
the SS capacitor is charged. Once the SS pin voltage exceeds the internal 1.23-V reference voltage, the error
amp is controlled by the reference instead of the SS pin. The SS pin voltage is clamped by an internal amplifier
at a level of 150 mV above the FB pin voltage. This feature provides a soft-start controlled recovery in the event
a severe overload pulls the output voltage (and FB pin) well below normal regulation but does not persist for 256
clock cycles.
Various sequencing and tracking schemes can be implemented using external circuits that limit or clamp the
voltage level of the SS pin. The SS pin acts as a non-inverting input to the error amplifier anytime SS voltage is
less than the 1.23-V reference. In the event a fault is detected (overtemperature, VCC undervoltage, hiccup
current limit), the soft-start capacitor will be discharged. When the fault condition is no longer present, a new soft-
start sequence will begin.
7.3.8 HO Output
The LM5118-Q1 contains a high-side, high-current gate driver and associated high voltage level shift. This gate
driver circuit works in conjunction with an internal diode and an external bootstrap capacitor. A 0.1-µF ceramic
capacitor, connected with short traces between the HB pin and HS pin is recommended for most circuit
configurations. The size of the bootstrap capacitor depends on the gate charge of the external FET. During the
off time of the buck switch, the HS pin voltage is approximately –0.5 V and the bootstrap capacitor is charged
from VCC through the internal bootstrap diode. When operating with a high PWM duty cycle, the buck switch will
be forced off each cycle for 400 ns to ensure that the bootstrap capacitor is recharged.
7.3.9 Thermal Protection
Internal Thermal Shutdown circuitry is provided to protect the integrated circuit in the event the maximum junction
temperature is exceeded. When activated, typically at 165°C, the controller is forced into a low power reset state,
disabling the output driver and the bias regulator. This protection is provided to prevent catastrophic failures from
accidental device overheating.
7.4 Device Functional Modes
Figure 10 shows how duty cycle effects the operational mode and is useful for reference in the following
discussions. Initially, only the buck switch is active and the buck duty cycle increases to maintain output
regulation as VIN decreases. When VIN is approximately equal to 15.5 V, the boost switch begins to operate with
a low duty cycle. If VIN continues to fall, the boost switch duty cycle increases and the buck switch duty cycle
decreases until they become equal at VIN = 13.2 V.
7.4.1 Buck Mode Operation: VIN > VOUT
The LM5118-Q1 buck-boost regulator operates as a conventional buck regulator with emulated current mode
control while VIN is greater than VOUT and the buck mode duty cycle is less than 75%. In buck mode, the LO
gate drive output to the boost switch remains low.
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Device Functional Modes (continued)
7.4.2 Buck-Boost Mode Operation: VIN ≊≊ VOUT
When VIN decreases relative to VOUT, the duty cycle of the buck switch will increase to maintain regulation.
Once the duty cycle reaches 75%, the boost switch starts to operate with a very small duty cycle. As VIN is
further decreased, the boost switch duty cycle increases until it is the same as the buck switch. As VIN is further
decreased below VOUT, the buck and boost switch operate together with the same duty cycle and the regulator
is in full buck-boost mode. This feature allows the regulator to transition smoothly from buck to buck-boost mode.
Note that the regulator can be designed to operate with VIN less than 4 V, but VIN must be at least 5 V
Figure 16 presents a timing illustration of the gradual transition from buck to buck-boost mode when the input
voltage ramps downward over a few switching cycles.
Figure 16. Buck (HO) and Boost (LO) Switch Duty Cycle vs. Time,
Illustrating Gradual Mode Change With Decreasing Input Voltage
7.4.3 High Voltage Start-Up Regulator
The LM5118-Q1 contains a dual-mode, high voltage linear regulator that provides the VCC bias supply for the
PWM controller and the MOSFET gate driver. The VIN input pin can be connected directly to input voltages as
high as 75 V. For input voltages below 10 V, an internal low dropout switch connects VCC directly to VIN. In this
supply range, VCC is approximately equal to VIN. For VIN voltages greater than 10 V, the low dropout switch is
disabled and the VCC regulator is enabled to maintain VCC at approximately 7 V. A wide operating range of 4 V
to 75 V (with a start-up requirement of at least 5 V) is achieved through the use of this dual mode regulator.
The output of the VCC regulator is current limited to 35 mA, typical. Upon power up, the regulator sources
current into the capacitor connected to the VCC pin. When the voltage at the VCC pin exceeds the VCC
undervoltage threshold of 3.7 V and the UVLO input pin voltage is greater than 1.23 V, the gate driver outputs
are enabled and a soft-start sequence begins. The gate driver outputs remain enabled until VCC falls below 3.5
V or the voltage at the UVLO pin falls below 1.13 V.
Vcc
Vin
Internal Enable Signal
3.5V
VIN and VCC
10V
7V
3.7V
21
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Device Functional Modes (continued)
In many applications, the regulated output voltage or an auxiliary supply voltage can be applied to the VCCX pin
to reduce the IC power dissipation. For output voltages between 4 V and 15 V, VOUT can be connected directly
to VCCX. When the voltage at the VCCX pin is greater than 3.85 V, the internal VCC regulator is disabled and
an internal switch connects VCCX to VCC, reducing the internal power dissipation.
In high voltage applications, take extrac care to ensure the VIN pin voltage does not exceed the absolute
maximum voltage rating of 76 V. During line or load transients, voltage ringing on the VIN line that exceeds the
absolute maximum rating can damage the IC. Both careful PCB layout and the use of quality bypass capacitors
located close to the VIN and GND pins are essential.
Figure 17. VIN and VCC Sequencing
7.4.4 Enable
The LM5118-Q1 contains an enable function which provides a very low input current shutdown mode. If the EN
pin is pulled below 0.5 V, the regulator enters shutdown mode, drawing less than 10 µA from the VIN pin.
Raising the EN input above 3 V returns the regulator to normal operation. The EN pin can be tied directly to the
VIN pin if this function is not needed. It must not be left floating. A 1-MΩpullup resistor to VIN can be used to
interface with an open-collector or open-drain control signal.
FB
VIN VCC
RT
HB
VOUT
GND
CSG
RAMP
SS
CS
LO
VOUT
HO
HS
COMP
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8 Application and Implementation
NOTE
Information in the following applications sections is not part of the TI component
specification, and TI does not warrant its accuracy or completeness. TI’s customers are
responsible for determining suitability of components for their purposes. Customers should
validate and test their design implementation to confirm system functionality.
8.1 Application Information
The LM5118-Q1 high voltage switching regulator features all of the functions necessary to implement an efficient
high voltage buck or buck-boost regulator using a minimum of external components. A buck-boost regulator can
maintain regulation for input voltages either higher or lower than the output voltage.
8.2 Typical Application
Figure 18. Simplified Application Schematic
RT = 6.4 x 109
f- 3.02 x 103
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Typical Application (continued)
8.2.1 Design Requirements
The procedure for calculating the external components is illustrated with the following design example. The
designations used in the design example correlate to the Figure 19. The design specifications are:
• VOUT = 12 V
• VIN = 5 V to 42 V
• f = 300 kHz
• Minimum load current (CCM operation) = 600 mA
• Maximum load current = 3 A
8.2.2 Detailed Design Procedure
8.2.2.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM5118-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
8.2.2.2 R7 = RT
RT sets the oscillator switching frequency. Generally speaking, higher operating frequency applications will use
smaller components, but have higher switching losses. An operating frequency of 300 kHz was selected for this
example as a reasonable compromise for both component size and efficiency. The value of RTcan be calculated
as follows:
(10)
therefore, R7 = 18.3 kΩ
OUT OUT IN(MIN) RIPPLE(BUCK-BOOST)
2(PEAK) IN(MIN) TOL
I (V V ) I
IV 2(1 L )
Ku
RIPPLE(BUCK)
OUT
1(PEAK) TOL
I
I
I2(1 L )
K
IN(MIN) OUT
OUT IN(MIN) RIPPLE
V (V )
L1 Buck-Boost Mode
(V V ) f I
u u
OUT IN(MAX) OUT
IN(MAX) RIPPLE
V (V V )
L1 Buck Mode
V f I
u u
26
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Typical Application (continued)
8.2.2.3 Inductor Selection, L1
The inductor value is determined based upon the operating frequency, load current, ripple current and the input
and output voltages. Refer to Figure 20 for details.
To keep the circuit in continuous conduction mode (CCM), the maximum ripple current IRIPPLE should be less
than twice the minimum load current. For the specified minimum load of 0.6 A, The maximum ripple current is 1.2
Ap-p. Also, the minimum value of L must be calculated both for a buck and buck-boost configurations. The final
value of inductance will generally be a compromise between the two modes. It is desirable to have a larger value
inductor for buck mode, but the saturation current rating for the inductor must be large for buck-boost mode,
resulting in a physically large inductor. Additionally, large value inductors present buck-boost mode loop
compensation challenges which will be discussed in the Error Amplifier Configuration section. For the design
example, the inductor values in both modes are calculated as:
(11)
where
• VOUT is the output voltage
• VIN(MAX) is the maximum input voltage
• f is the switching frequency
• IRIPPLE is the selected inductor peak to peak ripple current (1.2 A selected for this example)
• VIN(MIN) is the minimum input voltage
• (12)
The resulting inductor values are:
• L1 = 28 µH, Buck Mode
• L1 = 9.8 µH Buck-Boost mode
A 10-µH inductor was selected which is a compromise between these values, while favoring the buck-boost
mode. As illustrated in the compensation section below, the inductor value should be as low as possible to move
the buck-boost right-half-plane zero to a higher frequency. The ripple current is then rechecked with the selected
inductor value using Equation 11 and Equation 12,
IRIPPLE(BUCK) = 3.36 A (13)
IRIPPLE(BUCK-BOOST) = 1.17 A (14)
Because the inductor selected is lower than calculated for the Buck mode, the minimum load current for CCM in
buck mode is 1.68 A at maximum VIN.
With a 10-µH inductor, the worst case peak inductor currents can be estimated for each case, assuming a 20%
inductor value tolerance and assuming 80% efficiency of the converter.
(15)
where
•ηis efficiency
• LTOL is the inductor tolerance (16)
C15 = CRAMP = L x 10-6
2 x RSENSE
(BUCK-BOOST) IN(MIN) OUT RIPPLE(BUCK-BOOST)
OUT BUCK-BOOST
IN(MIN)
2.5 (1 M)
R13 V V I
I
10 K
V 2
˜
§ ·
˜ ˜ ˜
¨ ¸
¨ ¸
K
© ¹
(BUCK) RIPPLE(BUCK)
OUT BUCK
1.25(1 M)
R13 I
I
10 K
2
§ ·
˜ ˜
¨ ¸
K
© ¹
BUCK BOOST IN(MIN)
10
K 1 V
t
BUCK IN(MAX) OUT
10
K 1 V V
t
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Typical Application (continued)
For this example, Equation 15 and Equation 16 yield:
I1(PEAK) = 5.62 A (17)
I2(PEAK) = 13.4 A (18)
An acceptable current limit setting would be 6.7 A for buck mode because the LM5118-Q1 automatically doubles
the current limit threshold in buck-boost mode. The selected inductor must have a saturation current rating at
least as high as the buck-boost mode cycle-by-cycle current limit threshold, in this case at least 13.5 A. A 10-µH,
15-A inductor was chosen for this application.
8.2.2.4 R13 = RSENSE
To select the current sense resistor, begin by calculating the minimum K values for each mode using
Equation 19 and Equation 20. K represents the slope compensation of the controller and is different for each
mode, KBUCK and KBUCK-BOOST.
(19)
(20)
• KBUCK = 1.16
• KBUCK-BOOST = 3
Use Equation 21 and Equation 22 to calculate RSENSE for each mode of operation. A design margin, M, should
be selected between 10%-30% to allow for component tolerances. For this design M was selected to be 10%.
(21)
(22)
• R13(BUCK) = 19.75 mΩ
• R13(BUCK-BOOST) = 15.5 mΩ
An RSENSE value of no more than 15.5 mΩmust be used to ensure the required maximum output current in the
buck-boost mode. A standard value of 15 mΩwas selected for this design.
8.2.2.5 C15 = CRAMP
With the inductor value selected, the value of C3 necessary for the emulation ramp circuit is:
(23)
With the inductance value (L1) selected as 10 µH, the calculated value for CRAMP is 333 pF. A standard value of
330 pF was selected.
OUT
MAX OUT IN(MIN) RIPPLE(BUCK-BOOST)
OUT
IN(MIN)
V
ESR V V I
I
V 2
'
˜
OUT MAX OUT
MIN MAX
OUT IN(MIN) OUT
I D V
C with D
f V V V
u
u '
6OUT
IN(MIN) OUT
LIMIT(BUCK-BOOST)
50 10 V
2.5 C15 f (V V )
I10 R13
u u
u u
u
6OUT
IN(MAX)
LIMIT(BUCK)
50 10 V
1.25 C15 f V
I10 R13
u u
u u
u
28
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Typical Application (continued)
8.2.2.6 Inductor Current Limit Calculation
The current limit for each mode can be calculated using Equation 24 and Equation 26. If the peak current limit is
less than the calculated inductor peak current the R13 and C15 need to be recalculated. This can be done by
increasing the previous K values or M and reiterating the calculations.
(24)
ILIMIT(BUCK) = 7.795 A (25)
(26)
ILIMIT(BUCK-BOOST) = 14.29 A (27)
8.2.2.7 C9 - C12 = Output Capacitors
In buck-boost mode, the output capacitors C9 - C12 must supply the entire output current during the switch on-
time. For this reason, the output capacitors are chosen for operation in buck-boost mode, the demands being
much less in buck operation. Both bulk capacitance and ESR must be considered to ensure a given output ripple
voltage. Buck-boost mode capacitance can be estimated from:
(28)
ESR requirements can be estimated from:
(29)
For this example, with a ΔVOUT (output ripple) of 50 mV:
CMIN = 141 µF (30)
ESRMAX = 4.6 mΩ(31)
If hold-up times are a consideration, the values of the input and output capacitors must be increased
appropriately. Note that it is usually advantageous to use multiple capacitors in parallel to achieve the ESR value
required. Also, it is good practice to put a .1-µF to .47-µF ceramic capacitor directly on the output pins of the
supply to reduce high-frequency noise. Ceramic capacitors have good ESR characteristics, and are a good
choice for input and output capacitors. It should be noted that the effective capacitance of ceramic capacitors
decreases with dc bias. For larger bulk values of capacitance, a low ESR electrolytic is usually used. However,
electrolytic capacitors have poor tolerance, especially over temperature, and the selected value should be
selected larger than the calculated value to allow for temperature variation. Allowing for component tolerances,
the following values of COUT were chosen for this design example:
Two 180-µF Oscon electrolytic capacitors for bulk capacitance
Two 47-µF ceramic capacitors to reduce ESR
Two 0.47-µF ceramic capacitors to reduce spikes at the output.
IRMS(BUCK-BOOST) = 1 - D
IOUT D(1 -D)
IRMS(BUCK) = IOUT D(1 -D)
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Typical Application (continued)
8.2.2.8 D1
Reverse recovery currents degrade performance and decrease efficiency. For these reasons, a Schottky diode of
appropriate ratings should be used for D1. The voltage rating of the boost diode should be equal to VOUT plus
some margin. D1 conducts continually in buck mode and only when the buck switch is off in Buck-Boost mode.
8.2.2.9 D4
A Schottky type recirculating diode is required for all LM5118-Q1 applications. The near ideal reverse recovery
characteristics and low forward voltage drop are particularly important diode characteristics for high input voltage
and low output voltage applications. The reverse recovery characteristic determines how long the current surge
lasts each cycle when the buck switch is turned on. The reverse recovery characteristics of Schottky diodes
minimize the peak instantaneous power in the buck switch during the turnon transition. The reverse breakdown
rating of the diode should be selected for the maximum VIN plus some safety margin.
The forward voltage drop has a significant impact on the conversion efficiency, especially for applications with a
low output voltage. Rated current for diodes vary widely from various manufacturers. For the LM5118-Q1 this
current is user selectable through the current sense resistor value. Assuming a worst case 0.6-V drop across the
diode, the maximum diode power dissipation can be high. The diode should have a voltage rating of VIN and a
current rating of IOUT. A conservative design would at least double the advertised diode rating since
specifications between manufacturers vary. For the reference design a 100-V, 10-A Schottky in a D2PAK
package was selected.
8.2.2.10 C1 - C5 = Input Capacitor
A typical regulator supply voltage has a large source impedance at the switching frequency. Good-quality input
capacitors are necessary to limit the ripple voltage at the VIN pin while supplying most of the switch current
during the buck switch on-time. When the buck switch turns on, the current into the buck switch steps from zero
to the lower peak of the inductor current waveform, then ramps up to the peak value, and then drops to the zero
at turnoff. The RMS current rating of the input capacitors depends on which mode of operation is most critical.
(32)
This value is a maximum at 50% duty cycle which corresponds to VIN = 24 V.
(33)
Checking both modes of operation we find:
• IRMS(BUCK) = 1.5 A
• IRMS(BUCK-BOOST) = 4.7 A
Therefore C1-C5 should be sized to handle 4.7 A of ripple current. Quality ceramic capacitors with a low ESR
should be selected. To allow for capacitor tolerances, five 2.2-µF, 100-V ceramic capacitors will be used. If step
input voltage transients are expected near the maximum rating of the LM5118-Q1, a careful evaluation of the
ringing and possible spikes at the device VIN pin should be completed. An additional damping network or input
voltage clamp may be required in these cases.
8.2.2.11 C20
The capacitor at the VCC pin provides noise filtering and stability for the VCC regulator. The recommended value
of C20 should be no smaller than 0.1 µF, and should be a good-quality, low-ESR, ceramic capacitor. A value of 1
µF was selected for this design. C20 should be 10 x C8.
If operating without VCCX, then
fOSC x (QCBuck + Boost) + ILOAD(INTERNAL) (34)
must be less than the VCC current limit.
OFF IN
C21 R1 R3 R1 R3
t Ln 1 .98
R1 R3 V R3
ª º
˜ ˜
˜ ˜
« »
˜
¬ ¼
R3 = 1.23 x VIN(MIN) + 5 PA x R1 - 1.23
R1
- 1
R9
R8 1.23V
VOUT
=
tSS = 10 PA
C16 x 1.23V
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Typical Application (continued)
8.2.2.12 C8
The bootstrap capacitor between the HB and HS pins supplies the gate current to charge the buck switch gate at
turnon. The recommended value of C8 is 0.1 µF to 0.47 µF, and should be a good-quality, low-ESR, ceramic
capacitor. A value of 0.1 µF was chosen for this design.
8.2.2.13 C16 = CSS
The capacitor at the SS pin determines the soft-start time, that is, the time for the reference voltage and the
output voltage, to reach the final regulated value. The time is determined from:
(35)
and assumes a current limit > Iload + ICout
For this application, a C16 value of 0.1 µF was chosen which corresponds to a soft-start time of about 12 ms.
8.2.2.14 R8, R9
R8 and R9 set the output voltage level, the ratio of these resistors is calculated from:
(36)
For a 12-V output, the R8/R9 ratio calculates to 8.76. The resistors should be chosen from standard value
resistors and a good starting point is to select resistors within power ratings appropriate for the output voltage.
Values of 309 Ωfor R9 and 2.67 kΩfor R8 were selected.
8.2.2.15 R1, R3, C21
A voltage divider can be connected to the UVLO pin to set a minimum operating voltage VIN(UVLO) for the
regulator. If this feature is required, the easiest approach to select the divider resistor values is to choose a value
for R1 between 10 kΩand 100 kΩ, while observing the minimum value of R1 necessary to allow the UVLO
switch to pull the UVLO pin low. This value is:
R1 ≥1000 × VIN(MAX)
R1 ≥75 k in our example
R3 is then calculated from:
(37)
Because VIN(MIN) for our example is 5 V, set VIN(UVLO) to 4.0 V for some margin in component tolerances and
input ripple.
R1 = 75 k is chosen since it is a standard value.
R3 = 29.332 k is calculated from Equation 37. The 29.4 k value was used since it is a standard value.
Capacitor C21 provides filtering for the divider and the off-time of the hiccup duty cycle during current limit. The
voltage at the UVLO pin should never exceed 15 V when using an external set-point divider. It may be necessary
to clamp the UVLO pin at high input voltages.
Knowing the desired off time during hiccup current limit, the value of C21 is given by:
(38)
Notice that tOFF varies with VIN
In this example, C21 was chosen to be 0.1 µF. This will set the tOFF time to 723 µs with VIN = 12 V.
ESRzero = 2S x ESR x COUT
1
fRHPzero = RLOAD (1 - D)2
2S x L x D
MAX
P(MOD) LOAD OUT
1 D
f2 R C
S u u
DCGain(MOD) = RLOAD x VIN
10RS(VIN + 2VOUT)
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Typical Application (continued)
8.2.2.16 R2
A 1-M pullup resistor connected from the EN pin to the VIN pin is sufficient to keep enable in a high state if on-off
control is not used.
8.2.2.17 Snubber
A snubber network across the buck recirculating diode reduces ringing and spikes at the switching node.
Excessive ringing and spikes can cause erratic operation and increase noise at the regulator output. In the limit,
spikes beyond the maximum voltage rating of the LM5118-Q1 or the recirculating diode can damage these
devices. Selecting the values for the snubber is best accomplished through empirical methods. First, make sure
the lead lengths for the snubber connections are very short. Start with a resistor value between 5 and 20 Ω.
Increasing the value of the snubber capacitor results in more damping, however the snubber losses increase.
Select a minimum value of the capacitor that provides adequate clamping of the diode waveform at maximum
load. A snubber may be required for the boost diode as well. The same empirical procedure applies. Snubbers
were not necessary in this example.
8.2.2.18 Error Amplifier Configuration
8.2.2.18.1 R4, C18, C17
These components configure the error amplifier gain characteristics to accomplish a stable overall loop gain. One
advantage of current mode control is the ability to close the loop with only three feedback components, R4, C18
and C17. The overall loop gain is the product of the modulator gain and the error amplifier gain. The DC
modulator gain of the LM5118-Q1 is as follows:
(39)
The dominant, low frequency pole of the modulator is determined by the load resistance (RLOAD) and output
capacitance (COUT). The corner frequency of this pole is:
(40)
For this example, RLOAD = 4 Ω, DMAX = 0.705, and COUT = 454 µF, therefore:
fP(MOD) = 149 Hz (41)
DC Gain(MOD) = 4.598 = 13.25 dB (42)
Additionally, there is a right-half plane (RHP) zero associated with the modulator. The frequency of the RHP zero
is:
(43)
fRHPzero = 7.8 kHz (44)
The output capacitor ESR produces a zero given by:
(45)
ESRZERO = 76 kHz (46)
The RHP zero complicates compensation. The best design approach is to reduce the loop gain to cross zero at
about 25% of the calculated RHP zero frequency. The Type ll error amplifier compensation provided by R4, C18,
and C17 places one pole at the origin for high DC gain. The 2nd pole should be located close to the RHP zero.
The error amplifier zero (Equation 47) should be placed near the dominate modulator pole. This is a good
starting point for compensation. Refer to the on-line LM5118-Q1 Quick-Start calculator for ready to use equations
and more details.
fz = 2 x S x R4 x C18
1
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Typical Application (continued)
Components R4 and C18 configure the error amplifier as a type II configuration which has a DC pole and a zero
at:
(47)
C17 introduces an additional pole used to cancel high-frequency switching noise. The error amplifier zero
cancels the modulator pole leaving a single pose response at the crossover frequency of the loop gain if the
crossover frequency is much lower than the right half plane zero frequency. A single pole response at the
crossover frequency yields a very stable loop with 90 degrees of phase margin.
For the design example, a target loop bandwidth (crossover frequency) of 2.0 kHz was selected (about 25% of
the right-half-plane zero frequency). The error amplifier zero (fz) should be selected at a frequency near that of
the modulator pole and much less than the target crossover frequency. This constrains the product of R4 and
C18 for a desired compensation network zero to be less than 2 kHz. Increasing R4, while proportionally
decreasing C18 increases the error amp gain. Conversely, decreasing R4 while proportionally increasing C18
decreases the error amp gain. For the design example C18 was selected for 100 nF and R4 was selected to be
10 kΩ. These values set the compensation network zero at 159 Hz. The overall loop gain can be predicted as
the sum (in dB) of the modulator gain and the error amp gain.
If a network analyzer is available, the modulator gain can be measured and the error amplifier gain can be
configured for the desired loop transfer function. If a network analyzer is not available, the error amplifier
compensation components can be designed with the guidelines given. Step load transient tests can be
performed to verify acceptable performance. The step load goal is minimal overshoot with a damped response.
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Typical Application (continued)
8.2.3 Application Curves
The plots in Figure 21 through Figure 26 show the gain and phase diagrams of the design example. The overall
bandwidth is lower in a buck-boost application due the compensation challenges associated with the right-half-
plane zero. For a pure buck application, the bandwidth could be much higher. The LM5116 data sheet is a good
reference for compensation design of a pure buck mode regulator.
Figure 21. Modulator Gain and Phase - Buck Mode Figure 22. Modulator Gain and Phase - Buck-Boost Mode
Figure 23. Error Amplifier Gain and Phase - Buck Mode Figure 24. Error Amplifier Gain and Phase - Buck-Boost
Mode
Figure 25. Overall Loop Gain and Phase - Buck Mode Figure 26. Overall Loop Gain and Phase - Buck Boost
Mode
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9 Power Supply Recommendations
9.1 Thermal Considerations
The highest power dissipating components are the two power MOSFETs, the recirculating diode, and the output
diode. The easiest way to determine the power dissipated in the MOSFETs is to measure the total conversion
losses (PIN - POUT), then subtract the power losses in the Schottky diodes, output inductor and any snubber
resistors. An approximation for the recirculating Schottky diode loss is:
P = (1 – D) × IOUT × VFWD. (48)
The boost diode loss is
P = IOUT × VFWD. (49)
If a snubber is used, the power loss can be estimated with an oscilloscope by observation of the resistor voltage
drop at both turnon and turnoff transitions. The LM5118-Q1 package has an exposed thermal pad to aid power
dissipation. Selecting diodes with exposed pads will aid the power dissipation of the diodes as well. When
selecting the MOSFETs, pay careful attention to RDS(ON) at high temperature. Also, selecting MOSFETs with low
gate charge will result in lower switching losses.
HB
HO
VCC
CS
CSG
LM5118-Q1
VOUT
VOUT
LO
VIN
L1
D1
D2
Cout
Q2
Q1
HS
VBIAS
VCCX
HB
HO
VCC
CS
GND
LM5118-Q1
VOUT
LO
CSG
VIN
L1
D1
D2
Cout
Q2
Q1
HS
VCCX
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9.2 Bias Power Dissipation Reduction
Buck or Buck-boost regulators operating with high input voltage can dissipate an appreciable amount of power
while supplying the required bias current of the IC. The VCC regulator must step-down the input voltage VIN to a
nominal VCC level of 7 V. The large voltage drop across the VCC regulator translates into high power dissipation
in the VCC regulator. There are several techniques that can significantly reduce this bias regulator power
dissipation. Figure 27 and Figure 28 depict two methods to bias the IC, one from the output voltage and one from
a separate bias supply. In the first case, the internal VCC regulator is used to initially bias the VCC pin. After the
output voltage is established, the VCC pin bias current is supplied through the VCCX pin, which effectively
disables the internal VCC regulator. Any voltage greater than 4.0 V can supply VCC bias through the VCCX pin.
However, the voltage applied to the VCCX pin should never exceed 15 V. The voltage supplied through VCCX
must be large enough to drive the switching MOSFETs into full saturation.
Figure 27. VCC Bias From VOUT 4 V < VOUT < 15 V
Figure 28. VCC Bias With Additional Bias Supply
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Bias Power Dissipation Reduction (continued)
In a buck-boost regulator, there are two loops where currents are switched very fast. The first loop starts from the
input capacitors, and then to the buck switch, the inductor, the boost switch then back to the input capacitor. The
second loop starts from the inductor, and then to the output diode, the output capacitor, the recirculating diode,
and back to the inductor. Minimizing the PCB area of these two loops reduces the stray inductance and
minimizes noise and the possibility of erratic operation. A ground plane in the PCB is recommended as a means
to connect the input filter capacitors to the output filter capacitors and the PGND pins of the LM5118-Q1.
Connect all of the low current ground connections (CSS, RT, CRAMP) directly to the regulator AGND pin. Connect
the AGND and PGND pins together through topside copper area covering the entire underside of the device.
Place several vias in this underside copper area to the ground plane of the input capacitors.
Controller
VIN GND GND VOUT
Inductor
CIN
CIN
COUT
COUT
RSENSE
QBUCK
DBUCK
DBOOST
QBOOST
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10 Layout
10.1 Layout Guidelines
In a buck-boost regulator, there are two loops where currents are switched very fast. The first loop starts from the
input capacitors, and then to the buck switch, the inductor, the boost switch then back to the input capacitor. The
second loop starts from the inductor, and then to the output diode, the output capacitor, the re-circulating diode,
and back to the inductor. Minimizing the PCB area of these two loops reduces the stray inductance and
minimizes noise and the possibility of erratic operation. A ground plane in the PCB is recommended as a means
to connect the input filter capacitors to the output filter capacitors and the PGND pins of the LM5118-Q1.
Connect all of the low current ground connections (CSS, RT, CRAMP) directly to the regulator AGND pin. Connect
the AGND and PGND pins together through topside copper area covering the entire underside of the device.
Place several vias in this underside copper area to the ground plane of the input capacitors.
10.2 Layout Example
Figure 29. LM5118-Q1 Layout Example
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11 Device and Documentation Support
11.1 Device Support
11.1.1 Development Support
11.1.1.1 Custom Design With WEBENCH® Tools
Click here to create a custom design using the LM5118-Q1 device with the WEBENCH® Power Designer.
1. Start by entering the input voltage (VIN), output voltage (VOUT), and output current (IOUT) requirements.
2. Optimize the design for key parameters such as efficiency, footprint, and cost using the optimizer dial.
3. Compare the generated design with other possible solutions from Texas Instruments.
The WEBENCH Power Designer provides a customized schematic along with a list of materials with real-time
pricing and component availability.
In most cases, these actions are available:
• Run electrical simulations to see important waveforms and circuit performance
• Run thermal simulations to understand board thermal performance
• Export customized schematic and layout into popular CAD formats
• Print PDF reports for the design, and share the design with colleagues
Get more information about WEBENCH tools at www.ti.com/WEBENCH.
11.2 Trademarks
WEBENCH is a registered trademark of Texas Instruments.
All other trademarks are the property of their respective owners.
11.3 Electrostatic Discharge Caution
These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam
during storage or handling to prevent electrostatic damage to the MOS gates.
11.4 Glossary
SLYZ022 —TI Glossary.
This glossary lists and explains terms, acronyms, and definitions.
12 Mechanical, Packaging, and Orderable Information
The following pages include mechanical, packaging, and orderable information. This information is the most
current data available for the designated devices. This data is subject to change without notice and revision of
this document. For browser-based versions of this data sheet, refer to the left-hand navigation.
PACKAGE OPTION ADDENDUM
www.ti.com 19-Jun-2017
Addendum-Page 1
PACKAGING INFORMATION
Orderable Device Status
(1)
Package Type Package
Drawing Pins Package
Qty Eco Plan
(2)
Lead/Ball Finish
(6)
MSL Peak Temp
(3)
Op Temp (°C) Device Marking
(4/5)
Samples
LM5118Q1MH/NOPB ACTIVE HTSSOP PWP 20 73 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 LM5118
Q1MH
LM5118Q1MHX/NOPB ACTIVE HTSSOP PWP 20 2500 Green (RoHS
& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 LM5118
Q1MH
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance
do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may
reference these types of products as "Pb-Free".
RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.
Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based
flame retardants must also meet the <=1000ppm threshold requirement.
(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.
(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.
(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation
of the previous line and the two combined represent the entire Device Marking for that device.
(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish
value exceeds the maximum column width.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information
provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and
continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.
TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.
TAPE AND REEL INFORMATION
*All dimensions are nominal
Device Package
Type Package
Drawing Pins SPQ Reel
Diameter
(mm)
Reel
Width
W1 (mm)
A0
(mm) B0
(mm) K0
(mm) P1
(mm) W
(mm) Pin1
Quadrant
LM5118Q1MHX/NOPB HTSSOP PWP 20 2500 330.0 16.4 6.95 7.1 1.6 8.0 16.0 Q1
PACKAGE MATERIALS INFORMATION
www.ti.com 1-Jul-2017
Pack Materials-Page 1
*All dimensions are nominal
Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)
LM5118Q1MHX/NOPB HTSSOP PWP 20 2500 367.0 367.0 35.0
PACKAGE MATERIALS INFORMATION
www.ti.com 1-Jul-2017
Pack Materials-Page 2
MECHANICAL DATA
PWP0020A
www.ti.com
MXA20A (Rev C)
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and ISO 26262), TI is not responsible for any failure to meet such industry standard requirements.
Where TI specifically promotes products as facilitating functional safety or as compliant with industry functional safety standards, such
products are intended to help enable customers to design and create their own applications that meet applicable functional safety standards
and requirements. Using products in an application does not by itself establish any safety features in the application. Designers must
ensure compliance with safety-related requirements and standards applicable to their applications. Designer may not use any TI products in
life-critical medical equipment unless authorized officers of the parties have executed a special contract specifically governing such use.
Life-critical medical equipment is medical equipment where failure of such equipment would cause serious bodily injury or death (e.g., life
support, pacemakers, defibrillators, heart pumps, neurostimulators, and implantables). Such equipment includes, without limitation, all
medical devices identified by the U.S. Food and Drug Administration as Class III devices and equivalent classifications outside the U.S.
TI may expressly designate certain products as completing a particular qualification (e.g., Q100, Military Grade, or Enhanced Product).
Designers agree that it has the necessary expertise to select the product with the appropriate qualification designation for their applications
and that proper product selection is at Designers’ own risk. Designers are solely responsible for compliance with all legal and regulatory
requirements in connection with such selection.
Designer will fully indemnify TI and its representatives against any damages, costs, losses, and/or liabilities arising out of Designer’s non-
compliance with the terms and provisions of this Notice.
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