General Description
The MAX17007/MAX17008 are dual Quick-PWM™ step-
down controllers intended for general power generation
in battery-powered systems. The two switched-mode
power supplies (SMPSs) can also be combined to oper-
ate in a two-phase single-output mode. Constant on-
time Quick-PWM operation provides fast response to
load transients and handles wide input/output (I/O) volt-
age ratios with ease, while maintaining a relatively con-
stant switching frequency. The switching frequency can
be individually adjusted between 200kHz and 600kHz
with external resistors. Differential output current sens-
ing allows output sense-resistor sensing for an accurate
current limit, or lossless inductor direct-current resis-
tance (DCR) current sensing for lower power dissipation
while maintaining 0.7% output accuracy. Overvoltage
(MAX17007 only), undervoltage protection, and accu-
rate user-selectable current limits (15mV, 30mV, 45mV,
and 60mV) ensure robust operations.
The SMPS outputs can operate in skip mode or in ultra-
sonic mode for improved light-load efficiency. The ultra-
sonic mode eliminates audible noises by maintaining a
minimum switching frequency of 25kHz in pulse-
skipping mode.
The output voltage of SMPS1 can be dynamically
adjusted by changing the voltage at the REFIN1 pin.
The device includes a 0.5% accurate reference output
that can be used to set the REFIN1 voltage. An external
5V bias supply is required to power the internal circuitry
and its gate drivers.
Independent on/off controls with well-defined logic thresh-
olds and independent open-drain power-good outputs
provide flexible system configurations. To prevent current
surges at startup, the internal voltage target is slowly
ramped up from zero to the final target with a slew rate of
1.3mV/µs for SMPS1 at CSL1 and 0.65mV/µs for SMPS2
at FB2. To prevent the output from ringing off below
ground in shutdown, the internal voltage target is ramped
down from its previous value to zero with the same
respective slew rates. Integrated bootstrap switches
eliminate the need for external bootstrap diodes.
The MAX17007/MAX17008 are available in a space-
saving, 28-pin, 4mm x 4mm, thin QFN package with an
exposed backside pad.
Applications
Features
Dual Quick-PWM with Fast Transient Response
Automatic Dynamic REFIN1 Detection and
PGOOD1/Fault Blanking
Fixed and Adjustable Output Voltages
0.7% Output Accuracy Over Line and Load
OUT1: 0 to 2V Dynamic Output or Preset 1.05V
OUT2: 0.7V to 2V Range or Preset 1.5V
Resistor-Programmable Switching Frequency
Integrated BST Switches
Differential Current-Sense Inputs
Low-Cost DCR Sensing or Accurate Current-
Sense Resistors
Internally Coupled Current-Sense Compensation
Combinable Mode Supports High-Current
Dynamic Output Voltages
Selectable Forced-PWM, Pulse Skip, or Ultrasonic
Mode Operation
26V Maximum Input Voltage Rating
Independent Enable Inputs
Independent Power-Good Outputs
Overvoltage Protection (MAX17007 Only)
Undervoltage/Thermal Protection
Automatic Dynamic REFIN1 Detection and
PGOOD1/Fault Blanking
Voltage Soft-Start and Soft-Shutdown
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
________________________________________________________________
Maxim Integrated Products
1
MAX17007
MAX17008
18
THIN QFN
(4mm x 4mm)
4
17
5
16
6
15
1422
7
19
3
20
2
21
1223
1224
1125
1026
927
828
1
LX1
DH1
PGOOD1
EN1
CSH1
TOP VIEW
CSL1
REFIN1
LX2
DH2
PGOOD2
EN2
CSH2
CSL2
FB2
BST2
PGND
DL2
VDD
DL1
GND
BST1
REF
ILIM1
(CCI) ILIM2
VCC
SKIP
TON1
TON2
+
Pin Configuration
Ordering Information
19-3200; Rev 0; 2/08
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642,
or visit Maxim’s website at www.maxim-ic.com.
EVALUATION KIT
AVAILABLE
+
Denotes a lead-free package.
*Future product—contact factory for availability.
PART TEMP RANGE PIN-PACKAGE
MAX17007GTI+ -40°C to +105°C 28 Thin QFN
MAX17008GTI+* -40°C to +105°C 28 Thin QFN
Notebook Computers
Low-Power I/O Supplies
GPU Core Supplies
5V and 3.3V Supplies
2 to 4 Li+ Cells Battery-
Powered Devices
Quick-PWM is a trademark of Maxim Integrated Products, Inc.
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
2 _______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(VIN = 12V, VDD = VCC = EN1 = EN2 = 5V, VREFIN1 = 2V, SKIP = GND, TA= 0 to +85°C, unless otherwise noted. Typical values are
at TA= +25°C.)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
BST1, BST2 to GND ...............................................-0.3V to +34V
BST1, BST2 to VDD.................................................-0.3V to +28V
TON1, TON2 to GND..............................................-0.3V to +28V
VDD to GND..............................................................-0.3V to +6V
VDD to VCC ............................................................-0.3V to +0.3V
LX1 to BST1..............................................................-6V to +0.3V
LX2 to BST2..............................................................-6V to +0.3V
DH1 to LX1 ..............................................-0.3V to (VBST1 + 0.3V)
DH2 to LX2 ..............................................-0.3V to (VBST2 + 0.3V)
ILIM1, ILIM2, REF to GND ..........................-0.3V to (VCC + 0.3V)
CSH1, CSH2, CSL1, CSL2, FB2, REFIN1 to GND....-0.3V to +6V
EN1, EN2, SKIP, PGOOD1, PGOOD2 to GND.........-0.3V to +6V
DL1 to GND ................................................-0.3V to (VDD + 0.3V)
DL2 to PGND..............................................-0.3V to (VDD + 0.3V)
PGND to GND ......................................................-0.3V to + 0.3V
REF Short Circuit to GND ...........................................Continuous
Continuous Power Dissipation (TA= +70°C)
28-Pin TQFN T2844-1
(derate 20.8mW/°C above +70°C) ............................1667mW
Extended Operating Temperature Range .........-40°C to +105°C
Junction Temperature......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering 10s) ..................................+300°C
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
PWM CONTROLLER
Input Voltage Range VIN 4.5 26 V
Quiescent Supply Current
(VDD, VCC)IDD + ICC Output forced above regulation voltage,
EN1 = EN2 = 5V 1.7 2.5 mA
Shutdown Supply Current
(VDD, VCC)ISHDN EN1 = EN2 = GND, TA = +25°C 0.1 5 µA
RTON1 = RTON2 = 97.5k
(600kHz)
142
(-15%) 174 194
(+15%)
RTON1 = RTON2 = 200k
(300kHz)
305
(-10%) 336 368
(+10%)
On-Time (Note 1) tON1, tON2
VIN = 12V,
VCSL1 = VCSL2
= VCCI = 1.2V,
separate or
combined
mode
RTON1 = RTON2 = 302.5k
(200kHz)
425
(-15%) 500 575
(+15%)
ns
Minimum Off-Time tOFF(MIN) (Note 1) 250 330 ns
TON1, TON2, Shutdown Supply
Current
ITON1,
ITON2
EN1 = EN2 = GND, VTON1 = VTON2 = 26V,
VDD = 0V or 5V, TA = +25°C 0.01 1 µA
REFIN1 Voltage Range VREFIN1 (Note 2) 0 VREF V
FB2 Regulation Voltage VFB2 Adjustable mode 0.7 V
FB2 Input Voltage Range Preset mode 1.7 2.3 V
FB2 Combined-Mode Threshold Combined mode 3.8 VCC -
1V
VCC -
0.4 V
REFIN1 Dual Mode™
Switchover Threshold 3.8
VCC -
1V
VCC -
0.4 V
REFIN1, FB2 Bias Current IREFIN1,
IFB2
REFIN1 = 0.5V to 2V;
VFB2 = 0.7V, TA = +25°C -0.1 +0.1 µA
SMPS1 Voltage Accuracy VCSL1 Measured at CSL1, REFIN1 = VCC,
VIN = 2V to 26V, SKIP = VCC (Note 2) 1.043 1.05 1.057 V
SMPS2 Voltage Accuracy VCSL2 Measured at CSL2, FB2 = REF,
VIN = 2V to 26V, SKIP = VCC (Note 2) 1.489 1.5 1.511 V
Dual Mode is a trademark of Maxim Integrated Products, Inc.
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
_______________________________________________________________________________________ 3
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Load Regulation Error ILOAD = 0 to 3A, SKIP = VCC (Note 3) 0.1 %
Line Regulation Error VDD = 4.5V to 5.5V, VIN = 4.5V to 26V (Note 3) 0.25 %
CSL1 Soft-Start/-Stop Slew Rate SRSS1 Rising/falling edge on EN1 1.25 mV/µs
FB2 Soft-Start/-Stop Slew Rate SRSS2 Rising/falling edge on EN2 0.63 mV/µs
Dynamic REFIN1 Slew Rate SRDYN Rising edge on REFIN1 11.4 mV/µs
INTERNAL REFERENCE
Reference Voltage VREF VDD = 4.5V to 5.5V 1.990 2.000 2.010 V
Reference Lockout Voltage VREF(UVLO) Rising edge, hysteresis = 230mV 1.8 V
Reference Load Regulation IREF = -10µA to +100µA 1.980 2.015 mV
FAULT DETECTION
With respect to the internal target voltage
(error comparator threshold); rising edge;
hysteresis = 50mV
260 300 340 mV
Dynamic transition VREF +
0.30 V
SMPS1 Overvoltage Trip
Threshold and PGOOD1 Upper
Threshold
VOVP1,
VPG1_H
Minimum OVP threshold 0.7 V
SMPS2 Adjustable Mode
Overvoltage Trip Threshold and
PGOOD2 Upper Threshold
VOVP2,
VPG2_H
With respect to the internal target voltage
0.7V (error comparator threshold);
hysteresis = 50mV
120 150 180 mV
Output Overvoltage Fault
Propagation Delay tOVP CSL1/FB2 forced 25mV above trip threshold 5 µs
SMPS1 Undervoltage Protection
Trip Threshold and Lower
PGOOD1 Threshold
VUVP1,
VPG1_L
With respect to the internal target voltage
(error comparator threshold); falling edge;
hysteresis = 50mV
-240 -200 -160 mV
SMPS2 Undervoltage Protection
Trip Threshold and Lower
PGOOD2 Threshold
VUVP2,
VPG2_L
With respect to the internal target voltage
0.7V (error comparator threshold);
falling edge; hysteresis = 50mV
-130 -100 -70 mV
Output Undervoltage Fault
Propagation Delay tUVP CSL1/FB2 forced 25mV below trip threshold 110 205 310 µs
UVP falling edge, 25mV overdrive 5
OVP rising edge, 25mV overdrive 5
PGOOD_ Propagation Delay tPGOOD
Startup delay from regulation 100 200 300
µs
PGOOD_ Output Low Voltage ISINK = 3mA 0.4 V
PGOOD_ Leakage Current IPGOOD
CSL1 = REFIN1, FB2 = 0.7V (PGOOD_ high
impedance), PGOOD_ forced to 5V,
TA = +25°C
1 µA
Dynamic REFIN1 Transition Fault-
Blanking Threshold
Fault blanking initiated; REFIN1 deviation
from the internal target voltage (error
comparator threshold); hysteresis = 10mV
±50 mV
Thermal-Shutdown Threshold TSHDN Hysteresis = 15°C (Note 3) 160 °C
VCC Undervoltage Lockout
Threshold VUVLO(VCC) Rising edge, PWM disabled below this level,
hysteresis = 100mV 3.95 4.20 4.45 V
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V, VDD = VCC = EN1 = EN2 = 5V, VREFIN1 = 2V, SKIP = GND, TA= 0 to +85°C, unless otherwise noted. Typical values are
at TA= +25°C.)
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
4 _______________________________________________________________________________________
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V, VDD = VCC = EN1 = EN2 = 5V, VREFIN1 = 2V, SKIP = GND, TA= 0 to +85°C, unless otherwise noted. Typical values are
at TA= +25°C.)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
CURRENT LIMIT
CSH1, CSH2 0 2.3
Current-Sense Input Range CSL1, CSL2 0 2.3 V
Current-Sense Input (CSH_)
Leakage Current CSH_ = GND or VCC, TA = +25°C -0.2 +0.2 µA
Current-Sense Input (CSL_)
Leakage Current CSL_= CSL_ = 2V, TA = +25°C 1 µA
TA = +25°C 28 30 32
VCSH_ - VCSL_
ILIM1 = ILIM2 = REF TA = 0°C to +85°C 27 30 33
VCSH_ - VCSL_, ILIM1 = ILIM2 = VCC 56 60 64
VCSH_ - VCSL_, ILIM1 = ILIM2 = OPEN 42 45 48
Current-Limit Threshold (Fixed) VCSLIMT_
VCSH_ - VCSL_, ILIM1 = ILIM2 = GND 13 15 17
mV
Current-Limit Threshold
(Negative) VNEG VCSH_ - VCSL_,SKIP = VCC -1.2 x
VCSLIMIT_ mV
Current-Limit Threshold
(Zero Crossing) VZX VCSH_ - VCSL_,SKIP = GND or OPEN;
ILIM1 = ILIM2 = REF 1 mV
Ultrasonic Frequency SKIP = OPEN (3.3V); VCSL1 = VREFIN1 +
50mV; VCSL2 = VFB2 + 50mV 20 kHz
VCSL1 = VREF1 + 50mV 22 33 46
Ultrasonic Current-Limit
Threshold SKIP = OPEN (3.3V) VCSL2 = VFB2+ 50mV 18 30 46 mV
Current-Balance Amplifier (GMI)
Offset [V(CSH1,CSL1) - V(CSH2,CSL2)] at ICCI = 0 -3 +3 mV
Current-Balance Amplifier (GMI)
Transconductance
ICCI / [V(CSH1,CSL1) - V(CSH2,CSL2)];
VCCI = VCSL1 = VCSL2 = 0.5V to 2V, and
V(CSH_,CSL_) = -60.0mV to +60.0mV,
ILIM1 = GND
180 µS
GATE DRIVERS
Low state (pulldown) 1.7 4.0
DH1, DH2 Gate Driver
On-Resistance RON(DH) BST_ - LX_ forced
to 5V High state (pullup) 1.7 4.0
High state (pullup) 1.3 3.0
DL1, DL2 Gate Driver
On-Resistance RON(DL) Low state (pulldown) 0.6 2.5
DH1, DH2 Gate Driver
Source/Sink Current IDH DH_ forced to 2.5V, BST_ - LX_ forced to 5V 1.2 A
DL1, DL2 Gate Driver
Source Current IDL(SOURCE) DL_ forced to 2.5V 1 A
DL1, DL2 Gate Driver
Sink Current IDL(SINK) DL_ forced to 2.5V 2.4 A
DH_ low to DL high 10 25 40
Driver Propagation Delay DL_ low to DH high 15 30 45 ns
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
_______________________________________________________________________________________ 5
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V, VDD = VCC = EN1 = EN2 = 5V, VREFIN1 = 2V, SKIP = GND, TA= 0 to +85°C, unless otherwise noted. Typical values are
at TA= +25°C.)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
DL_ falling, CDL = 3nF 10 20
DL_ Transition Time DL_ rising, CDL = 3nF 10 20 ns
DH_ falling, CDH = 3nF 10 20
DH_ Transition Time DH_ rising, CDH = 3nF 10 20 ns
Internal BST_ Switch
On-Resistance RBST_ IBST_ = 10mA, VDD = 5V 6.5 11.0
LX_, BST_ Leakage Current VBST_ = VLX_ = 26V, TA = +25°C 5 µA
INPUTS AND OUTPUTS
EN1, EN2 Logic-Input Threshold EN1, EN2 rising edge,
hysteresis = 300mV/600mV (min/max) 1.20 1.70 2.20 V
Logic-Input Current EN1, EN2, TA = +25°C -0.5 +0.5 µA
High (5V) VCC -
0.4
Open (3.3V) 3.0 3.6
Ref (2.0V) 1.7 2.3
Quad-Level Input-Logic Levels SKIP, ILIM1, ILIM2
Low (GND) 0.4
V
Quad-Level Logic-Input Current SKIP, ILIM1, ILIM2 forced to GND or VCC,
TA = +25°C -2 +2 µA
ELECTRICAL CHARACTERISTICS
(VIN = 12V, VDD = VCC = EN1 = EN2 = 5V, VREFIN1 = 2V, SKIP = GND, TA= -40°C to +105°C, unless otherwise noted.) (Note 4)
PARAMETER SYMBOL CONDITIONS MIN MAX UNITS
PWM CONTROLLER
Input Voltage Range VIN 4.5 26 V
Quiescent Supply Current
(VDD, VCC)IDD + ICC Output forced above regulation voltage,
EN1 = EN2 = 5V 2.5 mA
RTON1 = RTON2 =
97.5k (600kHz) 142 194
RTON1 = RTON2 =
200k (300kHz) 305 368
On-Time (Note 1) tON1,
tON2
VIN = 12V,
VCSL1 = VCSL2 =
VCCI = 1.2V,
separate or
combined mode RTON1 = RTON2 =
302.5k (200kHz) 425 575
ns
Minimum Off-Time tOFF(MIN) (Note 1) 330 ns
REFIN1 Voltage Range VREFIN1 0 VREF V
FB2 Input Voltage Range Preset mode 1.7 2.3 V
FB2 Combined-Mode Threshold Combined mode 3.75 VCC -
0.4 V
REFIN, FB2 Bias Current IREFIN1,
IFB2 -0.1 +0.1 µA
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
6 _______________________________________________________________________________________
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V, VDD = VCC = EN1 = EN2 = 5V, VREFIN1 = 2V, SKIP = GND, TA= -40°C to +105°C, unless otherwise noted.) (Note 4)
PARAMETER SYMBOL CONDITIONS MIN MAX UNITS
REFIN1 Dual-Mode
Switchover Threshold 3.75
VCC -
0.4 V
SMPS1 Voltage Accuracy VCSL1 Measured at CSL1, REFIN1 = VCC;
VIN = 2V to 26V, SKIP = VCC (Note 2) 1.039 1.061 V
SMPS2 Voltage Accuracy VCSL2 Measured at CSL2, FB2 = REF;
VIN = 2V to 26V, SKIP = VCC (Note 2) 1.485 1.515 V
INTERNAL REFERENCE
Reference Voltage VREF VDD = 4.5V to 5.5V 1.985 2.015 V
FAULT DETECTION
SMPS1 Overvoltage Trip
Threshold and PGOOD1 Upper
Threshold
VOVP1,
VPG1_H
With respect to the internal target voltage
(error comparator threshold); rising edge;
hysteresis = 50mV
260 340 mV
SMPS2 Overvoltage Trip
Threshold and PGOOD2 Upper
Threshold
VOVP2,
VPG2_H
With respect to the internal target voltage
0.7V (error comparator threshold);
hysteresis = 50mV
120 180 mV
SMPS1 Undervoltage Protection
Trip Threshold and Lower
PGOOD1 Threshold
VUVP1,
VPG1_L
With respect to the internal target voltage
(error comparator threshold) falling edge;
hysteresis = 50mV
-240 -160 mV
SMPS2 Undervoltage Protection
Trip Threshold and Lower
PGOOD2 Threshold
VUVP2,
VPG2_L
With respect to the internal target voltage
0.7V (error comparator threshold)
falling edge; hysteresis = 50mV
-130 -70 mV
Output Undervoltage Fault
Propagation Delay tUVP REFIN1/FB2 forced 25mV below trip
threshold 80 320 µs
PGOOD_ Propagation Delay tPGOOD Startup delay from regulation 80 320 µs
PGOOD_ Output Low Voltage ISINK = 3mA 0.4 V
VCC Undervoltage Lockout
Threshold VUVLO(VCC) Rising edge, PWM disabled below this level;
hysteresis = 100mV 3.8 4.45 V
CURRENT LIMIT
CSH1, CSH2 0 2.3
Current-Sense Input Range CSL1, CSL2 0 2.3 V
Current-Limit Threshold (Fixed) VCSLIMT VCSH_ - VCSL_, ILIM1 = ILIM2 = REF 27 33 mV
Ultrasonic Frequency
SKIP = OPEN (3.3V);
VCSL1 = VREFIN1 + 50mV;
VCSL2 = VFB2 + 50mV
18 kHz
VCSL1 = VREF1 + 50mV 22 46
Ultrasonic Current-Limit
Threshold SKIP = OPEN (3.3V) VCSL2 = VFB2 + 50mV 18 46
mV
Current-Balance Amplifier (GMI)
Offset [V(CSH1,CSL1) - V(CSH2,CSL2)] at ICCI = 0 -3 +3 mV
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
_______________________________________________________________________________________ 7
Note 1: On-time and off-time specifications are measured from 50% point to 50% point at the DH pin with LX = GND, VBST = 5V, and
a 250pF capacitor connected from DH to LX. Actual in-circuit times might differ due to MOSFET switching speeds.
Note 2: The 0 to 0.5V range is guaranteed by design, not production tested.
Note 3: Not production tested.
Note 4: Specifications to TA= -40°C to +105°C are guaranteed by design, not production tested.
ELECTRICAL CHARACTERISTICS (continued)
(VIN = 12V, VDD = VCC = EN1 = EN2 = 5V, VREFIN1 = 2V, SKIP = GND, TA= -40°C to +105°C, unless otherwise noted.) (Note 4)
PARAMETER SYMBOL CONDITIONS MIN MAX UNITS
GATE DRIVERS
Low state (pulldown) 4.5
DH1, DH2 Gate Driver
On-Resistance RON(DH) BST_ - LX_ forced to
5V High state (pullup) 4.0
High state (pullup) 3
DL1, DL2 Gate Driver
On-Resistance RON(DL) Low state (pulldown) 2
DH_ low to DL high 8 42
Driver Propagation Delay DL_ low to DH high 12 48 ns
Internal BST_ Switch
On-Resistance RBST_ IBST_ = 10mA, VDD = 5V 11
INPUTS AND OUTPUTS
EN1, EN2 Logic-Input Threshold EN1, EN2 rising edge;
hysteresis = 300mV/600mV (min/max) 1.20 2.20 V
High (5V) VCC -
0.4
Open (3.3V) 3.0 3.6
Ref (2.0V) 1.7 2.3
Quad-Level Input Logic Levels SKIP, ILIM1, ILIM2
Low (GND) 0.4
V
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
8 _______________________________________________________________________________________
Typical Operating Characteristics
(Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, TA= +25°C, unless otherwise noted.)
SMPS2 1.5V EFFICIENCY
vs. LOAD CURRENT
MAX17007/8 toc01
LOAD CURRENT (A)
EFFICIENCY (%)
1010.1
50
60
70
80
90
100
40
20
30
10
0.01 100
6V
12V 20V
SKIP MODE
PWM MODE
SMPS2 1.5V EFFICIENCY
vs. LOAD CURRENT
MAX17007/8 toc02
LOAD CURRENT (A)
EFFICIENCY (%)
1010.1
50
60
70
80
90
100
40
20
30
10
0.01 100
SKIP MODE
PWM MODE
VIN = 12V
ULTRASONIC
MODE
SMPS2 1.5V OUTPUT VOLTAGE
vs. LOAD CURRENT
MAX17007/8 toc03
LOAD CURRENT (A)
OUTPUT VOLTAGE (V)
105
1.48
1.50
1.52
1.54
1.46
015
SKIP MODE
ULTRASONIC MODE
PWM
VIN = 12V
COMBINED 1.2V EFFICIENCY
vs. LOAD CURRENT
MAX17007/8 toc04
LOAD CURRENT (A)
EFFICIENCY (%)
1010.1
50
60
70
80
90
100
40
20
30
10
0.01 100
SKIP MODE
PWM MODE
6V
12V 20V
COMBINED 1.2V OUTPUT VOLTAGE
vs. LOAD CURRENT
MAX17007/8 toc05
LOAD CURRENT (A)
OUTPUT VOLTAGE (V)
16 20842412
1.19
1.20
1.21
1.22
1.18
028
SKIP MODE
PWM
VIN = 12V
SMPS2 SWITCHING FREQUENCY
vs. LOAD CURRENT
MAX17007/8 toc06
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
1010.1
200
250
300
350
150
50
100
0
0.01 100
SKIP MODE
PWM MODE
VIN = 12V
ULTRASONIC
MODE
SMPS2 SWITCHING FREQUENCY
vs. INPUT VOLTAGE
MAX17007/8 toc07
INPUT VOLTAGE (V)
SWITCHING FREQUENCY (kHz)
20424
250
300
350
200
02881216
VIN = 12V
SKIP = 5V
IOUT2 = 5A
IOUT2 = 0A
SMPS2 SWITCHING FREQUENCY
vs. TEMPERATURE
MAX17007/8 toc08
TEMPERATURE (°C)
SWITCHING FREQUENCY (kHz)
040-20 60
290
270
310
330
250
-40 1201008020
VIN = 12V
SKIP = 5V
IOUT2 = 5A
IOUT2 = 0A
SMPS2 MAXIMUM OUTPUT CURRENT
vs. INPUT VOLTAGE
MAX17007/8 toc09
INPUT VOLTAGE (V)
MAXIMUM OUTPUT CURRENT (A)
20424
12
11
13
14
10
02881216
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
_______________________________________________________________________________________
9
SMPS2 MAXIMUM OUTPUT CURRENT
vs. TEMPERATURE
MAX17007/8 toc10
TEMPERATURE (°C)
MAXIMUM OUTPUT CURRENT (A)
80400
11
12
13
14
10
-40 1206020-20 100
VIN = 12V
NO-LOAD SUPPLY CURRENT
vs. INPUT VOLTAGE
MAX17007/8 toc11
INPUT VOLTAGE (V)
SUPPLY CURRERT (IBIAS) (mA)
2016
4
8
12
16
0
6
10
14
2
424812
SKIP MODE
PWM MODE
ULTRASONIC MODE
EN1 = HIGH
EN2 = LOW
NO-LOAD INPUT CURRENT
vs. INPUT VOLTAGE
MAX17007/8 toc12
INPUT VOLTAGE (V)
INPUT CURRENT (mA)
2016
0.1
1
10
100
0.01
424812 1814 22610
SKIP MODE
PWM MODE
ULTRASONIC MODE
EN1 = HIGH
EN2 = LOW
REFERENCE VOLTAGE
vs. REFERENCE LOAD CURRENT
MAX17007/8 toc13
REFERENCE LOAD CURRENT (µA)
REFERENCE VOLTAGE (V)
8060
1.99
1.97
2.01
2.03
2.05
1.95
-20 10020 400
REFIN1 TO CSL1 OFFSET VOLTAGE
DISTRIBUTION
MAX17007/8 toc14
OFFSET VOLTAGE (mV)
SAMPLE PERCENTAGE (%)
3.0
60
50
70
80
20
10
30
40
90
0
-5.0 5.0-1.0 1.0-3.0
SAMPLE SIZE = 100
TA = +85°C
TA = +25°C
SMPS1 PRESET 1.05V
VOLTAGE DISTRIBUTION
MAX17007/8 toc15
SMPS1 VOLTAGE (mV)
SAMPLE PERCENTAGE (%)
1.053
60
50
70
80
20
10
30
40
90
0
1.045 1.0551.049 1.0511.047
SAMPLE SIZE = 100
TA = +85°C
TA = +25°C
SMPS2 PRESET 1.5V
VOLTAGE DISTRIBUTION
MAX17007/8 toc16
SMPS2 VOLTAGE (mV)
SAMPLE PERCENTAGE (%)
1.503
15
10
20
25
5
30
0
1.495 1.5051.499 1.5011.497
SAMPLE SIZE = 100
TA = +85°C
TA = +25°C
COMBINED-MODE CURRENT BALANCE
vs. LOAD CURRENT
MAX17007/8 toc17
LOAD CURRENT (A)
VCSH - VCSL (mV)
201552510
50
40
20
10
30
0
030
SMPS1
SMPS2
SOFT-START WAVEFORM
MAX17007/8 toc18
400µs/div
A
B
E
F
C
D
5V
A. EN1, EN2, 5V/div
B. REF, 2V/div
C. VOUT1, 1V/div
D. VOUT2, 1V/div
E. PGOOD1, 5V/div
F. PGOOD2, 5V/div
0
2V
0
5V
0
0
5V
0
1.05V
1.5V
0
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, TA= +25°C, unless otherwise noted.)
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
10 ______________________________________________________________________________________
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, TA= +25°C, unless otherwise noted.)
SMPS1 STARTUP WAVEFORM
(HEAVY LOAD)
MAX17007/8 toc19
200µs/div
A
B
E
F
G
C
D
5V
A. EN1, 5V/div
B. REF, 2V/div
C. VOUT1, 500mV/div
D. ILX1, 10A/div
E. PGOOD1, 10V/div
F. LX1, 10V/div
G. DL1, 10V/div
0
2V
5V
0
12V
0
5V
0
0
1.05V
0
8A
IOUT1 = 8A
SMPS1 STARTUP WAVEFORM
(LIGHT LOAD)
MAX17007/8 toc20
200µs/div
A
B
E
F
G
C
D
5V
A. EN1, 5V/div
B. REF, 2V/div
C. VOUT1, 500mV/div
D. ILX1, 5A/div
SKIP = 5V
IOUT1 = 2A
E. PGOOD1, 10V/div
F. LX1, 10V/div
G. DL1, 10V/div
0
2V
0
1.05V
0
2A
0
5V
0
5V
0
0
12V
SMPS1 SHUTDOWN WAVEFORM
MAX17007/8 toc21
200µs/div
A
B
E
F
G
C
D
5V
A. EN1, 5V/div
B. REF, 5V/div
C. VOUT1, 500mV/div
D. ILX1, 5A/div
E. PGOOD1, 10V/div
F. LX1, 10V/div
G. DL1, 10V/div
IOUT1 = 0.5A
SKIP = GND
0
2V
1.05V
0
0
0
5V
0
5V
0
12V
SMPS2 LOAD-TRANSIENT RESPONSE
(PWM MODE)
MAX17007/8 toc22
20µs/div
A
B
C
A. VOUT2, 50mV/div
B. ILX2, 10A/div
C. LX2, 10V/div
1.5V
10A
2A
12V
0
IOUT2 = 2A TO 10A TO 2A
SKIP = 5V
SMPS2 LOAD-TRANSIENT RESPONSE
(SKIP MODE)
MAX17007/8 toc23
20µs/div
A
B
C
A. VOUT2, 50mV/div
B. ILX2, 10A/div
C. LX2, 10V/div
1.5V
8A
0A
12V
0
IOUT2 = 0.5A TO 8.5A TO 0.5A
SKIP = GND
SMPS1 OUTPUT OVERLOAD WAVEFORM
MAX17007/8 toc24
200µs/div
A
B
E
C
D
IOUT1 = 2A TO 15A
A. VOUT1, 500mV/div
B. ILX1, 10A/div
C. LX1, 10V/div
1.05V
10A
2A
5V
0
5V
12V
0
0
D. PGOOD1, 5V/div
E. DL1 5V/div
SMPS1 OUTPUT OVERVOLTAGE
WAVEFORM
MAX17007/8 toc25
40µs/div
A
B
C
A. VOUT1, 1V/div
B. PGOOD1, 5V/div
C. DL1, 5V/div
1.05V
0
5V
0
5V
0
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 11
DYNAMIC OUTPUT VOLTAGE
TRANSITION (PWM MODE)
MAX17007/8 toc26
20µs/div
A
B
C
D
IOUT1 = 2A
REFIN1 = 1V
TO 1.2V TO 1V
1.2V
1V
0
2A
12V
0
5V
A. VOUT1, 100mV/div
B. ILX1, 10A/div
C. LX1, 10V/div
D. DL1, 5V/div
SKIP = 5V
DYNAMIC OUTPUT VOLTAGE
TRANSITION (SKIP MODE)
MAX17007/8 toc27
40µs/div
A
B
C
D
IOUT1 = 1A
1.2V
1V
0
0
12V
0
5V
A. VOUT1, 100mV/div
B. ILX1, 10A/div
C. LX1, 10V/div
D. DL1, 5V/div
REFIN1 = 1V
TO 1.2V TO 1V
SKIP = GND
DYNAMIC OUTPUT VOLTAGE TRANSITION
(SKIP MODE-FORCED TRANSITION)
MAX17007/8 toc28
20µs/div
A
B
C
D
IOUT1 = 3A
1.2V
1V
0
0
12V
0
5V
SKIP = REF
A. VOUT1, 100mV/div
B. ILX1, 10A/div
C. LX1, 10V/div
D. DL1, 5V/div
REFIN1 = 1V TO 1.2V TO 1V
Pin Description
Typical Operating Characteristics (continued)
(Circuit of Figure 1, VIN = 12V, VDD = 5V, SKIP = GND, TA= +25°C, unless otherwise noted.)
PIN NAME FUNCTION
1 REF
2V Reference Voltage Output. Bypass REF to GND with a 2.2nF ceramic capacitor. The reference can
source up to 100µA. Loading REF degrades output-voltage accuracy according to the REF load
regulation error (see theTypical Operating Characteristics). The reference shuts down when both EN1
and EN2 are low.
2 ILIM1
This four-level input determines the CSH1 to CSL1 current limit for SMPS1:
VDD (5V) = 60mV current limit
Open (3.3V) = 45mV current limit
REF (2V) = 30mV current limit
GND = 15mV current limit
In combined mode, ILIM1 sets the current-limit threshold for both sides.
This four-level input determines the CSH2 to CSL2 current limit for SMPS2:
VCC (5V) = 60mV current limit
Open (3.3V) = 45mV current limit
REF (2V) = 30mV current limit
GND = 15mV current limit
In combined mode, ILIM2 is the current balance integrator (CCI) output pin. Connect a capacitor
(CCCI) between CCI and the output. The CCI capacitor value depends on the ILIM1 setting based on
the following table:
ILIM1 CCCI at ILIM2 (pF)
VCC (5V) 120
Open (3.3V) 180
REF (2V) 220
3ILIM2
(CCI)
GND 470
4V
CC 5V Analog Supply Input. Bypass VCC from VDD using a 10_ resistor, and to analog ground using a
1µF ceramic capacitor.
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
12 ______________________________________________________________________________________
Pin Description (continued)
PIN NAME FUNCTION
5SKIP
Pulse-Skipping Control Input. This four-level input determines the mode of operation under normal
steady-state conditions and dynamic output-voltage transitions:
VDD (5V) = Forced-PWM operation
Open (3.3V) = Ultrasonic mode (without forced-PWM during transitions)
REF (2V) = Pulse-skipping mode (with forced-PWM during transitions)
GND = Pulse-skipping mode (without forced-PWM during transitions)
There are no dynamic transitions for SMPS2, so SKIP = 2V and SKIP = GND have the same pulse-
skipping behavior for SMPS2 without any forced-PWM transitions.
In combined mode, the ultrasonic mode is disabled, and the SKIP = OPEN (3.3V) setting is identical
to the SKIP = GND setting.
6 TON1
Frequency-Setting Input for SMPS1. An external resistor between the input power source and TON1
sets the switching period (TSW1) of SMPS1:
TSW1 = CTON (RTON1 + 6.5k)
where CTON = 16.26pF.
TON1 is high impedance in shutdown.
In combined mode, TON1 sets the switching period for both SMPS1 and SMPS2.
7 TON2
Frequency-Setting Input for SMPS2. An external resistor between the input power source and TON2
sets the switching period (TSW2) of SMPS2:
TSW2 = CTON (RTON2 + 6.5k)
where CTON = 16.26pF.
Set TON2 to a switching frequency different from TON1. A 10% to 30% difference in switching
frequency between SMPS1 and SMPS2 is recommended.
TON2 is high impedance in shutdown.
In combined mode, TON2 cannot be left open.
8 REFIN1
External Reference Input for SMPS1. REFIN1 sets the feedback regulation voltage of CSL1. SMPS1
includes an internal window comparator to detect REFIN1 voltage changes that are greater than
±50mV (typ), allowing the controller to blank PGOOD1 and the fault protection, and force the output
transition, if enabled. When REFIN1 is tied to VCC, SMPS1 regulates the output to 1.05V.
In combined mode, REFIN1 sets the feedback regulation voltage of the combined output.
9 CSL1
Output-Sense and Negative Current-Sense Input for SMPS1. When using the internal preset 1.05V
feedback divider (REFIN1 = VCC), the controller uses CSL1 to sense the output voltage. Connect to
the negative terminal of the current-sense element. Figure 14 describes two different current-sensing
options—using accurate sense resistors or lossless inductor DCR sensing.
10 CSH1
Positive Current-Sense Input for SMPS1. Connect to the positive terminal of the current-sense
element. Figure 14 describes two different current-sensing options—using accurate sense resistors
or lossless inductor DCR sensing.
11 EN1
Enable Control Input for SMPS1. Connect to VCC for normal operation. Pull EN1 low to disable
SMPS1. The controller slowly ramps down the output voltage to ground and after the target voltage
reaches 0.1V, the controller forces DL1 low. When both EN1 and EN2 are low, the device enters the
low-power shutdown state.
In combined mode, EN1 controls the combined SMPS output. EN2 is unused and must be grounded.
12 PGOOD1
Open-Drain Power-Good Output for SMPS1. PGOOD1 is low when the SMPS1 voltage is more than 200mV
below or 300mV above the target voltage, during soft-start, and in shutdown. After the SMPS1 soft-start
circuit has terminated, PGOOD1 becomes high impedance 200µs after the output is in regulation. PGOOD1
is blanked (forced high-impedance state) when a dynamic REFIN1 transition is detected.
13 DH1 High-Side Gate-Driver Output for SMPS1. DH1 swings from LX1 to BST1. DH1 is low in shutdown.
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 13
Pin Description (continued)
PIN NAME FUNCTION
14 LX1
Inductor Connection for SMPS1. Connect LX1 to the switched side of the inductor. LX1 serves as the
lower supply rail for the DH1 high-side gate driver.
15 BST1
Bootstrap Capacitor Connection for SMPS1. The MAX17007/MAX17008 include an internal boost
switch/diode connected between VDD and BST1. Connect to an external capacitor as shown in Figure 1.
16 GND Ground. Analog and power ground connection for the low-side gate driver of SMPS1.
17 DL1
Low-Side Gate Driver Output for SMPS1. DL1 swings from GND to VDD. DL1 is forced low after the shutdown
sequence has completed. DL1 is also forced high when an output overvoltage fault is detected, overriding any
negative current-limit condition that may be present. DL1 is forced low in VCC UVLO.
18 VDD 5V Driver Supply Input. Connect VDD to VCC through a 10 resistor. Bypass to ground through a 1µF or
greater ceramic capacitor. VDD is internally connected to the BST diodes and the low-side gate drivers.
19 DL2
Low-Side Gate-Driver Output for SMPS2. DL2 swings from PGND to VDD. DL2 is forced low after the
shutdown sequence has completed. DL2 is also forced high when an output overvoltage fault is
detected, overriding any negative current-limit condition that may be present. DL2 is forced low in VCC
UVLO.
20 PGND Power Ground for the Low-Side Gate Driver of SMPS2
21 BST2
Bootstrap Capacitor Connection for SMPS2. The MAX17007/MAX17008 include an internal boost switch/
diode connected between VDD and BST2. Connect to an external capacitor as shown in Figure 1.
22 LX2
Inductor Connection for SMPS2. Connect LX2 to the switched side of the inductor. LX2 serves as the
lower supply rail for the DH2 high-side gate driver.
23 DH2 High-Side Gate-Driver Output for SMPS2. DH2 swings from LX2 to BST2. DH2 is low in shutdown.
24 PGOOD2
Open-Drain Power-Good Output for SMPS2. PGOOD2 is low when the FB2 voltage is more than 100mV
below or 150mV above the target voltage, during soft-start, and in shutdown. After the SMPS2 soft-start
circuit has terminated, PGOOD2 becomes high impedance 200µs after the output is in regulation.
In combined mode, PGOOD2 is not used and can be left open.
25 EN2
SMPS2 Enable Input. Connect to VCC for normal operation. Pull EN2 low to disable SMPS2. The controller
slowly ramps down the output voltage to ground, and after the target voltage reaches 0.1V, the controller
forces DL2 low. When both EN1 and EN2 are low, the device enters the low-power shutdown state.
In combined mode, EN2 is not used and should be connected to GND.
26 CSH2
Positive Current-Sense Input for SMPS2. Connect to the positive terminal of the current-sense element.
Figure 14 describes two different current-sensing options—using accurate sense resistors or lossless
inductor DCR sensing.
27 CSL2
Output-Sense and Negative Current-Sense Input for SMPS2. When using the internal preset 1.5V
feedback divider (FB2 = REF), the controller uses CSL2 to sense the output voltage. Connect to the
negative terminal of the current-sense element. Figure 14 describes two different current-sensing
options—using accurate sense resistors or lossless inductor DCR sensing.
28 FB2
SMPS2 Feedback Input. Adjust the SMPS2 voltage with a resistive voltage-divider between SMPS2 output
and GND. Connect FB2 to REF for preset 1.5V output. Tie FB2 to VCC to configure the
MAX17007/MAX17008 for combined-mode operation.
EP PAD Exposed Backside Pad. Connect to analog ground.
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
14 ______________________________________________________________________________________
MAX17007
MAX17008
6
TON1
AGND
RREFIN1
RREFIN2
16
CVCC
1µF
R2
100kΩ
TO SYSTEM
POWER-GOOD
R9
10Ω
RTON1
220kΩ
RNTC1
10kΩ
R4
3.01kΩ
RTON2
180kΩ
CREF
2.2nF
RREFIN3
GND
PWR
PWR
PWR
AGND
PWR
AGND
2ILIM1
3ILIM2
(CCI)
5SKIP
11 EN1
25 EN2
1REF
RREFIN1 = 80.6kΩ
RREFIN2 = 121kΩ
RREFIN3 = 249kΩ8REFIN1
12 PGOOD1
24 PGOOD2
7
TON2
15
BST1
13
DH1
14
LX1
17
DL1
NH1
NL1
20
PGND
10
CSH1
9
CSL1
28
FB2
4
VCC
18
VDD
REF
REF
+3.3V
H = 1.0V
L = 1.2V
4-LEVEL SKIP PIN
REF
+5V
CONNECT TO REF FOR
FIXED 1.5V OUTPUT
*LOWER INPUT VOLTAGES REQUIRE
ADDITIONAL INPUT CAPACITANCE.
VIN
7V TO 20V
EP
POWER GROUND
ANALOG GROUND
R1
100kΩ
CVDD
2.2µF
C1
0.22µF
CBST1
0.1µF
PWR
DL1
CIN1
R3
1.5kΩ
COUT1
2 x 330µF
12mΩ
L1
1µH, 16A, 3mΩVOUT1
1.2V/1.0V, 12A
ILIM1
ILIM2
VCC
OPEN
REF
GND
R7
10Ω
C2
1nF
PWR
COUT1-CER
5 x 10µF
CERAMIC
VOUT2
1.5V, 12A
RNTC2
10kΩ
R6
3.01kΩ
PWR
AGND
PWR
21
BST2
23
DH2
22
LX2
19
DL2
NH2
NL2
26
CSH2
27
CSL2
VIN
7V TO 20V
C3
0.22µF
CBST2
0.1µF
PWR
DL2
CIN2
R5
1.5kΩ
COUT2
2 x 330µF
12mΩ
L2
1µH, 16A, 3mΩ
R8
10Ω
C4
1nF
PWR
COUT2-CER
5 x 10µF
CERAMIC
CURRENT
LIMIT
60mV
45mV
30mV
15mV
Figure 1. MAX17007/MAX17008 Separate-Mode Standard Application Circuit
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 15
VOUT1 = 1.0V/1.2V AT 12A
(FIGURE 1)
VOUT = 1.5V AT 12A
(FIGURE 1)
COMPONENT VIN = 7V to 20V
TON1 = 220 k (270kHz)
VIN = 7V to 20V
TON2 = 180 k (330kHz)
Input Capacitor
(per Phase)
(2x) 10µF, 25V
Taiyo Yuden TMK432BJ106KM
(2x) 10µF, 25V
Taiyo Yuden TMK432BJ106KM
Output Capacitor (2x) 330µF, 2.5V, 12m, C case
SANYO 2R5TPE330MCC2
(2x) 330µF, 2.5V, 12m, C case
SANYO 2R5TPE330MCC2
Inductor 1µH, 3.25m, 16A
Wurth Electronics 7443552100
1µH, 3.25m, 16A
Wurth Electronics 7443552100
Schottky Diode
2A, 30V Schottky diode (SMA)
Nihon EC21QS03L
Central Semiconductor
CMSH2-40M
2A, 30V Schottky diode (SMA)
Nihon EC21QS03L
Central Semiconductor
CMSH2-40M
High-Side MOSFET
Fairchild Semiconductor
(1x) FDS8690
8.6m/11.4m (typ/max)
Fairchild Semiconductor
(1x) FDS8690
8.6m/11.4m (typ/max)
Low-Side MOSFET
Fairchild Semiconductor
(1x) FDS8670
4.2m/5m (typ/max)
Fairchild Semiconductor
(1x) FDS8670
4.2m/5m (typ/max)
Table 1. Component Selection for Standard Applications
MANUFACTURER WEBSITE MANUFACTURER WEBSITE
AVX Corp. www.avxcorp.com Pulse Engineering www.pulseeng.com
BI Technologies www.bitechnologies.com Renesas Technology Corp. www.renesas.com
Central Semiconductor Corp. www.centralsemi.com SANYO Electric Co., Ltd. www.sanyodevice.com
Fairchild Semiconductor www.fairchildsemi.com Sumida Corp. www.sumida.com
International Rectifier www.irf.com Taiyo Yuden www.t-yuden.com
KEMET Corp. www.kemet.com TDK Corp. www.component.tdk.com
NEC Tokin Corp. www.nec-tokin.com TOKO America, Inc. www.tokoam.com
Panasonic Corp. www.panasonic.com Vishay/Siliconix www.vishay.com
Table 2. Component Suppliers
Detailed Description
The MAX17007/MAX17008 standard application circuit
(Figure 1) generates the 1V to 1.2V/12A and 1.5V/12A
chipset voltages in a notebook computer. The input sup-
ply range is 7V to 20V for the specific application. Table 1
lists component selections, while Table 2 lists the com-
ponent manufacturers. Figure 2 shows the combined-
mode standard application circuit and Figure 3 is the
MAX17007/MAX17008 functional diagram.
The MAX17007/MAX17008 contain two constant on-
time step-down controllers designed for low-voltage
power supplies. The two SMPSs can also be combined
to operate as a two-phase high-current single-output
regulator. Constant on-time Quick-PWM operation pro-
vides fast response to load transients and handles wide
I/O voltage ratios with ease, while maintaining a rela-
tively constant switching frequency. The switching fre-
quency can be adjusted between 200kHz and 600kHz
with external resistors. Differential output current sens-
ing allows output sense-resistor sensing for an accu-
rate current-limit, lossless inductor DCR current
sensing for lower power dissipation while maintaining
0.7% output accuracy. Overvoltage (MAX17007) and
undervoltage protection and accurate user-selectable
current limits (four different levels) ensure robust opera-
tions.
The MAX17007/MAX17008 feature a special combined-
mode configuration that allows higher current outputs
to be supported. A current-balance integrator maintains
equal currents in the two phases, improving efficiency
and power distribution.
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
16 ______________________________________________________________________________________
MAX17007
MAX17008
6
TON1
AGND
RREFIN1
RREFIN2
16
CVCC
1µF
PGOOD2 NOT USED
IIN COMBINED MODE
ILIM2 FUNCTIONS AS
CCI OUTPUT IN
COMBINED MODE
EN2 MUST BE
GROUNDED
R9
10Ω
RTON1
220kΩ
RNTC1
10kΩ
R4
3.01kΩ
RTON2
220kΩ
CREF
2.2nF
CCCI
220pF
RREFIN3
GND
PWR
PWR
PWR
AGND
PWR
AGND
2ILIM1
3ILIM2
(CCI)
5SKIP
11 EN1
25 EN2
1REF
RREFIN1 = 80.6kΩ
RREFIN2 = 121kΩ
RREFIN3 = 249kΩ8REFIN1
12 PGOOD1
24 PGOOD2
7
TON2
15
BST1
13
DH1
14
LX1
17
DL1
NH1
NL1
20
PGND
10
CSH1
9
CSL1
28
FB2
4
VCC
18
VDD
REF
VOUT
+3.3V
H = 1.0V
L = 1.2V
+5V
+5V
CONNECT TO 5V FOR
COMBINED MODE OPERATION
*LOWER INPUT VOLTAGES REQUIRE
ADDITIONAL INPUT CAPACITANCE.
VIN
7V TO 20V
EP
POWER GROUND
ANALOG GROUND
R1
100kΩ
CVDD
2.2µF
C1
0.22µF
CBST1
0.1µF
PWR
DL1
CIN1
R3
1.5kΩ
COUT1
4 x 330µF
12mΩ
L1
1µH, 16A, 3mΩ
VOUT1
1.2V/1.0V, 24A
ILIM
PIN
VCC
OPEN
REF
GND
R7
10Ω
C2
1nF
PWR
COUT1-CER
10 x 10µF
CERAMIC
RNTC2
10kΩ
R6
3.01kΩ
PWR
AGND
21
BST2
23
DH2
22
LX2
19
DL2
NH2
NL2
26
CSH2
27
CSL2
VIN
7V TO 20V
C3
0.22µF
CBST2
0.1µF
PWR
DL2
CIN2
R5
1.5kΩ
L2
1µH, 16A, 3mΩ
R8
10Ω
C4
1nF
CURRENT
LIMIT
CCCI
(pF)
60mV
45mV
30mV
15mV
120
180
220
470
Figure 2. MAX17007/MAX17008 Combined-Mode Standard Application Circuit
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 17
MAX17007
MAX17008
POWER-GOOD AND
FAULT PROTECTION 2
(FIGURE 13)
PWM CONTROLLER 2
(FIGURE 4)
CURRENT LIMIT 2
(FIGURE 8)
SMPS2 TARGET
DECODE
(FIGURE 9B)
MUX
2.0V
REF
PGOOD2
COMBINE
(FB2 = VCC)
FB2
CSH2
CSL2
ILIM2
EN2
LX2
CURRENT-
SENSE GAIN
VALLEY
CURRENT
LIMIT
DH2
BST2
PGND
DL2
VDD
CSL2
TARGET2
Gm
Gm
POWER-GOOD AND
FAULT PROTECTION 1
(FIGURE 13)
PWM CONTROLLER 1
(FIGURE 4)
CURRENT LIMIT 1
(FIGURE 8)
SMPS1 TARGET
DECODE
(FIGURE 9A)
MUX
PGOOD1
REF
VCC
REFIN1
CSH1
CSL1
ILIM1
SKIP
TON1
TON2
EN1
LX1
CURRENT-
SENSE GAIN
VALLEY
CURRENT
LIMIT
DH1
BST1
GND
DL1
CSL1
Gm
Gm
TARGET1
FAULT1
FAULT2
VDD
CURRENT
BALANCE
COMBINE
(FB2 = VCC)
COMBINE
(FB2 = VCC)
COMBINE
(FB2 = VCC)
Figure 3. MAX17007/MAX17008 Functional Diagram
+5V Bias Supply (VCC, VDD)
The MAX17007/MAX17008 require an external 5V bias
supply in addition to the battery. Typically, this 5V bias
supply is the notebook’s 95%-efficient 5V system sup-
ply. Keeping the bias supply external to the IC improves
efficiency and eliminates the cost associated with the 5V
linear regulator that would otherwise be needed to sup-
ply the PWM circuit and gate drivers. If stand-alone
capability is needed, the 5V supply can be generated
with an external linear regulator such as the MAX1615.
The 5V bias supply powers both the PWM controllers
and internal gate-drive power, so the maximum current
drawn depends on the external MOSFET’s gate capaci-
tance, and the selected switching frequency:
IBIAS = IQ+ fSW1QG(SMPS1) + fSW2QG(SMPS2)
= 4mA to 40mA (typ)
Bypass VCC with a 1µF or greater ceramic capacitor to
the analog ground. Bypass VDD with a 2.2µF or greater
ceramic capacitor to the power ground. VCC and VDD
should be separated with a 10Ωresistor (Figure 1).
2V Reference
The 2V reference is accurate to ±1% over temperature
and load, making REF useful as a precision system ref-
erence. Bypass REF to GND with a 2.2nF. The refer-
ence sources up to 100µA and sinks 10µA to support
external loads.
Combined-Mode Operation (FB2 = VCC)
Combined-mode operation allows the MAX17007/
MAX17008 to support even higher output currents by
sharing the load current between two phases, distribut-
ing the power dissipation over several power compo-
nents to improve the efficiency. The MAX17007/
MAX17008 are configured in combined mode by con-
necting FB2 to VCC. See Figure 2 for the combined-
mode standard application circuit.
Table 3 lists the pin function differences between com-
bined mode and separate mode. See the
Pin Description
table for additional details.
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
18 ______________________________________________________________________________________
PIN COMBINED MODE SEPARATE MODE
FB2 Connect to VCC to configure MAX17007/MAX17008 for
combined-mode operation
Connect to REF for preset 1.5V, or use a resistor-
divider to set the SMPS2 output voltage
REFIN1 Sets the combined output voltage—dynamic, fixed, and
preset voltages supported
Sets the SMPS1 output voltage—dynamic, fixed,
and preset voltages supported
EN1 Enables/disables combined output Enables/disables SMPS1
EN2 Not used; connect to GND Enables/disables SMPS2
PGOOD1 Power-good indicator for combined output voltage Power-good indicator for SMPS1
PGOOD2 Not used; can be left open Power-good indicator for SMPS2
TON1 Sets the per-phase switching frequency for both SMPSs Sets the switching frequency for SMPS1
TON2 Use the same resistor as TON1 Sets the switching frequency for SMPS2
ILIM1 Sets the per-phase current limit for both SMPSs Sets SMPS1 current limit
ILIM2 (CCI) Current-balance integrator output; connect a capacitor
from CCI to the output Sets SMPS2 current limit
SKIP Only three distinct modes of operation; ultrasonic mode
not supported Supports all four modes of operation
Table 3. Pin Function in Combined and Separate Modes
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 19
SMPS Detailed Description
Free-Running Constant-On-Time PWM
Controller with Input Feed-Forward
The Quick-PWM control architecture is a pseudo-fixed-
frequency, constant-on-time, current-mode regulator
with voltage feed-forward. This architecture relies on
the output filter capacitor’s ESR to act as a current-
sense resistor, so the output ripple voltage provides the
PWM ramp signal. The control algorithm is simple: the
high-side switch on-time is determined solely by a one-
shot whose pulse width is inversely proportional to
input voltage and directly proportional to output volt-
age. Another one-shot sets a minimum off-time (150ns
typ). The on-time one-shot is triggered if the error com-
parator is low, the low-side switch current is below the
valley current-limit threshold, and the minimum off-time
one-shot has timed out. Figure 4 is the PWM controller
block diagram.
On-Time One-Shot
The heart of the PWM core is the one-shot that sets the
high-side switch on-time. This fast, low-jitter, adjustable
one-shot includes circuitry that varies the on-time in
response to battery and output voltage. In independent
mode, the high-side switch on-time is inversely propor-
tional to the battery voltage as sensed by the TON1
and TON2 inputs, and proportional to the voltages on
CSL1 and CSL2 pins:
SMPS1 On-Time tON1 = TSW1 (VCSL1/VIN)
SMPS2 On-Time tON2 = TSW2 (VCSL2/VIN)
where TSW1 (switching period of SMPS1) is set by the
resistance between TON1 and VIN, TSW2 is set by the
resistance between TON2 and VIN. This algorithm
results in a nearly constant switching frequency despite
the lack of a fixed-frequency clock generator.
MAX17007
MAX17008
CSL OR
CCI
DH DRIVER
DL DRIVER
ON-TIME
COMPUTE
INTEGRATOR
(CCV)
TON
TRIG
ONE-SHOT
Q
ERROR
AMPLIFIER
INTERNAL
FB
ZERO
CROSSING
VALLEY
CURRENT
LIMIT
OV
FAULT
AMPLIFIED
CURRENT
SENSE
SLOPE
COMP
tON
R
SQ
R
S
Q
TARGET
tOFF(MIN)
TRIG
Q
Figure 4. PWM Controller Block Diagram
MAX17007/MAX17008
Switching Frequency
The MAX17007/MAX17008 feature independent resis-
tor-programmable switching frequencies for each
SMPS, providing flexibility for applications where one
SMPS operates at a lower switching frequency when
connected to a high-voltage input rail while the other
SMPS operates at a higher switching frequency when
connected to a lower voltage rail as a second-stage
regulator. Connect a resistor (RTON) between TON and
VIN to set the switching period TSW = 1/fSW:
TSW1 = CTON (RTON1 + 6.5kΩ)
TSW2 = CTON (RTON2 + 6.5kΩ)
where CTON = 16.26pF. A 97.5kΩto 302.5kΩcorre-
sponds to switching periods of 1.67µs (600kHz) to 5µs
(200kHz) for SMPS1 and SMPS2. High-frequency
(600kHz) operation optimizes the application for the
smallest component size, trading off efficiency due to
higher switching losses. This may be acceptable in
ultra-portable devices where the load currents are
lower and the controller is powered from a lower volt-
age supply. Low-frequency (200kHz) operation offers
the best overall efficiency at the expense of component
size and board space.
For continuous conduction operation, the actual switching
frequency can be estimated by:
where VDIS is the sum of the parasitic voltage drops in
the inductor discharge path, including synchronous
rectifier, inductor, and printed-circuit board (PCB)
resistances; VCHG is the sum of the resistances in the
charging path, including the high-side switch, inductor,
and PCB resistances; and tON is the on-time calculated
by the on-time block.
When operating in separate mode, it is recommended
that both SMPS switching frequencies be set apart by
10% to 30% to prevent the two sides from beating
against each other.
Combined-Mode On-Time One-Shot
In combined mode (FB2 = VCC), TON1 sets the on-
time, and hence the switching frequency, for both SMPS.
The on-time is programmed using the TON1 equation,
which sets the switching frequency per phase. The
effective switching frequency as seen on the input and
output capacitors is twice the per-phase frequency.
Combined-Mode Current Balance
In combined mode, the one-shot for SMPS2 varies the
on-time in response to the input voltage and the differ-
ence between the SMPS1 and SMPS2 inductor cur-
rents. The SMPS1 one-shot in combined mode behaves
the same way as it does in separate mode. As such,
SMPS2 regulates the current balance, while SMPS1
regulates the voltage.
Two identical transconductance amplifiers integrate the
difference between SMPS1 and SMPS2 current-sense
signals. The summed output is internally connected to
CCI, allowing adjustment of the integration time con-
stant with a compensation network (usually a capacitor)
connected between CCI and the output.
The resulting compensation current and voltage are
determined by the following equations:
ICCI = Gm[(VCSH1 - VCSL1) - (VCSH2 - VCSL2)]
VCCI = VOUT + ICCIZCCI
where ZCCI is the impedance at the CCI output. The
SMPS2 on-time one-shot uses this integrated signal
(VCCI) to set the SMPS2 high-side MOSFETs on-time.
When SMPS1 and SMPS2 current-sense signals
(VCSH1 - VCSL1 and VCSH2 - VCSL2) become unbal-
anced, the transconductance amplifiers adjust the
SMPS2 on-time, which increases or decreases the
SMPS2 inductor current until the current-sense signals
are properly balanced. In combined mode, the SMPS2
on-time is given by:
SMPS2 On-Time tON2 = TSW2 (VCCI/VIN)
SMPS Enable Controls (EN1, EN2)
EN1 and EN2 provide independent control of output
soft-start and soft-shutdown. This allows flexible control
of startup and shutdown sequencing. The outputs can
be started simultaneously, sequentially, or indepen-
dently. To provide sequential startup, connect EN of
one regulator to PGOOD of the other. For example, with
EN1 connected to PGOOD2, OUT1 soft-starts after
OUT2 is in regulation.
When configured in separate mode, the two outputs are
independent. A fault at one output does not trigger
shutdown of the other.
When configured in combined mode (FB2 = VCC), EN1
is the master control input that enables/disables the
combined output, while EN2 has no function and must
be connected to GND. The startup slew rate follows
that of SMPS1.
Toggle EN low to clear the overvoltage, undervoltage,
and thermal-fault latches.
fVV
tVV
SW OUT DIS
ON IN CHG
=+
+()
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
20 ______________________________________________________________________________________
Soft-Start
Soft-start begins when EN is driven high and REF is in
regulation. During soft-start, the output is ramped up
from 0V to the final set voltage at 1.3mV/µs slew rate for
SMPS1, and 0.65mV/µs for SMPS2, reducing the inrush
current and providing a predictable ramp-up time for
power sequencing:
The soft-start circuitry does not use a variable current
limit, so full output current is available immediately. The
respective PGOOD becomes high impedance approxi-
mately 200µs after the target voltage has been
reached. The MAX17007/MAX17008 automatically use
pulse-skipping mode during soft-start and use forced-
PWM mode during soft-shutdown, regardless of the
SKIP configuration.
For automatic startup, the battery voltage should be
present before VCC. If the controller attempts to bring
the output into regulation without the battery voltage
present, the fault latch trips. The controller remains shut
down until the fault latch is cleared by toggling EN or
cycling the VCC power supply below 0.5V.
Soft-Shutdown
Soft-shutdown begins when the system pulls EN low,
an output undervoltage fault, or a thermal fault. During
soft-shutdown, the respective PGOOD is pulled low
immediately and the output voltage ramps down with
the same startup slew rate for the respective outputs.
After the controller reaches the 0V target, the drivers
are disabled (DL_ and DH_ pulled low) and the internal
10Ωdischarge on CSL_ activated. The MAX17007/
MAX17008 shut down completely when both EN are
low—the reference turns off after both SMPSs have
reached the 0V target, and the supply current drops to
about 1µA (max).
Slowly discharging the output capacitors by slewing
the output over a long period of time (typically 0.5ms to
2ms) keeps the average negative inductor current low
(damped response), thereby preventing the negative
output-voltage excursion that occurs when the con-
troller discharges the output quickly by permanently
turning on the low-side MOSFET (underdamped
response). This eliminates the need for the Schottky
diode normally connected between the output and
ground to clamp the negative output-voltage excursion.
Modes of Operation
Forced-PWM Mode (
SSKKIIPP
= 5V)
The low-noise forced-PWM mode (SKIP = 5V) disables
the zero-crossing comparator, which controls the low-
side switch on-time. This forces the low-side gate-drive
waveform to constantly be the complement of the high-
side gate-drive waveform, so the inductor current
reverses at light loads while DH maintains a duty factor
of VOUT/VIN. The benefit of forced-PWM mode is to
keep the switching frequency fairly constant. However,
forced-PWM operation comes at a cost: the no-load 5V
bias current remains between 2mA to 5mA, depending
on the switching frequency.
The MAX17007/MAX17008 automatically use forced-
PWM operation during shutdown, regardless of the
SKIP configuration.
Automatic Pulse-Skipping Mode
(
SSKKIIPP
= GND or 2V)
In skip mode (SKIP = GND or 2V), an inherent automatic
switchover to PFM takes place at light loads. This
switchover is affected by a comparator that truncates
the low-side switch on-time at the inductor current’s
zero crossing. The zero-crossing comparator threshold
is set by the differential across CSL_ and CSH_.
DC output-accuracy specifications refer to the thresh-
old of the error comparator. When the inductor is in con-
tinuous conduction, the MAX17007/MAX17008 regulate
the valley of the output ripple, so the actual DC output
voltage is higher than the trip level by 50% of the output
ripple voltage. In discontinuous conduction (SKIP =
GND or 2V and IOUT < ILOAD(SKIP)), the output voltage
has a DC regulation level higher than the error-com-
parator threshold by approximately 1.5% due to slope
compensation. However, the internal integrator corrects
for most of it, resulting in very little load regulation.
When SKIP = 2V, the MAX17007/MAX17008 use forced-
PWM operation during all dynamic output-voltage transi-
tions until 100µs after the transition has been
completed—REFIN1 and the internal target are within
±50mV (typ) and an error-amplifier transition is detected.
Since SMPS2 does not support dynamic transitions, SKIP
= 2V and SKIP = GND have the same pulse-skipping
behavior without any forced-PWM transitions.
ttV
SR
V
mV s
START SHDN FB
SS
FB
22
2
2
2
065
===
./µ
tt
V
SR
V
mV s
START SHDN REFIN
SS
REFIN
11 1
1
1
13
== =
./µ
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 21
MAX17007/MAX17008
When SKIP is pulled to GND, the MAX17007/MAX17008
remain in pulse-skipping mode. Since the output is not
able to sink current, the timing for negative dynamic out-
put-voltage transitions depends on the load current and
output capacitance. Letting the output voltage drift down
is typically recommended in order to reduce the potential
for audible noise since this eliminates the input current
surge during negative output-voltage transitions. Figure 5
shows the pulse-skipping/discontinuous crossover point.
Ultrasonic Mode (
SSKKIIPP
= Open = 3.3V)
Leaving SKIP unconnected or connecting SKIP to 3.3V
activates a unique pulse-skipping mode with a mini-
mum switching frequency of 25kHz. This ultrasonic
pulse-skipping mode eliminates audio-frequency mod-
ulation that would otherwise be present when a lightly
loaded controller automatically skips pulses. In ultra-
sonic mode, the controller automatically transitions to
fixed-frequency PWM operation when the load reaches
the same critical conduction point (ILOAD(SKIP)) that
occurs when normally pulse skipping.
An ultrasonic pulse occurs when the controller detects
that no switching has occurred within the last 30µs.
Once triggered, the ultrasonic controller pulls DL high,
turning on the low-side MOSFET to induce a negative
inductor current (Figure 6). After the inductor current
reaches the negative ultrasonic current threshold, the
controller turns off the low-side MOSFET (DL pulled
low) and triggers a constant on-time (DH driven high).
When the on-time has expired, the controller reenables
the low-side MOSFET until the controller detects that
the inductor current dropped below the zero-crossing
threshold. Starting with a DL pulse greatly reduces the
peak output voltage when compared to starting with a
DH pulse.
The output voltage at the beginning of the ultrasonic
pulse determines the negative ultrasonic current thresh-
old, resulting in the following equations for SMPS1:
(SMPS1 adjustable mode)
(SMPS1 preset mode)
where VCSL1 > VREFIN1 in adjustable mode, VCSL1 >
1.05V in preset mode, and RCS1 is the current-sense
resistance seen across CSH1 to CSL1.
Similarly for SMPS2:
(SMPS2 adjustable mode)
(SMPS2 preset mode)
where VCSL2 > 0.7V in adjustable mode, VCSL2 > 1.5V
in preset mode, and RCS2 is the current-sense resis-
tance seen across CSH2 to CSL2.
In combined mode, ultrasonic mode setting is disabled,
and the SKIP = OPEN (3.3V) setting is identical to the
SKIP = GND setting.
VIRVV
ISONIC L CS CSL222 2
15 065..==
()
×−
VIR VV
ISONIC L CS CSL111 1
105 065..==−
()
×
VIRVV
ISONIC L CS REFIN CSL111 1 1
065.== −
()
×
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
22 ______________________________________________________________________________________
INDUCTOR CURRENT
ILOAD = IPEAK/2
ON-TIME0 TIME
IPEAK
L
VIN - VOUT
∆I
∆t=
Figure 5. Pulse-Skipping/Discontinuous Crossover Point
ON-TIME (tON)
ISONIC
0
ZERO-CROSSING
DETECTION
INDUCTOR
CURRENT
40µs (MAX)
Figure 6. Ultrasonic Waveform
Valley Current-Limit Protection
The current-limit circuit employs a unique “valley” cur-
rent-sensing algorithm that senses the inductor current
across the output current-sense element—inductor
DCR or current-sense resistor, which generates a volt-
age between CSH_ and CSL_. If the current exceeds
the valley current-limit threshold during the low-side
MOSFET conduction time, the PWM controller is not
allowed to initiate a new cycle. The valley current-limit
threshold is set by the four-level ILIM_ pin, with selec-
table limits of 15mV, 30mV, 45mV, and 60mV.
The actual peak current is greater than the valley cur-
rent-limit threshold by an amount equal to the inductor
ripple current (Figure 7). Therefore, the exact current-
limit characteristic and maximum load capability are a
function of the inductor value and battery voltage.
When combined with the undervoltage protection cir-
cuit, this current-limit method is effective in almost
every circumstance. See Figure 8.
In forced-PWM mode, the MAX17007/MAX17008 also
implement a negative current limit to prevent excessive
reverse inductor currents when VOUT is sinking current.
The negative current-limit threshold is set to approxi-
mately 120% of the positive current limit.
In combined mode, ILIM1 sets the per-phase current
limit for both phases.
MOSFET Gate Drivers (DH, DL)
The DH and DL drivers are optimized for driving mod-
erate-sized high-side, and larger low-side power
MOSFETs. This is consistent with the low duty factor
seen in notebook applications, where a large VIN -
VOUT differential exists. The high-side gate driver (DH)
sources and sinks 1.2A, and the low-side gate driver
(DL) sources 1.0A and sinks 2.4A. This ensures robust
gate drive for high-current applications. The DH floating
high-side MOSFET driver is powered by internal boost
switch charge pumps at BST, while the DL synchro-
nous-rectifier driver is powered directly by the 5V bias
supply (VDD).
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 23
INDUCTOR CURRENT
ILIMIT
ILOAD
0 TIME
IPEAK
ILIM(VAL) = ILOAD(MAX) 1- LIR
2
()
Figure 7. “Valley” Current-Limit Threshold Point
CSH
ILIM
CSL
SKIP
ZERO
CROSSING
VALLEY
CURRENT
LIMIT
QUAD-LEVEL
DECODE
CURRENT-
SENSE
GAIN
Figure 8. Current-Limit Block Diagram
MAX17007/MAX17008
Output Voltage
The MAX17007/MAX17008 feature preset and
adjustable output voltages for both SMPSs, and
dynamic output voltages for SMPS1. In combined
mode, the output voltage is set by REFIN1, and all fea-
tures for SMPS1 output-voltage configuration and
dynamic voltage changes apply to the combined out-
put. Figure 9 is the SMPS target decode block diagram.
Preset/Adjustable Output Voltages
(Dual-Mode Feedback)
Connect REFIN1 to VCC to set the SMPS1 voltage to
preset 1.05V. Connect FB2 to REF to set the SMPS2
voltage to preset 1.5V. The SMPS1 output voltage can
be adjusted up to 2V by changing REFIN1 voltage with-
out using an external resistive voltage-divider. The out-
put voltage of SMPS2 can be adjusted with an external
resistive voltage-divider between CSL2 and GND with
the center tap connected to FB2 (Figure 10). Choose
RFB2LO (resistance from FB2 to GND) to be approxi-
mately 10kΩand solve for RFBHI (resistance from CSL2
to FB2) using the equation:
The MAX17007/MAX17008 regulate the valley of the
output ripple, so the actual DC output voltage is higher
than the slope compensated target by 50% of the out-
put ripple voltage. Under steady-state conditions, the
MAX17007/MAX17008s’ internal integrator corrects for
this 50% output ripple voltage error, resulting in an out-
put-voltage accuracy that is dependent only on the off-
set voltage of the integrator amplifier provided in the
Electrical Characteristics
table.
Dynamic Output Voltages (REFIN1)
The MAX17007/MAX17008 regulate the output to the
voltage set at REFIN1. By changing the voltage at
REFIN1 (Figure 1), the MAX17007/MAX17008 can be
used in applications that require dynamic output volt-
age changes between two set points. For a step-volt-
age change at REFIN, the rate of change of the output
voltage is limited either by the internal 9.5mV/µs slew-
rate circuit or by the component selection—inductor
current ramp, the total output capacitance, the current
limit, and the load during the transition—whichever is
slower. The total output capacitance determines how
much current is needed to change the output voltage,
while the inductor limits the current ramp rate.
RRV
V
FB HI FB LO CSL
22 2
07 1=−
⎛
⎝
⎜⎞
⎠
⎟
.
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
24 ______________________________________________________________________________________
VCC - 1V
FB2
REF - 0.3V
TARGET1
COMBINE
(FB2 = VCC)
PRESET
(FB1 = VCC)
PRESET
(FB2 = REF)
(B) SMPS2 TARGET DECODE
TARGET2
REF (2.0V)
0.7V
1.5V
5R
8R
7R
REFIN1
(A) SMPS1 TARGET DECODE
TARGET1
REF (2.0V)
1.05V
9.5R
10.5R
VCC - 1V
Figure 9. SMPS Target Decode Block Diagram
MAX17007
MAX17008
DL2
GND
LX2
L2
FB2
CSL2
CSH2
RFB2LO
COUT2
RSENSE2
RFB2HI
NL2
Figure 10. Setting VOUT2 with a Resistive Voltage-Divider
Additional load current can slow down the output volt-
age change during a positive REFIN1 voltage change,
and can speed up the output voltage change during a
negative REFIN1 voltage change.
Automatic Fault Blanking (SMPS1)
When the MAX17007/MAX17008 detect that the internal
target and REFIN1 are more than ±50mV (typ) apart,
the controller automatically blanks PGOOD1, blanks the
UVP protection, and sets the OVP threshold to max REF
+ 300mV. The blanking remains until 1) the internal tar-
get and REFIN1 are within ±30mV of each other, and 2)
an edge is detected on the error amplifier signifying
that the output is in regulation. This prevents the system
or internal fault protection from shutting down the con-
troller during transitions. Figure 11 shows the dynamic
REFIN transition (SKIP = GND) and Figure 12 shows
the dynamic REFIN1 transition (SKIP = REF).
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 25
REFIN1
PGOOD1
INTERNAL
PWM CONTROL
BLANK HIGH-Z
SKIP
LX1
SET TO REF + 300mV
OVP1
VOUT1
INTERNAL TARGET1
ACTUAL VOUT1
BLANK HIGH-Z
NO PULSES: VOUT > VTARGET
DYNAMIC REFIN WINDOW
TARGET1 + 300mV
Figure 11. Dynamic REFIN1 Transition (SKIP = GND)
REFIN1
PGOOD1
INTERNAL
PWM CONTROL
BLANK HIGH-Z
LX1
OVP1
VOUT1
ACTUAL VOUT1
50mV
INTERNAL TARGET1
BLANK HIGH-Z
DYNAMIC REFIN WINDOW
SKIP SKIPPWMPWM
TARGET1 + 300mV TARGET1 + 300mV
SET TO REF +
300mV
Figure 12. Dynamic REFIN1 Transition (SKIP = REF)
MAX17007/MAX17008
Internal Integration
An integrator amplifier forces the DC average of the FB
voltage to equal the target voltage. This internal amplifier
integrates the feedback voltage and provides a fine
adjustment to the regulation voltage (Figure 4), allowing
accurate DC output-voltage regulation regardless of the
compensated feedback ripple voltage and internal slope-
compensation variation. The integrator amplifier has the
ability to shift the output voltage by ±140mV (typ).
The MAX17007/MAX17008 disable the integrator by
connecting the amplifier inputs together at the begin-
ning of all dynamic REFIN1 transitions done in pulse-
skipping mode. The integrator remains disabled until
20µs after the transition is completed (the internal target
settles) and the output is in regulation (edge detected
on the error comparator).
Power-Good Outputs (PGOOD)
and Fault Protection
PGOOD_ is the open-drain output that continuously
monitors the respective output voltage for undervoltage
and overvoltage conditions. The respective PGOOD_ is
actively held low in shutdown (EN_ = GND) during soft-
start and soft-shutdown. Approximately 200µs (typ)
after the soft-start terminates, PGOOD_ becomes high
impedance as long as the respective output voltage is
in regulation.
PGOOD1 goes low if the output voltage drops 200mV
below the target voltage (REFIN1 or fixed 1.05V), or
rises 300mV above the target voltage (REFIN1 or fixed
1.05V), or the SMPS1 controller is shut down.
In adjustable mode, PGOOD2 goes low if the feedback
voltage drops 100mV below the target voltage (0.7V), or
rises 150mV above the target voltage (0.7V), or the
SMPS2 controller is shut down. In preset mode (fixed
1.5V), the PGOOD2 thresholds are -200mV and +300mV.
For a logic-level PGOOD output voltage, connect an
external pullup resistor between PGOOD and VDD. A
100kΩpullup resistor works well in most applications.
See Figure 13.
Overvoltage Protection (OVP, MAX17007 Only)
When the internal feedback voltage rises above the
overvoltage threshold, the OVP comparator immediate-
ly pulls DH low and forces DL high, pulls PGOOD low,
sets the fault latch, and disables the faulted SMPS con-
troller. Toggle EN or cycle VCC power below the VCC
POR to clear the fault latch and restart the controller.
The overvoltage thresholds are +300mV for SMPS1
(fixed 1.05V and adjustable REFIN1), +300mV for
SMPS2 in preset mode (fixed 1.5V output), and +150mV
for SMPS2 in adjustable mode (0.7V feedback).
An OV fault on one side does not affect the other side.
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
26 ______________________________________________________________________________________
TARGET
+ VOVP
TARGET
- VUVP
CSL OR FB
FAULT
OVP ENABLED
(MAX17007 ONLY)
FAULT
LATCH
ONE
SHOT
200µs
POWER-GOOD
IN
CLK
OUT
OVP
UVP
EN
SOFT-START
COMPLETE
NOTE: ONLY THE MAX17007 HAS
OVP FUNCTION ENABLED.
Figure 13. Power-Good and Fault Protection
Undervoltage Protection (UVP)
When the feedback voltage drops below the undervolt-
age threshold, the controller immediately pulls PGOOD
low and triggers a 200µs one-shot timer. If the feed-
back voltage remains below the undervoltage fault
threshold for the entire 200µs, then the undervoltage
fault latch of the faulted SMPS is set and that SMPS
begins its shutdown sequence. When the internal target
voltage drops below 0.1V, the MAX17007/MAX17008
force DL low for the faulted SMPS. Toggle EN or cycle
VCC power below VCC POR to clear the fault latch and
restart the controller.
The undervoltage thresholds are -200mV for SMPS1
(fixed 1.05V and adjustable REFIN1), -200mV for
SMPS2 in preset mode (fixed 1.5V output), and -100mV
for SMPS2 in adjustable mode (0.7V feedback).
A UV fault on one side does not affect the other side.
Thermal-Fault Protection (T
SHDN
)
The MAX17007/MAX17008 feature a thermal-fault pro-
tection circuit. When the junction temperature rises
above +160°C, a thermal sensor activates the fault
latch, pulls PGOOD low, and shuts down the controller.
Both DL and DH are pulled low. Toggle EN or cycle
VCC power below VCC POR to reactivate the controller
after the junction temperature cools by 15°C.
VCC POR and UVLO
Each SMPS of the MAX17007/MAX17008 is enabled
when its respective EN is driven high. On the first rising
EN, the reference powers up first. Once the reference
exceeds its undervoltage lockout (UVLO) threshold
(~ 60µs), the internal analog blocks are turned on and
masked by a 140µs one-shot delay in order to allow the
bias circuitry and analog blocks enough time to settle
to their proper states. With the control circuitry reliably
powered up, the PWM controller begins switching. The
second rising EN, if controlled separately, also has the
140µs one-shot delay before its first DH pulse.
Power-on reset (POR) occurs when VCC rises above
approximately 3V, resetting the fault latch and preparing
the controller for operation. The VCC UVLO circuitry
inhibits switching until VCC rises above 4.25V. The con-
troller powers up the reference once the system enables
the controller, VCC exceeds 4.25V, and either EN is dri-
ven high. With the reference in regulation, the controller
ramps the output voltage to the target voltage with a
1.3mV/µs slew rate for SMPS1 and 0.65mV/µs for SMPS2.
If the VCC voltage drops below 4.25V, the controller
assumes that there is not enough supply voltage to make
valid decisions. To protect the output from overvoltage
faults, the controller shuts down immediately and forces
a high-impedance output (DL and DH pulled low).
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 27
MODE CONTROLLER STATE DRIVER STATE
Shutdown (EN_ = High to Low)
Output UVP (Latched)
Thermal Fault (Latched)
Voltage soft-shutdown initiated. Error
amplifier target slowly ramped down to
GND.
DL_ low and DH_ low after soft-shutdown
completed, internal 10 discharge on CSL_
activated.
(T 0 1V )
Output OVP (Latched)
Controller shuts down and internal target
slews down. Controller remains off until
EN_ toggled or VCC power cycled.
DL_ immediately forced high, DH_ pulled low
(high-side MOSFET disabled).
VCC UVLO Falling Edge
Controller shuts down and the internal
target slews down. Controller remains off
until VCC rises back above UVLO threshold.
DL_ low, DH_ low, internal 10 discharge on
CSL_ activated.
VCC UVLO Rising Edge SMPS controller enabled (assuming EN_
pulled high). DL_, DH_ switching.
VCC POR SMPS inactive. DL_ low.
Table 4. Fault Protection and Shutdown Operation
MAX17007/MAX17008
Quick-PWM Design Procedure
Firmly establish the input voltage range and maximum
load current before choosing a switching frequency
and inductor operating point (ripple-current ratio). The
primary design trade-off lies in choosing a good switch-
ing frequency and inductor operating point, and the fol-
lowing four factors dictate the rest of the design:
• Input voltage range: The maximum value
(VIN(MAX)) must accommodate the worst-case input
supply voltage allowed by the notebook’s AC
adapter voltage. The minimum value (VIN(MIN))
must account for the lowest input voltage after
drops due to connectors, fuses, and battery selec-
tor switches. If there is a choice at all, lower input
voltages result in better efficiency.
• Maximum load current: There are two values to
consider. The peak load current (ILOAD(MAX)) deter-
mines the instantaneous component stresses and fil-
tering requirements, and thus drives output
capacitor selection, inductor saturation rating, and
the design of the current-limit circuit. The continuous
load current (ILOAD) determines the thermal stress-
es and thus drives the selection of input capacitors,
MOSFETs, and other critical heat-contributing com-
ponents. Most notebook loads generally exhibit
ILOAD = ILOAD(MAX) x 80%.
• Switching frequency: This choice determines the
basic trade-off between size and efficiency. The
optimal frequency is largely a function of maximum
input voltage due to MOSFET switching losses that
are proportional to frequency and VIN2. The opti-
mum frequency is also a moving target due to rapid
improvements in MOSFET technology that are mak-
ing higher frequencies more practical.
• Inductor operating point: This choice provides
trade-offs between size vs. efficiency and transient
response vs. output noise. Low inductor values pro-
vide better transient response and smaller physical
size, but also result in lower efficiency and higher
output noise due to increased ripple current. The
minimum practical inductor value is one that causes
the circuit to operate at the edge of critical conduc-
tion (where the inductor current just touches zero
with every cycle at maximum load). Inductor values
lower than this grant no further size-reduction benefit.
The optimum operating point is usually found
between 20% and 50% ripple current.
Inductor Selection
The per-phase switching frequency and operating point
(% ripple current or LIR) determine the inductor value
as follows:
For example: ILOAD(MAX) = 15A, VIN = 12V, VOUT =
1.5V, fSW = 300kHz, 30% ripple current or LIR = 0.3:
Find a low-loss inductor having the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite
cores are often the best choice, although powdered
iron is inexpensive and can work well at 200kHz. The
core must be large enough not to saturate at the peak
inductor current (IPEAK):
In combined mode, ILOAD(MAX) is the per-phase maxi-
mum current, which is half the actual maximum load
current for the combined output.
Transient Response
The inductor ripple current impacts transient-response
performance, especially at low VIN - VOUT differentials.
Low inductor values allow the inductor current to slew
faster, replenishing charge removed from the output fil-
ter capacitors by a sudden load step. The amount of
output sag is also a function of the maximum duty fac-
tor, which can be calculated from the on-time and mini-
mum off-time. The worst-case output sag voltage can
be determined by:
where tOFF(MIN) is the minimum off-time (see the
Electrical Characteristics
table).
The amount of overshoot due to stored inductor energy
can be calculated as:
where NPH is the number of active phases per output.
NPH is 1 for separate mode, and NPH is 2 for com-
bined-mode operation.
VIL
NC V
SOAR
LOAD MAX
PH OUT OUT
≈
()
∆()
2
2
V
VT
Vt
SAG
OUT SW
IN OFF M
=
()
⎛
⎝
⎜⎞
⎠
⎟+LI
LOAD(MAX)
2
∆( IIN
OUT OUT IN OUT
IN SW O
CV VV
VTt
)
⎡
⎣
⎢⎤
⎦
⎥
−
⎛
⎝
⎜⎞
⎠
⎟−2FFF MIN()
⎡
⎣
⎢⎤
⎦
⎥
II LIR
PEAK LOAD MAX
=+
⎛
⎝
⎜⎞
⎠
⎟
()
12
LVV
kHz A
V
V
=−
××
⎛
⎝
⎜⎞
⎠
⎟⎛
⎝
12 1 5
300 15 0 3
15
12
.
.
.
⎜⎜ ⎞
⎠
⎟=097.µH
LVV
f I LIR
V
V
IN OUT
SW LOAD MAX
OUT
IN
=−
⎛
⎝
⎜⎞
⎠
⎟⎛
⎝
⎜⎞
⎠
⎟
()
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
28 ______________________________________________________________________________________
Setting the Valley Current Limit
The minimum current-limit threshold must be high
enough to support the maximum load current when the
current limit is at the minimum tolerance value. The val-
ley of the inductor current occurs at ILOAD(MAX) minus
half the ripple current; therefore:
where ILIMIT(LOW) equals the minimum current-limit
threshold voltage divided by the output sense element
(inductor DCR or sense resistor).
The four-level ILIM setting sets a valley current limit of
15mV, 30mV, 45mV, or 60mV across the CSH_ to CSL_
differential input.
Special attention must be made to the tolerance and
thermal variation of the on-resistance in the case of
DCR sensing. Use the worst-case maximum value for
RDCR from the inductor data sheet, and add some mar-
gin for the rise in RDCR with temperature. A good gen-
eral rule is to allow 0.5% additional resistance for each
°C of temperature rise, which must be included in the
design margin unless the design includes an NTC ther-
mistor in the DCR network to thermally compensate the
current-limit threshold.
The current-sense method (Figure 14) and magnitude
determine the achievable current-limit accuracy and
power loss. The sense resistor can be determined by:
RSENSE_ = VLIM_/ILIMIT_
For the best current-sense accuracy and overcurrent
protection, use a 1% tolerance current-sense resistor
between the inductor and output as shown in Figure
14a. This configuration constantly monitors the inductor
current, allowing accurate current-limit protection.
However, the parasitic inductance of the current-sense
resistor can cause current-limit inaccuracies, especially
when using low-value inductors and current-sense
resistors. This parasitic inductance (LESL) can be can-
celled by adding an RC circuit across the sense resis-
tor with an equivalent time constant:
Alternatively, low-cost applications that do not require
highly accurate current-limit protection can reduce the
overall power dissipation by connecting a series RC
circuit across the inductor (Figure 14b) with an equiva-
lent time constant:
and:
where RCS is the required current-sense resistance and
RDCR is the inductor’s series DC resistance. Use the
worst-case inductance and RDCR values provided by
the inductor manufacturer, adding some margin for the
inductance drop over temperature and load.
RL
CRR
DCR EQ
=×+
⎡
⎣
⎢⎤
⎦
⎥
1
1
1
2
RR
RR
R
CS DCR
=+
2
12
CR L
R
EQ EQ ESL
SENSE
=
II
N
LIR
LIMIT LOW LOAD MAX
PH
() ()
>−
⎛
⎝
⎜⎞
⎠
⎟
12
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 29
SENSE RESISTOR
L
MAX17007
MAX17008
COUT
INPUT (VIN)
CIN
CSL_
CSH_
PGND
DL_
DH_
LX_
CEQ
REQ
NH
NLDL
LESL RSENSE
CEQREQ = LESL
RSENSE
a) OUTPUT SERIES RESISTOR SENSING
Figure 14. Current-Sense Configurations (Sheet 1 of 2)
MAX17007/MAX17008
Output Capacitor Selection
The output filter capacitor must have low enough effec-
tive series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR
to satisfy stability requirements.
In core and chipset converters and other applications
where the output is subject to large-load transients, the
output capacitor’s size typically depends on how much
ESR is needed to prevent the output from dipping too
low under a load transient. Ignoring the sag due to
finite capacitance:
In low-power applications, the output capacitor’s size
often depends on how much ESR is needed to maintain
an acceptable level of output ripple voltage. The output
ripple voltage of a step-down controller equals the total
inductor ripple current multiplied by the output capacitor’s
ESR. The maximum ESR to meet ripple requirements is:
where fSW is the switching frequency.
With most chemistries (polymer, tantalum, aluminum,
electrolytic), the actual capacitance value required
relates to the physical size needed to achieve low ESR
and the chemistry limits of the selected capacitor tech-
nology. Ceramic capacitors provide low ESR, but the
capacitance and voltage rating (after derating) are
determined by the capacity needed to prevent VSAG
and VSOAR from causing problems during load tran-
sients. Generally, once enough capacitance is added
to meet the overshoot requirement, undershoot at the
rising load edge is no longer a problem (see the VSAG
and VSOAR equations in the
Transient Response
sec-
tion). Thus, the output capacitor selection requires
carefully balancing capacitor chemistry limitations
(capacitance vs. ESR vs. voltage rating) and cost.
Output Capacitor Stability Considerations
For Quick-PWM controllers, stability is determined by the
in-phase feedback ripple relative to the switching frequen-
cy, which is typically dominated by the output ESR. The
boundary of instability is given by the following equation:
where COUT is the total output capacitance, RESR is the
total ESR of the output capacitors, RCS is the current-
sense resistance, and ACS is the current-sense gain as
determined by the ILIM setting. ACS equals 2, 2.67, 4, and
8 for ILIM settings of 5V, 3.3V, 2V, and GND, respectively.
For a 300kHz application, the effective zero frequency
must be well below 95kHz, preferably below 50kHz. For
the standard application circuit with ceramic output
capacitors, the output ripple cannot be relied upon to
be in phase with the inductor current due to the low
ESR of the ceramic capacitors. Stability is mainly
dependent on the current-sense gain. With ILIM = 2V,
ACS = 4, and an effective current-sense resistance of
approximately 3.5mΩ, then the ESR zero works out to:
1/[2πx (2x330µF + 5 x 10µF) x 4 x 3.5mΩ] = 16kHz
This is well within the stability requirements.
RRAR
EFF ESR CS CS
=+
RfC
EFF SW OUT
≥1
2
f
RC
SW
EFF OUT
ππ
≥1
2
RVf L
VV V V
ESR IN SW
IN OUT OUT RIPPLE
≤−
()
⎡
⎣
⎢
⎢
⎤
⎦
⎥
⎥
RR V
I
ESR PCB STEP
LOAD MAX
+
()
≤∆()
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
30 ______________________________________________________________________________________
MAX17007
MAX17008
COUT
INPUT (VIN)
CIN
b) LOSSLESS INDUCTOR SENSING
FOR THERMAL COMPENSATION:
R2 SHOULD CONSIST OF AN NTC RESISTOR IN
SERIES WITH A STANDARD THIN-FILM RESISTOR.
CSL_
CSH_
PGND
DL_
DH_
LX_
CEQ
R1R2
NH
NLDL
L
INDUCTOR
RDCR
RCS = R2 = RDCR
R1 + R2
RDCR = L [ 1 + 1 ]
CEQ R1 R2
Figure 14. Current-Sense Configurations (Sheet 2 of 2)
When only using ceramic output capacitors, output
overshoot (VSOAR) typically determines the minimum
output capacitance requirement. Their relatively low
capacitance value can allow significant output over-
shoot when stepping from full-load to no-load condi-
tions, unless designed with a small inductance value
and high switching frequency to minimize the energy
transferred from the inductor to the capacitor during
load-step recovery.
Unstable operation manifests itself in two related but
distinctly different ways: double pulsing and feedback
loop instability. Double pulsing occurs due to noise on
the output or because the ESR is so low that there is
not enough voltage ramp in the output voltage signal.
This “fools” the error comparator into triggering a new
cycle immediately after the minimum off-time period
has expired. Double pulsing is more annoying than
harmful, resulting in nothing worse than increased out-
put ripple. However, it can indicate the possible pres-
ence of loop instability due to insufficient ESR. Loop
instability can result in oscillations at the output after
line or load steps. Such perturbations are usually
damped, but can cause the output voltage to rise
above or fall below the tolerance limits.
The easiest method for checking stability is to apply a
very fast zero-to-max load transient and carefully
observe the output voltage ripple envelope for over-
shoot and ringing. It can help to simultaneously monitor
the inductor current with an AC current probe. Do not
allow more than one cycle of ringing after the initial
step-response under/overshoot.
Input Capacitor Selection
The input capacitor must meet the ripple current
requirement (IRMS) imposed by the switching currents.
The IRMS requirements can be determined by the fol-
lowing equation for a single-phase application:
In combined mode, the input RMS current simplifies to:
where ILOAD is the combined output current of both
phases.
For most applications, nontantalum chemistries (ceram-
ic, aluminum, or OS-CON) are preferred due to their
resistance to inrush surge currents typical of systems
with a mechanical switch or connector in series with the
input. If the Quick-PWM controller is operated as the
second stage of a two-stage power-conversion system,
tantalum input capacitors are acceptable. In either con-
figuration, choose an input capacitor that exhibits less
than +10°C temperature rise at the RMS input current
for optimal circuit longevity.
Power-MOSFET Selection
Most of the following MOSFET guidelines focus on the
challenge of obtaining high load-current capability
when using high-voltage (> 20V) AC adapters. Low-
current applications usually require less attention.
The high-side MOSFET (NH) must be able to dissipate
the resistive losses plus the switching losses at both
VIN(MIN) and VIN(MAX). Calculate both of these sums.
Ideally, the losses at VIN(MIN) should be roughly equal to
losses at VIN(MAX), with lower losses in between. If the
losses at VIN(MIN) are significantly higher than the losses
at VIN(MAX), consider increasing the size of NH(reducing
RDS(ON) but with higher CGATE). Conversely, if the loss-
es at VIN(MAX) are significantly higher than the losses at
VIN(MIN), consider reducing the size of NH(increasing
RDS(ON) to lower CGATE). If VIN does not vary over a
wide range, the minimum power dissipation occurs
where the resistive losses equal the switching losses.
Choose a low-side MOSFET that has the lowest possible
on-resistance (RDS(ON)), comes in a moderate-sized
package (i.e., one or two 8-pin SOs, DPAK, or D2PAK),
and is reasonably priced. Make sure that the DL gate
driver can supply sufficient current to support the gate
charge and the current injected into the parasitic gate-
to-drain capacitor caused by the high-side MOSFET
turning on; otherwise, cross-conduction problems might
occur (see the
MOSFET Gate Drivers (DH, DL)
section).
MOSFET Power Dissipation
Worst-case conduction losses occur at the duty factor
extremes. For the high-side MOSFET (NH), the worst-
case power dissipation due to resistance occurs at the
minimum input voltage:
Generally, a small high-side MOSFET is desired to
reduce switching losses at high input voltages.
However, the RDS(ON) required to stay within package
power dissipation often limits how small the MOSFET
can be. Again, the optimum occurs when the switching
losses equal the conduction (RDS(ON)) losses. High-
side switching losses do not usually become an issue
until the input is greater than approximately 15V.
PD NH sistive V
VIR
OUT
IN MIN LOAD DS ON
(Re )
() ()
=⎛
⎝
⎜⎞
⎠
⎟
()
2
II
VVVV
RMS LOAD
IN OUT IN OUT
=⎛
⎝
⎜⎞
⎠
⎟−
()
22
II V VV I V VV
V
RMS
LOAD OUT IN OUT LOAD OUT IN OUT
IN
=−
()
+−
()
1
2112
222
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 31
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
32 ______________________________________________________________________________________
Calculating the power dissipation in high-side MOSFET
(NH) due to switching losses is difficult since it must
allow for difficult quantifying factors that influence the
turn-on and turn-off times. These factors include the
internal gate resistance, gate charge, threshold voltage,
source inductance, and PCB layout characteristics. The
following switching-loss calculation provides only a
very rough estimate and is no substitute for breadboard
evaluation, preferably including verification using a
thermocouple mounted on NH:
where COSS is the NHMOSFET’s output capacitance,
QG(SW) is the charge needed to turn on the NHMOS-
FET, and IGATE is the peak gate-drive source/sink cur-
rent (2.4A typ).
Switching losses in the high-side MOSFET can become
an insidious heat problem when maximum AC adapter
voltages are applied due to the squared term in the C x
VIN2x fSW switching-loss equation. If the high-side
MOSFET chosen for adequate RDS(ON) at low battery
voltages becomes extraordinarily hot when biased from
VIN(MAX), consider choosing another MOSFET with
lower parasitic capacitance.
For the low-side MOSFET (NL), the worst-case power
dissipation always occurs at maximum input voltage:
The worst case for MOSFET power dissipation occurs
under heavy overloads that are greater than
ILOAD(MAX), but are not quite high enough to exceed
the current limit and cause the fault latch to trip. To pro-
tect against this possibility, you can “over design” the
circuit to tolerate:
where IVALLEY(MAX) is the maximum valley current
allowed by the current-limit circuit, including threshold
tolerance and on-resistance variation. The MOSFETs
must have a good size heatsink to handle the overload
power dissipation.
Choose a Schottky diode (DL) with a forward voltage
low enough to prevent the low-side MOSFET body
diode from turning on during the dead time. Select a
diode that can handle the load current during the dead
times. This diode is optional and can be removed if effi-
ciency is not critical.
Boost Capacitors
The boost capacitors (CBST) must be selected large
enough to handle the gate-charging requirements of
the high-side MOSFETs. Typically, 0.1µF ceramic
capacitors work well for low-power applications driving
medium-sized MOSFETs. However, high-current appli-
cations driving large, high-side MOSFETs require boost
capacitors larger than 0.1µF. For these applications,
select the boost capacitors to avoid discharging the
capacitor more than 200mV while charging the high-
side MOSFETs’ gates:
where N is the number of high-side MOSFETs used for
one regulator, and QGATE is the gate charge specified
in the MOSFET’s data sheet. For example, assume (2)
IRF7811W n-channel MOSFETs are used on the high
side. According to the manufacturer’s data sheet, a sin-
gle IRF7811W has a maximum gate charge of 24nC
(VGS = 5V). Using the above equation, the required
boost capacitance would be:
Selecting the closest standard value, this example
requires a 0.22µF ceramic capacitor.
Applications Information
Minimum Input Voltage Requirements
and Dropout Performance
The output-voltage adjustable range for continuous-
conduction operation is restricted by the nonadjustable
minimum off-time one-shot. For best dropout perfor-
mance, use the slower (200kHz) on-time settings. When
working with low input voltages, the duty-factor limit
must be calculated using worst-case values for on- and
off-times. Manufacturing tolerances and internal propa-
gation delays introduce an error to the on-times. This
error is greater at higher frequencies. Also, keep in
mind that transient response performance of buck reg-
ulators operated too close to dropout is poor, and bulk
output capacitance must often be added (see the
Transient Response
section (the VSAG equation) in the
Quick-PWM Design Procedure
section).
CnC
mV F
BST =×=
224
200 024.µ
CNQ
mV
BST GATE
=×
200
II I
II LIR
LOAD VALLEY MAX INDUCTOR
VALLEY MAX LOAD MAX
=+
⎛
⎝
⎜⎞
⎠
⎟
=+
⎛
⎝
⎜⎞
⎠
⎟
()
() ()
∆
2
2
PD NL sistive V
VIR
OUT
IN MAX LOAD DS ON
(Re )
() ()
=−⎛
⎝
⎜⎞
⎠
⎟
⎡
⎣
⎢
⎢
⎤
⎦
⎥
⎥
()
12
PD NHSwitching V I f Q
I
IN MAX LOAD SW GSW
GATE
()
() ()
=⎛⎛
⎝
⎜⎞
⎠
⎟
+COSSVVf
IN MAX SW()
2
2
In a single-phase configuration, the absolute point of
dropout is when the inductor current ramps down dur-
ing the minimum off-time (∆IDOWN) as much as it ramps
up during the on-time (∆IUP). The ratio h = ∆IUP/
∆IDOWN is an indicator of the ability to slew the inductor
current higher in response to increased load and must
always be greater than 1. As h approaches 1—the
absolute minimum dropout point—the inductor current
cannot increase as much during each switching cycle,
and VSAG greatly increases unless additional output
capacitance is used. A reasonable minimum value for h
is 1.5, but adjusting this up or down allows trade-offs
between VSAG, output capacitance, and minimum
operating voltage. For a given value of h, the minimum
operating voltage can be calculated as:
where VCHG is the parasitic voltage drop in the charge
path (see the
On-Time One-Shot
section), and tOFF(MIN)
is from the
Electrical Characteristics
table. The absolute
minimum input voltage is calculated with h = 1.
If the calculated VIN(MIN) is greater than the required min-
imum input voltage, then reduce the operating frequency
or add output capacitance to obtain an acceptable VSAG.
If operation near dropout is anticipated, calculate VSAG to
be sure of adequate transient response.
Dropout Design Example:
VOUT = 1.5V
fSW = 300kHz
tOFF(MIN) = 250ns
VCHG = 150mV (10A load)
h = 1.5
Calculating again with h = 1 gives the absolute limit of
dropout:
Therefore, VIN must be greater than 1.78V, even with
very large output capacitance, and a practical input volt-
age with reasonable output capacitance would be 2.0V.
PCB Layout Guidelines
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. The switching
power stage requires particular attention. If possible,
mount all the power components on the top side of the
board with their ground terminals flush against one
another. Follow these guidelines for good PCB layout:
• Keep the high-current paths short, especially at the
ground terminals. This is essential for stable, jitter-
free operation.
• Connect all analog grounds to a separate solid
copper plane, which connects to the GND pin of
the Quick-PWM controller. This includes the VCC
bypass capacitor, REF bypass capacitors, REFIN1
components, and feedback compensation/dividers.
• Keep the power traces and load connections short.
This is essential for high efficiency. The use of thick
copper PCBs (2oz vs. 1oz) can enhance full-load
efficiency by 1% or more. Correctly routing PCB
traces is a difficult task that must be approached in
terms of fractions of centimeters, where a single
mΩof excess trace resistance causes a measur-
able efficiency penalty.
• Keep the high current, gate-driver traces (DL, DH,
LX, and BST) short and wide to minimize trace
resistance and inductance. This is essential for
high-power MOSFETs that require low-impedance
gate drivers to avoid shoot-through currents.
• When trade-offs in trace lengths must be made, it is
preferable to allow the inductor charging path to be
made longer than the discharge path. For example,
it is better to allow some extra distance between the
input capacitors and the high-side MOSFET than to
allow distance between the inductor and the low-
side MOSFET or between the inductor and the out-
put filter capacitor.
• Route high-speed switching nodes away from sensi-
tive analog areas (REF, REFIN1, FB2, CSH, and CSL).
Layout Procedure
1) Place the power components first, with ground ter-
minals adjacent (low-side MOSFET source, CIN,
COUT, and anode of the low-side Schottky). If possi-
ble, make all these connections on the top layer
with wide, copper-filled areas.
2) Mount the controller IC adjacent to the low-side
MOSFET. The DL gate traces must be short and
wide (50 mils to 100 mils wide if the MOSFET is 1in
from the controller IC).
VVmV
s kHz V
IN MIN()
.
(. . ) .=+
−××
⎡
⎣
⎢⎤
⎦
⎥=
1 5 150
1 0 25 1 0 300 178
µ
VVmV
s kHz V
IN MIN()
.
(. . ) .=+
−××
⎡
⎣
⎢⎤
⎦
⎥=
1 5 150
1 0 25 1 5 300 186
µ
VVV
ht f
IN MIN OUT CHG
OFF MIN SW
()
()
=−
−×
()
⎛
⎝
⎜
⎜
⎞
⎠
⎟
⎟
1
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
______________________________________________________________________________________ 33
MAX17007/MAX17008
3) Group the gate-drive components (BST capacitors,
VDD bypass capacitor) together near the controller IC.
4) Make the DC-DC controller ground connections as
shown in Figures 1 and 2. This diagram can be
viewed as having four separate ground planes: I/O
ground, where all the high-power components go;
the power ground plane, where the PGND pin and
VDD bypass capacitor go; the master’s analog
ground plane where sensitive analog components,
the master’s GND pin, and VCC bypass capacitor
go; and the slave’s analog ground plane where the
slave’s GND pin and VCC bypass capacitor go. The
master’s GND plane must meet the PGND plane
only at a single point directly beneath the IC.
Similarly, the slave’s GND plane must meet the
PGND plane only at a single point directly beneath
the IC. The respective master and slave ground
planes should connect to the high-power output
ground with a short metal trace from PGND to the
source of the low-side MOSFET (the middle of the
star ground). This point must also be very close to
the output capacitor ground terminal.
5) Connect the output power planes (VOUT and sys-
tem ground planes) directly to the output filter
capacitor positive and negative terminals with multi-
ple vias. Place the entire DC-DC converter circuit
as close to the load as is practical. See Figure 15.
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
34 ______________________________________________________________________________________
OUTPUT 1
SMPS1
COUT1
COUT1
COUT2
COUT2
CIN1
CIN2
SMPS2
OUTPUT 2
KELVIN SENSE VIAS
UNDER THE INDUCTOR
(SEE MAX17007 EVALUATION KIT)
VIA TO POWER
GROUND
VCC BYPASS
CAPACITOR
REF BYPASS
CAPACITOR
X-RAY VIEW.
IC MOUNTED
ON BOTTOM
SIDE OF PCB.
IC LAYOUT
CONNECT GND AND PGND THE
CONTROLLER AT ONE POINT
ONLY AS SHOWN
CONNECT THE
EXPOSED PAD TO
ANALOG GND
VIA TO ANALOG
GROUND
POWER STAGE LAYOUT (TOP SIDE OF PCB)
INDUCTOR
L1
INDUCTOR
L2
INPUT
POWER GROUND
+
INDUCTOR DCR SENSING
KELVIN SENSE VIAS TO
INDUCTOR PAD
CSL
CSH
Figure 15. PCB Layout Example
MAX17007/MAX17008
Dual and Combinable QPWM Graphics
Core Controllers for Notebook Computers
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
35
____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600
© 2008 Maxim Integrated Products is a registered trademark of Maxim Integrated Products, Inc.
Package Information
For the latest package outline information, go to
www.maxim-ic.com/packages.
PACKAGE TYPE PACKAGE CODE DOCUMENT NO.
28 TQFN T2844-1 21-0139
Chip Information
TRANSISTOR COUNT: 13,103
PROCESS: BiCMOS