General Description
The MAX16818 pulse-width modulation (PWM)
LED driver controller provides high-output-current
capability in a compact package with a minimum number
of external components. The MAX16818 is suitable for
use in synchronous and nonsynchronous step-down
(buck) topologies, as well as in boost, buck-boost, SEPIC,
and Cuk LED drivers. The MAX16818 is the first LED
driver controller that enables Maxim’s technology for
fast LED current transients of up to 20A/µs and 30kHz
dimming frequency.
This device utilizes average-current-mode control that
enables optimal use of MOSFETs with optimal charge and
on-resistance characteristics. This results in the minimized
need for external heatsinking even when delivering up
to 30A of LED current. True differential sensing enables
accurate control of the LED current. A wide dimming range
is easily implemented to accommodate an external PWM
signal. An internal regulator enables operation over a wide
input voltage range: 4.75V to 5.5V or 7V to 28V and above
with a simple external biasing device. The wide switching
frequency range, up to 1.5MHz, allows for the use of small
inductors and capacitors.
The MAX16818 features a clock output with 180° phase
delay to control a second out-of-phase LED driver to
reduce input and output filter capacitors size or to mini-
mize ripple currents. The MAX16818 offers programmable
hiccup, overvoltage, and overtemperature protection.
The MAX16818ETI+ is rated for the extended tempera-
ture range (-40°C to +85°C) and the MAX16818ATI+ is
rated for the automotive temperature range (-40°C to
+125°C). This LED driver controller is available in a lead-
free, 0.8mm high, 5mm x 5mm 28-pin TQFN package with
an exposed pad.
Applications
●Front Projectors/Rear Projection TVs
●Portable and Pocket Projectors
●LCD TVs and Display Backlight
Benets and Features
●Flexible Architecture Supports a Range of LED
LIghting Applications with Minimal Component Count
• Up to 30A Output Current
• 4.75V to 5.5V or 7V to 28V Input Voltage Range
• Average-Current-Mode Control
• True-Differential Remote-Sense Input
●Frequency Management Reduces EMI and
Interference with Other System Clocks
• Programmable Switching Frequency or External
Synchronization from 125kHz to 1.5MHz
• Clock Output for 180° Out-of-Phase Operation
●Integrated Protection Features and Thermally
Enhanced Package Improve System Reliability
• Output Overvoltage and Hiccup-Mode Overcurrent
Protection
• Thermal Shutdown
• Thermally Enhanced 28-Pin TQFN Package
+Denotes a lead(Pb)-free/RoHS-compliant package.
*EP = Exposed pad.
Package Information appears at end of data sheet.
PART TEMP RANGE PIN-PACKAGE
MAX16818ATI+ -40°C to +125°C 28 TQFN-EP*
MAX16818ETI+ -40°C to +85°C 28 TQFN-EP*
Q1
HIGH-FREQUENCY
PULSE TRAIN
C2
Q3
L1
R1
V
LED
C1
7V TO 28V
CSP
DL
DH
NOTE: MAXIM TOPOLOGY
PGND
EN
IN
ILIM
OVI
CLP
Q2
MAX16818
MAX16818 1.5MHz, 30A High-Efficiency, LED Driver
with Rapid LED Current Pulsing
19-0666; Rev 4; 5/15
Simplied Diagram
Ordering Information
EVALUATION KIT AVAILABLE
IN to SGND ...........................................................-0.3V to +30V
BST to SGND ........................................................-0.3V to +35V
BST to LX ................................................................-0.3V to +6V
DH to LX .................................... -0.3V to [(VBST - VLX_) + 0.3V]
DL to PGND ............................................. -0.3V to (VDD + 0.3V)
VCC to SGND ..........................................................-0.3V to +6V
VCC, VDD to PGND .................................................-0.3V to +6V
SGND to PGND ....................................................-0.3V to +0.3V
All Other Pins to SGND ............................ -0.3V to (VCC + 0.3V)
Continuous Power Dissipation (TA = +70°C)
28-Pin TQFN (derate 34.5mW/°C above +70°C) ......2758mW
Operating Temperature Range
MAX16818ATI+ ........................................... .-40°C to +125°C
MAX16818ETI+ .............................................. -40°C to +85°C
Maximum Junction Temperature .....................................+150°C
Storage Temperature Range ............................ -60°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
(VCC = 5V, VDD = VCC, TA = TJ = TMIN to TMAX, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
SYSTEM SPECIFICATIONS
Input Voltage Range VIN
7 28
V
Short IN and VCC together for 5V input
operation 4.75 5.50
Quiescent Supply Current IQEN = VCC or SGND, not switching 2.7 5.5 mA
LED CURRENT REGULATOR
SENSE+ to SENSE- Accuracy
(Note 2)
No load, VIN = 4.75V to 5.5V, fSW = 500kHz 0.594 0.6 0.606 V
No load, VIN = 7V to 28V, fSW = 500kHz 0.594 0.6 0.606
Soft-Start Time tSS 1024 Clock
Cycles
STARTUP/INTERNAL REGULATOR
VCC Undervoltage Lockout UVLO VCC rising 4.1 4.3 4.5 V
VCC Undervoltage Hysteresis 200 mV
VCC Output Voltage VIN = 7V to 28V, ISOURCE = 0 to 60mA 4.85 5.1 5.30 V
MOSFET DRIVERS
Output Driver Impedance RON Low or high output, ISOURCE/SINK = 20mA 1.1 3.0 W
Output Driver Source/Sink Current IDH,IDL 4 A
Nonoverlap Time tNO CDH/DL = 5nF 35 ns
OSCILLATOR
Switching Frequency Range 125 1500 kHz
Switching Frequency
fSW
RT = 500kW121 125 129
kHzSwitching Frequency RT = 120kW495 521 547
Switching Frequency RT = 39.9kW1515 1620 1725
Switching Frequency Accuracy 120kW ≤ RT ≤ 500kW-5 +5 %
40kW ≤ RT ≤ 120kW-8 +8
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2
MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Electrical Characteristics
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional operation of the device at these
or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to absolute maximum rating conditions for extended periods may affect
device reliability.
Absolute Maximum Ratings
(VCC = 5V, VDD = VCC, TA = TJ = TMIN to TMAX, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
CLKOUT Phase Shift q_CLKOUT With respect to DH, fSW = 125kHz 180 Degrees
CLKOUT Output Low Level VCLKOUTL ISINK = 2mA 0.4 V
CLKOUT Output High Level VCLKOUTH ISOURCE = 2mA 4.5 V
SYNC Input-High Pulse Width tSYNC 200 ns
SYNC Input Clock High Threshold VSYNCH 2.0 V
SYNC Input Clock Low Threshold VSYNCL 0.4 V
SYNC Pullup Current ISYNC_OUT VRT/SYNC = 0V 250 750 µA
SYNC Power-Off Level VSYNC_OFF 0.4 V
INDUCTOR CURRENT LIMIT
Average Current-Limit Threshold VCL CSP to CSN 24.0 26.9 28.2 mV
Reverse Current-Limit Threshold VCLR CSP to CSN -3.2 -2.3 -0.1 mV
Cycle-by-Cycle Current Limit CSP to CSN 60 mV
Cycle-by-Cycle Overload
Response Time VCSP to VCSN = 75mV 260 ns
Hiccup Divider Ratio LIM to VCM, no switching 0.547 0.558 0.565 V/V
Hiccup Reset Delay 200 ms
LIM Input Impedance LIM to SGND 55.9 kW
CURRENT-SENSE AMPLIFIER
CSP or CSN Input Resistance RCS 4 kW
Common-Mode Range VCMR(CS) VIN = 7V to 28V 0 5.5 V
Input Offset Voltage VOS(CS) 0.1 mV
Amplifier Gain AV(CS) 34.5 V/V
3dB Bandwidth f3dB 4 MHz
CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
Transconductance gm550 µS
Open-Loop Gain AVOL(CE) No load 50 dB
DIFFERENTIAL VOLTAGE AMPLIFIER FOR LED CURRENT (DIFF)
Common-Mode Voltage Range VCMR(DIFF) 0 +1.0 V
DIFF Output Voltage VCM VSENSE+ = VSENSE- = 0V 0.6 V
Input Offset Voltage VOS(DIFF) -1 +1 mV
Amplifier Gain AV(DIFF) 0.994 1 1.006 V/V
3dB Bandwidth f3dB CDIFF = 20pF 3 MHz
Minimum Output-Current Drive IOUT(DIFF) 4 mA
SENSE+ to SENSE- Input
Resistance RVS VSENSE- = 0V 50 100 kW
V_IOUT AMPLIFIER
Gain-Bandwidth Product VV_IOUT = 2.0V 4 MHz
3dB Bandwidth VV_IOUT = 2.0V 1 MHz
Output Sink Current 30 µA
Output Source Current 90 µA
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Electrical Characteristics (continued)
Note 1: Specifications at TA = +25°C are 100% tested. Specifications over the temperature range are guaranteed by design.
Note 2: Does not include an error due to finite error amplifier gain. See the Voltage-Error Amplifier (EAOUT) section.
(VCC = 5V, VDD = VCC, TA = TJ = TMIN to TMAX, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Maximum Load Capacitance 50 pF
V_IOUT Output to IOUT Transfer
Function RSENSE = 1mW, 100mV ≤ V_IOUT ≤ 5.5V 132.3 135 137.7 mV/A
Offset Voltage 1 mV
VOLTAGE-ERROR AMPLIFIER (EAOUT)
Open-Loop Gain AVOLEA 70 dB
Unity-Gain Bandwidth fGBW 3 MHz
EAN Input Bias Current IB(EA) VEAN = 2.0V -0.2 +0.03 +0.2 µA
Error Amplifier Output Clamping
Voltage VCLAMP(EA) With respect to VCM 883 930 976 mV
POWER-GOOD AND OVERVOLTAGE PROTECTION
PGOOD Trip Level VUV PGOOD goes low when VOUT is below this
threshold 87.5 90 92.5 %VOUT
PGOOD Output Low Level VPGLO ISINK = 4mA 0.4 V
PGOOD Output Leakage Current IPG PGOOD = VCC 1 µA
OVI Trip Threshold OVPTH With respect to SGND 1.244 1.276 1.308 V
OVI Input Bias Current IOVI 0.2 µA
ENABLE INPUT
EN Input High Voltage VEN EN rising 2.437 2.5 2.562 V
EN Input Hysteresis 0.28 V
EN Pullup Current IEN 13.5 15 16.5 µA
THERMAL SHUTDOWN
Thermal Shutdown Temperature rising 150 °C
Thermal Shutdown Hysteresis 30 °C
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Electrical Characteristics (continued)
(TA = +25°C, using Figure 5, unless otherwise noted.)
LOW-SIDE DRIVER (DL) SINK
AND SOURCE CURRENT
MAX16818 toc09
3A/div
100ns/div
CLOAD = 22nF
VIN = 12V
HIGH-SIDE DRIVER (DH) SINK
AND SOURCE CURRENT
MAX16818 toc08
2A/div
100ns/div
CLOAD = 22nF
VIN = 12V
DRIVER FALL TIME
vs. DRIVER LOAD CAPACITANCE
MAX16818 toc07
CAPACITANCE (nF)
tF (ns)
2116116
20
40
60
80
100
0
1
VIN = 12V
fSW = 250kHz
DH
DL
DRIVER RISE TIME
vs. DRIVER LOAD CAPACITANCE
MAX16818 toc06
CAPACITANCE (nF)
tR (ns)
2116116
20
40
60
80
100
0
1
VIN = 12V
fSW = 250kHz
DH
DL
VCC LOAD REGULATION
vs. INPUT VOLTAGE
MAX16818 toc05
VCC LOAD CURRENT (mA)
VCC (V)
125100755025
4.85
4.95
5.05
5.15
5.25
4.75
0 150
VIN = 12V
VIN = 5V
VIN = 24V
HICCUP CURRENT LIMIT vs. REXT
MAX16818 toc04
REXT (MΩ)
CURRENT LIMIT (A)
161284
23.5
24.0
24.5
25.0
25.5
26.0
23.0
0 20
VIN = 12V
fSW = 250kHz
R1 = 1mΩ
VOUT = 1.5V
CURRENT-SENSE THRESHOLD
vs. OUTPUT VOLTAGE
MAX16818 toc03
VOUT (V)
(VCSP - VCSN) (mV)
4321
26.5
27.0
27.5
28.0
28.5
29.0
26.0
0 5
VIN = 12V
fSW = 250kHz
SUPPLY CURRENT vs. TEMPERATURE
MAX16818 toc02
TEMPERATURE (°C)
SUPPLY CURRENT (mA)
603510-15
62
64
66
68
70
60
-40 85
VIN = 12V
fSW = 250kHz
CDL/CDH = 22nF
SUPPLY CURRENT (IQ) vs. FREQUENCY
MAX16818 toc01
FREQUENCY (kHz)
SUPPLY CURRENT (mA)
13001100900700500300
10
20
30
40
50
60
0
100 1500
EXTERNAL CLOCK
NO DRIVER LOAD
VIN = 12V
VIN = 24V
VIN = 5V
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Typical Operating Characteristics
(TA = +25°C, using Figure 5, unless otherwise noted.)
SYNC, CLKOUT, AND LX WAVEFORM
MAX16818 toc16
1µs/div
CLKOUT
5V/div
VIN = 12V
fSW = 250kHz
LX
10V/div
SYNC
5V/div
FREQUENCY vs. TEMPERATURE
MAX16818 toc15
TEMPERATURE (°C)
fSW (kHz)
603510-15
242
244
246
248
250
252
254
256
258
260
240
-40 85
VIN = 12V
FREQUENCY vs. RT
MAX16818 toc14
RT (kΩ)
fSW (kHz)
470
430
390
350
310
270
230
190
150
110
70
1000
10,000
100
30 510
VIN = 12V
LOW-SIDE DRIVER (DL) FALL TIME
MAX16818 toc13
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
LOW-SIDE DRIVER (DL) RISE TIME
MAX16818 toc12
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
HIGH-SIDE DRIVER (DH) FALL TIME
MAX16818 toc11
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
HIGH-SIDE DRIVER (DH) RISE TIME
MAX16818 toc10
2V/div
40ns/div
CLOAD = 22nF
VIN = 12V
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Typical Operating Characteristics (continued)
PIN NAME FUNCTION
1 PGND Power-Supply Ground
2, 7 N.C. No Connection. Not internally connected.
3 DL Low-Side Gate-Driver Output
4 BST Boost Flying Capacitor Connection. Reservoir capacitor connection for the high-side MOSFET driver
supply. Connect a ceramic capacitor between BST and LX.
5 LX Source Connection for the High-Side MOSFET
6 DH High-Side Gate-Driver Output. Drives the gate of the high-side MOSFET.
8, 22, 25 SGND Signal Ground. Ground connection for the internal control circuitry. Connect SGND and PGND
together at one point near the IC.
9 CLKOUT Oscillator Output. Rising edge of CLKOUT is phase-shifted from the rising edge of DH by 180°.
10 PGOOD Power-Good Output
11 EN
Output Enable. Drive high or leave unconnected for normal operation. Drive low to shut down the
power drivers. EN has an internal 15µA pullup current. Connect a capacitor from EN to SGND to
program the hiccup-mode duty cycle.
12 RT/SYNC
Switching-Frequency Programming and Chip-Enable Input. Connect a resistor from RT/SYNC to
SGND to set the internal oscillator frequency. Drive RT/SYNC to synchronize the switching frequency
with external clock.
13 V_IOUT Voltage Source Output Proportional to the Inductor Current. The voltage at V_IOUT = 135 x ILED x RS.
14 LIM Current-Limit Setting Input. Connect a resistor from LIM to SGND to set the hiccup current-limit
threshold. Connect a capacitor from LIM to SGND to ignore short output overcurrent pulses.
15 OVI
Overvoltage Protection. Connect OVI to DIFF. When OVI exceeds 12.7% above the programmed
output voltage, DH is latched low and DL is latched high. Toggle EN or recycle the input power to reset
the latch.
16 CLP Current-Error Amplifier Output. Compensate the current loop by connecting an RC network to ground.
17 EAOUT Voltage-Error Amplifier Output. Connect to the external compensation network.
18 EAN Voltage-Error Amplifier Inverting Input
19 DIFF Differential Remote-Sense Amplifier Output. DIFF is the output of a precision unity-gain amplifier
whose inputs are SENSE+ and SENSE-.
20 CSN Current-Sense Differential Amplifier Negative Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Pin Description
PIN NAME FUNCTION
21 CSP Current-Sense Differential Amplifier Positive Input. The differential voltage between CSN and CSP is
amplified internally by the current-sense amplifier (gain = 34.5) to measure the inductor current.
23 SENSE- Differential LED Current-Sensing Negative Input. SENSE- is used to sense the LED current. Connect
SENSE- to the negative side of the LED current-sense resistor.
24 SENSE+ Differential LED Current-Sensing Positive Input. SENSE+ is used to sense the LED current. Connect
SENSE+ to the positive side of the LED current-sense resistor.
26 IN Supply Voltage Connection. Connect IN to VCC for a +5V system.
27 VCC Internal +5V Regulator Output. VCC is derived from the IN voltage. Bypass VCC to SGND with 4.7µF
and 0.1µF ceramic capacitors.
28 VDD
Supply Voltage for Low-Side and High-Side Drivers. Connect a parallel combination of 0.1µF and 1µF
ceramic capacitors to PGND and a 1W resistor to VCC to filter out the high peak currents of the driver
from internal circuitry.
— EP Exposed Pad. Connect the exposed pad to a copper pad (SGND) to improve power dissipation.
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Pin Description (continued)
Figure 1. Typical Application Circuit for a Boost LED Driver (Nonsynchronous)
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
L1
VLED
C1 LED
STRING
R8
VIN
C2
D1
VIN
7V TO 28V
C10
C9
C8
C7
R12
R7
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C3
C6 C5 C4
R1
R2
R3
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Typical Application Circuits
Figure 2. Typical Application Circuit for an Input-Referred Buck-Boost LED Driver (Input: 7V to 28V, Output: 1 to 6 LEDs in Series)
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
VLED
C1
LED
STRING
1 TO 6
LEDS
R8
VIN
C2
L1
D1
VIN
7V TO 28V
C10
C9
C8
C7
R12
R7
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C3
C6 C5 C4
R1
R2
R3
VCC
VCC
RS+
RS- OUT
MAX4073T
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Typical Application Circuits (continued)
Figure 3. Typical Application Circuit for a SEPIC LED Driver
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
L1
L2
VLED
C2 LED
STRING
R8
VIN
C3
D1
VIN
7V TO 28V
C11
C10
C9
C8
R12
R7
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C4
C7 C6 C5
R1
R2
R3
C1
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Typical Application Circuits (continued)
Figure 4. Application Circuit for a Ground-Referred Buck-Boost LED Driver
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q2
Q3
Q1
L1
VLED
D1
C1
LED
STRING
R8
VIN
C2
C4
VIN
7V TO 18V
C11
C10
C9
C8
R12
R7
D2
R4 R5
R6
VCC
ON/OFF
R11
R10
PGND
N.C.
N.C.
DL
BST
LX
DH
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
VLED
C3
C7 C6 C5
R1
R2
R3
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Typical Application Circuits (continued)
Figure 5. Application Circuit for a Buck LED Driver
1
2
3
4
5
6
8
7
21
20
19
18
17
16
15
9
10
11
12
1314
22 23 24 25 26 27 28
Q1
L1
C1
LED
STRING
R6
VIN
C2
C4
VIN
7V TO 28V
C11
C10
C9
C8
R10
R5
D1
R3
R4
C3
VCC
ON/OFF
R9
R8
PGND
N.C.
N.C.
DL
BST
LX
DH
LIM V_IOUT RT/SYNC EN PGOOD CLKOUT SGND
SGND SENSE- SENSE+ SGND IN V
CC
V
DD
OVI
CLP
EAOUT
EAN
DIFF
CSN
CSP
VCC
MAX16818
C7 C6 C5
R1
R2
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Typical Application Circuits (continued)
Figure 6. MAX16818 Functional Diagram
2 x fS (V/s)
RAMP
RT/SYNC
CSP
CSN
SGND
SENSE-
SENSE+
CLP
LIM
IN
EN
VDD
BST
DH
LX
DL
PGND
PGOOD
AV = 34.5
AV = 4
100kΩ
126.7kΩ
PWM
COMPARATOR
0.5V x VCC
TO INTERNAL
CIRCUITS HICCUP MODE
CURRENT LIMIT
S
R
Q
Q
V_IOUT
gm = 500µS
DIFF
CLKOUT
CLK
CPWM
CEA
VCLAMP
HIGH
VCLAMP
LOW
CA
VCC
0.1 x VREF
N
+0.6V
VREF = 0.6V
VCM (0.6V)
OVP COMP
0.12 x VREF
LATCH
RAMP
GENERATOR
SOFT-
START
OSCILLATOR
CLEAR ON UVLO RESET OR
ENABLE LOW
OVP LATCH
DIFF
AMP
EAN
EAOUT
OVI
VEA
ERROR AMP
0.5 x VCLAMP
VCM
VCM
IS
VCC
Ct
RT
S
R
Q
Q
UVLO
POR
TEMP SENSOR
5V
LDO
REGULATOR
MAX16818
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Functional Diagram
Detailed Description
The MAX16818 is a high-performance average-
current-mode PWM controller for high-power, high-
brightness LEDs (HB LEDs). Average current-mode
control is the ideal method for driving HB LEDs. This
technique offers inherently stable operation, reduces
component derating and size by accurately controlling
the inductor current. The device achieves high efficiency
at high current (up to 30A) with a minimum number of
external components. The high- and low-side drivers
source and sink up to 4A for lower switching losses while
driving high-gate-charge MOSFETs. The MAX16818's
CLKOUT output is 180° out-of-phase with respect to the
high-side driver. CLKOUT drives a second MAX16818
LED driver out of phase, reducing the input-capacitor
ripple current.
The MAX16818 consists of an inner average current loop
representing inductor current and an outer voltage loop
voltage-error amplifier (VEA) that directly controls LED
current. The combined action of the two loops results in
a tightly regulated LED current. The inductor current is
sensed across a current-sense resistor. The differential
amplifier senses LED current through a sense resistor
in series with the LEDs and the resulting sensed volt-
age is compared against an internal 0.6V reference at
the error-amplifier input. The MAX16818 will adjust the
LED current to within 1% accuracy to maintain emitted
spectrum of the light in HB LEDs.
IN, VCC, and VDD
The MAX16818 accepts either a 4.75V to 5.5V or 7V
to 28V input voltage range. All internal control circuitry
operates from an internally regulated nominal voltage of
5V (VCC). For input voltages of 7V or greater, the internal
VCC regulator steps the voltage down to 5V. The VCC
output voltage is a regulated 5V output capable of
sourcing up to 60mA. Bypass the VCC to SGND with
4.7µF and 0.1µF low-ESR ceramic capacitors for high-
frequency noise rejection and stable operation.
The MAX16818 uses VDD to power the low-side and high-
side drivers. Isolate VDD from VCC with a 1W resistor and
put a 1µF capacitor in parallel with a 0.1µF capacitor to
ground to prevent high-current noise spikes created by
the driver from disrupting internal circuitry.
The TQFN is a thermally enhanced package and can
dissipate up to 2.7W. The high-power packages allow
the high-frequency, high-current converter to operate
from a 12V or 24V bus. Calculate power dissipation in
the MAX16818 as a product of the input voltage and the
total VCC regulator output current (ICC). ICC includes
quiescent current (IQ) and gate-drive current (IDD):
PD = VIN x ICC
ICC = IQ + [fSW x (QG1 + QG2)]
where QG1 and QG2 are the total gate charge of the
low-side and high-side external MOSFETs at VGATE =
5V, IQ is 3.5mA (typ), and fSW is the switching frequency
of the converter.
Undervoltage Lockout (UVLO)
The MAX16818 includes an undervoltage lockout with
hysteresis and a power-on-reset circuit for converter turn-
on. The UVLO rising threshold is internally set at 4.35V
with a 200mV hysteresis. Hysteresis at UVLO eliminates
chattering during startup.
Most of the internal circuitry, including the oscillator, turns
on when the input voltage reaches 4V. The MAX16818
draws up to 3.5mA of current before the input voltage
reaches the UVLO threshold.
Soft-Start
The MAX16818 has an internal digital soft-start for a
monotonic, glitch-free rise of the output current. Soft-start
is achieved by the controlled rise of the error amplifier
dominant input in steps using a 5-bit counter and a 5-bit
DAC. The soft-start DAC generates a linear ramp from 0
to 0.7V. This voltage is applied to the error amplifier at a
third (noninverting) input. As long as the soft-start voltage
is lower than the reference voltage, the system converges
to that lower reference value. Once the soft-start DAC
output reaches 0.6V, the reference takes over and the
DAC output continues to climb to 0.7V, assuring that it
does not interfere with the reference voltage.
Internal Oscillator
The internal oscillator generates a clock with the frequen-
cy proportional to the inverse of RT. The oscillator fre-
quency is adjustable from 125kHz to 1.5MHz with better
than 8% accuracy using a single resistor connected from
RT/SYNC to SGND. The frequency accuracy avoids the
over-design, size, and cost of passive filter components
like inductors and capacitors. Use the following equation
to calculate the oscillator frequency:
For 120kW ≤ RT ≤ 500kW:
10
TSW
6.25 x 10
R f
=
For 40kW ≤ RT ≤ 120kW
10
TSW
6.40 x 10
R f
=
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
The oscillator also generates a 2VP-P voltage-ramp
signal for the PWM comparator and a 180? out-of-phase
clock signal for CLKOUT to drive a second LED regulator
out-of-phase.
Synchronization
The MAX16818 can be easily synchronized by
connecting an external clock to RT/SYNC. If an external
clock is present, then the internal oscillator is disabled
and the external clock is used to run the device. If the
external clock is removed, the absence of clock for
32µs is detected and the circuit starts switching from the
internal oscillator. Pulling RT/SYNC to ground for at least
50µs disables the converter. Use an open-collector
transistor to synchronize the MAX16818 with the external
system clock.
Control Loop
The MAX16818 uses an average-current-mode control
scheme to regulate the output current (Figure 7). The
main control loop consists of an inner current loop for
controlling the inductor current and an outer current
loop for regulating the LED current. The inner current
loop absorbs the inductor pole reducing the order of the
outer current loop to that of a single-pole system. The
current loop consists of a current-sense resistor (RS),
a current-sense amplifier (CA), a current-error amplifier
(CEA), an oscillator providing the carrier ramp, and a
PWM comparator (CPWM) (Figure 7). The precision CA
amplifies the sense voltage across RS by a factor of 34.5.
The inverting input to the CEA senses the CA output. The
CEA output is the difference between the voltage-error
amplifier output (EAOUT) and the amplified voltage from
the CA. The RC compensation network connected to
CLP provides external frequency compensation for the
CEA. The start of every clock cycle enables the high-
side drivers and initiates a PWM on-cycle. Comparator
CPWM compares the output voltage from the CEA with
a 0V to 2V ramp from the oscillator. The PWM on-cycle
terminates when the ramp voltage exceeds the error
voltage. Compensation for the outer LED current loop
varies based upon the topology.
The MAX16818 outer LED current control loop
consists of the differential amplifier (DIFF AMP), reference
volt age, and VEA. The unity-gain differential amplifier
provides true differential remote sensing of the volt-
age across the LED current set resistor, RLS. The
differential amplifier output connects to the inverting input
(EAN) of the VEA. The DIFF AMP is bypassed and the
inverting input is available to the pin for direct feedback. The
noninverting input of the VEA is internally connected to
an internal precision reference voltage, set to 0.6V. The
VEA controls the inner current loop (Figure 6). A feedback
network compensates the outer loop using the EAOUT and
EAIN pins.
Figure 7. MAX16818 Control Loop
DRIVE
Z COMP
CPWM
CEA
CA
CLP
VREF + VCM = 1.2V
CSPCSN
CCFF
VIN
COUT
RLS
RS
LED
STRING
IL
CCF RCF
VEA
600mV
SENSE+
EAOUT
SENSE-
EAN
DIFF
MAX16818
DIFF
AMP
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Inductor Current-Sense Amplier
The differential current-sense amplifier (CA) provides a
DC gain of 34.5. The maximum input offset voltage of the
current-sense amplifier is 1mV and the common-mode
voltage range is 0 to 5.5V (IN = 7V to 28V). The
current-sense amplifier senses the voltage across a
current-sense resistor. The maximum common-mode
voltage is 3.6V when VIN = 5V.
Inductor Peak-Current Comparator
The peak-current comparator provides a path for
fast cycle-by-cycle current limit during extreme fault
conditions, such as an inductor malfunction (Figure 8).
Note the average current-limit threshold of 26.9mV still
limits the output current during short-circuit conditions.
To prevent inductor saturation, select an inductor with a
saturation current specification greater than the average
current limit. Proper inductor selection ensures that only
the extreme conditions trip the peak-current comparator,
such as an inductor with a shorted turn. The 60mV thresh-
old for triggering the peak-current limit is twice the full-scale
average current-limit voltage threshold. The peak-current
comparator has only a 260ns delay.
Current-Error Amplier (for Inductor Currents)
The MAX16818 has a transconductance current-error
amplifier (CEA) with a typical gm of 550µS and 320µA
output sink- and source-current capability. The current-
error amplifier output CLP serves as the inverting input
to the PWM comparator. CLP is externally accessible
to provide frequency compensation for the inner current
loops (Figure 7). Compensate CEA so the inductor current
negative slope, which becomes the positive slope to the
inverting input of the PWM comparator, is less than the
slope of the internally generated voltage ramp (see the
Compensation section).
PWM Comparator and R-S Flip-Flop
The PWM comparator (CPWM) sets the duty cycle for
each cycle by comparing the output of the current-error
amplifier to a 2VP-P ramp. At the start of each clock cycle,
an R-S flip-flop resets and the high-side driver (DH) goes
high. The comparator sets the flip-flop as soon as the
ramp voltage exceeds the CLP voltage, thus terminating
the on-cycle (Figure 8).
Figure 8. MAX16818 Phase Circuit
2 x fS (V/s)
RAMP
CLK
CSP
CSN
EAN
EAOUT
SHDN
CLP
VDD
BST
DH
LX
DL
PGND
AV = 34.5
PEAK-CURRENT
COMPARATOR
60mV
S
R
Q
Q
gm = 550µS
CPWM
SET
CLR
CEA
CA
MAX16818
VEA
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Differential Amplier
The DIFF AMP facilitates remote sensing at the load
(Figure 7). It provides true differential LED current
(through the RLS sense resistor) sensing while rejecting
the common-mode voltage errors due to high-current
ground paths. The VEA provides the difference between
the differential amplifier output (DIFF) and the desired
LED current-sense voltage. The differential amplifier has
a bandwidth of 3MHz. The difference between SENSE+
and SENSE- is regulated to 0.6V. Connect SENSE+ to
the positive side of the LED current-sense resistor and
SENSE- to the negative side of the LED current-sense
resistor (which is often PGND).
MOSFET Gate Drivers (DH, DL)
The high-side (DH) and low-side (DL) drivers drive
the gates of external n-channel MOSFETs (Figures
1–5). The drivers' 4A peak sink- and source-current
capability provides ample drive for the fast rise and fall
times of the switching MOSFETs. Faster rise and fall
times result in reduced cross-conduction losses. Due to
physical realities, extremely low gate charges and RDS(ON)
resistance of MOSFETs are typically exclusive of each
other. MOSFETs with very low RDS(ON) will have a
higher gate charge and vice versa. Choosing the high-side
MOSFET (Q1) becomes a trade-off between these two
attributes. Applications where the input voltage is much
higher than the output voltage result in a low duty cycle
where conduction losses are less important than switch-
ing losses. In this case, choose a MOSFET with very low
gate charge and a moderate RDS(ON). Conversely, for
applications where the output voltage is near the input
voltage resulting in duty cycles much greater than 50%,
the RDS(ON) losses become at least equal, or even more
important than the switching losses. In this case, choose
a MOSFET with very low RDS(ON) and moderate gate
charge. Finally, for the applications where the duty cycle
is near 50%, the two loss components are nearly equal,
and a balanced MOSFET with moderate gate charge and
RDS(ON) work best.
In a buck topology, the low-side MOSFET (Q2) typically
operates in a zero voltage switching mode, thus it does
not have switching losses. Choose a MOSFET with very
low RDS(ON) and moderate gate charge.
Size both the high-side and low-side MOSFETs to handle
the peak and RMS currents during overload conditions.
The driver block also includes a logic circuit that provides
an adaptive nonoverlap time to prevent shoot-through
currents during transition. The typical nonoverlap time
between the high-side and low-side MOSFETs is 35ns.
BST
The MAX16818 uses VDD to power the low- and high-
side MOSFET drivers. The high-side driver derives
its power through a bootstrap capacitor and VDD
supplies power internally to the low-side driver. Connect a
0.47µF low-ESR ceramic capacitor between BST and LX.
Connect a Schottky rectifier from BST to VDD. Keep the
loop formed by the boost capacitor, rectifier, and IC small
on the PCB.
Protection
The MAX16818 includes output overvoltage protection
(OVP). During fault conditions when the load goes
to high impedance (opens), the controller attempts
to maintain LED current. The OVP disables the
MAX16818 whenever the voltage exceeds the thresh-
old, protecting the external circuits from undesirable
voltages.
Current Limit
The VEA output is clamped to 930mV with respect to
the common-mode voltage (VCM). Average-current-
mode control has the ability to limit the average
current sourced by the converter during a fault condition.
When a fault condition occurs, the VEA output clamps to
930mV with respect to the common-mode voltage
(0.6V) to limit the maximum current sourced by the
converter to ILIMIT = 26.9mV/RS. The hiccup current
limit overrides the average current limit. The MAX16818
includes hiccup current-limit protection to reduce the
power dissipation during a fault condition. The hiccup
current-limit circuit derives inductor current information
from the output of the current amplifier. This signal is
compared against one half of VCLAMP(EA). With no
resistor connected from the LIM pin to ground, the
hiccup current limit is set at 90% of the full-load average
current limit. Use REXT to increase the hiccup current
limit from 90% to 100% of the full load average limit.
The hiccup current limit can be disabled by connecting
LIM to SGND. In this case, the circuit follows the average-
current-limit action during overload conditions.
Overvoltage Protection
The OVP comparator compares the OVI input to the
overvoltage threshold. A detected overvoltage event
latches the comparator output forcing the power stage
into the OVP state. In the OVP state, the high-side
MOSFET turns off and the low-side MOSFET latches on.
Connect OVI to the center tap of a resistor-divider from
VLED to SGND. In this case, the center tap is compared
against 1.276V. Add an RC delay to reduce the sensitiv-
ity of the overvoltage circuit and avoid nuisance tripping
of the converter. Disable the overvoltage function by
connecting OVI to SGND.
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Applications Information
Application Circuit Descriptions
This section provides some detail regarding the application
circuits in the Simplified Diagram and Figures 1–5. The
discussion includes some description of the topology as
well as basic attributes.
High-Frequency LED Current Pulser
The Simplified Diagram shows the MAX16818
providing high-frequency, high-current pulses to the LEDs.
The basic topology must be a buck, since the inductor
always connects to the load in that configuration (in
all other topologies, the inductor disconnects from the
load at one time or another). The design minimizes the
current ripple by oversizing the inductor, which allows
for a very small (0.01µF) output capacitor. When
MOSFET Q3 turns on, it diverts the current around the
LEDs at a very fast rate. Q3 also discharges the output
capacitor, but since the capacitor is so small, it does
not stress the MOSFET. Resistor R1 senses the LED/
Q3 current and there is no reaction to the short that Q3
places across the LEDs. This design is superior in
that it does not attempt to actually change the inductor
current at high frequencies and yet the current in the
LEDs varies from zero to full in very small periods of time.
The efficiency of this technique is very high. Q3 must be
able to dissipate the LED current applied to its RDS(ON)
at some maximum duty cycle. If the circuit needs to
control extremely high currents, use paralleled
MOSFETs. PGOOD is low during LED pulsed-current
operation.
Boost LED Driver
In Figure 1, the external components configure the
MAX16818 as a boost converter. The circuit applies the
input voltage to the inductor during the on-time, and
then during the off-time the inductor, which is in series
with the input capacitor, charges the output capacitor.
Because of the series connection between the input
voltage and the inductor, the output voltage can never
go lower than the input voltage. The design is
nonsynchronous, and since the current-sense resistor
connects to ground, the power supply can go to any output
voltage (above the input) as long as the components are
rated appropriately. R2 again provides the sense voltage
the MAX16818 uses to regulate the LED current.
Input-Referenced LED Driver
The circuit in Figure 2 shows a step-up/step-down
regulator. It is similar to the boost converter in Figure 1 in
that the inductor is connected to the input and the MOSFET
is essentially connected to ground. However, rather than
going from the output to ground, the LEDs span from the
output to the input. This effectively removes the boost-only
restriction of the regulator in Figure 1, allowing the voltage
across the LEDs to be greater than or less than the input
voltage. LED current sensing is not ground-referenced,
so a high-side current-sense amplifier is used to measure
current.
SEPIC LED Driver
Figure 3 shows the MAX16818 configured as a SEPIC
LED driver. While buck topologies require the output to
be lesser than the input, and boost topologies require
the output to be greater than the input, a SEPIC topology
allows the output voltage to be greater than, equal to,
or less than the input. In a SEPIC topology, the voltage
across C1 is the same as the input voltage, and L1 and L2
are the same inductance. Therefore, when Q1 conducts
(on-time), both inductors ramp up current at the same rate.
The output capacitor supports the output voltage during
this time. During the off-time, L1 current recharges C1 and
combines with L2 to provide current to recharge C2 and
supply the load current. Since the voltage waveform across
L1 and L2 are exactly the same, it is possible to wind both
inductors on the same core (a coupled inductor). Although
voltages on L1 and L2 are the same, RMS currents can be
quite different so the windings may have a different gauge
wire. Because of the dual inductors and segmented energy
transfer, the efficiency of a SEPIC converter is somewhat
lower than standard bucks or boosts. As in the boost
driver, the current-sense resistor connects to ground,
allowing the output voltage of the LED driver to exceed the rated
maximum voltage of the MAX16818.
Ground-Referenced Buck/Boost LED Driver
Figure 4 depicts a buck/boost topology. During the
on-time with this circuit, the current flows from the input
capacitor, through Q1, L1, and Q3 and back to the input
capacitor. During the off-time, current flows up through
Q2, L1, D1, and to the output capacitor C1. This topology
resembles a boost in that the inductor sits between the
input and ground during the on-time. However, during
the off-time the inductor resides between ground and the
output capacitor (instead of between the input and output
capacitors in boost topologies), so the output voltage can
be any voltage less than, equal to, or greater than the
input voltage. As compared to the SEPIC topology, the
buck/boost does not require two inductors or a series
capacitor, but it does require two additional MOSFETs.
Buck Driver with Synchronous Rectication
In Figure 5, the input voltage can go from 7V to 28V and,
because of the ground-based current-sense resistor, the
output voltage can be as high as the input. The synchro-
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
nous MOSFET keeps the power dissipation to a minimum,
especially when the input voltage is large when compared
to the voltage on the LED string. It is important to keep
the current-sense resistor, R1, inside the LC loop, so that
ripple current is available. To regulate the LED current, R2
creates a voltage that the differential amplifier compares
to 0.6V. If power dissipation is a problem in R2, add a
noninverting amplifier and reduce the value of the sense
resistor accordingly.
Inductor Selection
The switching frequencies, peak inductor current, and
allowable ripple at the output determine the value
and size of the inductor. Selecting higher switching
frequencies reduces the inductance requirement, but
at the cost of lower efficiency. The charge/discharge
cycle of the gate and drain capacitances in the
switching MOSFETs create switching losses. The
situation worsens at higher input voltages, since switching
losses are proportional to the square of the input voltage.
The MAX16818 can operate up to 1.5MHz, however for
VIN > +12V, use lower switching frequencies to limit the
switching losses.
The following discussion is for buck or continuous boost-
mode topologies. Discontinuous boost, buck-boost,
and SEPIC topologies are quite different in regards to
component selection.
Use the following equations to determine the minimum
inductance value:
Buck regulators:
INMAX LED LED
MIN INMAX SW L
( V V ) x V
L V x f x I
−
=∆
Boost regulators:
LED INMAX INMAX
MIN LED SW L
( V V ) x V
L V x f x I
−
=∆
where VLED is the total voltage across the LED string.
As a first approximation choose the ripple current, ∆IL,
equal to approximately 40% of the output current.
Higher ripple current allows for smaller inductors, but it
also increases the output capacitance for a given
voltage ripple requirement. Conversely, lower ripple
current increases the inductance value, but allows the
output capacitor to reduce in size. This trade-
off can be altered once standard inductance and
capacitance values are chosen. Choose inductors from the
standard surface-mount inductor series available from various
manufacturers.
For example, for a buck regulator and 2 LEDs in series,
calculate the minimum inductance at VIN(MAX) = 13.2V,
VLED = 7.8V, ∆IL = 400mA, and fSW = 330kHz:
Buck regulators:
MIN (13.2 7.8) x 7.8
L 24.2 H
13.2 x 330k x 0.4
−
= = µ
For a boost regulator with four LEDs in series, calculate
the minimum inductance at VIN(MAX) = 13.2V, VLED =
15.6V, ∆IL =400mA, and fSW = 330kHz:
Boost regulators:
MIN (15.6 13.2) x 13.2
L 15.3 H
15.6 x 330k x 0.4
−
= = µ
The average-current-mode control feature of the
MAX16818 limits the maximum peak inductor current
and prevents the inductor from saturating. Choose an
inductor with a saturating current greater than the
worst-case peak inductor current. Use the following
equation to determine the worst-case inductor current:
CL L
ILPEAK S
VI
R2
∆
= +
where RS is the inductor sense resistor and VCL =
0.0282V.
Switching MOSFETs
When choosing a MOSFET for voltage regulators,
consider the total gate charge, RDS(ON), power
dissipation, and package thermal impedance. The product
of the MOSFET gate charge and on-resistance is a figure
of merit, with a lower number signifying better performance.
Choose MOSFETs optimized for high-frequency switching
applications.
The average current from the MAX16818 gate-drive output
is proportional to the total capacitance it drives at DH and
DL. The power dissipated in the MAX16818 is proportional
to the input voltage and the average drive current. See the
IN, VCC, and VDD section to determine the maximum total
gate charge allowed from the combined driver outputs.
The gate-charge and drain-capacitance (CV2) loss, the
cross-conduction loss in the upper MOSFET due to finite
rise/fall times, and the I2R loss due to RMS current in
the MOSFET RDS(ON) account for the total losses in the
MOSFET.
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Buck Regulator
Estimate the power loss (PDMOS_) caused by the high-side
and low-side MOSFETs using the following equations:
MOS HI G DD SW
PD (Q x V x f )
−= +
IN OUT R F SW
2
DS(ON) RMS HI
V x I x (t t ) x f
2
(R x I )
−
+
+
where QG, RDS(ON), tR, and tF are the upper-switching
MOSFET's total gate charge, on-resistance at maximum
operating temperature, rise time, and fall time, respectively.
22
RMS HI VALLEY PK VALLEY PK D
I (I I I x I ) x 3
−= ++
For the buck regulator, D = VLEDs/VIN, IVALLEY =
(IOUT - ∆IL/2) and IPK = (IOUT +∆IL2).
MOS LO G DD SW
PD (Q x V x f )
−= +
2
DS(ON) RMS LO
22
RMS LO VALLEY PK VALLEY PK
(R x I )
(1 D)
I (I I I x I ) x 3
−
−−
= ++
For example, from the typical specifications in the
Applications Information section with VOUT = 7.8V, the
high-side and low-side MOSFET RMS currents are
0.77A and 0.63A, respectively, for a 1A buck regulator.
Ensure that the thermal impedance of the MOSFET
package keeps the junction temperature at least +25°C
below the absolute maximum rating. Use the following
equation to calculate the maximum junction temperature:
TJ = (PDMOS x θ
JA) + TA, where θJA and TA are the
junction-to-ambient thermal impedance and ambient
temperature, respectively.
To guarantee that there is no shoot-through from VIN
to PGND, the MAX16818 produces a nonoverlap time
of 35ns. During this time, neither high- nor low-side
MOSFET is conducting, and since the output inductor
must maintain current flow, the intrinsic body diode of the
low-side MOSFET becomes the conduction path. Since
this diode has a fairly large forward voltage, a Schottky
diode (in parallel to the low-side MOSFET) diverts
current flow from the MOSFET body diode because
of its lower forward voltage, which, in turn, increases
efficiency.
Boost Regulator
Estimate the power loss (PDMOS_) caused by the
MOSFET using the following equations:
FET G DD SW
IN OUT R F SW 2
DS(ON) RMS HI
22
RMS HI VALLEY PK VALLEY PK
PD (Q x V x f )
V x I x (t t ) x f (R x I )
2
D
I (I I I x I ) x 3
−
−
= +
+
+
= ++
For a boost regulator in continuous mode, D = VLEDs/
(VIN + VLEDs), IVALLEY = (IOUT - ∆IL/2) and IPK = (IOUT
+ ∆IL/2).
The voltage across the MOSFET:
VMOSFET = VLED + VF
where VF is the maximum forward voltage of the diode.
The output diode on a boost regulator must be rated to
handle the LED series voltage, VLED. It should also have
fast reverse-recovery characteristics and should handle
the average forward current that is equal to the LED
current.
Input Capacitors
For buck regulator designs, the discontinuous input
current waveform of the buck converter causes large ripple
currents in the input capacitor. The switching frequency,
peak inductor current, and the allowable peak-to-peak
voltage ripple reflected back to the source dictate the
capacitance requirement. Increasing switching frequency
or paralleling out-of-phase converters lowers the peak-to-
average current ratio, yielding a lower input capacitance
requirement for the same LED current. The input ripple
is comprised of ∆VQ (caused by the capacitor discharge)
and ∆VESR (caused by the ESR of the capacitor). Use low-
ESR ceramic capacitors with high-ripple-current capability
at the input. Assume the contributions from the ESR and
capacitor discharge are equal to 30% and 70%,
respectively. Calculate the input capacitance and ESR
required for a specified ripple using the following equation:
ESR
IN L
OUT
V
ESR I
I 2
∆
=∆
+
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MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Buck:
OUT
IN Q SW
I x D(1 D)
C V x f
−
=∆
where IOUT is the output current of the converter. For
example, at VIN = 13.2V, VLED = 7.8V, IOUT = 1A, ∆IL =
0.4A, and fSW = 330kHz, the ESR and input capacitance
are calculated for the input peak-to-peak ripple of 100mV
or less yielding an ESR and capacitance value of 25mW
and 10µF.
For boost regulator designs, the input-capacitor current
waveform is dominated by the inductor, a triangle wave
a magnitude of ∆IL. For simplicity's sake, the current
waveform can be approximated by a square wave with a
magnitude that is half that of the triangle wave. Calculate
the input capacitance and ESR required for a specified
ripple using the following equation:
ESR
IN L
V
ESR I
∆
=∆
Boost:
L
IN Q SW
I x D
2
C V x f
∆
=∆
Duty cycle, D, for a boost regulator is equal to (VOUT
- VIN)/VOUT. As an example, at VIN = 13.2V, VLED =
15.6V, IOUT = 1A, ∆IL = 0.4A, and fSW = 330kHz, the
ESR and input capacitance are calculated for the input
peak-to-peak ripple of 100mV or less yielding an ESR
and capacitance value of 250mW and 1µF, respectively.
Output Capacitor
For buck converters, the inductor always connects to the
load, so the inductance controls the ripple current. The
output capacitance shunts a fraction of this ripple
current and the LED string absorbs the rest. The capacitor
reactance (which includes the capacitance and ESR) and
the dynamic impedance of the LED diode string form a
conductance divider that splits the ripple current between
the LEDs and the capacitor. In many cases, the capacitor
is very large as compared to the ESR, and this divider
reduces to the ESR and the LED resistance.
Boost converters place a harsher requirement on the
output capacitors as they must sustain the full load
during the on-time of the MOSFET and are replenished
during the off-time. The ripple current in this case is the
full load current, and the holdup time is equal to the
duty cycle times the switching period.
Current Limit
In addition to the average current limit, the MAX16818
also has hiccup current limit. The hiccup current limit is
set to 10% below the average current limit to ensure that
the circuit goes in hiccup mode during continuous output
short circuit. Connecting a resistor from LIM to ground
increases the hiccup current limit, while shorting LIM to
ground disables the hiccup current-limit circuit.
Average Current Limit
The average-current-mode control technique of the
MAX16818 accurately limits the maximum output current.
The MAX16818 senses the voltage across the sense
resistor and limit the peak inductor current (IL-PK)
accordingly. The on-cycle terminates when the current-
sense voltage reaches 25.5mV (min). Use the
following equation to calculate the maximum current-sense
resistor value:
−
=
=
SOUT
3
R
S
0.0255
R I
0.75 x 10
PD R
where PDR is the dissipation in the series resistors.
Select a 5% lower value of RS to compensate for
any parasitics associated with the PCB. Also, select a
noninductive resistor with the appropriate power rating.
Hiccup Current Limit
The hiccup current-limit value is always 10% lower than
the average current-limit threshold, when LIM is left
unconnected. Connect a resistor from LIM to SGND
to increase the hiccup current-limit value from 90% to
100% of the average current-limit value. The average
current-limit architecture accurately limits the
average output current to its current-limit threshold. If the
hiccup current limit is programmed to be equal or above
the average current-limit value, the output current
does not reach the point where the hiccup current limit
can trigger. Program the hiccup current limit at least
5% below the average current limit to ensure that the
hiccup current-limit circuit triggers during overload. See the
Hiccup Current Limit vs. REXT graph in the Typical
Operating Characteristics.
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22
MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Compensation
The main control loop consists of an inner current loop
(inductor current) and an outer LED current loop. The
MAX16818 uses an average current-mode control scheme
to regulate the LED current (Figure 7). The VEA output
provides the controlling voltage for the current source.
The inner current loop absorbs the inductor pole reducing
the order of the LED current loop to that of a single-pole
system. The major consideration when designing the
current control loop is making certain that the inductor
downslope (which becomes an upslope at the output of
the CEA) does not exceed the internal ramp slope. This
is a necessary condition to avoid subharmonic oscillations
similar to those in peak current mode with insufficient slope
compensation. This requires that the resistance, RCF, at
the output of the CEA be limited, based on the following
equation (Figure 6)
Buck:
RAMP SW
CF V m S LED
V fL
RAgRV
××
≤×××
where VRAMP = 2V, gm = 550µS, and AV = 34.5.
SW
CF S LED
fL
R 105 RV
×
≤××
Boost:
( )
( )
RAMP SW
CF V m S LED IN
SW
CF S LED IN
V fL
RAgR V V
fL
R 105 RV V
××
≤××× −
×
≤××−
The crossover frequency of the inner current loop is
expressed:
Buck:
V m S IN CF
C_buck RAMP
AgRVR
fV 2L
××× ×
=× π×
When AV = 34.5, gm = 550µS, and VRAMP = 2V, this
becomes:
( )
S IN CF
C_buck
9.488mS V R V R
f2L
×× ×
=π×
Boost:
V m S LED CF
C_boost RAMP
AgRV R
fV 2L
××× ×
=× π×
( )
S LED CF
C_boost
9.488mS V R V R
f2L
×× ×
=π×
For adequate phase margin, place the zero formed by
RCF and CCZ not more than 1/3 to 1/5 of the crossover
frequency. The pole formed by RCF and CCP may not be
required in most applications but can be added to minimize
noise at a frequency at or above the switching frequency.
Power Dissipation
The TQFN is a thermally enhanced package and can
dissipate about 2.7W. The high-power package makes the
high-frequency, high-current LED driver possible to operate
from a 12V or 24V bus. Calculate power dissipation in the
MAX16818 as a product of the input voltage and the total
VCC regulator output current (ICC). ICC includes quiescent
current (IQ) and gate drive current (IDD):
[ ]
D IN CC
CC Q SW G1 G2
P V x I
I I f x (Q Q )
=
=++
where QG1 and QG2 are the total gate charge of the low-
side and high-side external MOSFETs at VGATE = 5V, IQ
is estimated from the Supply Current (IQ) vs. Frequency
graph in the Typical Operating Characteristics, and fSW
is the switching frequency of the LED driver. For boost
drivers, only consider one gate charge, QG1.
Use the following equation to calculate the maximum
power dissipation (PDMAX) in the chip at a given ambient
temperature (TA):
PDMAX = 34.5 x (150 - TA) mW.
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23
MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
PCB Layout Guidelines
Use the following guidelines to layout the switching
voltage regulator:
1) Place the IN, VCC, and VDD bypass capacitors close
to the MAX16818.
2) Minimize the area and length of the high current loops
from the input capacitor, upper switching MOSFET,
inductor, and output capacitor back to the input
capacitor negative terminal.
3) Keep short the current loop formed by the lower
switching MOSFET, inductor, and output capacitor.
4) Place the Schottky diodes close to the lower MOSFETs
and on the same side of the PCB.
5) Keep the SGND and PGND isolated and connect
them at one single point.
6) Run the current-sense lines CSP and CSN very close
to each other to minimize the loop area. Similarly, run
the remote voltage-sense lines SENSE+ and SENSE-
close to each other. Do not cross these critical signal
lines through power circuitry. Sense the current right
at the pads of the current-sense resistors.
7) Avoid long traces between the VDD bypass
capacitors, the driver output of the MAX16818, the
MOSFET gates, and PGND. Minimize the loop formed
by the VCC bypass capacitors, bootstrap diode,
bootstrap capacitor, the MAX16818, and the upper
MOSFET gate.
8) Distribute the power components evenly across the
board for proper heat dissipation.
9) Provide enough copper area at and around the
switching MOSFETs, inductor, and sense resistors to
aid in thermal dissipation.
10) Use wide copper traces (2oz) to keep trace
inductance and resistance low to maximize efficiency.
Wide traces also cool heat-generating components.
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
28 TQFN-EP T2855+3 21-0140 90-0023
28
27
26
25
24
23
22
8
9
10
11
12
13
14
15
16
17
18
19
20
21
7
6
5
432
1
MAX16818
TQFN
+
TOP VIEW
N.C.
PGND
DL
BST
LX
DH
N.C.
VDD
VCC
IN
SGND
SENSE+
SENSE-
SGND
CSP
CSN
DIFF
EAN
EAOUT
CLP
* EXPOSED PAD
OVI
LIM
V_IOUT
RT/SYNC
EN
PGOOD
CLKOUT
SGND
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24
MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
Package Information
For the latest package outline information and land patterns
(footprints), go to www.maximintegrated.com/packages. Note
that a “+”, “#”, or “-” in the package code indicates RoHS status
only. Package drawings may show a different suffix character, but
the drawing pertains to the package regardless of RoHS status.
Chip Information
PROCESS: BiCMOS
Pin Conguration
REVISION
NUMBER
REVISION
DATE DESCRIPTION PAGES
CHANGED
0 10/06 Initial release —
1 6/08 Replaced Compensation section and corrected Figure 4 12, 23
2 3/09 Updated formula in Inductor Selection section 20
3 4/14 No /V OPNs; removed automotive references from Applications section 1
4 5/15 Updated Benefits and Features and Package Information sections 1, 24
Maxim Integrated cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim Integrated product. No circuit patent licenses
are implied. Maxim Integrated reserves the right to change the circuitry and specications without notice at any time. The parametric values (min and max limits)
shown in the Electrical Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
Maxim Integrated and the Maxim Integrated logo are trademarks of Maxim Integrated Products, Inc.
MAX16818 1.5MHz, 30A High-Efciency, LED Driver
with Rapid LED Current Pulsing
© 2015 Maxim Integrated Products, Inc.
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25
Revision History
For pricing, delivery, and ordering information, please contact Maxim Direct at 1-888-629-4642, or visit Maxim Integrated’s website at www.maximintegrated.com.
