TDA4601 - Infineon Technologies

PWM-QR IC

TDA4601

Control IC for Switched-Mode

Power Supplies

Never stop thinking.

Power Management & Supply

Datasheet, V2.0, 1 Jun 1994

Edition 1994-06-01

Published by Infineon Technologies AG,

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TDA4601

Revision History: 1994-06-01 Datasheet

Previous Version:

Page Subjects (major changes since last revision)

The integrated circuit TDA 4601/D is designed for driving, controlling and protecting the switching

transistor in self-oscillating flyback converter power supplies as well as for protecting the overall

power supply unit. In case of disturbance, the rise of the secondary voltage is prevented. In addition

to the ICs application range including TV-receivers video tape recorders, hifi devices and active

loudspeakers, it can also be used in power supply units for professional applications due to its wide

control range and high voltage stability during increased load changes.

Type Ordering Code Package

TDA 4601 Q67000-A2379 P-SIP-9-1

Control ICs for Switched-Mode Power Supplies

Bipolar IC

TDA 4601

P-SIP-9-1

Features

●Direct control of the switching transistor

●Low start-up current

●Reversing linear overload characteristic

●Base current drive proportional to collector current

●Protective circuit in case of disturbance

Version 2.0 3 1 Jun 1994

Version 2.0 4 1 Jun 1994

TDA 4601

Block Diagram

Pin Definitions and Functions

Pin No. Function

1VREF output

2 Zero passage identification

3 Input control amplifier, overload amplifier

4 Collector current simulation

5 Connection for additional protective circuit

6 Ground (rigidly connected to substrate mounting plate)

7 DC-output for charging coupling capacitor

8 Pulse output - driving of switching transistor

9 Supply voltage

Version 2.0 5 1 Jun 1994

TDA 4601

Circuit Description

The TDA 4601 is designed for driving, controlling and protecting the switching transistor in flyback

converter power supplies during start-up, normal and overload operations as well as during

disturbed operation. In case of disturbance the drive of the switching transistor is inhibited and a

secondary voltage rise is prevented.

Start-Up

The start-up procedures (on-mode) include three consecutive operating phases as follows:

1. Build-Up of Internal Reference Voltage

The internal reference voltage supplies the voltage regulator and effects charging of the coupling

electrolytic capacitor connected to the switching transistor. Current consumption will remain at

I9< 3.2 mA with a supply voltage up to V9 approx. 12 V.

2. Enabling of Internal Voltage - Reference Voltage V1 = 4 V

Simultaneously with V9 reaching approx. 12 V, an internal voltage becomes available, providing

all component elements, with the exception of the control logic, with a thermally stable and

overload-resistant current supply.

3. Enabling of Control Logic

In conjunction with the generation of the reference voltage, the current supply for the control logic

is activated by means of an additional stabilization circuit. The integrated circuit is then ready for

operation.

The start-up phase above described are necessary for ensuring the charging of the coupling

electrolytic capacitor, which in turn supplies the switching transistor. Only then is it possible to

ensure that the transistor switches accurately.

Normal Operating Mode / Control Operating Mode

At the input of pin 2 the zero passages of the frequency provided by the feedback coil are registered

and forwarded to the control logic. Pin 3 (control input, overload and standby identification) receives

the rectified amplitude fluctuations of the feedback coil. The control amplifier operates with an input

voltage of approx. 2 V and a current of approx. 1.4 mA. Depending on the internal voltage reference,

the overload identification limits inconjunction with collector current simulator pin 4 the operating

range of the control amplifier. The collector current is simulated by an external RC-combination

present at pin 4 and internally set threshold voltages. The largest possible collector current

applicable to the switching transistor (point of return) increases in proportion to the increased

capacitance (10 nF). Thus the required operating range of the control amplifier is established. The

range of control lies between a DC-voltage clamped at 2 V and a sawtooth - shaped rising AC-

voltage, which can vary up to a max. amplitude of 4 V (reference voltage). During secondary load

reduction to approx. 20 W, the switching frequency is increased (approx. 50 kHz) at an almost

constant pulse duty factor (1:3). During additional secondary load decreases to approx. 1 W, the

switching frequency increases to approx. 70 kHz and pulse duty factor to approx. 1:11. At the same

time collector peak current is reduced to < 1 A.

Version 2.0 6 1 Jun 1994

TDA 4601

The output levels of the control amplifier as well as those of the overload identification and collector

current simulator are compared in the trigger and forwarded to the control logic. Via pin 5 it is

possible to externally inhibit the operations of the IC. The output at pin

pin 8 will be inhibited when voltages of ≤ – 0.1 are present at pin 5.

Flipflops for controlling the base current amplifier and the base current shut-down are set in the

control logic depending on the start-up circuit, the zero passage identification as well as on the

enabling by the trigger. The base current amplifier forwards the sawtoothspahed V4 voltage to the

output of pin 8. A current feedback with an external resistor (R = 0.68 Ω) is present between pin 8

and pin 7. The applied value of the resistor determines the max. amplitude of the base driving

current for the switching transistor.

Protective Operating Mode

The base current shut-down activated by the control logic clamps the output of pin 7 to 1.6 V. As a

result, the drive of the switching transistor is inhibited. This protective measure is enabled if the

supply voltage at pin 9 reaches a value ≤6.7 V or if voltages of

≤ – 0.1 are present at pin 5.

In case of short-circuits occurring in the secondary windings of the switched-mode power supply,

the integrated circuit continuously monitors the fault conditions. During secondary, completely load-

free operation only a small pulse duty factor is set. As a result the total power consumption of the

power supply is held at N= 6 ... 10 W during both operating modes. After the output has been

inhibited for a voltage supply of ≤6.7 V, the reference voltage (4 V) is switched off if the voltage

supply is further reduced by ∆V9= 0.6 V.

Protective Operating Mode at Pin 5 in Case of Disturbance

The protection against disturbances such as primary undervoltages and/or secondary over-

voltages (e.g. by changes in the component parameters for the switched-mode power supply) is

realized as follows:

Protective Operating Mode with Continuous Fault Condition Monitoring

In case of disturbance the output pulses at pin 8 are inhibited by falling below the protective

threshold V5, with a typical value of V1/2. As a result current consumption is reduced (I9≥14 mA

at V9=10V).

With a corresponding high-impedance start-up resistor *), supply voltage V9 will fall below the

minimum shut-down threshold (5.7 V) for reference voltage V1.V1 will be switched off and current

consumption is further reduced to I9≤3.2 mA at V9≤10 V.

Because of these reductions in current consumption, the supply voltage can rise again to reach the

switch-on threshold of V9≥12.3 V. The protective threshold at pin 5 is released and the power

supply is again ready for operation.

VREF

2

------------

VREF

2

------------

Version 2.0 7 1 Jun 1994

TDA 4601

In case of continuing problems of disturbance (V5≤V1/2 – 0.1 V) the switch-on mode is interrupted

by the periodic protective operating mode described above, i.e. pin 8 is inhibited and V9 is falling,

etc.

Switch-On in the Wide Range Power Supply (90 Vac to 270 Vac)

(application circuit 2)

Self-oscillating flyback-converters designed as wide range power supplies require a power source

independent of the rectified line voltage for TDA 4601. Therefore the winding polarity of winding

11/13 corresponds to the secondary side of the flyback converter transformer. Start-up is not as

smooth as with an immediately available supply voltage, because TDA 4601 has to be supplied by

the start-up circuit until the entire secondary load has been charged. This leads to long switch-on

times, especially if low line voltages are applied.

However, the switch-on time can be shortened by applying the special start-up circuit (dotted line).

The uncontrolled phase of feedback control winding 15/9 is used for activating purposes.

Subsequent to activation, the transistor T1 begins to block when winding 11/13 generates the

current supply for TDA 4601. Therefore, the control circuit cannot be influenced during operation.

Version 2.0 8 1 Jun 1994

TDA 4601

Absolute Maximum Ratings

Parameter Symbol Limit Values Unit

min. max.

Supply voltage V9020V

Voltages

Reference output V106V

Zero passage identification V2– 0.6 0.6 V

Control amplifier V303V

Collector current simulation V408V

Blocking input V508V

Base current cut-off point V70V9V

Base current amplifier output V80V9V

Currents

Zero passage identification Il i2 –5 5 mA

Control amplifier Il 3 – 3 3 mA

Collector current simulation Il 4 05mA

Blocking input Il 5 05mA

Base current cut-off point IQ 7 – 1 1.5 A

Base current amplifier output IQ 8 – 1.5 0 A

Junction temperature Tj125 ˚C

Storage temperature range Tstg – 40 125 ˚C

Thermal resistances:

system-air TDA 4601

system-case TDA 4601 Rth SA

Rth SC

70

15 K/W

K/W

system-air 1) TDA 4601-D

system-case 2) TDA 4601-D Rth SA

Rth SA1

60

44 K/W

K/W

Version 2.0 9 1 Jun 1994

TDA 4601

1) Case soldered on PC-board without cooling surface

2) Case soldered on PC-board with copper-clad 35 µm layer, cooling surface 25 cm2

3) Rth SA1 = 44 K/W and PV = 1 W

Operating Range

Supply voltage V97.8 18 V

Case temperature TDA 4601 TC085˚C

Ambient temperature range 3) TDA 4601-D TA070˚C

Absolute Maximum Ratings (cont’d)

Parameter Symbol Limit Values Unit

min. max.

Version 2.0 10 1 Jun 1994

TDA 4601

*) DC-component only

Characteristics

TA = 25 ˚C

according to measurement circuit 1 and diagram

Parameter Symbol Limit Values Unit

min. typ. max.

Start Operation

Current consumption (V1 not yet

switched on)

V9 = 2 V

V9 = 5 V

V9 = 10 V

I9

1.5

2.4

0.5

2.0

3.2

mA

Switching point for V1V911.0 11.8 12.3 V

Normal Operation

V9 = 10 V; Vcont = – 10 V; Vclock = ± 0.5 V; f = 20 kHz

duty cycle 1:2 after switch-on

Current consumtion

Vcont = – 10 V

Vcont = 0 V I9

I9

110

50 135

75 160

100 mA

mA

Reference voltage

I1 < 0.1 mA

I1 < 5 mA V1

V1

4.0

4.0 4.2

4.2 4.5

4.5 V

V

Temperature coeffiecient of

reference voltage TC1–10

– 3 1/K

Control voltage Vcont = 0 V V32.3 2.6 2.9 V

Collector current simulation voltage

Vcont = 0 V

Vcont = 0 V/– 10 V V4*)

∆V4*)1.8

0.3 2.2

0.4 2.5

0.5 V

V

Clamping voltage V56.0 7.0 8.0 V

Output voltages

Vcont = 0 V

Vcont = 0 V/– 10 V

VQ7 *)

VQ8 *)

∆VQ8

2.7

1.6

3.3

3.4

2.0

4.0

2.4

V

Feedback voltage V2*)0.2 V

Version 2.0 11 1 Jun 1994

TDA 4601

The cooling conditions have to be optimized with regard to maximum ratings

(TA;Tj;Rth JC ;Rth SA).

Protective Operation

V9 = 10 V; Vcont = – 10 V; Vclock =  0.5 V; f = 20 kHz;

duty cycle 1:2

Parameter Symbol Limit Values Unit

min. typ. max.

Current consumption

V5 < 1.9 V I914 22 28 mA

Switch-off voltage

V5 < 1.9 V VQ 7 1.3 1.5 1.8 V

Switch-off voltage

V5 < 1.9 V V41.8 2.1 2.5 V

Blocking input

Blocking voltage

Vcont = 0 V V5V

Supply voltage blocked for V8

Vcont = 0 V V46.7 7.4 7.8 V

V1 off (with further reduction of V9)∆V90.3 0.6 1.0 V

Characteristics

TA = 25 ˚C; according to measurement circuit 2

Parameter Symbol Limit Values Unit

min. typ. max.

Switching time (secondary voltage) tON 350 450 ms

Voltage variation S3 = closed

∆N3 = 20 W ∆V2 sec 100 500 mV

Voltage variation S2 = closed

∆N3 = 15 W ∆V2 sec 500 1000 mV

Standby operation S1 = open

secondary useful load = 3 W ∆V2 sec

f70 20

75 30

–V

kHz

V1

2

------ 0.1–V1

2

------

Version 2.0 12 1 Jun 1994

TDA 4601

Circuit Diagram

Version 2.0 13 1 Jun 1994

TDA 4601

Test and Measurement Circuit 1

Test Diagram: Overload Operation

Version 2.0 14 1 Jun 1994

TDA 4601

Application Circuit 2

Wide range from 80 to 270 Vac

Version 2.0 15 1 Jun 1994

TDA 4601

Notes on Application Circuit 2

Wide Range SMPS

Filtering of the rectified AC-voltage has been increased up to 470µF to ensure a constant and hum-

free supply at Vline = 80 Vac. The stabilized phase is tapped for supplying the IC. In order to ensure

good start-up conditions for the SMPS in the low voltage range, the non-stabilized phase of winding

13/15 is used as a starting aid (BD 139), which is turned off after start-up by means of Z-diode C12.

In comparison to the 220 Vac standard circuit, however, the collector-emitter circuit had to be

altered to improve the switching behavior of BU 208 for the entire voltage range (80 to 270 Vac).

Diode BY 231 is necessary to prevent inverse operation of BU 208 and may be integrated for

switching times with a secondary power < 75 W (BU 208 D).

Compared to the IC TDA 4600-2, the TDA 4601 has been improved in turn-off during under-voltage

at pin 5. The TDA 4601 is additionally provided with a differential amplifier input at pin 5, enabling

precise turn-off at the output of pin 8 accompanied by hysteresis. For wide range SMPS, TDA 4601

is recommendable instead of TDA 4600-2. If a constant quality standard equal to that of the

standard circuit is to be maintained, wide range SMPS (80 to 270 Vac) with secondary power of

120 W can only be implemented at the expense of time.

Version 2.0 16 1 Jun 1994

TDA 4601

Supplements to Application Circuit 2

Efficiency versus Output Power

Version 2.0 17 1 Jun 1994

TDA 4601

Supplements to Application Circuit 2

Load Characteristics V2sec = f (I2sec)

Output Voltage V2sec (line change)

Version 2.0 18 1 Jun 1994

TDA 4601

Further Applications

Application Circuit 3

Version 2.0 19 1 Jun 1994

TDA 4601

Notes on Application Circuit 3

Fully Insulated, Clamp-contacted PTC-Thermistor Suitable for SMPS-Applications at

Increased Start-Up Currents

The newly developed PTC-thermistor Q63100-P2462-J29 is designed for applications in SMPS as

well as in various other electronic circuits, which, for example, receive the supply voltage directly

from the rectified line voltage and require an increased current during turn-on. Used in the flyback

converter power supply of TV-sets, an application proved millions of times over, the new PTC-

thermistor in the auxiliary circuit branch has resulted in a power saving of no less than 2 W. This

increase in efficiency has a highly favorable effect on the standby operation of TV-sets.

The required turn-on current needs only 6 to 8 s until the operating temperature of the PTC-

thermistor is reached. Low thermal capacitance of the PTC-thermistor allows the circuit to be

operated again after no more than 2 s. Another positive feature is the improved short-circuit

strength. The clamp contacts permit more or less unlimited switching operations and thus

guarantee high reliability. A flame-retardant plastic package and small dimensions are additional

advantages of this newly developed PTC-thermistor.

Technical Data

Parameter Symbol Limit Values Unit

Breakdown voltage at TA = 60 ˚C VBD rms 350 V

Resistance at TA = 25 ˚C R25 5kΩ

Resistance tolerance ∆R25 25 %

Trip current (typ.) IK20 mA

Residual current at VA max IR2mA

Max. application voltage Vop max rms 265 V

Reference temperature (typ.) TREF 190 ˚C

Temperature coefficient (typ.) TC26 %/K

Max. operating current Imax 0.1 A

Storage temperature range Tstg – 25 to 125 ˚C

Version 2.0 20 1 Jun 1994

TDA 4601

Application Circuit 4

Version 2.0 21 1 Jun 1994

TDA 4601

Notes on Application Circuit 4

Improved Load Control and Short-Circuit Characteristics

Turn-on is the same as for circuit 3.

To make the price more attractive, switching transistor BU 508 A was selected.

To ensure optimum standby conditions, the capacitance between pins 2 and 3 was increased to

100 pF.

Z- diode C6.2 transfers control voltage ∆Vcont directly to pin 3 resulting in improved load control.

Design and coupling conditions of various flyback transformers were sometimes a reason for

overshoot spectra, which, despite the RC-attenuating element 33 Ωx 22 nF and the 10-kΩ resistor,

even penetrated across the feedback winding 9/15 to the zero passage indicator input (pin 2) and

activated double and multiple pulses in the IC. Double and multiple pulses, however, lead to

magnetic saturation in the flyback transformer and thus increase the risk of damaging the switched-

mode power supply.

The larger the quantities of power to be passed, the more easily overshoots are generated. This

can be observed around the point of return. The switched-mode power supply, however, reduces

its own power to a minimum in all cases of overload or short-circuit. A series resonant circuit, whose

resonance corresponds to the transformer’s selfoscillation, was created by combination of the

4.7-µH inductance and the 22-nF capacitance. This resonant circuit short-circuits overshoots via a

33-Ω resistor.

f1

2πLC

----------------- 500 kHz≈=





Version 2.0 22 1 Jun 1994

TDA 4601

Application Circuit 5

Version 2.0 23 1 Jun 1994

TDA 4601

Notes on Application Circuit 5

Highly Stable Secondary Side

Power supplies for commercial purposes require highly constant low voltages and high currents

which, on the basis of the flyback converter principle, can be realized only under certain conditions,

but, on the other hand, are implemented for economical reasons. An electrically isolated flyback

converter with a highly stable secondary side must receive the control information from this

secondary side. There are only two possibilities for meeting this requirement: either through a

transformer which is magnetically isolated from the flyback converter or by means of an

optocoupler. The development of CNY 17 has enabled the manufacture of a component suitable for

electrical isolation and characterized by high reliability and long-term stability.

The IC TDA 4601-D is the sucessor of the TDA 4600-D. It is compatible with its predecessor in all

operational functions and in the control of a self-oscillating flyback converter. Pin 3 is the input for

the control information, where the latter is compared with the reference voltage prevailing at pin 1

and the control from the optocoupler and subsequently transformed into a frequency/pulse width

control.

The previous feedback and control information winding is not necessary. The feedback information

(zero passage) is obtained from winding 3/4 - supply winding. The time constant chain 330 Ω/3.3 nF

and 330 Ω/2.2 nF was implemented in series with 150 µH to prevent interference at pin 2. The LC-

element forms a series resonant circuit for overshoots of the flyback converter and short-circuits

them.

Version 2.0 24 1 Jun 1994

TDA 4601

Application Circuit 6

Version 2.0 25 1 Jun 1994

TDA 4601

Notes on Application Circuit 6

Wide Range Plug SMPS up to 30 W

Due to their volume and weight, plug SMPS have so far been limited to a restricted primary voltage

and a secondary power of no more than 6 W.

The line-isolated wide range flyback converter presented here has a variable frequency and is

capable of producing a secondary power of 30 W. It is characterized by a compact design with an

approx. weight of 400 g. The entire line voltage range of 90 to 260 Vac is stabilized to ±1.5 % on

the secondary side. Load fluctuations between 0.1 and 2 A are regulated to within 5 %. The output

(secondary side) is overload, short-circuit, and openloop proof.

Version 2.0 26 1 Jun 1994

TDA 4601

Application Circuit 7

Version 2.0 27 1 Jun 1994

TDA 4601

Notes on Application Circuit 7

Wide Range SMPS with Reducing Peak Collector Current ICBU208

for Rising Line Voltage

(variable point of return)

Wide range SMPS have to be dimensioned at line voltages of 90 to 260 Vac. The difference

between the maximum collector current IC BU 208 max and the largest possible limit current

IC BU 208 limit which causes magnetic saturation of the flyback transformer and flows through the

primary inductance winding 5/7 is to be determined at Vacmin (IC BU 208 limit ≥1.2 x IC BU 208 max).

Then, the transmissible power of the flyback transformer and its value at Vacmax is to be determined.

In the standard circuit the collector current IC BU 208 max is almost constant at the point of return

independent of the line voltage. The transmissible power on the secondary side, however,

increases at the point of return in proportion to the rising rectified line voltage applied (figures 1 and

2).

In the wide range SMPS a line voltage ratio of 270/90 = 3/1 is obtained, causing doubling of the

transmissible power on the secondary side, i.e. in the wide range SMPS a far too large flyback

transformer had to be implemented.

The point of return protecting the SMPS against overloads or short circuits, is derived from the time

constant at pin 4 t4

= 270 kΩ x 4.7 nF. Thus, the largest possible pulse width is determined.

With the introduction of the 33-kΩ resistor this time constant is reduced as a function of the control

voltage applied to winding 13/15, rectified by diode BY 360 and filtered by the 1-µF capacitance,

which means that the pulse time becomes shorter. By means of the Z-diode

C

18 the line voltage

level can be defined at which the influence of the time constant correction becomes noticeable. The

change in the rectified voltage of winding 13/15 is proportional to the change in the rectified line

voltage.

At the point of return ICBU208

the peak collector current has been reduced with the aid of the given

values from 5.2 A at 90 Vac to 3.3 A at 270 Vac. The transmissible power at the point of return

remains stable between 125 and 270 Vac due to the set activation point of the point of return

correction (unbroken curve in figure 2).

Version 2.0 28 1 Jun 1994

TDA 4601

Load Characteristics

Figure 1

Figure 2

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