LM3478QMM/NOPB - NATIONAL SEMICONDUCTOR

LM3478

-Q1

COUT

100µF, 10 V

VOUT= 5 V, 2 A

CIN

100 µF, 6.3 V

0.025 Ÿ

IRF7807 60 k

RF1

D MBRD340

10 µH

VIN = 3.3 V (±10%)

RSN

CSN

0.01 µF

RFA

40 k

CC22 nF

20 k

4.7 k RF2

ISEN

COMP

AGND PGND

FA/SD

VIN

Product

Folder

Order

Now

Technical

Documents

Tools &

Software

Support &

Community

An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

LM3478Q-Q1

SNVSAX8 –APRIL 2018

LM3478Q-Q1 High-Efficiency Low-Side N-Channel Controller for Switching Regulator

1 Features

1• LM3478Q-Q1 is AEC-Q100 Qualified and

Manufactured on an Automotive Grade Flow

• 8-lead VSSOP-8

• Internal Push-Pull Driver With 1-A Peak Current

Capability

• Current Limit and Thermal Shutdown

• Frequency Compensation Optimized With a

Capacitor and a Resistor

• Internal Soft Start

• Current Mode Operation

• Undervoltage Lockout With Hysteresis

• Create a Custom Design Using the LM3478 with

the WEBENCH Power Designer

2 Applications

• Distributed Power Systems

• Battery Chargers

• Offline Power Supplies

• Telecom Power Supplies

• Automotive Power Systems

• Wide Supply Voltage Range of 2.97 V to 40 V

• 100-kHz to 1-MHz Adjustable Clock Frequency

• ±2.5% (Over Temperature) Internal Reference

• 10-µA Shutdown Current (Over Temperature)

3 Description

The LM3478Q-Q1 is a versatile Low-Side N-Channel

MOSFET controller for switching regulators. It is

suitable for use in topologies requiring a low side

MOSFET, such as boost, flyback, SEPIC, etc.

Moreover, the LM3478Q-Q1 can be operated at

extremely high switching frequency in order to reduce

the overall solution size. The switching frequency of

the LM3478Q-Q1 can be adjusted to any value

between 100 kHz and 1 MHz by using a single

external resistor. Current mode control provides

superior bandwidth and transient response, besides

cycle-by-cycle current limiting. Output current can be

programmed with a single external resistor.

The LM3478Q-Q1 has built in features such as

thermal shutdown, short-circuit protection, over

voltage protection, etc. Power saving shutdown mode

reduces the total supply current to 5 µA and allows

power supply sequencing. Internal soft-start limits the

inrush current at start-up.

Device Information(1)

PART NUMBER PACKAGE BODY SIZE (NOM)

LM3478Q-Q1 VSSOP (8) 3.00 mm x 3.00 mm

(1) For all available packages, see the orderable addendum at

the end of the datasheet.

Typical High Efficiency Step-Up (Boost) Converter

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Table of Contents

1 Features.................................................................. 1

2 Applications ........................................................... 1

3 Description............................................................. 1

4 Revision History..................................................... 2

5 Pin Configuration and Functions......................... 3

6 Specifications......................................................... 4

6.1 Absolute Maximum Ratings ...................................... 4

6.2 ESD Ratings - LM3478Q-Q1 ................................... 4

6.3 Recommended Operating Conditions....................... 4

6.4 Thermal Information.................................................. 5

6.5 Electrical Characteristics........................................... 5

6.6 Typical Characteristics.............................................. 7

7 Detailed Description............................................ 11

7.1 Overview................................................................. 11

7.2 Functional Block Diagram....................................... 12

7.3 Feature Description................................................. 12

7.4 Device Functional Modes........................................ 15

8 Application and Implementation ........................ 16

8.1 Application Information............................................ 16

8.2 Typical Applications ................................................ 16

9 Power Supply Recommendations...................... 28

10 Layout................................................................... 28

10.1 Layout Guidelines ................................................. 28

10.2 Layout Example .................................................... 29

11 Device and Documentation Support................. 30

11.1 Custom Design with WEBENCH Tools................. 30

11.2 Receiving Notification of Documentation Updates 30

11.3 Documentation Support ....................................... 30

11.4 Trademarks........................................................... 30

11.5 Electrostatic Discharge Caution............................ 30

11.6 Glossary................................................................ 30

12 Mechanical, Packaging, and Orderable

Information........................................................... 30

4 Revision History

DATE REVISION NOTES

April 2018 * Initial release of SNVSAX8 literature number

for LM3478Q-Q1. See LM3478 device

literature number SNVS085 for the

commercial-grade device Revision History

LM3478

-Q1

4 5

ISEN

COMP

AGND PGND

FA/SD

VIN

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

5 Pin Configuration and Functions

DGK Package

VSSOP-8

Top View

Pin Functions

PIN I/O DESCRIPTION

NAME NO.

ISEN 1 I Current sense input pin. Voltage generated across an external sense resistor is fed into this pin.

COMP 2 I Compensation pin. A resistor, capacitor combination connected to this pin provides compensation for the

control loop.

FB 3 I Feedback pin. The output voltage should be adjusted using a resistor divider to provide 1.26 V at this pin.

AGND 4 G Analog ground pin.

PGND 5 G Power ground pin.

DR 6 O Drive pin. The gate of the external MOSFET should be connected to this pin.

FA/SD 7 I Frequency adjust and Shutdown pin. A resistor connected to this pin sets the oscillator frequency. A high

level on this pin for longer than 30 µs will turn the device off. The device will then draw less than 10µA from

the supply.

VIN 8 P Power Supply Input pin.

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

6 Specifications

6.1 Absolute Maximum Ratings

over operating free-air temperature (unless otherwise noted) (1)

MIN MAX UNIT

Input Voltage 45 V

FB Pin Voltage –0.4< V V FB < 7 V

FA/SD Pin Voltage –0.4 < VFA/SD VFA/SD< 7 V

Peak Driver Output Current

(<10µs) 1 A

Power Dissipation Internally Limited

Junction Temperature +150 °C

Lead Temperature Vapor Phase (60 s) 215 °C

Infrared (15 s) 260 °C

DR Pin Voltage –0.4 ≤VDR VDR ≤8 V

ISEN Pin Voltage 500 mV

Tstg Storage temperature −65 150 °C

(1) AEC Q100-002 indicates HBM stressing is done in accordance with the ANSI/ESDA/JEDEC JS-001 specification.

6.2 ESD Ratings - LM3478Q-Q1 VALUE UNIT

V(ESD) Electrostatic discharge Human body model (HBM), per AEC Q100-002(1) ±2000 V

Charged device model (CDM), per

AEC Q100-011 Other pins ±750

Corner pins (1, 4, 5, and 8) ±750

6.3 Recommended Operating Conditions

over operating free-air temperature range (unless otherwise noted) MIN NOM MAX UNIT

Supply Voltage 2.97 ≤VIN VIN ≤40 V

Junction Temperature Range −40 ≤TJTJ≤+125 °C

Switching Frequency 100 ≤FSW FSW ≤1 MHz

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

(1) For more information about traditional and new thermal metrics, see the IC Package Thermal Metrics application report, SPRA953.

6.4 Thermal Information

THERMAL METRIC(1) LM3478Q-Q1

UNITDGK

8 PINS

RθJA Junction-to-ambient thermal resistance 157.2 °C/W

RθJC(top) Junction-to-case (top) thermal resistance 49.9 °C/W

RθJB Junction-to-board thermal resistance 77.1 °C/W

ψJT Junction-to-top characterization parameter 4.7 °C/W

ψJB Junction-to-board characterization parameter 75.8 °C/W

RθJC(bot) Junction-to-case (bottom) thermal resistance N/A °C/W

(1) The voltage on the drive pin, VDR is equal to the input voltage when input voltage is less than 7.2 V. VDR is equal to 7.2 V when the

input voltage is greater than or equal to 7.2 V.

(2) The limits for the maximum duty cycle can not be specified since the part does not permit less than 100% maximum duty cycle

operation.

(3) For this test, the FA/SD pin is pulled to ground using a 40-K resistor.

(4) For this test, the FA/SD pin is pulled to 5 V using a 40-K resistor.

6.5 Electrical Characteristics

Unless otherwise specified, VIN = 12V, RFA = 40kΩ, TJ= 25°C

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

VFB Feedback Voltage VCOMP = 1.4V, 2.97 ≤VIN ≤40V 1.2416 1.26 1.2843 V

VCOMP = 1.4V, 2.97 ≤VIN ≤40V, −40°C

≤TJ≤125°C 1.228 1.292

ΔVLINE Feedback Voltage Line

Regulation 2.97 ≤VIN ≤40V 0.001 %/V

ΔVLOAD Output Voltage Load

Regulation IEAO Source/Sink ±0.5 %/A

VUVLO Input Undervoltage

Lock-out 2.85 V

−40°C ≤TJ≤125°C 2.97

VUV(HYS) Input Undervoltage

Lock-out Hysteresis 170 mV

−40°C ≤TJ≤125°C 130 210

Fnom Nominal Switching

Frequency RFA = 40KΩ400 kHz

RFA = 40KΩ,−40°C ≤TJ≤125°C 350 440

RDS1 (ON) Driver Switch On

Resistance (top) IDR = 0.2A, VIN= 5V 16

RDS2 (ON) Driver Switch On

Resistance (bottom) IDR = 0.2A 4.5

VDR (max) Maximum Drive Voltage

Swing(1) VIN < 7.2V VIN V

VIN ≥7.2V 7.2

Dmax Maximum Duty Cycle(2) 100%

Tmin (on) Minimum On Time 325 ns

−40°C ≤TJ≤125°C 210 600

ISUPPLY Supply Current (non-

switching) See (3) 2.7 mA

See (3),−40°C ≤TJ≤125°C 3.3

IQQuiescent Current in

Shutdown Mode

VFA/SD = 5V (4), VIN = 5V 5 µA

VFA/SD = 5V (4), VIN = 5V, −40°C ≤TJ≤

125°C 10

VSENSE Current Sense

Threshold Voltage VIN = 5V 135 156 180 mV

VIN = 5V, −40°C ≤TJ≤125°C 125 190

VSC Short-Circuit Current

Limit Sense Voltage VIN = 5V 343 mV

VIN = 5V, −40°C ≤TJ≤125°C 250 415

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Electrical Characteristics (continued)

Unless otherwise specified, VIN = 12V, RFA = 40kΩ, TJ= 25°C

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

(5) The over-voltage protection is specified with respect to the feedback voltage. This is because the over-voltage protection tracks the

feedback voltage. The overvoltage protection threshold is given by adding the feedback voltage, VFB to the over-voltage protection

specification.

(6) The FA/SD pin should be pulled to VIN through a resistor to turn the regulator off. The voltage on the FA/SD pin must be above the

maximum limit for Output = High to keep the regulator off and must be below the limit for Output = Low to keep the regulator on.

VSL Internal Compensation

Ramp Voltage VIN = 5V 92 mV

VIN = 5V, −40°C ≤TJ≤125°C 52 132

VSL ratio VSL/VSENSE 0.30 0.49 0.70

VOVP Output Over-voltage

Protection (with respect

to feedback voltage) (5)

VCOMP = 1.4V 32 50

VCOMP = 1.4V, −40°C ≤TJ≤125°C 25

VSSOP Package 78

VSSOP Package, −40°C ≤TJ≤125°C 85

VOVP(HYS) Output Over-Voltage

Protection Hysteresis(5) VCOMP = 1.4V 60 mV

VCOMP = 1.4V, −40°C ≤TJ≤125°C 20 110

Gm Error Amplifier

Transconductance

VCOMP = 1.4V, IEAO = 100µA

(Source/Sink) 600 800 1000 µS

VCOMP = 1.4V, IEAO = 100µA

(Source/Sink), −40°C ≤TJ≤125°C 365 1265

AVOL Error Amplifier Voltage

Gain

VCOMP = 1.4V, IEAO = 100µA

(Source/Sink) 38 V/V

VCOMP = 1.4V, IEAO = 100µA

(Source/Sink), −40°C ≤TJ≤125°C 26 44

IEAO Error Amplifier Output

Current (Source/ Sink)

Source, VCOMP = 1.4V, VFB = 0V 80 110 140 µA

Source, VCOMP = 1.4V, VFB = 0V, −40°C

≤TJ≤125°C 50 180

Sink, VCOMP = 1.4V, VFB = 1.4V −100 −140 −180 µA

Sink, VCOMP = 1.4V, VFB = 1.4V, −40°C

≤TJ≤125°C −85 −185

VEAO Error Amplifier Output

Voltage Swing Upper Limit, VFB = 0V, COMP Pin =

Floating 2.2 V

Upper Limit, VFB = 0V, COMP Pin =

Floating, −40°C ≤TJ≤125°C 1.8 2.4

Lower Limit, VFB = 1.4V 0.56 V

Lower Limit, VFB = 1.4V, −40°C ≤TJ≤

125°C 0.2 1.0

TSS Internal Soft-Start Delay VFB = 1.2V, VCOMP = Floating 4 ms

TrDrive Pin Rise Time Cgs = 3000pf, VDR = 0 to 3V 25 ns

TfDrive Pin Fall Time Cgs = 3000pf, VDR = 0 to 3V 25 ns

VSD

Shutdown threshold (6) Output = High 1.27 V

Output = High, −40°C ≤TJ≤125°C 1.4

Output = Low 0.65 V

Output = Low, −40°C ≤TJ≤125°C 0.3

ISD Shutdown Pin Current VSD = 5V −1µA

VSD = 0V +1

IFB Feedback Pin Current 15 nA

TSD Thermal Shutdown 165 °C

Tsh Thermal Shutdown

Hysteresis 10 °C

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

6.6 Typical Characteristics

Unless otherwise specified, VIN = 12V, TJ= 25°C.

Figure 1. IQvs Input Voltage (Shutdown) Figure 2. ISupply vs Input Voltage (Non-Switching)

Figure 3. ISupply vs VIN (Switching) Figure 4. Switching Frequency vs RFA

Figure 5. Frequency vs Temperature Figure 6. Drive Voltage vs Input Voltage

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Typical Characteristics (continued)

Unless otherwise specified, VIN = 12V, TJ= 25°C.

Figure 7. Current Sense Threshold vs Input Voltage Figure 8. COMP Pin Voltage vs Load Current

Figure 9. Efficiency vs Load Current (3.3-V Input and 12-V

Output) Figure 10. Efficiency vs Load Current (5-V Input and 12-V

Output)

Figure 11. Efficiency vs Load Current (9-V Input and 12-V

Output) Figure 12. Efficiency vs Load Current (3.3-V Input and 5-V

Output)

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

Typical Characteristics (continued)

Unless otherwise specified, VIN = 12V, TJ= 25°C.

Figure 13. Error Amplifier Gain Figure 14. Error Amplifier Phase

Figure 15. COMP Pin Source Current vs Temperature Figure 16. Short Circuit Sense Voltage vs Input Voltage

Figure 17. Compensation Ramp vs Compensation Resistor Figure 18. Shutdown Threshold Hysteresis vs Temperature

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Typical Characteristics (continued)

Unless otherwise specified, VIN = 12V, TJ= 25°C.

Figure 19. Duty Cycle vs Current Sense Voltage

PWM Comparator resets

the RS latch

PWM

Comparator

Blank-Out prevents false

reset

325 ns Blank-Out time

Oscillator Sets

the RS Latch

92 mV

typ

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

7 Detailed Description

7.1 Overview

TheLM3478Q-Q1 device uses a fixed frequency, Pulse Width Modulated (PWM) current mode control

architecture. The Functional Block Diagram shows the basic functionality. In a typical application circuit, the peak

current through the external MOSFET is sensed through an external sense resistor. The voltage across this

resistor is fed into the ISEN pin. This voltage is fed into the positive input of the PWM comparator. The output

voltage is also sensed through an external feedback resistor divider network and fed into the error amplifier

negative input (feedback pin, FB). The output of the error amplifier (COMP pin) is added to the slope

compensation ramp and fed into the negative input of the PWM comparator. At the start of any switching cycle,

the oscillator sets the RS latch using the switch logic block. This forces a high signal on the DR pin (gate of the

external MOSFET) and the external MOSFET turns on. When the voltage on the positive input of the PWM

comparator exceeds the negative input, the RS latch is reset and the external MOSFET turns off.

The voltage sensed across the sense resistor generally contains spurious noise spikes, as shown in Figure 20.

These spikes can force the PWM comparator to reset the RS latch prematurely. To prevent these spikes from

resetting the latch, a blank-out circuit inside the IC prevents the PWM comparator from resetting the latch for a

short duration after the latch is set. This duration is about 325 ns and is called the blanking interval and is

specified as minimum on-time in the Electrical Characteristics section. Under extremely light-load or no-load

conditions, the energy delivered to the output capacitor when the external MOSFET in on during the blanking

interval is more than what is delivered to the load. An over-voltage comparator inside theLM3478Q-Q1 prevents

the output voltage from rising under these conditions. The over-voltage comparator senses the feedback (FB pin)

voltage and resets the RS latch. The latch remains in reset state until the output decays to the nominal value.

Figure 20. Basic Operation of the PWM Comparator

VIN

FA/SD

COMP

PGND

Under Voltage

Lockout

7.2V

LDO

internal Vcc

QDRIVER

Fixed Frequency

Detect Oscillator

PWM

Isen

AGND

Softstart

logic

325mV Short Circuit

Comparator

Vfb+Vovp OVP

1.26V Reference

internal slope

compensation

THERMAL

LIMIT

(165°C)

slope compensation

ramp adjust current

source

Error

Amplifier

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

7.2 Functional Block Diagram

7.3 Feature Description

7.3.1 Overvoltage Protection

The LM3478Q-Q1 has over voltage protection (OVP) for the output voltage. OVP is sensed at the feedback pin

(pin 3). If at anytime the voltage at the feedback pin rises to VFB+ VOVP, OVP is triggered. See Electrical

Characteristics section for limits on VFB and VOVP.

OVP will cause the drive pin to go low, forcing the power MOSFET off. With the MOSFET off, the output voltage

will drop. TheLM3478Q-Q1 begins switching again when the feedback voltage reaches VFB + (VOVP - VOVP(HYS)).

See Electrical Characteristics for limits on VOVP(HYS).

OVP can be triggered if the unregulated input voltage crosses 7.2 V, the output voltage will react as shown in

Figure 21. The internal bias of the LM3478Q-Q1 comes from either the internal LDO as shown in the block

diagram or the voltage at the Vin pin is used directly. At Vin voltages lower than 7.2 V the internal IC bias is the

Vin voltage and at voltages above 7.2V the internal LDO of the LM3478Q-Q1 provides the bias. At the switch

over threshold at 7.2 V a sudden small change in bias voltage is seen by all the internal blocks of the LM3478Q-

Q1. The control voltage shifts because of the bias change, the PWM comparator tries to keep regulation. To the

PWM comparator, the scenario is identical to a step change in the load current, so the response at the output

voltage is the same as would be observed in a step load change. Hence, the output voltage overshoot here can

also trigger OVP. The LM3478Q-Q1 will regulate in hysteretic mode for several cycles, or may not recover and

simply stay in hysteretic mode until the load current drops or Vin is not crossing the 7.2 V threshold anymore.

Note that the output is still regulated in hysteretic mode.

VSEN Sn

PWM Comparator

Waveforms

Time

Voltage

'I0

'I1

'I2

VIN (V)

VFB (V)

7.2V

OVP

(1.31V)

1.26V

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

Feature Description (continued)

Depending on the requirements of the application, there is some influence one has over this effect. The threshold

of 7.2 V can be shifted to higher voltages by adding a resistor in series with VIN. In case VIN is right at the

threshold of 7.2 V, the threshold could cross over and over due to some slight ripple on VIN. To minimize the

effect on the output voltage one can filter the VIN pin with an RC filter.

Figure 21. The Feedback Voltage Experiences an Oscillation if the Input Voltage crosses the 7.2-V

Internal Bias Threshold

7.3.2 Slope Compensation Ramp

The LM3478Q-Q1 uses a current mode control scheme. The main advantages of current mode control are

inherent cycle-by-cycle current limit for the switch and simpler control loop characteristics. It is also easy to

parallel power stages using current mode control since current sharing is automatic. However, current mode

control has an inherent instability for duty cycles greater than 50%, as shown in Figure 22.

A small increase in the load current causes the switch current to increase by ΔI0. The effect of this load change

is ΔI1.

The two solid waveforms shown are the waveforms compared at the internal pulse width modulator, used to

generate the MOSFET drive signal. The top waveform with the slope Seis the internally generated control

waveform VC. The bottom waveform with slopes Snand Sfis the sensed inductor current waveform VSEN.

Figure 22. Sub-Harmonic Oscillation for D>0.5 and Compensation Ramp to Avoid Sub-Harmonic

Oscillation

Sub-harmonic Oscillation can be easily understood as a geometric problem. If the control signal does not have

slope, the slope representing the inductor current ramps up until the control signal is reached and then slopes

down again. If the duty cycle is above 50%, any perturbation will not converge but diverge from cycle to cycle

and causes sub-harmonic oscillation.

It is apparent that the difference in the inductor current from one cycle to the next is a function of Sn, Sfand Seas

shown in Equation 1.

LM3478-

ISEN

RSEN

RSL

'In =Sf - Se

Sn + Se'In-1

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Feature Description (continued)

(1)

Hence, if the quantity (Sf- Se)/(Sn+ Se) is greater than 1, the inductor current diverges and sub-harmonic

oscillation results. This counts for all current mode topologies. The LM3478Q-Q1 has some internal slope

compensation VSL which is enough for many applications above 50% duty cycle to avoid sub-harmonic

oscillation .

For boost applications, the slopes Se, Sfand Sncan be calculated with Equation 2,Equation 3, and Equation 4.

Se= VSL x fs(2)

Sf= Rsen x (VOUT - VIN)/L (3)

Sn= VIN x Rsen/L (4)

When Seincreases, then the factor that determines if sub-harmonic oscillation will occur decreases. When the

duty cycle is greater than 50%, and the inductance becomes less, the factor increases.

For more flexibility, slope compensation can be increased by adding one external resistor, RSL, in the ISEN's path.

Figure 23 shows the setup. The externally generated slope compensation is then added to the internal slope

compensation of the LM3478Q-Q1. When using external slope compensation, the formula for Sebecomes:

Se= (VSL + (K x RSL)) x fs(5)

A typical value for factor K is 40 µA.

The factor changes with switching frequency. Figure 24 is used to determine the factor K for individual

applications and Equation 6 gives the factor K.

K = ΔVSL / RSL (6)

It is a good design practice to only add as much slope compensation as needed to avoid sub-harmonic

oscillation. Additional slope compensation minimizes the influence of the sensed current in the control loop. With

very large slope compensation the control loop characteristics are similar to a voltage mode regulator which

compares the error voltage to a saw tooth waveform rather than the inductor current.

Figure 23. Adding External Slope Compensation Figure 24. External Slope Compensation

ΔVSL vs RSL

LM3478

-Q1

FA/SD

10 k

RFA

>1.3 V

MOSFET State

On-Normal Operation

OFF - Shutdown

LM3478

-Q1

FA/SD

RFA

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

Feature Description (continued)

7.3.3 Frequency Adjust/Shutdown

The switching frequency of the LM3478Q-Q1 can be adjusted between 100 kHz and 1 MHz using a single

external resistor. This resistor must be connected between FA/SD pin and ground, as shown in Figure 25. To

determine the value of the resistor required for a desired switching frequency, refer to Typical Characteristics or

use Equation 7:

RFA = 4.503 x 1011 x fS- 1.26 (7)

Figure 25. Frequency Adjust

The FA/SD pin also functions as a shutdown pin. If a high signal (>1.35 V) appears on the FA/SD pin, the

LM3478Q-Q1 stops switching and goes into a low current mode. The total supply current of the IC reduces to

less than 10 µA under these conditions. Figure 26 shows implementation of the shutdown function when

operating in frequency adjust mode. In this mode a high signal for more than 30 us shuts down the IC. However,

the voltage on the FA/SD pin should be always less than the absolute maximum of 7 V to avoid any damage to

the device.

Figure 26. Shutdown Operation in Frequency Adjust Mode

7.3.4 Short-Circuit Protection

When the voltage across the sense resistor measured on the ISEN pin exceeds 343 mV, short circuit current limit

protection gets activated. A comparator inside the LM3478Q-Q1 reduces the switching frequency by a factor of 5

and maintains this condition until the short is removed. In normal operation the sensed current will trigger the

power MOSFET to turn off. During the blanking interval the PWM comparator will not react to an over current so

that this additional 343 mV current limit threshold is implemented to protect the device in a short circuit or severe

overload condition.

7.4 Device Functional Modes

The device is set to run as soon as the input voltage crosses above the UVLO set point and at a frequency set

according to the FA/SD pin pull-down resistor. If the FA/SD pin is pulled high, the LM3478Q-Q1 enters shut-down

mode.

LM3478

-Q1

COUT

100µF, 10 V

VOUT= 5 V, 2 A

CIN

100 µF, 6.3 V

0.025 Ÿ

IRF7807 60 k

RF1

D MBRD340

10 µH

VIN = 3.3 V (±10%)

RSN

CSN

0.01 µF

RFA

40 k

CC22 nF

20 k

4.7 k RF2

ISEN

COMP

AGND PGND

FA/SD

VIN

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

8 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component

specification, and TI does not warrant its accuracy or completeness. TI’s customers are

responsible for determining suitability of components for their purposes. Customers should

validate and test their design implementation to confirm system functionality.

8.1 Application Information

The LM3478Q-Q1 may be operated in either the continuous conduction mode (CCM) or the discontinuous

current conduction mode (DCM). The following applications are designed for the CCM operation. This mode of

operation has higher efficiency and usually lower EMI characteristics than the DCM.

8.2 Typical Applications

8.2.1 Typical High Efficiency Step-Up (Boost) Converter

Figure 27. Typical High Efficiency Step-Up (Boost) Converter Schematic

The boost converter converts a low input voltage into a higher output voltage. The basic configuration for a boost

converter is shown in Figure 28. In the CCM (when the inductor current never reaches zero at steady state), the

boost regulator operates in two states. In the first state of operation, MOSFET Q is turned on and energy is

stored in the inductor. During this state, diode D is reverse biased and load current is supplied by the output

capacitor, COUT.

In the second state, MOSFET Q is off and the diode is forward biased. The energy stored in the inductor is

transferred to the load and the output capacitor. The ratio of the switch on time to the total period is the duty

cycle D as shown in Equation 8.

D = 1 - (Vin / Vout) (8)

Including the voltage drop across the MOSFET and the diode the definition for the duty cycle is shown in

Equation 9.

D = 1 - ((Vin - Vq)/(Vout + Vd)) (9)

Vdis the forward voltage drop of the diode and Vqis the voltage drop across the MOSFET when it is on.

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

Typical Applications (continued)

A. First Cycle Operation

B. Second Cycle of Operation

Figure 28. Simplified Boost Converter

8.2.1.1 Design Requirements

To properly size the components for the application, the designer needs the following parameters: input voltage

range, output voltage, output current range, and required switching frequency. These four main parameters affect

the choices of component available to achieve a proper system behavior.

For the power supply, the input impedance of the supply rail should be low enough that the input current

transient does not drop below the UVLO value. The factors determining the choice of inductor used should be

the average inductor current, and the inductor current ripple. If the switching frequency is set high, the converter

can be operated with very small inductor values. The maximum current that can be delivered to the load is set by

the sense resistor, RSEN. Current limit occurs when the voltage generated across the sense resistor equals the

current sense threshold voltage, VSENSE. Also, a resistor RSL adds additional slope compensation, if required.

The following sections describe the design requirements for a typical LM3478Q-Q1 boost application.

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 Custom Design with WEBENCH Tools

Click here to create a custom design using the LM3478Q-Q1 device with the WEBENCH®Power Designer.

1. Start by entering your VIN, VOUT and IOUT requirements.

2. Optimize your design for key parameters like efficiency, footprint and cost using the optimizer dial and

compare this design with other possible solutions from Texas Instruments.

3. WEBENCH Power Designer provides you with a customized schematic along with a list of materials with real

time pricing and component availability.

4. In most cases, you will also be able to:

– Run electrical simulations to see important waveforms and circuit performance,

– Run thermal simulations to understand the thermal performance of your board,

– Export your customized schematic and layout into popular CAD formats,

– Print PDF reports for the design, and share your design with colleagues.

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Typical Applications (continued)

5. Get more information about WEBENCH tools at www.ti.com/webench.

8.2.1.2.2 Power Inductor Selection

The inductor is one of the two energy storage elements in a boost converter. Figure 29 shows how the inductor

current varies during a switching cycle. The current through an inductor is quantified using Equation 10, which

shows the relationship of L, ILand VL.

(10)

The important quantities in determining a proper inductance value are IL(the average inductor current) and ΔIL

(the inductor current ripple). If ΔILis larger than IL, the inductor current will drop to zero for a portion of the cycle

and the converter will operate in the DCM. All the analysis in this datasheet assumes operation in the CCM. To

operate in the CCM, the following condition must be met by using Equation 11.

(11)

Choose the minimum IOUT to determine the minimum inductance value. A common choice is to set ΔILto 30% of

IL. Choosing an appropriate core size for the inductor involves calculating the average and peak currents

expected through the inductor. Use Equation 12,Equation 13, and Equation 14 to the peak inductor current in a

boost converter.

ILPEAK = Average IL(max) +ΔIL(max) (12)

Average IL(max) = Iout / (1-D) (13)

ΔIL(max) = D x Vin / (2 x fsx L) (14)

An inductor size with ratings higher than these values has to be selected. If the inductor is not properly rated,

saturation will occur and may cause the circuit to malfunction.

The LM3478Q-Q1 can be set to switch at very high frequencies. When the switching frequency is high, the

converter can be operated with very small inductor values. The LM3478Q-Q1 senses the peak current through

the switch which is the same as the peak inductor current as calculated in the previous equation.

t (s)

IL_AVG

ISW_AVG

D*Ts Ts

IL (A)

ID (A)

VIN LVV OUTIN 

LVV OUTIN -

(a)

(b)

ISW (A)

(C)

ID_AVG

=IOUT_AVG

t (s)

D*Ts Ts

VIN

'iL

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

Typical Applications (continued)

Figure 29. Inductor Current and Diode Current

ISWLIMIT = IOUT

(1-D) +(D x VIN)

(2 x fS x L)

RSEN = VSENSE - (D x VSL)

ISWLIMIT

0.000 0.100 0.200 0.300 0.400 0.500

CURRENT SENSE VOLTAGE (V)

100

120

DUTY CYCLE (%)

FS =

250 kHz

VSENSE

VSL

FS = 500 kHz

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Typical Applications (continued)

8.2.1.2.3 Programming the Output Voltage

The output voltage can be programmed using a resistor divider between the output and the FB pin. The resistors

are selected such that the voltage at the FB pin is 1.26 V. Pick RF1 (the resistor between the output voltage and

the feedback pin) and RF2 (the resistor between the feedback pin and ground) can be selected using the

following equation,

RF2 = (1.26 V x RF1) / (Vout - 1.26 V) (15)

A 100-pF capacitor may be connected between the feedback and ground pins to reduce noise.

8.2.1.2.4 Setting the Current Limit

The maximum amount of current that can be delivered to the load is set by the sense resistor, RSEN. Current limit

occurs when the voltage that is generated across the sense resistor equals the current sense threshold voltage,

VSENSE. When this threshold is reached, the switch will be turned off until the next cycle. Limits for VSENSE are

specified in the electrical characteristics section. VSENSE represents the maximum value of the internal control

signal VCS as shown in Figure 30. This control signal, however, is not a constant value and changes over the

course of a period as a result of the internal compensation ramp (VSL). Therefore the current limit threshold will

also change. The actual current limit threshold is a function of the sense voltage (VSENSE) and the internal

compensation ramp:

RSEN x ISWLIMIT = VCSMAX = VSENSE - (D x VSL) (16)

Where ISWLIMIT is the peak switch current limit, defined by Equation 17.

Figure 30. Current Sense Voltage vs Duty Cycle

Figure 30 shows how VCS (and current limit threshold voltage) change with duty cycle. The curve is equivalent to

the internal compensation ramp slope (Se) and is bounded at low duty cycle by VSENSE, shown as a dotted line.

As duty cycle increases, the control voltage is reduced as VSL ramps up. The graph also shows the short circuit

current limit threshold of 343 mV (typical) during the 325 ns (typical) blanking time. For higher frequencies this

fixed blanking time obviously occupies more duty cycle, percentage wise. Since current limit threshold varies with

duty cycle, the use Equation 17 to select RSEN and set the desired current limit threshold:

(17)

The numerator of Equation 17 is VCS, and ISWLIMIT using Equation 18.

(18)

RSEN = VSENSE

Vo - Vi

L x fS x D

ISWLIMIT +

ISWLIMIT = VSENSE - (D x(VSL + 'VSL))

RSEN

RSL > 40 PA

RSEN x (Vo - 2VIN)

2 x fS x L - VSL

RSEN < 2 x VSL x fS x L

Vo - (2 x VIN)

RSEN = VSENSE - (D x VSENSE x VSLratio)

ISWLIMIT

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

Typical Applications (continued)

To avoid false triggering, the current limit value should have some margin above the maximum operating value,

typically 120%. Values for both VSENSE and VSL are specified in Electrical Characteristics. However, calculating

with the limits of these two specs could result in an unrealistically wide current limit or RSEN range. Therefore,

Equation 19 is recommended, using the VSL ratio value given in Electrical Characteristics.

(19)

RSEN is part of the current mode control loop and has some influence on control loop stability. Therefore, once

the current limit threshold is set, loop stability must be verified. As described in the slope compensation section,

Equation 20 must hold true for a current mode converter to be stable.

Sf- Se< Sn+ Se(20)

To verify that this equation holds true, use Equation 21.

(21)

If the selected RSEN is greater than this value, additional slope compensation must be added to ensure stability,

as described in the section below.

8.2.1.2.5 Current Limit with External Slope Compensation

RSL is used to add additional slope compensation when required. It is not necessary in most designs and RSL

should be no larger than necessary. Select RSL according to Equation 22.

(22)

Where RSEN is the selected value based on current limit. With RSL installed, the control signal includes additional

external slope to stabilize the loop, which will also have an effect on the current limit threshold. Therefore, the

current limit threshold must be re-verified, as illustrated in Equation 23,Equation 24, and Equation 25 below.

VCS = VSENSE – (D x (VSL +ΔVSL)) (23)

Where ΔVSL is the additional slope compensation generated as discussed in the slope compensation ramp

section and calculated using Equation 24.

ΔVSL = 40 µA x RSL (24)

This changes the equation for current limit (or RSEN) as shown in Equation 25.

(25)

The RSEN and RSL values may have to be calculated iteratively in order to achieve both the desired current limit

and stable operation. In some designs RSL can also help to filter noise on the ISEN pin.

If the inductor is selected such that ripple current is the recommended 30% value, and the current limit threshold

is 120% of the maximum peak, a simpler method can be used to determine RSEN.Equation 26 below will provide

optimum stability without RSL, provided that the above 2 conditions are met.

(26)

8.2.1.2.6 Power Diode Selection

Observation of the boost converter circuit shows that the average current through the diode is the average load

current, and the peak current through the diode is the peak current through the inductor. The diode should be

rated to handle more than its peak current. The peak diode current can be calculated using Equation 27.

ID(Peak) = IOUT/ (1−D) + ΔIL(27)

RdrOn

tLH = Vdr - Vgsth

Qgs

Qgd + x

PSW = ILmax x Vout

2x fSW x (tLH + tHL)

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Typical Applications (continued)

Thermally the diode must be able to handle the maximum average current delivered to the output. The peak

reverse voltage for boost converters is equal to the regulated output voltage. The diode must be capable of

handling this voltage. To improve efficiency, a low forward drop schottky diode is recommended.

8.2.1.2.7 Power MOSFET Selection

The drive pin of the LM3478Q-Q1 must be connected to the gate of an external MOSFET. The drive pin (DR)

voltage depends on the input voltage (see Typical Characteristics). In most applications, a logic level MOSFET

can be used. For very low input voltages, a sub logic level MOSFET should be used. The selected MOSFET has

a great influence on the system efficiency. The critical parameters for selecting a MOSFET are:

1. Minimum threshold voltage, VTH(MIN)

2. On-resistance, RDS(ON)

3. Total gate charge, Qg

4. Reverse transfer capacitance, CRSS

5. Maximum drain to source voltage, VDS(MAX)

The off-state voltage of the MOSFET is approximately equal to the output voltage. Vds(max) must be greater

than the output voltage. The power losses in the MOSFET can be categorized into conduction losses and

switching losses. RDS(ON) is needed to estimate the conduction losses, Pcond:

Pcond = I2x RDS(ON) x D (28)

The temperature effect on the RDS(ON) usually is quite significant. Assume 30% increase at hot.

For the current I in Equation 28 the average inductor current may be used.

Especially at high switching frequencies the switching losses may be the largest portion of the total losses.

The switching losses are very difficult to calculate due to changing parasitics of a given MOSFET in operation.

Often the individual MOSFET's data sheet does not give enough information to yield a useful result. Equation 29

and Equation 30 give a rough idea how the switching losses are calculated:

(29)

(30)

8.2.1.2.8 Input Capacitor Selection

Due to the presence of an inductor at the input of a boost converter, the input current waveform is continuous

and triangular as shown in Figure 29. The inductor ensures that the input capacitor sees fairly low ripple currents.

However, as the input capacitor gets smaller, the input ripple goes up. The RMS current in the input capacitor is

given using Equation 31.

(31)

The input capacitor should be capable of handling the RMS current. Although the input capacitor is not as critical

in a boost application, low values can cause impedance interactions. Therefore a good quality capacitor should

be chosen in the range of 10 µF to 20 µF. If a value lower than 10 µF is used, then problems with impedance

interactions or switching noise can affect the LM3478Q-Q1. To improve performance, especially with Vin below 8

volts, it is recommended to use a 20 Ohm resistor at the input to provide an RC filter. The resistor is placed in

series with the VIN pin with only a bypass capacitor attached to the VIN pin directly (see Figure 31). A 0.1-µF or

1-µF ceramic capacitor is necessary in this configuration. The bulk input capacitor and inductor will connect on

the other side of the resistor at the input power supply.

VIN VIN

RIN

CIN

CBYPASS

LM3478

-Q1

LM3478Q-Q1

www.ti.com

SNVSAX8 –APRIL 2018

Typical Applications (continued)

Figure 31. Reducing IC Input Noise

8.2.1.2.9 Output Capacitor Selection

The output capacitor in a boost converter provides all the output current when the inductor is charging. As a

result it sees very large ripple currents. The output capacitor should be capable of handling the maximum RMS

current. Equation 32 shows the RMS current in the output capacitor.

(32)

Where

(33)

The ESR and ESL of the capacitor directly control the output ripple. Use capacitors with low ESR and ESL at the

output for high efficiency and low ripple voltage. Surface mount tantalums, surface mount polymer electrolytic,

polymer tantalum, or multi-layer ceramic capacitors are recommended at the output.

For applications that require very low output voltage ripple, a second stage LC filter often is a good solution. Most

of the time it is lower cost to use a small second Inductor in the power path and an additional final output

capacitor than to reduce the output voltage ripple by purely increasing the output capacitor without an additional

LC filter.

8.2.1.2.10 Compensation

For detailed explanation on how to select the right compensation components to attach to the compensation pin

for a boost topology, please see AN-1286 Compensation For The LM3748 Boost Controller SNVA067.

8.2.1.3 Application Curves

Figure 32. Efficiency vs Load Current (9-V In and 12-V Out) Figure 33. Efficiency vs Load Current (3.3-V In and 5-V

Out)

LM3478

-Q1

RF2

60 k

VOUT = 5 V, 1 A

CIN

15 µF, 35 V

0.05 Ÿ

IRF7807

10 µH

VIN = 3.3 V to 24 V

RSN

RFA

40 k

20 k

4.7 k RF1

ISEN

COMP

AGND PGND

FA/SD

VIN

CSN

1 nF D

MBRS130LT3

10 µH

1 µF, ceramic

LM3478Q-Q1

SNVSAX8 –APRIL 2018

www.ti.com

Typical Applications (continued)

8.2.2 Typical SEPIC Converter

Figure 34. Typical SEPIC Converter

Since the LM3478Q-Q1 controls a low-side N-Channel MOSFET, it can also be used in SEPIC (Single Ended

Primary Inductance Converter) applications. An example of a SEPIC using the LM3478Q-Q1 is shown in

Figure 34. Note that the output voltage can be higher or lower than the input voltage. The SEPIC uses two

inductors to step-up or step-down the input voltage. The inductors L1 and L2 can be two discrete inductors or

two windings of a coupled inductor since equal voltages are applied across the inductor throughout the switching

cycle. Using two discrete inductors allows use of catalog magnetics, as opposed to a custom inductor. The input

ripple can be reduced along with size by using the coupled windings for L1 and L2.

Due to the presence of the inductor L1 at the input, the SEPIC inherits all the benefits of a boost converter. One

main advantage of a SEPIC over a boost converter is the inherent input to output isolation. The capacitor CS

isolates the input from the output and provides protection against a shorted or malfunctioning load. Hence, the

SEPIC is useful for replacing boost circuits when true shutdown is required. This means that the output voltage

falls to 0V when the switch is turned off. In a boost converter, the output can only fall to the input voltage minus a

diode drop.

The duty cycle of a SEPIC is given using Equation 34.

(34)

In Equation 34, VQis the on-state voltage of the MOSFET, Q, and VDIODE is the forward voltage drop of the

diode.

8.2.2.1 Design Requirements