November 1997 - Linear Technology

LINEAR TECHNOLOG

NOVEMBER 1997 VOLUME VII NUMBER 4

, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,

FilterCAD, LinearView, Micropower SwitcherCAD and SwitcherCAD are trademarks of Linear Technology Corporation.

MICROWIRE is a trademark of National Semiconductor Corp. Other product names may be trademarks of the companies

that manufacture the products.

New 16-Bit SO-8 DAC

Has 1LSB Max INL and DNL

Over Industrial Temperature

by Jim Brubaker

and William C. Rempfer

New generations of industrial

systems are moving to 16 bits and

hence require high performance 16-bit

data converters. The new LTC1595/

LTC1596 16-bit DACs from LTC

provide the easiest to use, most cost

effective, highest performance solution

for industrial and instrumentation

applications. The LTC1595/LTC1596

are serial input, 16-bit, multiplying

current output DACs. Features of the

new DACs include:

❏±1LSB maximum INL and DNL

over the industrial temperature

range

❏Ultralow, 1nV-s glitch impulse

❏±10V output capability

❏Small SO-8 package (LTC1595)

❏Pin-compatible upgrade for

industry-standard 12-bit DACs

(DAC8043/8143 and AD7543)

Nice Features

of the 16-Bit DACs

The LTC1595/LTC1596 use precision

thin-ﬁlm resistors in a modiﬁed R/2R

architecture to provide a CMOS cur-

rent output DAC, as shown in Figure

1. The two DACs have SPI/MICRO-

WIRE™ compatible serial interfaces

and draw only 10µA from a single 5V

supply. They generate precision 0V–

10V or ±10V outputs using a single or

dual external op amp. The LTC1596

has an asynchronous clear input and

both devices have power-on reset.

16-Bit Accuracy Over

Temperature without

Autocalibration

In the past, only autocalibrated DACs

could achieve 16-bit (1LSB) accuracy

over temperature. Not only did this

require cumbersome calibration over-

head for the user but the DACs also

had to be recalibrated every time the

temperature changed because of the

DAC’s poor linearity drift. In addition,

the highly complicated autocal-

ibration circuitry made the DACs very

large and very expensive.

Now there is a better choice. The

LTC1595/LTC1596’s ultralow linear-

ity drift (well below ±0.2LSB from

–40°C to 85°C) allows the DACs to be

factory trimmed and to hold their

16-bit accuracy over time and tem-

perature. This provides a cleaner,

easier, more cost-effective solution

for precision systems.

Figure 2 shows the typical integral

nonlinearity (INL) and differential

nonlinearity (DNL) of the LTC1595.

The outstanding 0.25LSB typical val-

ues and the very low drift allow a

maximum 1LSB speciﬁcation to be

guaranteed over the extended indus-

trial temperature range.

continued on page 3

IN THIS ISSUE…

COVER ARTICLE

New 16-Bit SO-8 DAC

Has 1LSB Max INL and DNL

Over Industrial Temperature ......... 1

Jim Brubaker

and William C. Rempfer

Issue Highlight .............................. 2

LTC

in the News ........................... 2

DESIGN FEATURES

The LT

1374: New 500kHz, 4.5A

Monolithic Buck Converter ............ 5

Karl Edwards

The LT1581 Low Dropout Regulator

Provides 10 Amps of Output Current

..................................................... 8

Todd Owen

The LT1370: New 500kHz, 6A

Monolithic Boost Converter ......... 11

Karl Edwards

New LTC1504: Flexible, Efficient

Synchronous Switching Regulator

Can Source or Sink 500mA ......... 14

Dave Dwelley

Low Dropout Regulator Driver

Handles Fast Load Transients and

Operates on Single 3V–10V Input

................................................... 16

Lenny Hsiu

LT1579 Battery-Backup Regulator

Provides Uninterruptible Power ... 18

Todd Owen

High Efficiency Distributed

Power Converter Features

Synchronous Rectification .......... 21

Dale Eagar

DESIGN IDEAS

............................................. 25–32

(complete list on page 25)

DESIGN INFORMATION

Reference Squeezes More Performance

from Less Package ...................... 33

John Wright

LTC1659, LTC1448: Smallest Rail-to-

Rail 12-Bit DACs Have Lowest Power

................................................... 34

Hassan Malik

LTC1197/LTC1199: New Micropower

MSOP 10-Bit ADC Samples at 500ksps

................................................... 35

Guy Hoover and Marco Pan

New Device Cameos ..................... 37

Design Tools ................................ 39

Sales Offices ............................... 40

Linear Technology Magazine • November 1997

EDITOR’S PAGE

Issue Highlights

The subject of our cover article this

issue is the LTC1595/LTC1596 serial

input, 16-bit, multiplying current

output DACs. These parts feature

±1LSB maximum INL and DNL over

the industrial temperature range,

ultralow, 1nV-s glitch impulse and

±10V output capability. They are pin-

compatible with industry-standard

12-bit DACs.

The remainder of this issue’s Design

Features section introduces a variety

of new power products:

LT1374 is a new 4.5A buck con-

verter designed to meet the needs of

higher current and voltage applica-

tions. With its 500kHz operating

frequency and integral 80mΩ switch,

only a few surface mount compo-

nents are required to produce a

complete switching regulator. The

LT1374 features current mode con-

trol, external synchronization and a

low current (20µA typical) shutdown

mode.

Another new 500kHz switcher

introduced in this issue is the LT1370,

a 6A boost converter. At 65mΩ

on-resistance and 42V maximum

switch voltage, the LT1370 can be

used in a wide range of output voltage

and current applications. LT1370 fea-

tures include current mode operation,

external synchronization and a low

current shutdown mode (12µA typi-

cal). The LT1370 will operate in all

standard switching conﬁgurations,

including boost, buck, flyback, for-

ward, inverting and SEPIC.

For even higher currents, we intro-

duce the LT1339, the buck/boost

converter that “needs no steroids.”

The LT1339 is ideal for power levels

ranging from tens of watts to tens of

kilowatts. It has an innovative slope-

compensation function that allows

the circuit designer freedom in con-

trolling both the slope and offset of

the slope-compensation ramp. Addi-

tionally, the LT1339 has an average

current limit loop that yields a con-

stant output current limit, regardless

of input and/or output voltage.

Yet another new switcher is the

LTC1504 8-pin step-down switching

regulator. It consists of a 200kHz

ﬁxed frequency, voltage-feedback,

buck-mode switching regulator con-

troller and a pair of 1.5Ω power

switches in an 8-pin SO package. The

LTC1504 improves functionality by

integrating a synchronous rectiﬁer

on-chip, maximizing efﬁciency and

minimizing external parts count while

allowing the output to both sink and

source current: it can source or sink

up to 500mA with input voltages from

3.3V to 10V and output voltages as

low as 1.26V.

A new low dropout NPN regulator,

the LT1581, can provide up to 10

amps of output current. By using

separate supplies for the control cir-

cuitry and the drive to the NPN output

transistor, the dropout on the pass

transistor is reduced to the satura-

tion voltage of the transistor. This

makes it is easy to obtain dropout

voltages as low as 430mV at 10A of

load current. The part is fully protected

from overcurrent and overtempera-

ture conditions.

Another low dropout regulator

debuting this month is the LT1579

“smart regulator.” The LT1579 is a

dual input, single output, low drop-

out regulator that provides an

uninterruptible output voltage from

two independent input voltage sources

on a priority basis. It is capable of

providing 300mA from either input at

a dropout voltage of 0.4V.

The LT1573 is the latest low drop-

out voltage regulator driver from

Linear Technology. The LT1573 is

designed to provide a low cost solu-

tion to applications requiring high

current, low dropout and fast

transient response. The LT1573 is

available as an adjustable regulator

with an output range of 1.27V to

6.8V, and with ﬁxed output voltages

of 2.5V, 2.8V and 3.3V. Output accu-

racy is better than 1% to meet the

critical regulation requirements of fast

microprocessors.

LTC in the News…

LTC Achieves Record

Quarterly Bookings,

Sales and Profits

“This was a strong summer quarter

for us, as we achieved record quarterly

bookings, sales and profits,” explains

Robert Swanson, president and CEO,

based on Linear Technology’s latest

quarterly sales and earnings report.

“We grew cash by roughly $63 million

and now have over $500 million in

cash and short-term investments.

Demand was strong for our products,

particularly in the communications

and computer end markets. This

demand should help us continue to

grow in the upcoming quarter.”

These comments are based on

results announced on October 14,

1997. LTC had net sales for its first

quarter ended September 28 of

$109,802,000—an increase of 22%

over net sales of the $90,063,000 for

the first quarter of the previous year.

The Company also reported record net

income for the quarter of $40,643,000

or $0.51 per share, an increase of 30%

over last year of $31,358,000 or $0.40

per share, reported for the first quar-

ter of last year. A cash dividend of

$0.06 will be paid on November 12,

1997 to shareholders of record on

October 24, 1997.

The Design Ideas section of this

issue features two battery-related

applications: a single-cell Li-Ion

battery supervisor based on the

LT1496 op amp and the LT1634 pre-

cision reference, a low profile 800mA

Li-Ion charger using the LT1510-5

switch-mode battery charger. Also

included are a high voltage/high cur-

rent bench supply, a switcher that

generates two bias voltages without a

transformer, a single-supply random

code generator and a low distortion,

low power HDSL driver.

In addition, we present design

information on the LT1460 voltage

reference in the SOT-23 package, the

LTC1448 and LTC1659 12-bit rail-to-

rail DACs and the LTC1197/LTC1199

10-bit ADCs.

We conclude with a quintet of new

device cameos.

Linear Technology Magazine • November 1997

DESIGN FEATURES

Ultralow 1nV-s Glitch

Glitches in a DAC’s output when it

updates can be a big problem in pre-

cision applications. Usually, the

worst-case glitch occurs when the

DAC output crosses midscale. The

LTC1595/LTC1596’s new proprietary

deglitcher reduces the output glitch

impulse to 1nV-s, which is ten times

lower than any other 16-bit indus-

trial DAC. (1nV-s is equivalent to a

glitch of 7LSBs and 1µs duration.) In

addition, the deglitcher makes the

glitch impulse uniform for any code.

Figure 3 shows the output glitch for a

midscale transition with a 0V to 10V

output range.

Tiny Footprint: SO-8

Previous 16-bit DACs have had large

packages that required large board

space. The new LTC1595 breaks this

trend with an SO-8 pinout. With the

addition of a single output op amp

(MSOP or SO-8) and a voltage refer-

ence, a complete 16-bit voltage output

DAC can be constructed in a fraction

of the space of previous products.

Pin-for-Pin Upgrade to

Industry Standard 12-Bit DACs

An especially helpful feature for some

users is the ease of upgrade to these

new 16-bit devices. They directly

replace the popular 12-bit DAC8043/

8143 and AD7543. Because sensitiv-

ity of the LTC1595/LTC1596 INL to

op amp offset voltage is reduced by

ﬁve times compared to the 12-bit

devices, most systems can be easily

upgraded to true 16-bit resolution

and linearity without requiring more

precise op amps.

0V–10V and ±10V

Output Capability

Precision 0V–10V Outputs

with One Op Amp

Figure 4 shows the circuit for a 0V–

10V output range. The DAC uses an

external reference and a single op

amp in this conﬁguration. This cir-

cuit can also perform 2-quadrant

multiplication where the reference

input is driven by a ±10V input signal

112k 7k

112k

56k

112k

56k

112k56k56k56k

DECODER

D15

(MSB) D13

D14 D12 D11 D0

(LSB)

CLK IN

LOAD

REF

OUT1

GND

SRI

1595_01.EPS

DAC REGISTER

56k56k56k56k

INPUT 16-BIT SHIFT REGISTER

CLK

• • •

DIGITAL INPUT CODE

–0.5

INTEGRAL NONLINEARITY (LSB)

–0.3

–0.4

–0.2

0.2

0.3

0.4

0.5

–0.1

0.1

16384 32768

1595_2a.eps

49152 65535

DIGITAL INPUT CODE

–0.5

–0.3

–0.4

–0.2

0.2

0.3

0.4

0.5

–0.1

0.1

16384 32768 49152 65535

DIFFERENTIAL NONLINEARITY (LSB)

TIME (µs)

01234

OUTPUT VOLTAGE (mV)

1595 03 .eps

–10

+10

COMPETITOR'S DAC

LTC1595/LTC1596

1nV-s TYP

Figure 1. The LTC1595/LTC1596 16-bit CMOS DACs use a precision thin-ﬁlm modiﬁed R/2R architecture to provide unsurpassed accuracy and

stability. They provide a pin-compatible upgrade for industry-standard 12-bit CMOS DACs (DAC8043/8143 and AD7543).

Figure 2. The outstanding INL and DNL

(typically less than 0.25LSB) and very low

linearity drift allow a maximum 1LSB spec to

be guaranteed over industrial temperature.

Figure 3. The proprietary deglitcher reduces

the output glitch to less than 1nV-s, which is

ﬁve times less than any other 16-bit

industrial DAC. Further, the deglitcher makes

the glitch impulse uniform, independent of

code.

LTC1595, continued from page 1

Linear Technology Magazine • November 1997

DESIGN FEATURES

and V

OUT

swings from 0V to –V

REF

The full-scale accuracy of the circuit

is very precise because it is deter-

mined by precision-trimmed internal

resistors. The power dissipation of

the circuit is set by the op amp dissi-

pation and the current drawn from

the DAC reference input (7k nomi-

nal). The supply current of the DAC

itself is less than 10µA.

An advantage of the LTC1595/

LTC1596 is the ability to choose the

output op amp to optimize the accu-

racy, speed, power and cost of the

application. Using an LT1001 pro-

vides excellent DC precision, low noise

and low power dissipation (90mW

total for Figure 4’s circuit). For higher

speed, an LT1007 or LT1122 can be

used. The LT1122 will provide set-

tling to 1LSB in 3µs for a full-scale

transition. Figure 5 shows the 3µs

settling performance obtained with

VDD VREF

LTC1595

RFB

GND

VIN

CLOCK

DATA

LOAD

CLK

SRI

OUT1

33pF

VOUT

1595_04.EPS

–

LT1001

µP

0.1µF

REF

LTC1595

GND

REF

–10V TO 10V

CLK

SRI

OUT1

OUT

(–V

REF

REF)

1595_05.EPS

–

1/2 LT1112

R3

20k

R1

10k

R2

20k

–

1/2 LT1112

33pF

the LT1122. The feedback capacitor

in Figure 4 ensures stability. In higher

speed applications, it can be used to

optimize transient response. In slower

applications, the capacitor can be

increased to reduce glitch energy and

provide ﬁltering.

Precision ±10V Outputs

with a Dual Op Amp

Figure 6 shows a bipolar, 4-quadrant

multiplying application. The refer-

ence input can vary from –10V to 10V

and V

OUT

swings from –V

REF

to +V

REF

If a ﬁxed 10V reference is used, a

precision ±10V bipolar output will

result.

Unlike the unipolar circuit of Fig-

ure 4, the bipolar gain and offset will

depend on the matching of the exter-

nal resistors. A good way to provide

good matching and save board space

is to use a pack of matched 20k

resistors (the 10k unit is formed by

placing two 20k resistors in parallel).

The LT1112 dual op amp is an

excellent choice for high precision,

low power applications that do not

require high speed. The LT1124 will

provide faster settling. Again, with op

amp selection the user can optimize

the speed, power, accuracy and cost

of the application.

Conclusion: Time to Upgrade

The new LTC1595/LTC1596 provide

an excellent opportunity for users to

upgrade lower resolution systems to

16 bits, cleanly, easily and cost effec-

tively. They give outstanding accuracy

over temperature, ultralow glitch im-

pulse, small footprint and low cost.

For these reasons, they will improve

the performance of industrial and

instrumentation systems.

Figure 4. With a single external op amp, the DAC performs

2-quadrant multiplication with ±10V input and 0V to –V

REF

output. With a ﬁxed –10V reference, it provides a precision 0V–

10V unipolar output.

Figure 6. With a dual op amp, the DAC performs 4-quadrant multiplication.

With a ﬁxed 10V reference, it provides a ±10V bipolar output.

Figure 5. When used with an LT1122 (in the circuit of

Figure 4), the LTC1595/LTC1596 can settle in 3µs to a

full-scale step. The top trace shows the output swinging

from 0V to 10V. The bottom trace shows the gated

settling waveform settling to 1LSB (1/3 of a division)

in 3µs.

VOUT

5V/DIV

GATED VOUT

500µV/DIV

1µs/DIV

Linear Technology Magazine • November 1997

DESIGN FEATURES

The LT1374: New 500kHz, 4.5A

Monolithic Buck Converter by Karl Edwards

Introduction

Expanding on the current range of

500kHz switchers, Linear Technol-

ogy introduces a new 4.5A buck

converter, the LT1374. The LT1374 is

the big brother of the 1.5A LT1376.

Designed to meet the needs of higher

current and voltage applications, it

maintains high efﬁciency using an

on-chip 80mΩ switch. With its 500kHz

operating frequency and integral

switch, only a few external, surface

mount components are required to

produce a complete switching regula-

tor. All the features found on the

LT1376 have been retained, includ-

ing current mode control, external

synchronization and a low current

(typically 20µA) shutdown mode.

Improvements have been made to

reduce start-up headroom and switch-

ing noise. A novel power device layout

makes it possible to ﬁt a high speed,

bipolar, 80mΩ switch into a surface

mount SO-8 package. The LT1374 is

also available in DD and TO-220 pack-

ages for higher power applications.

LT1374 Features

❏Constant 500kHz switching

frequency

❏80mΩ high speed switch

❏20µA shutdown current

❏Uses all surface mount

components

❏Cycle-by-cycle current limiting

❏Available in SO-8 package

Circuit Description

The LT1374 is a constant-frequency,

current mode buck converter. As

shown in Figure 1, an internal clock

and two feedback loops control the

power switch. In addition to the nor-

mal output feedback error ampliﬁer,

a current sense ampliﬁer monitors

switch current on a cycle-by-cycle

basis. The feedback error ampliﬁer

output, V

, is compared to the switch-

current sense output to control the

trip point of the power switch. This

removes the additional 90° phase shift

that occurs with voltage-controlled

systems, easing frequency compen-

sation and giving a faster transient

response. Nonlinear slope compen-

sation is added to the current sense

signal to prevent subharmonic oscil-

–

INPUT

2.9V BIAS

REGULATOR

500kHz

OSCILLATOR

FREQUENCY

SHIFT CIRCUIT

GND

1374 BD

SLOPE COMP

0.01Ω

BIAS INTERNAL

CURRENT

SENSE

AMPLIFIER

VOLTAGE GAIN = 20

SYNC

SHUTDOWN

COMPARATOR

CURRENT

COMPARATOR

ERROR

AMPLIFIER

= 2000µMho

FOLDBACK

CURRENT

LIMIT

CLAMP

BOOST

RS

FLIP-FLOP DRIVER

CIRCUITRY

S

R



0.9V

LOCKOUT

COMPARATOR

0.4V

3.5µA

Q1

POWER

SWITCH

2.38V 2.42V

–

SHDN

Figure 1. LT1374 block diagram

Linear Technology Magazine • November 1997

DESIGN FEATURES

lation at higher duty cycles. The slope

amplitude has been set to stabilize

inductance values as low as 2µH. The

feedback pin, FB, provides several

overload protection functions. As feed-

back falls below 1.7V, a voltage clamp

is gradually applied to the V

pin,

reducing current limit. At power-up,

this performs a soft-start function.

When FB is below 1V, as in the case of

a shorted output, the oscillator fre-

quency is reduced to 100kHz. Because

the minimum on-time for the switch

remains the same, minimum duty

cycle is effectively reduced by a factor

of ﬁve. This lowers power dissipation

in both the LT1374 and the catch

diode, D1, as shown in Figure 2.

High switch efﬁciency is attained

by driving the switch with a voltage

that is higher than the input voltage,

allowing the switch to saturate. This

boosted voltage is generated on the

BOOST pin with an external capaci-

tor, C2, and diode, D2 as shown in

Figure 2.

The LT1374 has been improved by

reducing minimum start-up voltage

at low output currents, as shown in

Figure 4. At low output currents,

there is not enough energy in the

inductor at switch-off to drive the

switch-node capacitance down to

ground. In this condition, the mini-

mum start-up voltage is the unboosted

voltage drop across the output switch.

Improvements made to the drive cir-

cuitry of the LT1374 have reduced

this drop from 2.5V on the LT1376 to

1.2V.

Additionally, the minimum boost

voltage required to fully saturate the

switch has been reduced to 2.3V.

Current used by the boost circuit

appears as an efﬁciency loss; supply-

ing this from a lower voltage, via the

boost diode, D2, improves efﬁciency.

For most applications with outputs

above 3V, the conﬁguration shown in

Figure 2 will be optimal. Converters

with lower output voltages should

use the input or an alternative supply

for the boost diode.

Switch-transition time has impor-

tant effects on the usability and overall

efﬁciency of the converter. A fast

switch-transition time gives high efﬁ-

ciency, but it also creates EMI and

RFI. Slowing the transition time of the

switch reduces noise but lowers efﬁ-

ciency. The LT1374 operates at higher

currents than the LT1376, so the

same transition time would result in

more noise. Because the low on-

resistance of the LT1374 reduces I

losses in the switch, a small increase

in switching transition losses is an

acceptable trade-off to reduce noise.

The switching time for the LT1374 is

24ns, approximately 50% slower than

that of the LT1376. Most of the

LT1374’s circuitry operates from an

internally regulated 2.9V bias line.

By default, the bias regulator draws

power from the input pin, but if the

BIAS pin is connected to an external

source higher than 3V (normally V

OUT

bias current will be drawn from this

source, improving efﬁciency.

An external clock signal (up to

900kHz) can be fed into the SYNC pin

to increase the internal oscillator fre-

quency or synchronize it to a system

clock. The SYNC function is defeated

during start-up, when FB is low, to

allow correct operation of the reduced

frequency, shorted-output protection

circuit. The shutdown pin has two

functions, undervoltage lockout and

shutdown. Below 2.38V, the switch-

ing function is disabled. With the

addition of an external divider to the

input (R1 and R2 in Figure 2), an

undervoltage lockout can be imple-

mented. Below 0.4V, a complete

shutdown of the internal circuitry

occurs, reducing supply current to

20µA.

4.5A in an SO-8

The LT1374 is available in several

packages, two of which, the 7-lead

DD and the SO-8, are surface mount.

Thermal considerations are always

important when operating surface

mounted power devices without addi-

tional heat sinking. The output switch

of the LT1374 has been designed to

minimize power dissipation from both

on-resistance and base-drive current.

The SO-8 package has a fused ground

lead. This lead has a solid connection

to the tab to which the die is attached,

providing low thermal resistance out

of the package. Adding copper circuit

board area around this pin can reduce

junction-to-ambient thermal resis-

tance to 80°C/W. This allows the use

of the SO-8 packaged LT1374 in

applications that would have previ-

ously required a TO-220. For example,

a motor drive may require 4A at start-

up, but only 2.5A when running. The

SO-8 packaged LT1374 can provide

the full 4.5A of switch current required

to supply the 4A peak load. It could

supply this current for several sec-

onds due to the long thermal time

BOOST

LT1374-5

OUTPUT**

5V/4.25A

 * RIPPLE CURRENT RATING > I

OUT

/2

** L1 = COILTRONICS UP2-4R7; (561) 241-7876

 INCREASE L1 TO 10 µH FOR LOAD CURRENTS ABOVE 3.5A AND TO 20µH ABOVE 4A

INPUT

6V TO 25V

1374_02.EPS

C2

0.27µF



3.3nF

D1

MBRS330T3

C1

100µF, 10V

SOLID

TANTALUM

C3*

10µF TO

50µF

D2

1N914

L1**

5µH

BIAS

GND V

SHDN

R1

56k

R2

33k

LOAD CURRENT (A)

EFFICIENCY (%)

100

1.5 2.5

1374_03

0.5 1.0 2.0 3.0 3.5

Figure 2. 5V buck converter Figure 3. Efﬁciency: 10V in, 5V out

Linear Technology Magazine • November 1997

DESIGN FEATURES

constant of the SO-8 package. Calcu-

lating the temperature rise for 2.5A of

continuous current using the circuit

in Figure 2, a 12V to 5V converter, we

see that power losses due to the switch

resistance, switch drive current and

quiescent current are 0.57W, 0.1W

and 0.04W, respectively, totaling

0.71W. On a well laid-out PCB using

the SO-8 package, this power dissi-

pation would result in only a 57°C

rise of die temperature, allowing the

part to operate at an ambient tem-

perature of 77°C. If 4A of continuous

current is required, use the surface

mounted DD package (θ

= 30°C/W).

For even higher power, the TO-220

package (θ

= 4°C/W) may be used.

Application:

5V/4.25A Buck Converter

With its 25V input and 4.5A mini-

mum switch current, the LT1374 will

ﬁt into a wide range of applications.

Figure 2 shows a typical buck con-

verter with a 6V to 25V input range, a

5V output and 4.25A of output cur-

rent capability. Due to the low

on-resistance of the switch, efﬁciency

remains high over a wide range of

currents, as shown in Figure 3. To

reduce power dissipation, both the

BIAS pin and boost circuit are sup-

plied from the 5V output.

Several factors, including maxi-

mum current, core and copper losses,

size and cost, affect the choice of

inductor, L1. A high value, high cur-

rent inductor gives the highest output

current with the lowest ripple, at the

expense of a large physical size and

cost. Lower inductance values tend to

be physically smaller, have higher

current ratings and are cheaper, but

output ripple current, and hence

ripple voltage, increases.

The input capacitor, C3, experi-

ences very high ripple currents, up to

OUT

/2, so low ESR tantalum capaci-

tors are needed. At 4.25A output

current, two capacitors in parallel are

required to meet the ripple current

requirement. The ripple current in

the output capacitor, C1, is lower, but

its ESR still needs to be low to limit

output voltage ripple. The voltage drop

across the catch diode, D1, has a

signiﬁcant effect on overall converter

efﬁciency, especially at higher input

voltages when the switch duty cycle is

low. Its ability to survive short-circuit

conditions may increase its power

rating. For good electrical perfor-

mance, D1 must be placed close to

the LT1374. The power dissipated in

D1 will raise the PC board’s tempera-

ture around the LT1374. This must

be taken into account when modeling

or taking bench measurements of die

temperature.

The loop compensation capacitor,

, produces a pole in the frequency

response at 240Hz. Unity-gain phase

margin can be further improved with

the addition of a resistor, typically 2k,

in series with C

, adding a zero to the

frequency response. This, however,

can cause a large-signal subharmonic

problem in the loop. The output ripple

voltage feeds back through the error

ampliﬁer to the V

pin, changing the

current trip point of the next cycle.

This changes the voltage ripple at the

output, and the loop is closed. Adding

LOAD CURRENT (mA)

5.0

INPUT VOLTAGE (V)

5.5

6.0

6.5

7.0

7.5 LT1376

LT1374

8.0

10 100 1000

1374_04

1374 F06

HIGH

FREQUENCY

CIRCULATING

PATH

LOAD

SWITCH NODE

a second capacitor directly from the

pin to ground to form a pole at one-

ﬁfth the switching frequency solves

the problem.

PCB Layout

All high current, high speed circuits

require careful layout to obtain opti-

mum performance. When laying out

the PCB, keep the trace length around

the high frequency switching compo-

nents, shown in Figure 5, as short as

possible. This minimizes the EMI and

RFI radiation from the loop created by

this path. These traces have a para-

sitic inductance of approximately

20nH/inch, which can cause an

additional problem at higher operat-

ing voltages. At switch-off, the current

ﬂowing in the trace inductance causes

a voltage spike. This is in addition to

the input voltage across the switch

transistor. At higher currents, the

additional voltage can potentially

cause the output switching transis-

tor to exceed its absolute maximum

voltage rating.

Conclusion

With its 80mΩ on-resistance, the

LT1374 makes a very compact, low

parts count, high current DC/DC

converter without the need for sepa-

rate control and power devices. The

wide selection of package outlines

provides a solution for most minia-

ture, low proﬁle applications. A 25V,

500kHz, 4.5A switch gives the ﬂex-

ibility for a wide range of applications.

The LT1374 complements and extends

the current range of Linear’s DC/DC

converters.

Figure 4. Minimum input voltage with

5V output

Figure 5. High speed switch path

Linear Technology Magazine • November 1997

DESIGN FEATURES

Introduction

Improvements in microprocessors and

logic devices continue to increase their

speed and power. Gate dimensions

continue to decrease along with the

supply voltages required to power

them. Unfortunately, the supply cur-

rent requirements for these new logic

devices are heading in the opposite

direction: a larger number of gates

driven at higher clock frequencies

requires higher supply current. In

addition, the load transients are larger

and faster. Current steps from sev-

eral hundred milliamperes to several

amperes in tens of nanoseconds are

common.

Lower supply voltages, higher sup-

ply currents and faster load transients

require tighter speciﬁcations for regu-

lators. Input/output differential

voltages tend to be under 1V, supply

currents can be up to 10A and tran-

sitions from no load to full load can

occur in tens of nanoseconds. Low

dropout PNP-based regulators can

meet input/output differential speci-

ﬁcations, but their transient load

performance is less than stellar. Con-

ventional NPN-based regulators can

meet the higher load current speciﬁ-

cations and transient needs, but

dropout voltage tends to be above 1V.

In addition, this higher dropout volt-

age speciﬁcation requires the use of

higher input supply voltages, leading

to increased power dissipation in the

regulator.

The LT1581

Solves Many Problems

The new LT1581 NPN regulator can

meet the needs of many new designs.

By using separate supplies for the

control circuitry and the drive to the

NPN output transistor, the dropout

on the pass transistor is reduced to

the saturation voltage of the transis-

tor. With this approach, it is easy to

obtain dropout voltages as low as

The LT1581 Low Dropout Regulator

Provides 10 Amps of Output Current

by Todd Owen

220mV at 4A, 310mV at 7A or 430mV

at 10A of load current. The current

requirement for the control circuitry

is approximately 1/150 of the load

current, or about 70mA for a 10A

load. This control voltage must be at

least 1.2V greater than the output to

achieve these low dropout speciﬁca-

tions. Maximum supply voltage to the

power input is 7V and the maximum

accuracy has been improved to a guar-

anteed 0.6% tolerance. The part can

easily hold 1% output accuracy over

temperature, guaranteed, while

operating with an input/output dif-

ferential voltage of well under 1V.

Careful design has eliminated any

supply sequencing issues associated

with a dual-supply system. The out-

put is inhibited until both input

supplies are operating. If the control

voltage reaches a quiescent state ﬁrst,

the output current will be limited to a

few milliamperes until the power input

voltage stabilizes. If the power input

comes up ﬁrst, the output will not

turn on at all until the control voltage

reaches a quiescent state. The output

can never turn on unregulated. The

LT1581 can also be operated as a

single-supply device by tying the con-

trol and power inputs together.

Dropout in single supply operation is

essentially the same as would be

expected in a conventional NPN-based

low dropout regulator.

All of these features add together

to make a part with excellent static

and dynamic speciﬁcations. Addition-

ally, the low dropout voltage solves a

problem many designers have been

faced with: power dissipation. Lower-

ing the input/output differential

voltage translates directly to reduced

power dissipation in the regulator.

Care must be exercised regarding

board layout and thermal manage-

ment if the full performance capability

of the LT1581 is to be achieved.

Layout Considerations

Board layout becomes more critical

at higher current levels. Running too

much current through a narrow trace

can turn the trace into a fuse. Even

with a wide trace, the resistance pre-

sented can cause problems. For 1 oz.

copper, a 0.10 inch wide trace that is

0.7 inches long can easily handle up

Lower supply voltages,

higher supply currents and

faster load transients

require tighter

speciﬁcations for

regulators. Input/output

differential voltages tend to

be under 1V, supply

currents can be up to 10A

and transitions from no

load to full load can occur

in tens of nanoseconds.

control voltage is 13V. The part is

fully protected from overcurrent and

overtemperature conditions. Fixed

output voltage and adjustable ver-

sions are available in a 7-pin TO-220

package.

Both ﬁxed and adjustable versions

of the LT1581 have remote-sense pins,

permitting very accurate regulation

of output voltage at the load, where it

counts, rather than at the regulator.

Typical load regulation is only 1mV

for a 2.5V output over a 100mA to 10A

range. The adjust pin is brought out

on the ﬁxed voltage version of the

part. This allows for the addition of a

small bypass capacitor on the feed-

back divider, greatly improving

transient response. A capacitor value

of 0.1µF to approximately 1µF will

generally provide optimum transient

response. Additionally, reference

Linear Technology Magazine • November 1997

DESIGN FEATURES

to 10A of current. However, the same

trace has a resistance of 3mΩ. This

adds 30mV to the overall dropout

voltage of the regulator, and without

remote load sensing, adds a 30mV

load regulation error to the output.

Remote load sensing can virtually

eliminate the problems associated

with the parasitic trace resistance,

but cannot remove the added dropout

voltage. Adding 30mV to the dropout

voltage constitutes an increase of 7%.

This additional drop reduces the avail-

able headroom, and can reduce the

maximum available output current.

Additional consideration must be

given to power dissipation in the cop-

per trace. With 10A of current and

30mV of drop across the trace, 0.3W

of power is being dissipated in the

trace alone. This can add to the over-

all problem of system-level thermal

management. It is recommended that

a good reference source be consulted

to determine minimum required trace

widths.

High frequency current transients

also necessitate careful attention to

board layout. Load current steps

contain higher order frequency com-

ponents that must be handled by the

output decoupling network until the

regulator throttles to the new load

current level. Capacitors are not ideal

elements and contain parasitic resis-

tance and inductance that dominate

the change in output voltage at the

beginning of a load transient step.

Using capacitors with low ESR, low

ESL and good high frequency charac-

teristics is critical in high speed

applications. The location of the

decoupling network is critical to tran-

sient performance.

For DC operation, the designer’s

main concern will be parasitic resis-

tance. Consideration must be given

to all high current paths to minimize

the parasitic resistance and induc-

tance present in the input, output

and ground paths for the regulator.

Maximize the trace widths for all high

currents paths in the layout. The best

solution requires a multilayer board.

With one layer devoted to power and

a separate layer for the return path,

narrow copper traces are replaced

with entire copper planes. With power

and ground planes, parasitic resis-

tance is reduced to nearly zero. An

additional beneﬁt is the elimination

of the power dissipation in copper

traces, easing system-level thermal

management.

When considering the transient

performance, higher order frequency

components must not be neglected.

Capacitors with low ESR and low ESL

are a necessity. Transient load cur-

rents through the ESR of a capacitor

shows up as an instantaneous droop

in output voltage (∆V = ∆I • ESR).

When load current steps occur, para-

sitic inductance tries to limit the rate

of change of the current. The result is

a voltage spike at the start of the load

transient. This spike is deﬁned by V =

L • dI/dt, where L is the ESL of the

capacitors and their interconnecting

traces. These transient effects are

illustrated in Figure 1. Decoupling

capacitors should be placed as close

to the load as possible, with multiple

capacitors used to distribute and

reduce overall parasitics. Any narrow

traces will increase parasitic induc-

tance. Using power planes can reduce

these effects, but will not eliminate

them entirely. Other possible prob-

lems can occur when using vias to

connect capacitors to inner plane lay-

ers, as they have parasitic inductance

and resistance as well. The use of

multiple vias is highly recommended

to reduce parasitics. Also, be sure to

minimize the distance between pads

and vias.

Thermal Management

for the LT1581

Even at the minimum differential volt-

ages, power dissipation in the LT1581

can reach several watts. Without

adequate heat sinking, maximum

rated junction temperature can eas-

ily be exceeded. Excessive junction

temperature will reduce the reliabil-

ity of the part. By using an adequate

heat sink, junction temperature can

ESR

EFFECTS

1581_01.eps

ESL

EFFECTS CAPACITANCE

EFFECTS

POINT AT WHICH REGULATOR

TAKES CONTROL

SLOPE, =

V

t∆I

1581_02.eps

C2

220µF

10V

CONT

POWER

= 3.3V

CONTROL

= 5V

RTN

SENSE

OUT

ADJ

R1

110Ω

OUT

= 2.5V

LT1581

C3

22µF

25V

C4

0.33µF

R2

110Ω

C1

100µF

10V

100µF

10V

1µF

25V

10X

MICROPROCESSOR

SOCKET

Figure 1. Transient effects

Figure 2. The LT1581 delivers 2.5V from 3.3V at up to 9A.

Linear Technology Magazine • November 1997

DESIGN FEATURES

be maintained well under maximum,

ensuring reliable operation.

Power dissipation is made up of

two main components: the power in

the output transistor and the power

in the drive circuit. The power in the

additional control circuitry is negli-

gible at a maximum of 25mW.

To calculate the total power dissi-

pation, use the following formula:

PTOTAL = POUTPUT + PDRIVE

where POUTPUT =

(VPOWER – VOUT)(IOUT)

and PDRIVE =

(VCONTROL – VOUT)(ICONTROL)

CONTROL

is a function of output

current and is equal to between

OUT

/150 (typ) and I

OUT

/58 (max).

The electrical characteristics sec-

tion of the LT1581 data sheet provides

separate thermal resistances and

maximum junction temperatures for

both the control circuitry and the

power transistor. These speciﬁcations

have been developed to more accu-

rately reﬂect device temperature and

ensure safe operating temperatures.

Maximum junction temperature must

be calculated for both sections to

ensure that both thermal limits are

met.

Junction-to-case thermal resis-

tance is speciﬁed from the IC junction

to the bottom of the case directly

below the die. This is the lowest resis-

tance path for heat ﬂow. Proper

mounting is required to ensure the

best possible thermal ﬂow from this

area of the package to the heat sink.

Please consult Linear Technology’s

“Mounting Considerations for Power

Semiconductors,” 1990 Linear Appli-

cations Handbook, Volume 1, pages

RR3–1 to RR3–20. Note that the case

of the LT1581 is electrically connected

to the output.

The following example illustrates

how to calculate the heat sink size

needed to ensure reliable operation of

the LT1581. The circuit in Figure 2 is

used with the following assumptions:

VCONTROL (max continuous) = 5.25V

(5V + 5%),

VPOWER (max continuous) = 3.465V

(3.3V + 5%),

VOUT = 2.5V, IOUT = 8A, TA = 40°C,

θCASE–HEAT SINK = 1°C/W

(with thermal compound).

The following speciﬁcations are

from the LT1581 data sheet:

RθJC(CONTROL) = 0.65°C/W,

RθJC(POWER) = 2.5°C/W,

TJ(MAX) = 125°C for the control

section, 150°C for the power

section.

Power dissipation under these condi-

tions is equal to:

PDRIVE = (VCONTROL – VOUT)(ICONTROL)

ICONTROL = IOUT/58 = 8A/58 = 138mA

PDRIVE = (5.25V – 2.5V)(138mA) =

380mW

POUTPUT = (VPOWER – VOUT)(IOUT)

= (3.465V – 2.5V)(8A) = 7.72W

Total Power Dissipation = 8.1W

Junction temperature will be equal

to:

TJ = TA + PTOTAL (θHEAT SINK + θCASE –

HEAT SINK + θJC)

For the control section:

TJ(MAX) = 125°C = 40°C +

8.1W(θHEAT SINK + 1°C/W +

0.65°C/W)

θHEAT SINK + 1.65°C/W = (125°C –

40°C)/8.1W

θHEAT SINK(MAX) = 8.8°C/W

For the power section:

TJ(MAX) = 150°C = 40°C +

8.1W(θHEAT SINK + 1°C/W + 2.5°C/W)

θHEAT SINK + 3.5°C/W = (150°C –

40°C)/8.1W

θHEAT SINK(MAX) = 10°C/W

As shown, the maximum thermal

resistance for the heat sink must be

below 8.8°C/W to ensure reliable

operation from the LT1581. This

resistance value can be used to specify

the heat sink and airﬂow require-

ments needed to operate the LT1581

below speciﬁed thermal limits.

Conclusion

The LT1581 can be used for a variety

of applications where high current

and low dropout voltage are a must. It

can deliver up to 10A of output cur-

rent at a typical dropout voltage of

430mV. A remote-sense pin virtually

eliminates load regulation problems.

Supply-sequencing issues have been

eliminated. The LT1581’s excellent

transient performance makes it suit-

able for high current applications

where load current transients occur

at high frequency. The low dropout

voltage reduces the amount of input/

output differential voltage required

and simultaneously reduces the power

dissipation in the regulator. When

consideration is given to board layout

and thermal management, perfor-

mance can be maximized.

Fink, D.G. and D. Christiansen: Electronic

Engineers’ Handbook (Second Edition). McGraw

Hill, 1982. Figure 7–75.

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for

the latest information

on LTC products, 

visit

www.linear-tech.com

Linear Technology Magazine • November 1997

DESIGN FEATURES

The LT1370: New 500kHz, 6A

Monolithic Boost Converter by Karl Edwards

Introduction

Complementing and expanding on the

current LT1371/LT1372 family of

500kHz switchers, Linear Technol-

ogy introduces the LT1370, a 6A boost

converter. At 65mΩ on-resistance, 42V

maximum switch voltage and 500kHz

switching frequency, the LT1370 can

be used in a wide range of output

voltage and current applications.

The high efﬁciency switch is

included on the die, along with the

oscillator, control and protection cir-

cuitry necessary for a complete

switching regulator. This part com-

bines the convenience and low parts

count of a monolithic solution with

the switching capabilities of a dis-

crete power device and controller. All

the features found on the LT1371/

LT1372 have been retained on the

LT1370, including current mode

operation, external synchronization

and low current shutdown mode

(12µA typical). Only a few surface

mount components are needed to

complete a small, high efﬁciency DC/

DC converter. The LT1370 will oper-

ate in all the standard switching

conﬁgurations, including boost, buck,

ﬂyback, forward, inverting and SEPIC.

LT1370 Features

❏6A Minimum switch-current limit

❏Constant 500kHz switching

frequency

❏65mΩ high speed switch

❏12µA shutdown current

❏2.7V minimum input operating

voltage

❏Cycle-by-cycle current limiting

❏Available in DD and TO-220

packages

❏42V switch voltage

Circuit Description

The LT1370 is a current mode

switcher. This means that switch duty

cycle is directly controlled by the

switch current rather than by the

output voltage. Referring to Figure 1,

the switch is turned on at the start of

each oscillator cycle. It is turned off

when the switch current reaches a

predetermined level. The switch-off

current threshold is controlled by an

output voltage error ampliﬁer. This

technique has several advantages.

First, it responds quickly to input

voltage variations, unlike voltage mode

switchers, which have notoriously

poor line-transient response. Second,

it reduces the 90° phase shift at mid-

frequencies in the energy storage

inductor. This greatly simpliﬁes

closed-loop frequency compensation

under widely varying input voltage

and output load conditions. Finally,

it allows simple pulse-by-pulse cur-

rent limiting to provide maximum

switch protection under output over-

load or short-circuit conditions. A low

dropout internal regulator provides a

2.3V supply to all internal circuitry.

This low dropout design allows the

input voltage to vary from 2.7V to 25V

with virtually no change in device

performance. An internal 500kHz

oscillator is the basic clock for all

timing. It turns the output on via logic

and driver circuitry. Special adaptive

antisaturation circuitry detects the

onset of saturation in the power switch

and adjusts driver current instanta-

neously to limit switch saturation.

This minimizes driver dissipation and

provides very rapid turn off of the

switch.

A 1.245V bandgap reference biases

the positive input of the error ampli-

ﬁer. The negative input of the ampliﬁer

is brought out for positive output

voltage sensing. The error ampliﬁer

has nonlinear transconductance to

reduce output overshoot on start-up

or overload recovery. When the feed-

back voltage exceeds the reference by

40mV, error ampliﬁer transconduct-

ance increases by a factor of ten. This

allows the current-trip threshold, set

by V

, to be lowered faster, reducing

overshoot. The feedback input also

–

NFBA

NFB

S/S

100k

50k

0.005Ω

–

GND

1370_01.EPS

GND SENSE

1.245V

REF

5:1 FREQUENCY

SHIFT

OSCSYNC

SHUTDOWN

DELAY AND RESET LOW DROPOUT

2.3V REG ANTI-SAT

LOGIC DRIVER

SWITCH

–

≈ 20

COMP

Figure 1. LT1370 block diagram

Linear Technology Magazine • November 1997

DESIGN FEATURES

invokes oscillator frequency shifting,

which helps protect components dur-

ing overload conditions. When the

feedback voltage drops below 0.6V,

the oscillator frequency is reduced

5:1, to 100kHz. Lower switching fre-

quency, with the same minimum on

time, reduces the minimum switch

duty cycle and short-circuit current

in the inductor.

Similar to the LT1371, error ampli-

ﬁer circuitry allows the LT1370 to

directly regulate negative output volt-

ages. The NFB pin regulates at –2.49V

while the ampliﬁer’s output inter-

nally drives the FB pin to 1.245V.

This architecture, which uses the

same main error ampliﬁer, avoids

duplication of functions and main-

tains ease of use. The error signal

developed at the ampliﬁer output is

brought out externally to the V

pin.

The V

pin has three different func-

tions. It is used for frequency

compensation, current limit adjust-

ment and soft starting. During normal

regulator operation, this pin is in the

voltage range of 1V to 1.9V. The error

ampliﬁer is a current output (g

) type,

so this voltage can be externally

clamped to lower the current limit.

Likewise, a capacitor-coupled exter-

nal clamp provides soft start. Switch

duty cycle goes to zero if the V

pin is

pulled below the control threshold

(typically 1V), placing the LT1370 in

an idle mode.

The S/S pin has two functions,

synchronization and shutdown. The

internal oscillator can be synchro-

nized to a higher frequency by applying

a TTL square wave to this pin. This

allows the part to be synchronized to

a system clock. If the S/S pin is held

low, after a short delay the LT1370

will enter shutdown mode. In shut-

down mode, all internal circuitry is

disabled, reducing supply current to

12µA. An internal pull-up ensures

start-up when the S/S pin is left

open.

5V to 12V Boost Converter

Figure 2 shows a typical 5V to 12V

boost application. The feedback

divider network has been selected to

give the desired output voltage. As

long as R2 is less than 7k, FB input

bias current can be ignored. The

inductor needs to be chosen carefully

to meet both peak and average cur-

rent values. The output capacitor can

see high ripple currents—often, as in

this application, higher than the ripple

rating of a single capacitor. This

requires the use of two surface mount

tantalums in parallel; both capaci-

tors should be of the same value and

manufacturer. The input capacitor

does not have to endure such high

ripple currents and a single capacitor

will normally sufﬁce. The catch diode,

D1, must be rated for the output

voltage and average output current.

The compensation capacitor, C2, nor-

mally forms a pole in the 2Hz to 20Hz

range, with a series resistor, R3, to

add a zero at 1kHz to 5kHz.

A second capacitor, C3, is some-

times required to prevent erratic

switching. Ripple current in the output

capacitor’s ESR causes voltage ripple.

This feeds back through the error

amp to the V

pin, changing the cur-

rent-trip threshold cycle-to-cycle. The

problem appears as subharmonic

oscillation. Adding C3, typically one-

tenth the value of the main

compensation capacitor, reduces the

loop gain at the switching frequency,

preventing the oscillation.

The ground return from the com-

pensation network must be separate

from the high current switch ground.

If drops in the ground trace due to

switch current cause the V

pin to

dip, premature switch-off will occur.

This effect appears as poor load regu-

lation. A solution to this is to return

the compensation network to the FB

pin. The S/S pin in this example is

driven by a logical on/off signal, a low

input forcing the LT1370 into its 12µA

shutdown mode. Figure 3 shows the

overall converter efﬁciency. Note that

peak efﬁciency is over 90%; efﬁciency

stays above 86% at the device’s maxi-

mum operating current.

Positive-to-Negative

Converter

The NFB (negative feedback) pin allows

negative output regulators to be

designed with direct feedback. In the

circuit shown in Figure 4, a 2.7V to

13V input, –5V output converter, the

output is monitored by the NFB pin

and a simple divider network. No com-

plex level shifting or unusual

grounding techniques are required.

The regular FB pin is left open circuit

and the divider network, R2, R3, is

calculated based on the –2.49V NFB

reference voltage and 30µA of input

current. The switch-clamp diodes, D2

and D3, prevent the leakage spike

from the transformer, T1, from

LT1370

GND

1370_02.EPS

S/S

L1*



C1**

22µF

25V

C4**

22µF

25V

× 2

C2

0.047µFC3

0.0047µF



R3

R2

6.19k

R1

53.6k

OUT† 

12V

D1

MBRD835L

OFF

*

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UP2-4R7 (4.7µH)

UP4-100 (10µH)

AVX TPSD226M025R0200



+ +

L1

4.7µH

10µH

OUT



1.8A

2.0A

†

MAX I

OUT

OUTPUT CURRENT (A)

EFFICIENCY (%)

100

0.5 1.0 1.5 2.0

1370_03

Figure 2. 5V to 12V boost converter Figure 3. 12V output efﬁciency

Linear Technology Magazine • November 1997

DESIGN FEATURES

exceeding the switch’s absolute maxi-

mum voltage rating. The Zener voltage

of D2 must be higher than the output

voltage, but low enough that the sum

of input voltage and clamp voltage

does not exceed the switch voltage

rating.

5V SEPIC Converter

Figure 5 shows a SEPIC converter.

One of the advantages of the SEPIC

topology is that the input voltage can

range from below to above the output

voltage. In Figure 5, the input voltage

range is from 4V to 9V, with a 5V

output. The magnetic coupling of

inductors L1A and L1B is not critical

for operation, but generally they are

wound on the same core. C2 couples

LT1370



2.7V TO 13V

*PULSE PE-53719

(619) 674-8100

GND

NFB

1370_04.EPS

S/S

D2

P6KE-15A

D3

1N4148

D1

MBRD835L

C1

100µF

C2

0.047µF

C3

0.0047µFR1

R3

2.49k

R2

2.49k

–V

OUT†

–5V

C4

100µF

× 2

OFF



3V

5V

OUT



1.75A

2.25A

3.0A

T1*

•

†

MAX I

OUT



the inductors together and eliminates

the need for a switch snubber net-

work. C2 must have a very low ESR,

because the ripple current is equal to

/2. Its capacitance value is not

critical and has no signiﬁcant effect

on loop stability. The voltage across

C2 is equal to the input. A 4.7µF, 50V

ceramic will work in most SEPIC

applications. The S/S pin is used as

a logical on/off signal. In the off state,

there is no leakage to the output, and

only 12µA leakage from the input.

Thermal Performance

Due to the very low on-resistance of

the LT1370 switch and its driver efﬁ-

ciency, the surface mount DD package

version can be used over its full cur-

LT1370

GND



4V TO 9V

1370_05.EPS

S/S

C1

33µF

20V

C4

0.047µF

C5

0.0047µF

R1

R3

6.19k

R2

18.7k

1%



OUT†

C3

100µF

10V

× 2

OFF

L1A*

6.8µH

•

L1B*

6.8µH

C2

4.7µF

C1 = AVX TPSD 336M020R0200

C2 = TOKIN 1E475ZY5U-C304

C3 = AVX TPSD107M010R0100

BH ELECTRONICS 501-0726

(612) 894-9590

INPUT VOLTAGE MAY BE GREATER OR

LESS THAN OUTPUT VOLTAGE



D1

MBRD835L



4V

5V

7V

OUT



1.8A

2A

2.6A

2.9A

†

MAX I

OUT

*



rent range. With good layout, the DD

package can have a thermal resis-

tance as low as 20°C/W to 30°C/W,

junction-to-ambient. In the boost con-

verter shown in Figure 2, with an

input voltage of 5V and an output

voltage of 12V at 2A, power dissipa-

tion in the LT1370 is 2.2W. This causes

a 50°C to 60°C rise in die temperature

above ambient. Maximum junction

temperature is 125°C, so this appli-

cation would be able to operate up to

a 70°C ambient.

Conclusion

With its low resistance switch, 6A

operating current and 500kHz

operation, the LT1370 is ideal for

small, low parts count, high current

applications. Its high switching fre-

quency eliminates the need for bulky

magnetics and capacitors. Compared

to a separate control device and power

switch, the LT1370’s monolithic

approach simpliﬁes the design effort

required to implement a complete

DC/DC converter. The choice of TO-

220 and surface mount DD package,

42V switch, and ﬂexibility of switch-

ing topologies makes the LT1370 ideal

for a wide range of applications. The

LT1370 extends Linear Technology’s

LT1371 and LT1372 family of 500kHz,

monolithic switchers all the way up to

6A.

Figure 4. Positive-to-negative converter with direct feedback

Figure 5. Two Li-Ion cells to 5V SEPIC converter

Linear Technology Magazine • November 1997

DESIGN FEATURES

MAX

SHDN

GND

SENSE

SS COMP

LTC1504-3.3

SHUTDOWN



5V C

IN

22µF

IMAX

**

68k



1000pF

*

O.1µF

OUT

100µF

EXT

47µH

OUT

3.3V/500mA

C

IN

= AVX TPSC226M016R0375

C

OUT

= SANYO 16CV100GX

L

EXT

 = SUMIDA CD54-470

 *OPTIONAL: DELETE TO DISABLE SOFT START

 **OPTIONAL: DELETE TO DISABLE CURRENT LIMIT

1504_01.EPS

MAX

SHDN

GND

SENSE

SS COMP

LTC1504-3.3

SHUTDOWN



IMAX

68k



0.01µF



220pF



O.1µF

OUT

3.3V/500mA

C

IN

= AVX TPSC226M016R0375

C

OUT

= AVX TPSE107M016R0125

L

EXT

 = SUMIDA CD54-470

1504_02EPS



7.5k

IN

22µF

OUT

100µF

EXT

47µH

New LTC1504: Flexible, Efficient

Synchronous Switching Regulator Can

Source or Sink 500mA by Dave Dwelley

Introduction

The LTC1504 is a new addition to

LTC’s family of 8-pin step-down

switching regulators. It consists of a

200kHz ﬁxed frequency, voltage-feed-

back, buck-mode switching regulator

controller and a pair of 1.5Ω power

switches in an 8-pin SO package. The

LTC1504 also includes a synchro-

nous rectiﬁer on-chip, maximizing

efﬁciency and minimizing external

parts count while allowing the output

to both sink and source current: it

can source or sink up to 500mA with

input voltages from 3.3V to 10V and

output voltages as low as 1.26V. The

LT1504 can achieve 100% duty cycle

at the output switch, maximizing

dropout performance with low input-

to-output voltage differentials. The

LTC1504 includes an onboard

precision reference and user-

programmable current-limit and

soft-start circuits, allowing imp-

lementation of full-featured power

conversion circuits with a minimum

of external components. The archi-

tecture is optimized for maximum

efﬁciency at loads above 50mA and

does not include a light-load Burst

Mode™ circuit. This penalizes efﬁ-

ciency at very light loads but allows

the device to seamlessly shift between

sourcing and sinking current, opening

up a whole new class of applications.

A micropower shutdown mode is

included. The diminutive SO-8 pack-

age minimizes the amount of space

the LTC1504 ﬁlls while allowing

adequate thermal dissipation for

500mA load current levels. The

LTC1504 allows previously impos-

sible (or at least awkward) tasks to be

completed with ease.

Minimum

Component-Count Circuits

Figure 1 shows a fully functional

LTC1504 5V to 3.3V regulator,

including current limit and soft-start,

using the ﬁxed-output LTC1504-3.3

and only six external components.

The on-chip synchronous rectiﬁer

eliminates the need for an external

catch diode, and the internal cur-

rent-limit circuit requires only a single

resistor to program the maximum

output current. Efﬁciency is above

90% with load currents between 50mA

and 200mA, peaking at 92% at 100mA

and remaining above 82% all the way

to the maximum 500mA load. Cur-

rent limit is set at 500mA in this

example; it can be reduced by lower-

ing the value of R

IMAX

. C

sets the

start-up time at approximately 25ms.

Both the current limit resistor (R

IMAX

)

and soft-start capacitor (C

) are

optional; either can be deleted and its

pin left ﬂoating if the functionality is

not needed. An LTC1504 circuit can

be constructed with just four com-

ponents: input and output capacitors,

an external inductor and a compen-

sation capacitor.

The circuit in Figure 1 relies on the

ESR of the output capacitor to main-

tain loop stability with just a single

capacitor at the COMP pin. Figure 1

uses a surface mount electrolytic

capacitor with about 400mΩ ESR. A

low ESR tantalum output capacitor

can improve the transient response at

the output but requires a more com-

plex compensation network at the

Figure 1. Minimum parts count 5V–3.3V converter Figure 2. Improved transient response

Linear Technology Magazine • November 1997

DESIGN FEATURES

IMAX SHDN

SS COMP

VCC

GND

10µF

CERAMIC

TERMPWR

7.5k

0.01µF

220pF

15k

12k

COUT = AVX TPSC107M006R0150

LEXT = SUMIDA CD54-470 1540_03.EPS

LTC1504

110Ω

110Ω18

TO

27

LINES

IMAX SHDN

VCC

GND

SENSE

SS COMP

LTC1504

COUT

100µF

LEXT 47µH

Figure 3. SCSI-2 active terminator Figure 4. 5V supply splitter

MAX

SHDN

SS COMP

GND

10µF

CERAMIC

7.5k

0.01µF

220pF

11.8k

12.1k

OUT

= TAJC476M016R

EXT

= SUMIDA CDRH73-470 (LOWER RIPPLE/HIGHER EFFICIENCY)

 CDRH73-220 (FASTER TRANSIENT RESPONSE)

1540_04.EPS

LTC1504

MAX

SHDN

GND

SENSE

SS COMP

LTC1504

SPLIT SUPPLY

2.5V ±500mA

OUT

47µF

EXT



47µH (22µH)*

COMP pin (Figure 2). There is a

tradeoff to be made here: the mini-

mum component count solution is

the simplest and uses the least

expensive components but pays a pen-

alty in transient response. The low

ESR circuit in Figure 2 has improved

transient response and actually uses

less board space: the tantalum out-

put capacitor is smaller than the

electrolytic device used in Figure 1

and the additional compensation com-

ponents are tiny 0603 surface mount

devices. Unfortunately, the tantalum

capacitor used in Figure 2 costs sig-

niﬁcantly more than the electrolytic

device in Figure 1. You don’t get some-

thing for nothing.

Note that the input bypass capaci-

tor in both Figures 1 and 2 is an AVX

TPS type, a relatively costly surge-

tested tantalum capacitor. This is a

small, surface mount device that has

a surge current rating adequate to

support the 500mA maximum load

current of the LTC1504. Buck regula-

tors (like the LTC1504) inherently

draw large RMS currents from the

input bypass capacitor, and the

capacitor type chosen must be capable

of withstanding this current without

overheating. Different input bypass

capacitors can be used if their ripple

current rating is sufﬁcient to with-

stand the expected demand (as a rule

of thumb, one-half the maximum ex-

pected output current). Just as you

wouldn’t specify a 1/4W resistor to

dissipate 1W of power, neither should

you specify an inadequate input by-

pass capacitor for a buck regulator.

Tiny, expensive 10µF ceramic capaci-

tors work well, as do larger, cheaper

electrolytic capacitors with speciﬁed

ripple-current ratings. Circuits using

electrolytic input bypass devices of-

ten require an additional 0.1µF

ceramic capacitor right next to the

LTC1504 to keep the input capacitor

ESR under control. As with all switch-

ing regulator circuits, layout is critical

to obtaining maximum performance;

if in doubt, contact the LTC Applica-

tions Department for component

selection and layout advice.

Sink/Source Capability

Improves SCSI Terminators

and Supply Splitters

Figure 3 shows an adjustable-output

LTC1504 connected as a 2.85V regu-

lator for use as a SCSI terminator.

The ability of the LTC1504 circuit to

sink current makes it ideal for use in

terminator applications, where the

load is just as likely to be putting

current into the regulator as taking it

out. The synchronous-buck architec-

ture of the LTC1504 allows it to shift

cleanly between sourcing and sink-

ing current, making it ideal for such

applications. The high efﬁciency mini-

mizes the power drawn from the

TERMPWR supply and minimizes the

heat generated by the terminator

circuit. Finally, the small number of

tiny external components required

minimizes the space used by the ter-

minator circuit. As in Figure 2, a low

ESR output capacitor is used along

with an optimized compensation net-

work to improve output transient

response and maintain maximum

data ﬁdelity.

Substituting a different set of feed-

back resistors (Figure 4) creates a 5V

supply splitter, which creates a 2.5V

“ground” to allow analog circuitry to

operate from split supplies. Op amp

circuits and data converters like to

operate from dual supplies, and the

sink/source capability of the LTC1504

allows load currents to be returned

directly to the 2.5V “ground” supply.

The fast transient response of the

LTC1504 keeps the generated

“ground” voltage clean even as the

load switches rapidly between sinking

and sourcing current. Additionally,

the circuit in Figure 4 is signiﬁcantly

more efﬁcient than the traditional

resistor-divider method of supply

splitting at higher power levels.

Conclusion

The LTC1504 brings the beneﬁts of

synchronous rectiﬁcation to LTC’s

family of 8-pin switching regulators.

It provides a full-featured, high

efﬁciency, 500mA supply in a mini-

mum-size, minimum-parts-count

implementation. The LTC1504 is a

natural ﬁt in terminator circuits, sup-

ply-splitters and other applications

that require the output to both source

and sink current. Other uses include

distributed power, microprocessor

secondary-voltage supplies, low volt-

age, battery powered devices and

anywhere else a high efﬁciency, low

parts count, 500mA step-down sup-

ply is required.

Linear Technology Magazine • November 1997

DESIGN FEATURES

Low Dropout Regulator Driver Handles

Fast Load Transients and Operates on

Single 3V–10V Input by Lenny Hsiu

Introduction

The LT1573 is the latest low dropout

voltage regulator driver from Linear

Technology. The LT1573 is designed

to provide a low cost solution to appli-

cations requiring high current, low

dropout and fast transient response.

When combined with an external PNP

power transistor, this device provides

up to 5A of load current with dropout

voltages as low as 0.35V. The LT1573’s

circuitry is designed for extremely

fast transient response. This greatly

reduces the bulk storage capacitance

required when the regulator is used

in applications with fast, high cur-

rent load transients.

To keep cost and complexity low,

the LT1573 uses a new, time-delayed

latching current protection technique

that requires no external current

sense resistor. Base-drive current to

the external PNP is limited for instan-

taneous protection and a time-delayed

latch protects the regulator from con-

tinuous short circuits. The latch

time-out period can be varied by an

external capacitor. Guaranteed mini-

mum available base-drive current to

the external PNP is 250mA. An exter-

nal resistor can be added to reduce

the available base-drive current and

thereby limit the regulator output

current. The LT1573 is equipped with

an active-high shutdown and a

thermal shutdown function. The shut-

down function can be used to reset

the overcurrent latch. The thermal

shutdown function can be used to

protect the PNP power transistor if it

is thermally coupled to the LT1573.

The LT1573 is available as an

adjustable regulator with an output

range of 1.27V to 6.8V and with ﬁxed

output voltages of 2.5V, 2.8V and

3.3V. Output accuracy is better than

1% to meet the critical regulation

requirement of fast microprocessors.

A special 8-pin fused-lead surface

mount package is used to minimize

regulator footprint and provide

adequate heat sinking.

Basic Regulator Circuit

The basic regulator circuit is shown

in Figure 1. The adjustable-output

LT1573 senses the regulator output

voltage from its feedback pin via the

output voltage divider and drives the

base of the external PNP transistor to

maintain the regulator output at the

speciﬁed value. For ﬁxed-output ver-

sions of the LT1573, the regulator

output voltage is sensed from the

feedback pin via an internal voltage

divider. The resistor R

is required for

the overcurrent latch-off function. R

is also used to limit the drive current

available to the external PNP transis-

tor and to limit the power dissipation

in the LT1573. Limiting the drive

current to the external PNP transistor

will limit the output current of the

regulator, thereby minimizing the

stress on the regulator circuit under

overload conditions.

External PNP Transistor

Selection Criteria

The selection of an appropriate exter-

nal PNP transistor depends on the

regulator application speciﬁcations.

The critical PNP transistor selection

criteria include:

1. The maximum output current of

the PNP transistor

2. The dropout voltage at the

maximum output current

3. The gain-bandwidth product, f

of the transistor

The PNP transistor must be able to

supply the speciﬁed maximum regu-

lator output current to be qualiﬁed

for the regulator application. The V

saturation voltage of the transistor at

the maximum output current deter-

mines the dropout voltage of the

circuit. The dropout voltage deter-

mines the minimum regulator input

voltage for a certain speciﬁed output

voltage. The gain-bandwidth prod-

uct, f

, of the transistor determines

how fast the voltage regulator can

follow an output load change without

losing voltage regulation.

The D45H11 from Motorola and

the KSE45H11TU from Samsung can

Figure 1. 3.3V, 5A microprocessor supply

LOAD

R1

1.6k

R2

OUT1

OUT

MOTOROLA

D45H11



50Ω

D

24Ω

100pF

TIME

***

IN

4.5V–5.5V

OUT

= 1.265 • (1 + R1/R2)



100µF

TANT

OUT2

GND

LT1573

DRIVE

OUT



COMP

SHDN

†



LATCH

*

**

***

†

FOR T <45˚C, C

OUT1

= 24 × 1µF Y5V CERAMIC SURFACE MOUNT CAPACITORS

FOR T >45˚C, C

OUT1

= 24 × 1µF X7R CERAMIC SURFACE MOUNT CAPACITORS

PLACE C

OUT1

IN THE MICROPROCESSOR SOCKET CAVITY

OUT2

= 220µF CHIP TANTALUM

TIME

= 0.5µF FOR 100ms LATCH-OFF TIME AT ROOM TEMPERATURE

SHDN (ACTIVE HIGH) SHOULD BE TIED TO GROUND IF NOT USED

Linear Technology Magazine • November 1997

DESIGN FEATURES

be used in all LT1573 regulator cir-

cuits with current ratings up to 5A.

The D45H11 can supply 5A of output

current with dropout voltage as low

as 0.35V. The gain-bandwidth prod-

uct, f

, of the D45H11 is typically

40MHz; this enables a regulator com-

posed of this PNP transistor and an

LT1573 to handle load changes of

several amps in a few hundred nano-

seconds with a minimum amount of

output capacitance.

Dropout Voltage

The dropout voltage of an LT1573-

based regulator circuit is determined

by the V

saturation voltage of the

discrete external PNP transistor when

it is driven with a base current equal

to the available drive current of the

LT1573. The LT1573 is guaranteed to

sink 250mA of base current (440mA

typical). The available drive current of

the LT1573 can be reduced by adding

a resistor, R

in Figure 1, in series

with the drive pin. Table 1 lists some

useful operating points for the

D45H11 from Motorola. These points

were empirically determined using a

sample of devices.

Selecting R

Resistor R

can be used to limit the

available drive current to the external

PNP transistor. In order to select R

the user should ﬁrst choose the value

of drive current that will give the

required value of output current and

dropout voltage. For circuits using a

D45H11 as a pass transistor, this can

be done using Table 1. For circuits

using transistors other than the

D45H11, the user must characterize

the transistor to determine the drive-

current requirements for the speciﬁed

output current and dropout voltage.

In general, it is recommended that

the user choose the lowest value of

drive current that will satisfy the out-

put current requirements. This will

minimize the stress on circuit compo-

nents during overload conditions.

The value of R

can be calculated

with the following formula:

= ( V

– V

DRIVE

)/(I

DRIVE

+ I

)

where:

= the minimum input voltage to

the circuit

= the maximum emitter/base

voltage of the PNP pass transistor

DRIVE

= the minimum PNP base

current required

B = the current through R

= V

DRIVE

= the Drive pin saturation

voltage when the drive pin

current equals (I

DRIVE

+ I

Overcurrent Latch-Off

In addition to limiting the base-drive

current, R

is included in the circuit

for the overcurrent protection latch-

off function. There is a minimum

value for this resistance. It is calcu-

lated by the formula above, with the

drive current I

DRIVE

set to the guaran-

teed minimum available drive current

(= 250mA) from LT1573. Under some

conditions, R

can be replaced with a

short. This is possible in circuits where

an overload is unlikely and the input

voltage and drive requirements are

low. If R

is not included in the circuit,

the regulator is protected against over-

current conditions only by the

thermal-shutdown function. After the

value of R

is determined, a certain

amount of base-drive current is

available to the external PNP. An

overcurrent or output short-circuit

condition will demand a base-drive

current greater than the LT1573 can

supply. The internal drive transistor

will saturate. This overcurrent condi-

tion will trigger a time-out latch to

turn off the regulator system. The

time-out period is determined by an

external capacitor connected between

the LATCH and GND pins. The time-

out period is equal to the time it takes

for the capacitor to charge from 0V to

the latch threshold. If the overcurrent

or output short condition persists

longer than the time-out period, the

regulator will be shut down. Other-

wise, the regulator will function

normally. The latch can be reset by

recycling input power, by grounding

the latch pin or by putting the device

into shutdown.

Shutdown Function

The regulator can be shut down and

the overcurrent latch can be reset at

the same time by pulling the SHDN

pin voltage higher than the shutdown

threshold (about 1.3V). The regulator

will restart itself if the SHDN pin is

pulled below the shutdown thresh-

old. The shutdown pin voltage can be

higher than the input voltage. When

the shutdown pin voltage is higher

than 2V, the shutdown pin current

increases and is limited by a 20k

resistor.

Compensation

Figure 1 shows a microprocessor

power supply based on the LT1573

and the D45H11 PNP transistor, us-

ing twenty-four 1µF surface mount

ceramic capacitors in parallel with

one 220µF surface mount tantalum

capacitor at the output. To improve

the transient response to regulator

output-load variation of this circuit, a

capacitor in series with a resistor can

be inserted between the OUTPUT and

COMP pins; a 100pF capacitor in

series with a 1k resistor is

recommended. In theory, the output

capacitor forms the a pole in the

regulator system. An internal com-

pensation capacitor forms another

pole. The external compensation ca-

evirD tnerruC tuptuO tnerruC

Tlacipy

tuporD

egatloV

Am02A1V02.0

Am02A2V05.0

Am04A2V52.0

Am04A3V05.0

Am06A3V52.0

Am06A4V07.0

Am08A4V54.0

Am001A4V53.0

Am001A5V07.0

Am051A5V04.0

Am002A5V53.0

Am051A6V56.0

Am002A6V54.0

Am052A7V05.0

Table 1. Dropout voltage of the D45H11

continued on page 24

Linear Technology Magazine • November 1997

DESIGN FEATURES

Introduction

Many products, such as critical data

acquisition systems and process con-

trollers, need to operate continuously

even when primary power is inter-

rupted. Primary power for these

systems may come from a wall adapter,

a battery-charging unit powered from

the line or a removable battery. For a

line-powered circuit, a backup bat-

tery may provide reserve power until

line power is restored. In the case of a

battery-charging unit, the battery

automatically provides power during

a line outage. With removable batter-

ies, a secondary backup battery is

needed to maintain power when the

main battery is discharged or removed.

These circuits have several com-

mon requirements, regardless of the

power source. Most important is the

need for the regulated voltage to be

provided to critical circuitry at all

times. The switch from one input

source to another should be seam-

less, without signiﬁcant change in

the output. The input source that

provides power to the circuit ﬁrst can

also be important (for example, the

main battery should provide power

before the backup battery, regardless

of which has the higher terminal volt-

age). Additionally, some form of power

monitoring circuitry may be needed

to help in power management.

Smart Regulator

Makes the Grade

Designed for a multitude of applica-

tions, the LT1579 is a dual input,

single output, low dropout regulator

that provides an uninterruptible out-

put voltage from two independent

input voltage sources on a priority

basis. All power supplied to the load

is drawn from the primary input (V

IN1

)

until the device senses that the pri-

mary source is failing. At this point,

the LT1579 smoothly switches from

the primary input to the secondary

input (V

IN2

) to maintain output regu-

LT1579 Battery-Backup Regulator

Provides Uninterruptible Power

by Todd Owen

lation. The LT1579 is capable of pro-

viding 300mA from either input at a

dropout voltage of 0.4V. Total quies-

cent current is 50µA: 45µA from the

primary input source, 2µA from the

secondary input source, and an addi-

tional 3µA from the higher voltage of

the two.

A block diagram of the LT1579 is

shown in Figure 1. A single error

ampliﬁer controls both output stages

so regulation remains tight regard-

less of which input is providing power.

Threshold levels for the error ampli-

ﬁer and low-battery detectors are set

by the internal 1.5V reference. Out-

put voltage is set by an internal

resistor divider for ﬁxed voltage parts

and by an external divider for adjust-

able parts. Internal bias circuitry

powers the reference, error ampliﬁer,

output driver controls, logic ﬂags and

low-battery comparators.

The LT1579 is designed to main-

tain regulation even if one of the

inputs is instantaneously removed. If

the primary input is supplying load

current, removal and insertion of the

secondary input creates no notice-

able transient at the output. In this

case, the LT1579 continues to supply

current from the primary; no switch-

ing is required. If the primary input

source is unplugged while it is sup-

plying load current, the LT1579 must

quickly switch to the secondary

source. In this case, the LT1579 sees

the input capacitor as a rapidly dis-

charging battery. If it discharges too

quickly, the LT1579 does not have

sufﬁcient time to switch over without

a large transient occurring at the

–

BIASCOMP

1579_01.eps

OUTPUT

DRIVER

CONTROL

DROPOUT

DETECT

BIAS CURRENT

CONTROL

1.5V REFERENCE

E/A

LB1

ADJ

BACKUP

DROPOUT

LB01

LB02

VOUT

VIN1

VIN2

LBI1

LBI2

WARNING

FLAGS

LB2

SHDN

INTERNAL

RESISTOR DIVIDER

FOR FIXED-VOLTAGE

DEVICES ONLY

Figure 1. LT1579 block diagram

Linear Technology Magazine • November 1997

DESIGN FEATURES

output. The input capacitor must be

large enough to supply load current

during the transition from primary to

secondary input. Plugging the pri-

mary input back in creates a smaller

transient on the output because both

inputs are present to supply current

during the transition. Figure 2 shows

a typical output transient using 10µF

input and output capacitors and a

100mA load. Peak output deviation is

less than 50mV. Proportionally larger

values for input and output capaci-

tors are needed to limit peak deviations

on the output when delivering larger

load currents.

The LT1579 incorporates two

independent low-battery comparators

and two status ﬂags that provide

information to power management cir-

cuitry. The status ﬂags show which

input is supplying power to the load

and provide an early warning against

the loss of output regulation. A Sec-

ondary Select pin can be used to force

the switchover from the primary input

to the secondary. This active low logic

pin reduces power draw from the pri-

mary input to only 2µA. This provides

excellent protection for batteries that

are sensitive to deep-discharge condi-

tions, such as Li-Ion cells. The part

can be put into a low power shutdown

state where all bias currents and device

functions are turned off and all logic

ﬂags are high impedance. In shut-

down, the quiescent current drops to

a total of only 7µA: 2µA from the

primary input, 2µA from the second-

ary input and 3µA from the higher of

the two input voltages.

Internal protection circuitry in the

LT1579 guards against a number of

fault conditions. Protection circuitry

1579_03.eps

5V

300mA

R1

2.7M 100k 100k 100k 100k

1µF4.7µF

0.01µF

R2

R3

2.7M

1µF6V

R4

TO POWER MANAGEMENT

LT1579-5

GND

IN1

LBI1

IN2

LBI2

OUT

LBO1

LBO2

BACKUP

DROPOUT

SHDN

BIASCOMP

4.8V

100mA

IN1

IN2

IN1

100mA

IN2

LB01

BACKUP

LB02

DROPOUT

OUT

BCDE

1579_04.eps

Figure 2. Output deviation during removal

and replacement of primary input source Figure 3. LT1579 basic application

Figure 4. Basic application timing diagram

NPUT #1: 6V

OUTPUT

50mV/DIV

REMOVE

INPUT

REPLACE

INPUT

includes thermal-overload protection,

reverse-battery protection, input-

to-input current protection, reverse-

current protection and overcurrent

protection. If maximum junction tem-

perature is exceeded, the LT1579 will

shut off to prevent internal damage. If

either input is reversed, reverse cur-

rent ﬂow will be limited to less than

1mA. Inputs are isolated from one

another. No current will ﬂow between

inputs, regardless of their relative

voltages. If the output is held high

while the inputs are grounded, reverse

currents are limited to 7µA. Current-

limit protection is designed to protect

the device if the output is shorted to

ground. With the output shorted to

ground, current will be drawn from

the primary input until it is dis-

charged. No current is drawn from

the secondary input until the primary

input is discharged.

Circuit Examples

The basic application of the LT1579 is

shown in Figure 3. It uses two inde-

pendent voltage sources for the inputs.

These voltage sources may be batter-

ies, wall adapters or any other DC

source. The low-battery comparators

are conﬁgured to give a low output if

either input voltage drops below 5.5V.

The trip points can be adjusted by

changing the values of the divider

resistors (R1 and R2 for LB1, R3 and

R4 for LB2). All logic outputs (LBO1,

LBO2, BACKUP and DROPOUT) are

open-collector outputs that require

an external pull-up resistor. They are

capable of sinking 20µA at a maxi-

mum output voltage of 0.32V, which

is useful for driving both CMOS and

TTL logic families. For driving LED’s,

all logic outputs can sink 5mA at a

maximum output voltage of 1.2V.

Figure 4 is the timing diagram for

the basic circuit. No time scale is

shown for the timing diagram be-

cause actual discharge rates are a

function of the load current and the

type of batteries used. The timing

diagram is meant as a tool to help in

understanding the LT1579’s basic

operation.

Five milestones are noted on the

timing diagram. Time A is where the

primary input voltage drops enough

to trip the low-battery detector, LB1.

Linear Technology Magazine • November 1997

DESIGN FEATURES

The trip threshold for LB1 is set at

5.5V, slightly above the dropout volt-

age of the primary input. At time B,

the BACKUP ﬂag goes low, signaling

the beginning of the transition from

the primary source to the secondary

source. Between times B and C, the

input current makes a smooth transi-

tion from V

IN1

to V

IN2

. By time C, the

primary battery has exhausted most

of its useful charge. The primary input

will still deliver a small amount of

current to the load, diminishing as

the primary input voltage drops. By

time D, the secondary battery has

dropped to a low enough voltage to

trip the second low-battery detector,

LB2. The trip threshold for LB2 is also

set at 5.5V, slightly above where the

secondary input reaches dropout. At

time E, both inputs are low enough to

cause the LT1579 to enter dropout,

with the DROPOUT ﬂag signaling the

impending loss of output regulation.

Some interesting things can be

noted on the timing diagram. The

amount of current available from a

given input is determined by the

input/output voltage differential. As

the primary voltage drops, the amount

of current drawn from the input also

drops, slowing discharge of the bat-

BIASCOMP

1579_05.eps

LT1579-5

GND

IN1 VOUT

5V/300mA

LBI1

LBO2

SHDN

LBO1

R2

2.7M

R9

1.5M

RESET

C1

1µF

C2

1µF

C3

4.7µF

C4

0.01µF

R1

R3

R4

10M

D1D2

R10

MAIN GOOD

IN1

IN2

LBI2

R6

2.7M

D4

5.1V

1N751A

D1–D3 = 1N4148

C5

0.1µFVCC

GND

R5

R8

330k

R7

IN2

OUT

BACKUP

1/4

74C02 1/4

74C02

1/4

74C02

DROPOUT NC

Figure 5. Added SR latch shuts the LT1579 off when both low-battery detectors are tripped.

tery. Dropout-detection circuitry will

maintain the maximum current draw

from the input for the given input/

output voltage differential, based on

the impedance of the pass transistor.

In the case shown, this causes the

current drawn from the primary to

approach zero, although it never

reaches that point. Note that the pri-

mary begins to supply signiﬁcant

current again when the secondary

input drops low enough to cause a

loss in output regulation. This occurs

because the input/output voltage dif-

ferential of the primary input increases

as the output voltage drops. The

LT1579 will automatically maximize

the power drawn from the inputs to

maintain the highest possible output

voltage.

A ﬁnal circuit example is shown in

Figure 5. This circuit has a few notable

changes from the basic application.

First, the Secondary Select pin is con-

nected directly to LBO1. When the

primary input voltage drops below

the threshold level for LB1, the com-

parator output will pull the Secondary

Select pin low. This forces the device

to switch completely over to the sec-

ondary input, limiting the discharge

voltage of the cells. Second, the logic

gates used form an SR latch. When

both batteries are below the threshold

level for their respective compara-

tors, the latch will be set, forcing the

part into shutdown. The latch is reset

by pulling up on the RESET node,

allowing the part to come out of

shutdown.

The series resistance of a battery

can cause its terminal voltage to rise

as its current decreases. This effect

can reset the low-battery detector and

cause the LT1579 to oscillate between

the primary and secondary inputs. To

combat this, the low-battery com-

parators have up to 18mV of built-in

hysteresis at the input to the com-

parator (LBI1, LBI2). The hysteresis

is determined by the amount of load

current on the comparator output. At

no load, the comparator hysteresis is

zero, increasing to a maximum of

18mV for load currents above 20µA.

For the pull-up resistor shown, load

current on the output of the com-

parator is 5µA, so hysteresis will be

5mV. With the values shown for resis-

tor divider R2/R3, this translates to

19mV of hysteresis at the primary

input of the LT1579. Additional hys-

teresis can be added by connecting

continued on page 24

Linear Technology Magazine • November 1997

DESIGN FEATURES

High Efficiency Distributed

Power Converter Features

Synchronous Rectification

Introduction

The advent of the switching regulator

has greatly reduced the size, weight

and volume of power conversion cir-

cuitry, while improving both the speed

of response and efﬁciency. With the

output voltage requirements going

ever lower and currents ever higher,

close scrutiny is applied to the loss

mechanisms of the power converter.

The loss mechanisms are caused by

the circuit resistances (input capaci-

tor ESR, power switch on-resistance,

DC and AC resistance of the inductor,

resistance of any current-sense ele-

ments, resistance in the output diode

and ESR in the output capacitors)

multiplied by the squares of their

respective currents, and by ﬁxed for-

ward-voltage losses in the output

diode, multiplied by the diode cur-

rent. As output current goes up, the

ﬁrst thing to do is minimize all losses

caused by resistance (because the

power is proportional to I

R). It is easy

to minimize resistance because we

have available very low ESR capa-

citors, low on-resistance MOSFETs

and low series-resistance inductors.

We have controllers that place a very

small voltage across the current sense

resistors. Because we do such a good

job of dropping the resistive loss

mechanisms, the output diode for-

ward-voltage drop becomes the

greatest loss mechanism. This is how

the mandate for synchronous rectiﬁ-

cation comes about. Synchronous

rectiﬁcation is achieved by replacing

the output diode with a low on-resis-

tance switch. With synchronous

rectiﬁcation, efﬁciencies are higher,

and, more importantly, power dissi-

pated in the switching power supply

is lower, often eliminating the need

for heat sinks and/or fans.

Introducing the LT1339

The LT1339 is the buck/boost con-

verter that needs no steroids. As a

full-featured switching controller, the

LT1339 incorporates the features

needed for system-level solutions. The

unfortunate lack of such features in

most PWM converters forces design-

ers to grope for handfuls of jellybean

components. The LT1339 has an

innovative slope-compensation func-

tion that allows the circuit designer

freedom in controlling both the slope

and offset of the slope-compensation

ramp. Additionally, the LT1339 has

an average current limit loop that

yields a constant output current limit,

regardless of input and/or output

voltage. The LT1339’s RUN pin is

actually the input to a precision com-

parator, giving the designer freedom

to select an undervoltage lockout point

and hysteresis appropriate for the

design. The SYNC and SS (soft-start)

pins allow simple solutions to sys-

tem-level design considerations. Like

all Linear Technology controllers, the

LT1339 has anti-shoot-through cir-

cuitry that ensures the robustness

that is demanded in real-world appli-

cations for medium and high power

conversion.

For input voltages ranging from

12V to 48V and output voltages rang-

ing from 1.3V to 36V, the LT1339 is a

simple, robust solution to your power-

conversion problems. The LT1339 is

ideal for power levels ranging from

tens of watts to tens of kilowatts. The

LT1339 is straightforward and

remarkably easy to use. This is one

power converter that’s not afraid of

20A, 50A or even 150A of load current.

Distributed Power

Figure 1 details a typical low voltage

buck converter. This circuit has a V

range of 10V to 18V with conﬁgurable

output current and voltage. This

simple circuit delivers 250W of load

power into a 5V load while maintain-

ing efﬁciencies in the mid-nineties.

Higher Input Voltages

The circuit shown in Figure 1 is lim-

ited to 20V because of the maximum

rating (Abs Max) of the LT1339 V

pin. The input voltage can be extended

above 20V by inserting a 10V Zener

diode where the asterisk (*) is shown

in Figure 1. This will extend the input

voltage of Figure 1’s circuit up to 30V

(the Abs Max rating of the MOSFETs).

1 8

C2

0.1µF

C1

1µF

REF



*SEE TEXT

1339_01.EPS

SYNC

AVG



REF

SENSE



SENSE

–



TG

TS

13

2

4

5

7

19

18

17 20

11

12

RUN

REF



SLOPE

SGND

12V

BOOST

D1 1N914

LT1339

PGND

COMP



2200pF

COMP



4.7k

R1

100k

10V TO

18V



15k

AVG

2200pF

1500pF

C3

1µF

OUTPUT



2200µF

6.3V

OS-CON

5V

50A

Q1

IRL3803

Q2–Q5

IRL3103D2

×4

L1

10µH

50A



SENSE



0.002Ω

D3

1N5817

D2

1N5817

INPUT



1000µF

16V ×2

OS-CON



3K

1.66K

1.25K

450Ω

40Ω

OUT



5V

3.3V

2.8V

1.8V

1.3V

SENSE



0.01Ω

0.005Ω

0.002Ω

LIMIT



10A

20A

50A

Figure 1. 10V–18V in, 5V/50A out buck converter

by Dale Eagar

Linear Technology Magazine • November 1997

DESIGN FEATURES

Blame it

on the Physicists

As the input voltage approaches 30V,

the bottom MOSFETs will begin to

exhibit “phantom turn-on.” This

phenomenon is driven by the instan-

taneous voltage step on the drain, the

ratio of C

MILLER

to C

INPUT

, and yields

localized gate voltages above V

, the

threshold voltage of the bottom MOS-

FET. To defeat the physicists, we add

3.3V of negative offset to the bottom

gate drive, effectively making the

threshold of the bottom MOSFETs

3.3V harder to reach (see Figure 2).

This offset is provided by the 3.3V

Zener, 1µF capacitor, 10k resistor

and the 1N914 diode preceding the

gate of the bottom MOSFETs.

1 8

C3

0.1µF

C1

1µF

R1

10k

D3 1N914

1339_02.EPS

SYNC

AVG



REF

SENSE



SENSE

–



TG

TS

13

2

4

5

7

19

18

11

12

RUN

REF



SLOPE

SGND

12V

BOOST

D1

1N914 48V

LT1339

PGND

C2

2200pF

COMP



4.7k

R1

100k

12V

(SUPPLIED SEPARATELY)



15k

AVG

2200pF



1500pF

C3

1µF

C4

1µF

OUT



2200µF 

6.3V

OS-CON

×4

5V

50A

Q1–Q2

IRFZ44

×2

Q3–Q6

IRFZ44

×4

L1

10µH 50A

FB

SENSE



0.002Ω

Q7

FMMT619

Q9

FMMT619

Q10

FMMT720

Q8

FMMT720

REF



C5

47µF

17 20

INPUT



1500µF

63V

×4



18.2K

8.66K

3K

1.66k

1.25k

OUT



24V

12V

5V

3.3V

2.8V

SENSE



0.01Ω

0.005Ω

0.002Ω

LIMIT



10A

20A

50A

12V D2 3.3V

D4–D14

3A ×10



15V–25V



1000µF 35V

×2

R13

D1

12V

Q7 FZT849

C10

0.1µF

C11

22µF

35V

C1

0.1µF

R1

100k



47k

R2

R3

560Ω



230pF C

AVG



2.2nF

R4

33k

C4

220pF

U3

CNY17-3

SENSE



0.02Ω

1/2W

D2

IN4148

C3

1µF

12V

D4

MURS

120

D3 

MURS120

Q3 Si4450

Q4 Si4450

C3

3.3µFR10

10Ω

R11

10Ω

R9

10Ω

R14 10Ω

L1 7µH

Q6



T2**

T1*

R8

R12 1k

D6 1N914

D5

1N914

C7 1µF

C5

1µF

C9 0.1µF

C6

1µF

R5

3.3Ω

U2

LT1431

COLL REF

GND-S GND-F

V+



2.49k

REF



2.49k

R7 1k

R6 560Ω

OUT



220µF 10V

OS-CON

OUT



5V/6A

11

12

20

19

SENSE



SENSE

–



BOOST

TG

16



BG



2



4



3

6



5

9

14

REF



SLOPE



CT

SS



AVG



FB

PHASE

REF

SYNC SGND PGND

U1

LT1339

1815

RUN 12V

13 17

ISOLATION

BARRIER

PRIMARY

GROUND SECONDARY

GROUND

Q1, Q2 = SUD50N03

L1 = 15 TURNS AWG20 77130-A7

T1 = POWER TRANSFORMER

T2 = GATE-DRIVE TRANSFORMER

(SEE FIGURE 4 FOR DETAILS)

*

Si4539DY

Figure 2. 48V in, 5V/50A out, high power buck converter

Figure 3. Galvanically isolated synchronous forward converter (See Figure 4 for details of T1 and T2.)

Linear Technology Magazine • November 1997

DESIGN FEATURES

The Synchronous

Forward Converter

The forward converter can be thought

of as a buck converter with a

transformer ratio. The transformer

performs one or more of the following

functions:

1. Increases the duty factor in

converters with high transforma-

tion ratios, improving efﬁciency

2. Provides Galvanic isolation from

input to output

3. Allows the converter to operate in

the buck-boost mode, where the

input voltage can be either

higher or lower than the output

voltage

Figure 3 details a Galvanically iso-

lated LT1339 synchronous forward

converter. Operating at its rated load

of 6V at 5A, this circuit achieves 87%

efﬁciency with a 15V input and 85%

efﬁciency with a 24V input. Figure 4

shows details of the transformers used

in Figure 3’s circuit.

The Synchronous

Boost Converter

The LT1339 becomes a synchronous

boost controller when the PHASE pin

is grounded. Figure 5 details a 250W

boost converter that outputs 28V at

9A from a 5V supply.

Conclusion

The LT1339, like the many other

members of the Linear Technology

power conversion family, is full fea-

tured and well suited to the power

conversion needs found in the real

world. This robust workhorse is ﬁnd-

ing itself designed into countless

applications, including power distri-

bution, telecom, automotive and even

military applications.

SECONDARY, 9 TURNS TRIFILAR 26AWG

PRIMARY, 9 TURNS TRIFILAR 26AWG

2MIL

POLYESTER

FILM

1500VDC ISOLATION

TUCK TAPE ENDS

T1: PHILIPS EFD20-3F3 CORE

Lp = 93µH, Al = 1150nH/T

(NO GAP)

T2: COILTRONIX VP1-1400 (500V ISOLATION)

1339 04 .eps

BOOST 12V

Q1–Q2

IRF3205 ×2

Q3–Q6

IRF3205 ×4

Q8 FMMT619

Q7 FMMT720

C3

1µF

C4

1µF

Q10

FMMT720

Q9

FMMT619

L1

40µH

SENSE



0.002Ω

INPUT



220µF 6.3V

×4

IN

5V/60A

R2 100Ω

R3 100Ω

12V

12V (SUPPLIED SEPARATELY)

D1

1N914

R1

100k

C7

47µF

16V

OUT



2200µF

35V ×6



27k R

REF



1.2k



10k

C1

1µF



2200pF

COMP



7.5k

AVG



2.2nF

COMP



1.5nF



10µF

C6

0.1µF

SENSE

+

SENSE

–

REF



PHASE

CT

C

AVG



REF



SLOPE

RUN

U1

LT1339

SYNC PGND SGND

12

11

10

3

7

5

9



2

20 17

115 8

OUT



28V/8.5A

L1 = 12T 4× AWG12 ON 77437-A7

C5

10pF

Figure 5. This 5V to 28V synchronous boost converter limits input current at 60A (DC).

Figure 4. Transformer details of Figure 3’s circuit

Authors can be contacted

at (408) 432-1900

Linear Technology Magazine • November 1997

DESIGN FEATURES

D1 and R4. The values shown will give

an additional 200mV of hysteresis.

When LBO1 and LBO2 are high

impedance and either input is greater

than 6.5V, the logic-ﬂag voltages can

be above the maximum voltage rat-

ing. Internal clamps on the logic ﬂags

limit the output voltage to approxi-

mately 6.5V and the pull-up resistor

values shown will limit the current

into the logic ﬂags to less than the

maximum current rating.

Conclusion

The LT1579 can provide a continu-

ous regulated output voltage to critical

circuits from any of a number of dif-

ferent input sources. It will provide

up to 300mA of output current at a

dropout voltage of 0.4V. Should the

primary input fail, the device switches

seamlessly to the secondary input,

maintaining output regulation. A

single error ampliﬁer controls both

output stages so regulation remains

tight regardless of which input is pro-

viding power. The LT1579 can handle

instantaneous removal of either one

of its inputs without losing regula-

tion. System power management is

aided by two status ﬂags, which pro-

vide information about which input is

providing power and signal the loss of

output regulation. Two independent

low-battery comparators can be used

to monitor input voltages. Also, an

external pin can be used to force the

switch to the secondary input. Total

quiescent current of the LT1579 is

50µA, dropping to a mere 7µA in its

low power shutdown state. Internal

circuitry guards against a number of

fault conditions, including current

limit, thermal limit and reverse volt-

ages, protecting sensitive circuitry and

inputs. Whether the application is

simple or complex, the LT1579 is

truly a “smart” regulator.

LT1579, continued from page 20

pacitor and resistor form a zero that

adds phase margin to the regulator

system to prevent high frequency os-

cillation. The LT1573 has an internal

pole at approximately 5kHz. An ex-

ternal compensation zero between

10kHz and 100kHz is usually required

to stabilize the regulator. The zero

frequency is primarily determined by

the compensation capacitor and can

be roughly calculated by the following

equation:

ZERO

= (40kHz) 10pF ≤ C

COMP

≤ 100pF

30pF



COMP

(pF)

A compensation resistor between

1k and 10k is suggested. A compen-

sation resistor of 5k works for most

cases. In some cases, a larger value

compensation resistor in the sug-

gested range is needed to stop

oscillation above 1MHz. In some cases,

the output capacitor may have enough

equivalent series resistance (ESR) to

generate the required zero, and the

external compensation zero may not

be needed.

Output Capacitor

The LT1573 is designed to be used

with an external PNP transistor with

a high gain-bandwidth product, f

, to

make a regulator with very fast tran-

sient response; this can minimize the

size of the output capacitor. For a

regulator consisting of an LT1573

and a D45H11, a single 10µF surface

mount ceramic capacitor at the out-

put is enough to allow the regulator to

handle varying output loads of up to

5A in a few hundred nanoseconds

and to remain stable with a 30pF

external compensation capacitor in

series with a 7.5k resistor between

the OUTPUT and COMPENSATION

pins. If tighter voltage regulation is

needed, more capacitance can be

added to the regulator output. If more

capacitance is added to the output,

the bandwidth of the regulator will be

lowered. A larger compensation

capacitor may be needed to lower the

frequency of the compensation zero

to avoid high frequency oscillation.

Equal-value output capacitors with

different ESR can have different tran-

sient responses. High frequency

response will be strongly affected by

parasitics in the output capacitors

and board layout. Some experimen-

tation with the external compensation

will be required for optimum results.

Typical Applications

and Experimental

Transient Response

A voltage regulator made with an

LT1573 can be used as a micropro-

cessor power source, post regulator

for switching power supplies, high

efﬁciency, high accuracy linear regu-

lator or ultralow dropout regulator. A

typical application circuit for a micro-

processor power supply is shown in

Figure 1. Figure 2 shows the output

voltage transient of the circuit when

the output load varies between 0.2A

and 5A. About 80mV of transient

deviation from 3.3V output (2.5%)

can be observed.

Conclusion

The LT1573 provides a low cost solu-

tion to high current, fast transient,

low dropout voltage regulator appli-

cations such as fast microprocessor

power supplies. As a low dropout

regulator, the LT1573 can provide

load currents of up to 5A with drop-

out voltages as low as 0.35V. Because

of its fast transient capability, the

LT1573 cuts down the regulator cost

by requiring much less bulk storage

capacitance for tight voltage

regulation.

10µs/DIV

50mV/DIV

2.5A/DIV

Figure 2. Transient response for a 0.2A

to 5A step

LT1573, continued from page 17

Authors can be contacted

at (408) 432-1900

Linear Technology Magazine • November 1997

DESIGN IDEAS

Recently introduced precision

products from Linear Technology

allow designers to implement high

precision applications at super-

micropower levels. Among these

devices are the LT1496 quad preci-

sion input/rail-to-rail output op amp

and the LT1634 precision shunt volt-

age reference, which operate at only

1.5µA and 10µA, respectively. Even

at such low power levels, precision

performance is not compromised. The

LT1496 features 475µV maximum

input offset voltage and 1nA maxi-

mum input bias current. The LT1634

achieves 0.05% initial accuracy and

25ppm/°C maximum temperature

drift.

Figure 1 shows a single-cell Li-Ion

battery supervisory circuit. The build-

ing blocks of this circuit are the

LT1496 precision op amp and LT1634

voltage reference. The useful region

of operation of a single-cell Li-Ion

battery is between 4.2V and 3V. The

cell voltage drops fairly quickly below

3V. System operation below this volt-

age can be erratic. Although Li-Ion

battery use is becoming widespread,

it is costly to damage the battery. The

supervisory circuit protects the bat-

tery from overcharging and/or

overdraining and prevents the bat-

tery voltage from falling out of its

operating region. For instance, the

LT1496 operates down to 2.2V,

ensuring that circuit operation is

maintained when the battery voltage

falls below 3V.

The Li-Ion battery is monitored via

a voltage divider off the battery volt-

age (node A). The divided voltage is fed

into the positive inputs of compara-

tors A2 and A3 and compared to the

threshold voltages of 1.75V and 1.25V,

respectively. These voltages are

selected so that the minimum battery

charge voltage is 3V and the maxi-

mum is 4.2V. The LT1634 1.25V

reference is buffered by op amp A1.

The constant 1.25V across R2 creates

a 1µA constant current, so that the

output of A1 is ampliﬁed to 1.75V.

This output drives R

to provide con-

stant bias current for the LT1634.

Depending on the battery voltage,

the circuit is in one of the three states,

as shown in Table 1.

The voltage at node A is compared

to the two threshold voltages to deter-

mine the state of the circuit. For

instance, when node A reaches or

exceeds 1.75V (battery voltage reaches

4.2V), the outputs of A2 and A3 will

swing to the positive rail, terminating

the charger and connecting the load

to the battery. When node A falls

between 1.25V and 1.75V (battery

voltage between 3V and 4.2V), the

output of A2 swings low, turning the

charger on, while the output of A3

stays high, leaving the load connected.

When node A falls below 1.25V (bat-

tery voltage less than 3.0V), the output

of A2 stays low, keeping the charger

Single-Cell Li-Ion Battery Supervisor

by Albert Lee

–

A1

1/4 LT1496

A2

1/4 LT1496

A3

1/4 LT1496 A4

1/4 LT1496



175k

LT1634

1.250V

R1

500k

0.1%

R2

1.25M

0.1%

BAT



10M



10M

BATTERY

R3

1.75M

0.1%

R4

1.25M

0.1%

CHARGER

TO LOAD



1M

SW*

10µA

1µA

1.75V

1.25V

OFF

D1, D2 = 1N458

R1–R4 = CAR6 SERIES IRC (512) 992-7900

TP0610L for 50mA LOAD

Figure 1. Single-cell Li-Ion battery supervisory circuit

DESIGN IDEAS

Single-Cell Li-Ion Battery

Supervisor ...............................25

Albert Lee

800mA Li-Ion Battery Charger

Occupies Less Volume than

Two Stacked Quarters ............. 27

Fran Hoffart

Single-Supply

Random Code Generator .......... 29

Richard Markell

A Low Distortion, Low Power,

Single-Pair HDSL Driver Using

the LT1497 ............................. 30

George Feliz and Adolfo Garcia

100V, 2A Constant-Voltage/

Constant-Current Bench Supply

................................................31

Mitchell Lee

Switcher Generates Two Bias

Voltages without Transformer

................................................32

Jeff Witt

Linear Technology Magazine • November 1997

DESIGN IDEAS

on. The output of A3 will also swing

low, which, in turn, will cause the

output of A4 to go high, turning off

FET SW that disconnects the load

from the battery.

If node A were to bounce around at

either threshold voltage, the circuit

would bounce between states. To

avoid this problem, hysteresis is

added via the resistor and diode net-

works connected between the outputs

of A2 and A3 and their positive in-

puts. Figure 2 shows the behavior of

BAT

vs node A entering the trip points

with hysteresis. When V

BAT

rises to

4.2V (node A increases to 1.75V), op

amp A2’s output will switch from low

to high, causing current to ﬂow

through R

. The additional current

will raise node A by an amount

∆VA

HYS1

, which will clearly put the

circuit in state 3. The circuit will not

exit state 3 until V

BAT

falls to ∆V

HYS1

(310mV for the circuit shown) below

4.2V, which will cause node A to fall

back to the upper trip point of 1.75V

(point 1 of Figure 2). Similarly, when

BAT

drops below 3V (node A falls

below 1.25V), op amp A3’s output will

switch low, causing current to con-

duct through R

. This will drag node

A an amount ∆VA

HYS2

below 1.25V,

which will put the circuit in state 1.

The circuit will not exit state 1 until

the battery voltage is charged to an

amount ∆V

HYS2

(149mV for circuit

shown) above 3V (point 2). This will

bring node A back up to the lower trip

BAT

(V)

1.25

1.75

3.0 4.2

∆V

AHYS1

∆V

HYS1

∆V

HYS2

∆V

AHYS2

(V)

point, 1.25V, bringing the circuit out

of state 1. The amount of hysteresis

desired can be calculated using the

following formulas:

High Trip Point:

VBAT = (R3 • (VOHMIN + VBE +

1.75V)/RH1 + 1.75V • (R3/R4) +

1.75V)/(1 + R3/RH1)

∆VHYS1 = 4.2V – VBAT

Low Trip Point:

H2 = (1.25V – V

OLMAX

– V

)/R

∆VHYS2 = IRH2 • R3

where:

OHMIN

= output voltage swing high

(LT1496)

OLMAX

= output voltage swing low

(LT1496)

= diode voltage of 1N458

Using an automobile analogy, if

the LT1496 op amp is the transmis-

sion of the circuit (switching from one

state to the next), the LT1634 voltage

reference is the engine. It not only

generates the threshold voltages, but

also the amount of error that the

circuit will have. How much accuracy

and error you get depends on the car

you drive. Maximum input offset volt-

age and input bias current for the

LT1496 are 475µV and 1nA, respec-

tively. The LT1634 is a 0.05% initial

accuracy, 25ppm/°C tempco, 10µA

precision shunt reference. Its 1.250V

output voltage will appear at the input

of A3 with an accuracy of 0.088%

(initial accuracy + input offset volt-

age). R1 and R2 being 0.1% resistors,

the worst-case ratio error will be 0.2%.

The worst-case voltage error across

R1 will then be 0.2% or 1mV. This

error on the 1.75V threshold voltage

is 0.057%. Similarly, error at 1.75V

due to worst-case 2nA input bias

current is 0.057%. Total worst-case

error at 1.75V will be 0.202%.

BAT

error contributed by the volt-

age divider branch will consist of

three terms: resistor matching, op

amp input bias current and input

offset voltage. The amount of error is

different at the two trip points when

BAT

is 3V or 4.2V. Similar calcula-

tions as above result in 0.328% when

BAT

= 3V and 0.268% when V

BAT

4.2V. Therefore, total battery voltage

error at either trip points is better

than 0.47%. Since only the ratios of

R1 to R2 and R3 to R4 are critical,

precision matched resistors with ten

times better performance can be used

to reduce the overall error by 33%.

This supervisory circuit demon-

strates unparalleled performance

achievable only with Linear

Technology’s supermicropower pre-

cision devices. The supervisory circuit

consumes only 20µA. Battery voltage

monitoring and control accuracy is

better than 0.5%.

Figure 2. V

BAT

vs V

with hysteresis

etatSV

TAB

AedoN2AtuptuO3AtuptuO4AtuptuOsutatS

1V3<V52.1<woLwoLhgiH ,ffodaoL

egrahc

etats

V<V3

V2.4< V<V52.1

V57.1< woLhgiHwoL ,nodaoL

egrahc

etats

3V2.4>V57.1>hgiHhgiHwoL ,nodaoL

egrahc

detanimret

Table 1. Circuit states

Authors can be contacted

at (408) 432-1900

Linear Technology Magazine • November 1997

DESIGN IDEAS

800mA Li-Ion Battery Charger

Occupies Less Volume than

Two Stacked Quarters by Fran Hoffart

Each new generation of cell phones,

PDAs, portable instruments and other

handheld devices is invariably more

powerful, smaller and, most likely,

thinner than the last. The circuit

shown in Figure 1 is designed to

charge one or two Lithium-Ion cells at

currents up to 800mA, with all com-

ponents equal to or less than 2.2mm

(0.086 inches) tall. Using 0.031 inch

PC board material, the total circuit

thickness for this charger is 3.4mm

(0.136in) or the thickness of two

quarters. The complete 800mA

constant-current/constant-voltage

charger, including the PC board,

occupies less volume than two quar-

ters. This compact, low proﬁle

construction is ideal for cell phones

or other applications where circuit

height is restricted.

LT1510-5CGN High

Efﬁciency 500kHz Switch

Mode Battery Charger IC

The charger consists of an LT1510

constant-voltage/constant-current

PWM IC, which includes an onboard

1.5A switch. The LT1510 is available

in either 200kHz or 500kHz versions;

the higher frequency version allows

lower value, smaller-sized inductors

to be used. An internal 0.5% refer-

ence allows precision battery-voltage

programming and a current program-

ming pin allows a single resistor, PWM

signal or a programming current from

a DAC to control the charging cur-

rent. Also included are undervoltage

lockout and a low quiescent current

sleep mode that is activated when

input power is removed.

The internal NPN switch achieves

low saturation voltage by using a

bootstrapped base-drive technique.

A low boost voltage is bootstrapped to

the switch pin to generate a base-

drive voltage that is greater than the

input voltage. The ideal boost voltage

is from 3V to 5V. If a single lithium cell

is being charged, the battery voltage

can be used for the boost voltage, but

for two or more cells, a separate 3V

source is recommended for maximum

efﬁciency.

Fused-Lead Package Offers

Lower Thermal Resistance

The LT1510-5 is available in a spe-

cially constructed 16-lead plastic

SSOP package that has the die-attach

paddle connected (fused) directly to

the four corner leads and ﬁts in the

same area as an SO-8 package. This

low proﬁle fused-lead package pro-

vides a lower thermal resistance by

conducting much of the heat gener-

ated by the die through the copper

leads to the PC board copper. To take

advantage of the improved thermal

properties of this fused-lead package,

it is important to provide as much PC

board copper around the package

leads as is practical. Back-side cop-

per and internal copper layers

interconnected by feed-through vias

all contribute to the overall effective-

ness of the PC board used as a heat

sink. Other heat-producing surface

mount components, such as Schottky

diodes and the inductor, also rely on

the PC board copper to conduct heat

away from the components.

Although the actual component

area required for the charger can be

as low as 0.72in

, it is very likely that

additional copper area may be needed

to maintain component temperatures

at safe levels. The amount of copper

area required depends on factors such

as maximum ambient temperature,

airﬂow, input voltage and charging

current. The LT1510-5 can be

programmed for higher charging cur-

rents, but package and PC board

thermal constraints will determine

the maximum output current.

Selecting Thin Components

Some components are inherently thin

(resistors, ceramic capacitors, diodes

and transistors) but others are more

difﬁcult to make thin (inductors;

R5



11.0k

0.5%

R6



1.02k

0.5%

R4



4.99k

0.5%

TO V

Q1

2N7002

BAT

= 2.465V 1 +R5 + R6



()

BAT

= 2000 2.465V



()

L1

TP3-100

10µH

DI

MBRM

140T3

D2

MBRM140T3



C2

0.22µF

C1



10µF

D3 

MMBD914LT1 

GND

BOOST

GND

OVP

SENSE

GND

PROG

BAT

GND

LT1510-5

= 12V–20V

C4

0.1µF

R2

300

C3

1µF

C5

22µF

R1

6.19k



BAT

= 8.4V

Li-ION

BATTERY

(2 CELLS)

BAT

= 800mA

IBAT

R3

DI 1510 01.eps

Figure 1. Compact, low proﬁle, constant-current/constant-voltage charger for Li-Ion batteries

Linear Technology Magazine • November 1997

DESIGN IDEAS

electrolytic and tantalum capacitors).

The inductor used in this design is a

very thin Coiltronics 10µH gapped

toroid made of ferrite material, mea-

suring 2.2mm (0.086in) in thickness.

Other thicknesses, from 1.8mm to

3mm, are also available in the Coil-

tronics Thin-Pac™ line of inductors.

High dielectric-constant ceramic

chip capacitors are used for C1

through C4. C1 is a Y5U type, selected

because of its small size and high

RMS current capability; X7R type

material is selected for C2 through C4

because of its better temperature and

voltage coefﬁcients. The output

capacitor, C5, is a special polymer

aluminum electrolytic surface mount

capacitor with a package height of

1.8mm. Available from Panasonic, this

solid electrolyte capacitor features

small size, low ESR, high RMS cur-

rent rating and long life.

Even standard SO packages for

ICs are thicker than the 2.2mm limit

required for this design; fortunately,

the LT1510-5 is available in an SSOP

package (GN package), which has a

maximum height of only 1.75mm

(0.069in.).

D1 and D2 are Motorola PowerMite

1A, 40V Schottky diodes with a maxi-

mum package height of 1.15mm.

Although these diodes are very small,

they are rated for 1A, provided there

is sufﬁcient PC board copper sur-

rounding the leads to provide a heat

path for the power dissipated in the

diodes. D3 is a small-signal silicon

diode in a SOT-23 package with a

maximum package height of 1.1mm.

Also available in a SOT-23 package is

ecnerefeR rotangiseDytitnauQrebmuNtraPnoitpircseDrodneVenohP

1C1 TZ601E1E05RCHT 01µ%02,V52,F

cimareCU5Y nocraM0002-696)748(

2C1 A1TAM422C36021 22.0µ%02,V52,F

cimareCR7X XVA1115-282)702(

3C1 TAM501CZ5080 1µ%02,V01,F

cimareCR7X XVA1115-282)702(

4C1 A1TAM401G55080 1.0µ%02,V05,F

cimareCR7X XVA1115-282)702(

5C1 R022B1DCFEE

22µ%02,V5.21,F

remyloP

munimulA

citylortcelE

cinosanaP0665-549)804(

2D,1D2 3T041MRBMykttohcSV04,A1alorotoM7442-144)008(

3D1 1TL419DBMMnociliSV001A2.0alorotoM7442-144)008(

1L1 001-3PT01µcaP-nihTHscinortlioC6787-142)165(

1Q1 2007N2 32-TOS

lennahC-N

TEFSOM xeteZ0017-345)615(

1R1 %1.k91.6

rotsiseRpihC CRI0097-299)215(

2R1 003Ω%5,

rotsiseRpihC CRI0097-299)215(

3R1 %5,k1

rotsiseRpihC CRI0097-299)215(

4R1 %5.0,k99.4rotsiseRpihC CRI0097-299)215(

5R1 %5.0,k0.11rotsiseRpihC CRI0097-299)215(

6R1 %5.0,k20.1rotsiseRpihC CRI0097-299)215(

1U1 NGC5-0151TLCIregrahCyrettaBCTL0091-234)804(

MOSFET Q1, which is used as an

output-divider disconnect switch to

prevent the divider current from dis-

charging the battery when input power

is removed.

Charger Operation

A typical charge proﬁle for a dis-

charged Li-Ion battery is an initial

constant-current charge at 800mA

until the battery voltage rises to the

programmed voltage. It then changes

to a constant-voltage charge, with the

charging current gradually decreas-

ing to near 0mA as the battery

approaches full charge. If complete

charge termination is required, pull-

ing the V

pin low or sinking zero

current from the program pin stops

the charge current. These signals

could be supplied by an external timer

or microprocessor.

When the input power is removed,

the LT1510-5 goes into a low quies-

cent current (3µA) sleep mode, with

this current coming from the battery.

This low battery drain current allows

the battery to remain connected to

the charger for an extended period of

time without appreciably discharging

the battery. Additional battery-drain

current can result from reverse leak-

age current in the Schottky catch

diode D1. Many Schottky diodes have

relatively high leakage currents, so

care must be exercised in their

selection.

Refer to the LT1510 data sheet for

complete product speciﬁcations and

to design notes DN111 and DN124

and application note AN68 for addi-

tional application information.

Conclusion

A compact, low proﬁle Lithium-Ion

battery charger using all surface

mount components has been

described. With a total circuit height

of only 3.2mm (0.136in.) including

PC board, this circuit can charge one

or two Lithium-Ion cells at currents

up to 800mA. Other battery chemis-

tries can also be charged, although

charge termination circuitry will be

required if fast charging is used.

Table 1. Low-proﬁle components used in Figure 1’s circuit

Thin-Pac is a trademark of Coiltronics, Inc. PowerMite is

a registered trademark of Motorola Corp.

Linear Technology Magazine • November 1997

DESIGN IDEAS

Introduction

With the proliferation of satellite

receivers, cable systems and Internet

commerce, there is a need for secure

encryption. A pseudorandom code

generator that ﬁlled the bill several

years ago is now considered

“hackable” by many knowledgeable

people.

Presented here is a truly random

code generator that operates from a

single supply. The circuit allows

operation from a single 5V supply

with a minimum of adjustments.

The circuit produces random ones

and zeroes by comparing a stream of

random noise generated in a Zener

diode to a reference voltage level. If

the threshold is correctly set and the

time period is long enough, the noise

will consist of a random but equal

number of samples above and below

threshold.

That Fuzz is Noise

The circuit shown in Figure 1 is the

random noise generator. Optimum

noise performance is obtained from a

1N753A Zener diode, which has a 6.2

volt Zener “knee.” The diode is used to

generate random noise. We have found

that optimum noise output for this

diode occurs at the “knee” of the I-V

curve, where the Zener just starts to

limit voltage to 6.2 volts.

Operating a 6.2V Zener from a 5V

supply required some thought. Obvi-

ously, some type of voltage boosting

scheme was needed to provide the

diode with the 8V or more that it

requires in this circuit. U1, an

LTC1340 low noise, voltage-boosted

varactor driver, provides 9.2V at 20µA

from an input of 5V. This Zener cur-

rent is the optimal for noise output

from the diode (at 20µA the output is

about 20mV

P-P

The 1M and 249k resistors bias the

input to operational ampliﬁer U2 to

1.25V to match the input common

mode range of comparator U3. The

1µF capacitor provides an AC path

for the noise. Note: be careful where

you place any additional capacitors

in this part of the circuit or the noise

may be unintentionally rolled off. This

is one circuit where noise is desirable.

U2 is an LT1215 23MHz, 50V/µs,

dual operational ampliﬁer that can

operate from a single supply. It is

used as a wideband, gain-of-eleven

ampliﬁer to amplify the noise from

the Zener diode; the second op amp in

U2 is unused. U3, an LT1116 high

speed, ground-sensing comparator,

receives the noise at its positive input.

A threshold is set at the negative

comparator input and the output is

adjusted via the 2k potentiometer for

an equal number of ones and zeroes.

Single-Supply

Random Code Generator by Richard Markell

–

CP



SHDN

PGND

CC

OUT

AGND

1

2

3

4

8

7

6

U1

LTC1340

0.1µF

1000pF

9.2V

1N753A

6.2V

1µF

249k 1.25V

U2

LT1215 U3

LT1116

1µF

470k

47k

2k

10 TURN

10µF

CLOCK IN

0.1µF

1µF 

TANT

10pF

465 LE

82-LEVEL

OUTPUT

0.1µF

10 1

U4

74HC373

~20mV

P-P



NOISE

GND

1µF1µF

Figure 1. Single-supply random code generator

continued on page 36

Linear Technology Magazine • November 1997

DESIGN IDEAS

level of –72dBc. For a data rate of

1.544Mbps and 2-bit-per-symbol

encoding, the fundamental frequency

of operation is 392kHz.

The LT1497 provides such low dis-

tortion because it operates at only a

fraction of its output current capabil-

ity and is well within its voltage swing

limitations. There are other LTC

ampliﬁers that can achieve this per-

formance, but at the expense of higher

power dissipation or a larger package.

Performance

The circuit of Figure 1 was evaluated

for harmonic distortion with a 400kHz

sine wave and an output level of

5.6V

P-P

into 135Ω. Figure 2 shows

that the second harmonic is –72.3dB

relative to the fundamental for the

135Ω load. Third harmonic distortion

is not critical, because received sig-

nals are heavily ﬁltered before being

digitized by an A/D converter. Perfor-

mance with a 50Ω load (to simulate

more challenging test loops) is slightly

better at –75dB. The output signal

was attenuated to obtain maximum

sensitivity of the HP4195A network

analyzer used for the measurements.

With multicarrier applications such

as discrete multitone modulation

A Low Distortion, Low Power,

Single-Pair HDSL Driver Using the LT1497

by George Feliz and Adolfo Garcia

Introduction

High speed digital subscriber line

(HDSL) interfaces support full-duplex

data rates up to 1.544Mbps over

12,000 feet using two standard 135Ω

twisted-pair telephone wires. The high

data rate is achieved with a combina-

tion of encoding 2 bits per symbol

using two-binary, one-quaternary

(2B1Q) modulation, and sophisticated

digital signal processing to extract

the received signal. This performance

is possible only with low distortion

line drivers and receivers. In addi-

tion, the power dissipation of the

transceiver circuitry is critical because

it may be loop-powered from the cen-

tral ofﬁce over the twisted pair. Lower

power dissipation also increases the

number of transceivers that can

placed in a single, non-forced–air

enclosure. Single-pair HDSL requires

the same performance as two-pair

HDSL over a single twisted pair and

operates at twice the fundamental

2B1Q symbol rate. In HDSL systems

that use 2B1Q line coding, the signal

passband necessary to carry a data

rate of 1.544Mbps is 392kHz. This

signal rate will be used to quantify the

performance of the LT1497 in this

article.

Low Distortion Line Driver

The circuit of Figure 1 transmits sig-

nals over a 135Ω twisted pair through

a 1:1 transformer. The LT1497 dual

125mA, 50MHz current feedback

ampliﬁer was chosen for its ability to

cleanly drive heavy loads, while con-

suming a modest 7mA maximum

supply current per ampliﬁer in a ther-

mally enhanced SO-8 package. The

driver ampliﬁers are conﬁgured in

gains of two (A1) and minus one (A2)

to compensate for the attenuation

inherent in the back-termination of

the line and to provide differential

drive to the transformer. The trans-

mit power requirement for HDSL is

13.5dBm (22.4mW) into 135Ω, corre-

sponding to a 1.74V

RMS

signal. Since

2B1Q modulation is a 4-level pulse

amplitude modulated signal, the crest

factor (peak to RMS) of this signal is

1.61. Thus, a 13.5dBm, 2B1Q modu-

lated signal yields 5.6V

P-P

across the

135Ω load. The corresponding output

signal current is ±20.7mA peak. This

modest drive level increases for vary-

ing line conditions and is tested with

a standardized collection of test loops

that can have line impedances as low

as 25Ω. The LT1497’s high output

current and voltage swing drive the

135Ω line at the required distortion

A2

1/2 LT1497

–

560Ω



A1

1/2 LT1497

560Ω



560Ω



560Ω

68.1Ω



135Ω



OUT

5.6V

P-P

68.1Ω



1.1*

–5V

*MIDCOM 671-7807

(800) 643-2661

DI 1497 01.eps

–50

–100

FREQUENCY (kHz)

2HD

AMPLITUDE (dBm)

DI 1497 02

100 200 300 400 500 600 700 800 900 1000 1100

Figure 1. LT1497 HDSL driver

Figure 2. Harmonic distortion of Figure 1’s

circuit with a 400kHz sine wave and an

output level of 5.6V

P-P

into 135Ω

continued on page 32

Linear Technology Magazine • November 1997

DESIGN IDEAS

Most engineering labs are well

stocked with low voltage, moderate

current power supplies, but higher

voltage supplies capable of several

amperes of output current are hard

to find. I solved this problem in my lab

by building the supply shown in

Figure 1.

The circuit is based on U1, an

LT1270 high efficiency switching regu-

lator configured in a SEPIC topology,

which allows the output to be ad-

justed higher or lower than the input

voltage. Operation is similar to that of

a flyback converter, but the primary

and secondary windings are coupled

100V, 2A, Constant-Voltage/Constant-

Current Bench Supply by Mitchell Lee

together by capacitor C1. This allows

the primary and secondary windings

to share current, reducing copper

loss; it also eliminates the snubbing

circuitry and losses found in flyback

converters.

The converter is designed to oper-

ate from an input of 40V to 60V,

supplied by a line transformer, diode

bridge and filter capacitor (not shown).

Output voltage is linearly adjustable

from zero to 100V via potentiometer

R20.

The current is limited by two inde-

pendent loops. The first current limit

loop is user controlled over a range of

zero to 8A by setting potentiometer

R21. This setting does not interact

with changes in output voltage. A

second current limit loop limits the

maximum available current as a func-

tion of voltage (components R1–R5

and U2), minimizing component

stress. Under any given operating

condition, the lower of the two loops

takes control. Maximum available

output current is highest at low out-

put voltage settings (about 8A), and

decreases to 2A at 100V output.

–

3.3k

2N6387

IRF450

1N5817

10Ω

100Ω

U1

LT1270

680µF

100V

×2

15V 56µF

35V

1µF



22V

2N2907

2.2k

100Ω

0.33µF

U3B

1/2 LT1215

U3A

1/2 LT1215

100Ω

0.01µF10k

15k

20k

10k

0.1µF

R20

10k

1µF

LT1034CZ-2.5

R1

3.9k R2

3.9k

U2B

1/2 LT1413

U2A

1/2 LT1413

1N4148 1

0.1µF1k

3.9k

R21, 1k

R3

4.5k R4

3.9k R5

2.7M

0.03Ω, 2W

4.7k

0.1µF

1N4148

2.2k

10µF

200V

FILM CAP

10µF

200V

FILM CAP

L1

20µH

MUR1560

2.2k 1000pF

MUR120

MBR745

C1*

10µF, 100V

FILM CAP

OUT



0V–100V

100k

IN

40V–60V

15k

T1 =PRIMARY: 57 TURNS 20AWG

 SECONDARY: 57 TURNS 20AWG

 MPP 55076 MAG INC CORE

L1 = 18 TURNS 18AWG

 55380-A2 MAG INC CORE



3.9V



Figure 1. 100V/2A Constant Voltage/Constant Current Bench Supply

Jesus Rosales played a significant role in the

design, building and testing of this circuit.

Linear Technology Magazine • November 1997

DESIGN IDEAS

–50

–100

FREQUENCY (Hz)

AMPLITUDE (dBm)

DI 1497 03

100 200  300 400 500 600

3IMD3IMD

third-order intermodulation products

are well below –72dB. With a 50Ω

load, performance is within 1dB–2dB

of that with the 135Ω load.

Conclusion

The circuit presented provides out-

standing distortion performance in

an SO-8 package with remarkably

low power dissipation. It is ideally

suited for single pair digital subscriber

line applications, especially for remote

terminals.

(DMT) becoming as prevalent as

single-carrier applications, another

important measure of ampliﬁer

dynamic performance is 2-tone inter-

modulation. This evaluation is a

valuable tool to gain insight to ampli-

ﬁer linearity when processing more

than one tone at a time.

For this test, two sine waves at

300kHz and 400kHz were used with

levels set to obtain 5.6V

P-P

across the

135Ω load. Figure 3 shows that the Figure 3. 2-tone intermodulation for Figure

1’s circuit

HDSL Driver, continued from page 30

LCD displays and CCD imaging

circuits in today’s portable products

require several bias voltages from a

10V to 20V input at a few mA. When

symmetric bipolar bias supplies are

needed, the negative supply can be

generated with a discrete charge pump

operating from the power switch of

the boost regulator that generates

the positive supply. However, an asym-

metric bipolar supply is typically

required: for example 20V and –10V

for LCD displays or 15V and –7.5V for

CCDs. One possible solution is to add

a linear regulator to the negative out-

put; this adds cost and greatly reduces

the efficiency of the switcher. Another

possibility is a 2-output flyback cir-

cuit, but the added cost and bulk of a

Switcher Generates Two Bias Voltages

without Transformer by Jeff Witt

transformer make this solution un-

appealing. The circuit in Figure 1

avoids these penalties, producing 20V

at 5mA and –10V at 5mA from 3.3V

with 73% efficiency. The circuit uses

standard surface mount parts.

The LT1316, a micropower Burst

Mode switching regulator with an in-

tegrated 0.6A power switch, operates

in an ordinary boost circuit to gener-

ate the 20V (V

OUT1

) set by resistor

divider R1 and R2. An internal com-

parator at the FB pin regulates the

output by gating the LT1316’s oscil-

lator. A charge pump (C2 and

associated diodes) coupled to the

LT1316’s switch pin generates the

negative output voltage. This nega-

tive output (V

OUT2

) is monitored by

the LT1316’s low-battery detector

through the resistor divider R3 and

R4, using the positive 20V output as

a reference. When the negative out-

put falls below 10V, the low-battery

detector output (LBO pin and lowest

trace in Figure 2) turns Q1 on, en-

abling the charge pump and charging

output capacitor C4. Note that the

switch pin jumps between ground

and ~10V during this period. Once

the negative output has been charged

enough to overcome the low-battery

detector’s hysteresis, Q1 turns off

and the switch pin is free to fly to 20V,

charging the positive output.

This circuit can also operate di-

rectly from two alkaline or NiCd cells.

Slightly higher peak currents are nec-

essary; change R5, which determines

the peak switch current of the LT1316,

to 6.8kΩ and change L1 to 15µH.



3.3V

C1

33µF

10V

R5

10k

L1

47µF

C2

1µF

35V

C4

3.3µF 35V

C3

3.3µF

35V

R4

590k

Q1

2N7002

82k

R3

150k

R2

64.9k

150pF



R1

BAT54

LT1316

FB

LBO

LBI

8

1

SET

GND

7SHDN

OUT1



20V/5mA

OUT2



–10V/5mA

L1 = COILCRAFT DO1608C-473

C1 = AVX TAJB336M035R

C2 = AVX TAJA105M035R

C3, C4 = AVX TAJB335M035R

Figure 1. By gating the charge pump, this circuit generates a regulated

negative output with a magnitude different from that of the positive output. Figure 2. Voltage waveforms of Figure 1’s circuit

OUT1

200mV/DIV

(AC COUPLED)

SW PIN 20V/DIV

OUT2

1V/DIV

(AC COUPLED)

LBO PIN 5V/DIV

0.1ms/DIV

Linear Technology Magazine • November 1997

DESIGN INFORMATION

Reference Squeezes More Performance

from Less Package by John Wright

Introduction

The small outline transistor (SOT)

package is used extensively for space

sensitive, surface mount, discrete-

transistor applications. Recently,

several manufacturers have put volt-

age references into SOT packages,

but the current crop sacriﬁces per-

formance to ﬁt into this stingy

package. Simply put, the package

and die area are so small that tech-

niques used to design good references

need modiﬁcation. Today’s applica-

tions demand more performance in

smaller PC board space and the size

of the SOT is ideal—the SOT footprint

is 4.7 times smaller than the 8-pin SO

package footprint. The challenge is to

make a quality reference small enough

to ﬁt into the SOT package.

Voltage Reference or

Precision Regulator

The LT1460S3 is a series reference

that provides supply current and

power dissipation advantages over

shunt references, which must idle

their entire load current to operate.

This new reference uses curvature

compensation to obtain low tempera-

ture coefﬁcient, and laser-trimmed

precision thin-ﬁlm resistors to achieve

high output accuracy. The man-

ufacturing process used is a

stepper-based, high speed bipolar pro-

cess that was selected for its density.

The LT1460S3 uses a proprietary trim

algorithm at wafer sort that guaran-

tees less than 20ppm/°C temperature

coefﬁcient (typically 10ppm/°C). The

temperature coefﬁcient (TC) is affected

by package stress during assembly,

widening the distribution slightly. The

result is three grades, two with

guaranteed TC ≤ 20ppm/°C. This per-

formance is something of a milestone,

because the highest grade TC avail-

able in SOT heretofore was 50ppm/°C,

and that is the lowest grade available

on the LT1460S3. Low stress thin-

ﬁlm resistors and special layout

techniques were used to achieve 0.2%

maximum tolerance without the use

of gel coat to protect the surface from

stress. This combination of low TC

and high accuracy makes the

LT1460S3 capable of 8 bits of abso-

lute accuracy over temperature

without a system calibration.

The LT1460S3 supplies up to 20mA

of output drive, making it ideal for

precision regulator applications. It is

stable with any value output capaci-

tor, including zero, and this can be

helpful in applications where PC board

space is at a premium, or when fast

settling is demanded. An output

capacitor used for stability slows down

the reference transient response. The

LT1460S3 settles in 1µs to 0.2% for a

100µA pulse with no output capaci-

tor and in just 2µs for a 1mA pulse.

Measure It on a PC Board

Soldering a SOT reference onto a PC

board is a harsh thing to do. The

reference that leaves Linear Tech-

nology is within the data sheet

speciﬁcation, but the thermal shock

of being soldered onto a PC board

causes the reference to shift. The

problem is that the short, stiff leads

do not absorb stress and it is trans-

ferred to the die. Low stress thin-ﬁlm

resistors and careful layout tech-

niques in the LT1460S3 reduce

solder-related shift to typically 0.02%.

A graph of the output voltage after IR

reﬂow solder is shown in Figure 1.

After the part is soldered onto a PC

board, the output voltage should not

drift with time.

When an instrument manufacturer

sends out a PC board for duty in the

ﬁeld, it has had extensive testing,

perhaps a calibration, and its voltage

reference is stable. Over time, how-

ever, the reference output voltage may

change. This long-term drift is caused

mainly by differential stress between

the IC package and the PC board.

This output shift is a nonlinear func-

tion of time, and is normally expressed

in ppm/√kh. What this means is that

the shift in the ﬁrst one thousand

hours is more than that in the subse-

quent one thousand hours. Figure 2

shows a plot of long-term drift of the

LT1460S3 soldered to a PC board.

This part was measured in a 30°C

oven that was controlled to ±1°C. The

long-term drift is the trend line cre-

ated by this plot; over one thousand

hours, this typical part moved 50ppm,

or about 125µV for the LT1460S3-2.5.

Hysteresis in the output voltage is

another package-stress phenomenon.

A reference output voltage does not

return to exactly the same 25°C value

after it has been at different tempera-

tures. This is because the stress on

OUTPUT VOLTAGE (V)

2.490

DISTRIBUTION (%)

1460_01.EPS

02.500 2.510

LT1460HC LIMITS

TIME (HOURS)

0 100 200 300 400 500 600 700 800 900 1000

DRIFT (PPM)

1460_02

–20

–10

–40

–30

–60

–50

–70

–80

TREND LINE

Figure 1. Typical distribution of SOT-23

LT1460HC V

OUT

after IR reﬂow solder Figure 2. LT1460S3 long-term drift

Linear Technology Magazine • November 1997

DESIGN INFORMATION

the die changes with temperature and

never returns to exactly the same

state in which it started. Hysteresis is

roughly an exponential function of

the temperature excursion away from

25°C and depends on whether the IC

was previously at a higher or lower

temperature. The output voltage is

always measured at 25°C, but the IC

is cycled over its speciﬁed temperature

range before successive measure-

ments. The LT1460S3 has a typical

hysteresis of 100ppm for a 0°C to

70°C excursion. Hysteresis increases

to 250ppm with a –40°C to 85°C tem-

perature change because larger stress

is associated with a larger tempera-

ture range. The initial accuracy

speciﬁcation of 0.2% includes a mar-

gin for hysteresis. This is because it is

not possible to know what tempera-

ture changes the part may be

subjected to on its way to the cus-

tomer. Because hysteresis is increased

by wider temperature excursions, the

part may shift out of tolerance if is

stored outside of the speciﬁed tem-

perature range.

Summary

The LT1460S3’s high output current,

low temperature coefﬁcient and

excellent, yet conservative accuracy

speciﬁcation set a new standard for

SOT voltage references. This new ref-

erence is available in 2.5V and 5V

output voltages. For even tighter tol-

erance and lower temperature

coefﬁcient, the LT1460 is also avail-

able in MSOP, SO, PDIP and 3-lead

TO-92 packages.

LTC1659, LTC1448: Smallest Rail-to-Rail

12-Bit DACs Have Lowest Power

In this age of portable electronics,

power and size are the primary con-

cerns of most designers. The LTC1659

and the LTC1448 are rail-to-rail, 12-

bit, voltage output DACs that address

both of these concerns. The LTC1659

is a single DAC in an MSOP-8 pack-

age that draws only 250µA from a 3V

or 5V supply, whereas the LTC1448

is a dual DAC in an SO-8 package

that draws 450µA from a 3V or 5V

supply.

Figure 1 shows a convenient way

to use the LTC1659 in a digital con-

trol loop where 12-bit resolution is

required. The output of the LTC1659

will swing from 0V to V

REF

, because

there is a gain of one from the REF pin

to V

OUT

at full-scale. Because the out-

put can only swing up to V

, V

REF

should be less than or equal to V

prevent the loss of codes and degra-

dation of PSRR near full-scale.

To obtain full dynamic range, the

REF pin can be connected to the

supply pin, which can be driven from

a reference to guarantee absolute

accuracy (see Figure 2). The LT1236

is a precision 5V reference with an

input range of 7.2V to 40V. In this

conﬁguration, the LTC1659 has a wide

output swing of 0V to 5V. The LTC1448

can be used in a similar configuration

where dual DACs are needed.

by Hassan Malik

µP

1659_01

2.7V to 5.5V

CONTROL

VOLTAGE

(0V TO V

REF

)

REF

≤ V

REF

CLK V

OUT

LTC1659

CS/LD GND

µP

CONTROL

VOLTAGE

(OV TO 5V)



(7.2V to 40V)

REF

CLK V

OUT

LTC1659

CS/LD GND

1659_02



IN OUT

GND

LT1236

0.1µF

Figure 1. 12-bit DAC for digital control loop

Figure 2. 12-bit DAC with wide output swing

Authors can be contacted

at (408) 432-1900

Linear Technology Magazine • November 1997

DESIGN INFORMATION

Introduction

The LTC1197 and LTC1199 10-bit

serial ADCs, recently released by Lin-

ear Technology, offer small size, low

power operation and fast sample rates

with good AC and DC performance.

These parts are ideal for low power,

high speed and/or compact designs.

In this article, we will examine the

features and performance of the

LTC1197/LTC1199 that make these

parts excellent choices for such new

designs.

Features

Smallest Size (MSOP)

The LTC1197/LTC1199 are among

the smallest ADCs available. The serial

interface allows the LTC1197/

LTC1199 to be offered in 8-pin pack-

ages. The MSOP package (an SO is

also available) reduces the small foot-

print even further. These are some of

the ﬁrst ADCs available in the MSOP

package, which is about half the size

of the SO-8.

Flexible Inputs

The LTC1197 has a single differential

input with an adjustable reference.

The adjustable reference input allows

the full scale to be reduced to as low

as 200mV. This translates to an LSB

size of only 200µV. Combined with

the high impedance of the analog

input, this allows direct digitization

of low level transducer outputs, which

can save board space and the cost of

a gain stage.

With its software-selectable

2-channel MUX, the LTC1199 is ca-

pable of measuring either one

differential or two single-ended in-

puts. Both parts have a built-in

sample-and-hold.

Serial I/O

Using either 3- or 4-wire serial inter-

faces, the LTC1197/LTC1199 are

hardware and software compatible

with both SPI and MICROWIRE pro-

tocols. This compatibility is achieved

with no additional circuitry, allowing

easy interface to many popular

processors.

3V or 5V Supplies

The LTC1197/LTC1199 are 5V parts

= 4V–9V for the LTC1197 and

4V–6V for the LTC1199). Also avail-

able for use in 3V systems are the

LTC1197L and the LTC1199L (V

2.7V–4V). Designed for use in mixed-

supply systems, the digital inputs of

these devices can be taken above the

voltage without damaging the

ADC. This is useful in systems where

the ADC is running at a lower supply

voltage than the processor. If the ADC

is running at a higher supply voltage

than the processor, the ADC serial

data output voltage can easily be de-

creased to a level appropriate for the

processor.

To summarize, the LTC1197/

LTC1199/LTC1197L/LTC1199L are

a very ﬂexible group of parts. They are

capable of providing a designer with a

small footprint, one or two channels,

single-ended or differential inputs,

adjustable reference voltage,

SPI-/MICROWIRE-compatible serial

I/O and 3V or 5V operation.

Performance

Micropower Performance with

Auto Shutdown at Full Speed

Now let’s take a look at the perfor-

mance of the LTC1197/LTC1199.

Running continuously, the LTC1197L

consumes only 2.2mW at the maxi-

mum sampling rate (25mW for the

LTC1197). You can reduce the power

consumption dramatically, as shown

in Figure 1, simply by lowering the

sampling rate. The formula for calcu-

lating power consumption is:

PD = VCC • ICC • tCONV • fS

where P

is the power consumption,

is the supply voltage, I

is the

supply current while the conversion

is occurring, t

CONV

is the conversion

time and f

is the sample rate. As you

can see from the formula, lowering f

reduces the power consumption lin-

early. It is also important to minimize

CONV

by clocking the ADC at its maxi-

mum rate during the conversion.

Although the ADC draws slightly more

power at high clock rates, the total

power is less because the device is on

for a shorter period of time. For

example, the 2.2mW is cut to 10.4µW

by reducing the sampling rate from

210kHz to 1kHz. The calculation for

power dissipation is as follows:

PD = 2.7V • 0.8mA • 4.8µs •

1kHz = 10.4µW

High Speed Capability

Even though the LTC1197/LTC1199

are capable of micropower operation,

they are able to convert at rates of up

LTC1197/LTC1199: New Micropower

MSOP 10-Bit ADC Samples at 500ksps

SAMPLING FREQUENCY (kHz)

0.01

SUPPLY CURRENT (µA)

100

1000

10000

100

1197_01

0.1 0.1 110 1000

= 5V, 

CLK

= 7.2MHz

= 2.7V, 

CLK

= 3.5MHz

Figure 1. Supply current (I

) vs sampling

frequency f

by Guy Hoover and Marco Pan

Linear Technology Magazine • November 1997

to 500kHz (210kHz for LTC1197L).

These parts can also digitize fast in-

put signals up to the Nyquist

frequency (250kHz for the LTC1197)

with over nine effective number of

bits (ENOBs).

Good DC and AC Specs

The DC speciﬁcations of these parts

are very good. Linearity (both INL and

DNL) is typically 0.3LSB with a maxi-

mum spec of 1LSB. Offset is speciﬁed

at 2LSBs (Max) and gain error is

speciﬁed at 4LSBs (Max). These speci-

ﬁcations are guaranteed over the full

temperature range of the part. Both

commercial and industrial tempera-

ture range versions are available.

AC performance is equally impres-

sive. S/(N + D) is typically 60dB (58dB

for the L version). THD is typically

–64dB (–60dB for the L version) and

the peak harmonic or spurious noise

is typically –68dB (–63dB for the L

version). The L version has slightly

degraded speciﬁcations due to the

smaller dynamic range and the higher

internal noise of the ADC at its lower

operating current.

Conclusion

We have seen that the LTC1197 and

LTC1199 have a small footprint and

are capable of micropower operation.

These parts have a versatile serial

interface that is SPI/MICROWIRE

compatible. The adjustable reference

input, 2-channel, software-selectable

MUX and 5V or 3V operation add to

the versatility of this ADC family.

When this versatility is combined with

the high conversion rate and good DC

and AC performance, you can see

why these ADCs are good choices for

low power, high speed and/or com-

pact designs.

The 5k resistor and the 10µF capaci-

tor provide limited hysteresis so that

the adjustment of the potentiometer

is not as critical. Latch U4, a 74HC373,

ensures that the output remains

latched throughout one clock period.

The circuit’s output is taken from

U4’s Q0 output.

Some Thoughts on

Automatic Threshold

Adjustment

Several circuit designers have asked

about threshold adjustment without

manual knobs or potentiometers. One

way to implement this would be to

have the microprocessor count the

number of ones and zeroes over a

given time period and adjust the

threshold (perhaps via a digital pot) to

produce the required density of ones.

A more “analog” method of adjust-

ing threshold might be to implement

an integrator with reset. This circuit

integrates the number of ones and

zeroes over time to produce a zero

result for an adjustment that pro-

duces equal numbers of ones and

zeroes. Again, a digital pot could be

used to adjust threshold, with the

threshold being decreased for the case

of “not enough ones” and increased

for the case of “too many ones.”

After many more conversations

with the “cyber illuminati,” the circuit

in Figure 2 was devised. This circuit

can be used to replace the pot shown

in the dashed box in Figure 1. In

operation, an LT1004-2.5 is used as a

reference at the front end of a preci-

sion voltage divider string. A series of

voltages is generated along the divider

string and a jumper is used to con-

nect this voltage to a buffer and then

to the negative input of the LT1116

comparator. As was the case with the

2k pot, the voltage at pin 2 (the nega-

tive input of the comparator) sets the

threshold for the comparator. The

selection of voltage taps on the resistor

–

12k 1%

(1.30V)



499 1%

(1.25V)

499 1%

(1.20V)

499 1%

(1.15V)

499 1%

(1.10V)



11k 1%

JUMPER SELECTS

THRESHOLD VOLTAGE

LT1004

-2.5

15k

0.01µF

1µF

LT1490

1µF

TO LT1116

PIN 2 (FIGURE 1)

GROUND PINS 5 AND 6

string is arbitrary; they were selected

to allow a good adjustment range

(deﬁned as allowing jumper adjust-

ment to 50% ones and 50% zeros) for

a sample of ten 1N753A Zener diodes

used to produce noise. The jumper

could (and probably should) be re-

placed with analog switches controlled

by a microprocessor in medium- to

high-volume applications.

Figure 2. Jumper selects threshold for Figure 1’s circuit

Random Code, continued from page 29

CONCLUSIONS

Linear Technology Magazine • November 1997

NEW DEVICE CAMEOS

LTC1660 Micropower, Octal,

10-Bit DAC with Sleep Mode

The LTC1660 sets a new standard in

DAC density by integrating eight high

quality addressable 10-bit DACs in a

single, tiny 16-pin package, no larger

than an SO-8: required board area is

only 0.006in

per DAC.

Operating on a single 2.7V–5.5V

supply rail, the LTC1660 draws just

60µA per DAC (480µA for all eight),

for true micropower performance.

Sleep mode further reduces total sup-

ply-plus-reference current to just 1µA.

The LTC1660 is guaranteed mono-

tonic over temperature—DNL error is

typically ±0.2LSB (±0.5LSB Max).

Each of the rail-to-rail output ampli-

ﬁers can source or sink up to 5mA.

The outputs swing to within a few

millivolts of either supply rail when

unloaded and have an equivalent out-

put resistance of 80Ω when driving a

load to the rails. The output ampliﬁ-

ers are stable driving capacitive loads

up to 1000pF.

The 3-wire serial interface uses a

16-bit input word comprising four

address/control bits, ten input-code

bits, and two don’t-care bits. Asyn-

chronous CLR, power-on reset and

daisy-chain capability are also pro-

vided. It is possible to keep one or

more chips in a daisy chain in con-

tinuous Sleep mode by giving the

Sleep instruction to these chips each

time the active chips in the chain are

updated.

Ultralow supply current, power-

saving Sleep mode and extremely

compact size make the LTC1660 ideal

for battery-powered applications; its

straightforward usability, high per-

formance and low cost make it an

excellent choice as a general-purpose

converter.

LTC1234: Low Dropout

Regulator with Comparator

and Shutdown

The LTC1234 combines a micropower,

positive, low dropout linear regulator

and a low-battery comparator. With

only 20µA quiescent current, typical

dropout voltages are 250mV for 5V

output with 500mV of load, or 150mV

for 3.3V with 250mA of load. An inter-

nal 750mA P-channel pass transistor

draws no base current, allowing the

device to draw less than 35µA of

quiescent current over the extended

temperature range, independent of

load current.

Protection circuitry protects

against output short circuits, ther-

mal overloads and reverse current

when the input potential falls below

the output.

A micropower comparator with an

open drain output can be used for

system or battery monitoring. The SD

pin allows the regulator to be shut

down while the comparator and ref-

erence remain alive. The quiescent

current drops down to 5µA in shut-

down mode.

The LTC1234 is available in the

8-pin SO wide package.

LTC1068-XX:

New Family of Quad

Universal, Switched Capacitor

Filter Building Blocks

The LTC1068-XX consist of four iden-

tical, low noise, switched capacitor

ﬁlter building blocks. Each building

block, together with three to ﬁve

resistors and an external clock, pro-

vides 2nd order ﬁlter functions such

as lowpass, bandpass, highpass and

notch.

High precision quad 2nd order,

dual 4th order or single 8th order

ﬁlters can be designed with an

LTC1068-XX. All these devices are

fully supported by the new Filter-

CAD™ for Windows

ﬁlter design

software.

An external clock tunes the center

frequency of each ﬁlter building block.

The clock-to-center frequency ratio is

internally set to 25:1 (LTC1068-25),

50:1 (LTC1068-50) or 200:1

(LTC1068-200). The clock-to-center

frequency ratio can also be modiﬁed

with external resistors. The internal

sampling rate of all the LTC1068-XX

devices is twice the clock frequency.

This allows the frequency of input

signals to approach twice the clock

frequency before aliasing occurs. The

LTC1068-XX is designed to comple-

ment the LTC1068, which features a

100:1 clock-to-center frequency ratio.

The LTC1068-200, with its internal

400:1 sampling-rate-to-center-fre-

quency ratio behaves almost like a

continuous time ﬁlter; it is recom-

mended for non-band-limited ﬁlter

circuits such as highpass and notch.

Notch ﬁlters with 80dB notch depth

can be easily realized.

The LTC1068-50 is aimed at ﬁlter

applications, especially lowpass and

bandpass, requiring single 3V to 5V

supply and low power consumption.

The LTC1068-50 consumes 3mA

typical.

The LTC1068-25 lowers its inter-

nal clock-to-center frequency ratio so

it can achieve 200kHz cutoff frequen-

cies with a 5MHz clock.

The LTC1068-XX family is avail-

able in 28-pin SSOP. Demo boards

are also available to qualiﬁed

customers.

A customized version of the

LTC1068-XX in a 16-lead SO with

internal thin-ﬁlm resistors can be

ordered. Please contact LTC market-

ing for details.

New Device Cameos

Linear Technology Magazine • November 1997

NEW DEVICE CAMEOS

LTC1541/LTC1542:

Micropower Op Amp,

Comparator and Reference

The LTC1541/LTC1542 combine a

micropower ampliﬁer, comparator and

bandgap reference (LTC1541 only) in

an 8-pin package. These parts operate

from single supplies of 2.5V to 12.6V

or dual supplies of ±1.25V to ±6.3V,

with a typical supply current of 6µA.

Both the op amp and comparator

feature a common mode input voltage

range that extends from the negative

supply to within 1.3V of the positive

supply. The output stages swing from

rail-to-rail. The comparator’s invert-

ing input is internally connected to

the reference output (LTC1541) or

bonded out without the reference

(LTC1542).

The reference output voltage is 1.2V

±1% over the extended temperature

range (–40°C to 85°C). The output can

drive a bypass capacitor of up to

0.01µF without oscillation. It can also

source up to 2mA and sink up to 20µA.

The op amp is internally compen-

sated to be unity-gain stable, with a

puts. A highpass response is avail-

able if one of the resistors is replaced

with an external capacitor. Center

frequency is internally trimmed to

±0.5% and is very stable with tem-

perature. A shutdown pin puts the

chip into “zero-power” shutdown

state.

Operating from single or dual power

supplies of 4.75V to 10.5V total, the

LTC1562 is designed for applications

where dynamic range is critical, such

as receivers, data modems and DSP

antialiasing or reconstruction. With

±5V supplies and Q = 1, a 2nd order

section delivers a typical signal-to-

noise ratio (SNR) of 103dB, total

harmonic distortion of –96dB and DC

offset of 2mV. In applications that

ﬁlter signals of wide amplitude range,

it is possible to use the LTC1562’s

gain-setting resistor for gain control,

which reduces the input noise at

higher gains and extends the dynamic

range (maximum signal to minimum

noise) to 118dB.

The LTC1562 is offered in 16-pin

DIP and 20-pin SSOP packages.

For further information on any

of the devices mentioned in this

issue of Linear Technology, use

the reader service card or call

the LTC literature service

number:

1-800-4-LINEAR

Ask for the pertinent data sheets

and Application Notes.



for

the latest information

on LTC products, 

visit

www.linear-tech.com

typical gain bandwidth of 10kHz and

a typical slew rate of 8V/ms. The

comparator has ±3mV of internal hys-

teresis to ensure clean output

switching even with slow moving input

signals.

The LTC1541/LTC1542 are avail-

able in 8-pin SO and MSOP packages.

LTC1562 Quad

Universal Active Filter

with SNR >100dB

The LTC1562 is a new low noise, low

distortion, DC-accurate, continuous-

time ﬁlter block with rail-to-rail inputs

and outputs, optimized for 50kHz–

200kHz applications (other cutoff

frequency ranges are available by cus-

tom order). It contains four matched,

independent 2nd order ﬁlter sections,

which can be cascaded in any combi-

nation, such as one 8th order or two

4th order ﬁlters. Each section’s re-

sponse is programmed with three

external resistors, for center fre-

quency, Q and gain, and each section

provides lowpass and bandpass out-

Linear Technology Magazine • November 1997

DESIGN TOOLS

Applications on Disk

Noise Disk — This IBM-PC (or compatible) program

allows the user to calculate circuit noise using LTC op

amps, determine the best LTC op amp for a low noise

application, display the noise data for LTC op amps,

calculate resistor noise and calculate noise using specs

for any op amp. Available at no charge

SPICE Macromodel Disk — This IBM-PC (or compat-

ible) high density diskette contains the library of LTC

op amp SPICE macromodels. The models can be used

with any version of SPICE for general analog circuit

simulations. The diskette also contains working circuit

examples using the models and a demonstration copy

of PSPICE™ by MicroSim. Available at no charge

SwitcherCAD™ — The SwitcherCAD program is a pow-

erful PC software tool that aids in the design and

optimization of switching regulators. The program can

cut days off the design cycle by selecting topologies,

calculating operating points and specifying compo-

nent values and manufacturer’s part numbers. 144

page manual included. $20.00

SwitcherCAD supports the following parts: LT1070

series: LT1070, LT1071, LT1072, LT1074 and LT1076.

LT1082. LT1170 series: LT1170, LT1171, LT1172 and

LT1176. It also supports: LT1268, LT1269 and LT1507.

LT1270 series: LT1270 and LT1271. LT1371 series:

LT1371, LT1372, LT1373, LT1375, LT1376 and

LT1377.

Micropower SwitcherCAD™ — The MicropowerSCAD

program is a powerful tool for designing DC/DC con-

verters based on Linear Technology’s micropower

switching regulator ICs. Given basic design param-

eters, MicropowerSCAD selects a circuit topology and

offers you a selection of appropriate Linear Technology

switching regulator ICs. MicropowerSCAD also per-

forms circuit simulations to select the other components

which surround the DC/DC converter. In the case of a

battery supply, MicropowerSCAD can perform a bat-

tery life simulation. 44 page manual included. $20.00

MicropowerSCAD supports the following LTC micro-

power DC/DC converters: LT1073, LT1107, LT1108,

LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,

LT1300, LT1301 and LT1303.

Technical Books

1990 Linear Databook, Vol I —This 1440 page collec-

tion of data sheets covers op amps, voltage regulators,

references, comparators, ﬁlters, PWMs, data conver-

sion and interface products (bipolar and CMOS), in

both commercial and military grades. The catalog

features well over 300 devices. $10.00

1992 Linear Databook, Vol II — This 1248 page

supplement to the 1990 Linear Databook is a collection

of all products introduced in 1991 and 1992. The

catalog contains full data sheets for over 140 devices.

The 1992 Linear Databook, Vol II is a companion to the

1990 Linear Databook, which should not be discarded.

$10.00

1994 Linear Databook, Vol III —This 1826 page

supplement to the 1990 and 1992 Linear Databooks is

a collection of all products introduced since 1992. A

total of 152 product data sheets are included with

updated selection guides. The 1994 Linear Databook

Vol III is a companion to the 1990 and 1992 Linear

Databooks, which should not be discarded. $10.00

1995 Linear Databook, Vol IV —This 1152 page

supplement to the 1990, 1992 and 1994 Linear Da-

tabooks is a collection of all products introduced since

1994. A total of 80 product data sheets are included

with updated selection guides. The 1995 Linear Data-

book Vol IV is a companion to the 1990, 1992 and 1994

Linear Databooks, which should not be discarded.

$10.00

1996 Linear Databook, Vol V —This 1152 page supple-

ment to the 1990, 1992, 1994 and 1995 Linear

Databooks is a collection of all products introduced

since 1995. A total of 65 product data sheets are

included with updated selection guides. The 1996

Linear Databook Vol V is a companion to the 1990,

1992, 1994 and 1995 Linear Databooks, which should

not be discarded. $10.00

1990 Linear Applications Handbook, Volume I —

928 pages full of application ideas covered in depth by

40 Application Notes and 33 Design Notes. This cata-

log covers a broad range of “real world” linear circuitry.

In addition to detailed, systems-oriented circuits, this

handbook contains broad tutorial content together

with liberal use of schematics and scope photography.

A special feature in this edition includes a 22-page

section on SPICE macromodels. $20.00

1993 Linear Applications Handbook, Volume II —

Continues the stream of “real world” linear circuitry

initiated by the 1990 Handbook. Similar in scope to the

1990 edition, the new book covers Application Notes

40 through 54 and Design Notes 33 through 69.

References and articles from non-LTC publications

that we have found useful are also included. $20.00

1997 Linear Applications Handbook, Volume III —

This 976 page handbook maintains the practical outlook

and tutorial nature of previous efforts, while broaden-

ing topic selection. This new book includes Application

Notes 55 through 69 and Design Notes 70 through

144. Subjects include switching regulators, measure-

ment and control circuits, ﬁlters, video designs,

interface, data converters, power products, battery

chargers and CCFL inverters. An extensive subject

index references circuits in LTC data sheets, design

notes, application notes and

Linear Technology

maga-

zines. $20.00

Interface Product Handbook — This 424 page hand-

book features LTC’s complete line of line driver and

receiver products for RS232, RS485, RS423, RS422,

V.35 and AppleTalk

applications. Linear’s particular

expertise in this area involves low power consumption,

high numbers of drivers and receivers in one package,

mixed RS232 and RS485 devices, 10kV ESD protec-

tion of RS232 devices and surface mount packages.

Available at no charge

Power Solutions Brochure — This 84 page collection

of circuits contains real-life solutions for common

power supply design problems. There are over 88

circuits, including descriptions, graphs and perfor-

mance speciﬁcations. Topics covered include battery

chargers, PCMCIA power management, microproces-

sor power supplies, portable equipment power supplies,

micropower DC/DC, step-up and step-down switching

regulators, off-line switching regulators, linear regula-

tors and switched capacitor conversion.

Available at no charge

High Speed Ampliﬁer Solutions Brochure —

This 72 page collection of circuits contains real-life

solutions for problems that require high speed

ampliﬁers. There are 82 circuits including descrip-

tions, graphs and performance speciﬁcations. Topics

covered include basic ampliﬁers, video-related appli-

cations circuits, instrumentation, DAC and photodiode

ampliﬁers, ﬁlters, variable gain, oscillators and current

sources and other unusual application circuits.

Available at no charge

Data Conversion Solutions Brochure — This 52 page

collection of data conversion circuits, products and

selection guides serves as excellent reference for the

data acquisition system designer. Over 60 products

are showcased, solving problems in low power, small

size and high performance data conversion applica-

tions—with performance graphs and speciﬁcations.

Topics covered include ADCs, DACs, voltage refer-

ences and analog multiplexers. A complete glossary

deﬁnes data conversion speciﬁcations; a list of se-

lected application and design notes is also included.

Available at no charge

Telecommunications Solutions Brochure — This 72

page collection of circuits, new products and selection

guides covers a wide variety of products targeted for

the telecommunications industry. Circuits solving real

life problems are shown for central ofﬁce switching,

cellular phone, base station and other telecom applica-

tions. New products introduced include high speed

ampliﬁers, A/D converters, power products, interface

transceivers and ﬁlters. Reference material includes a

telecommunications glossary, serial interface stan-

dards, protocol information and a complete list of key

application notes and design notes.

Available at no charge

continued on page 40

DESIGN TOOLS

Acrobat is a trademark of Adobe Systems, Inc. AppleTalk

is a registered trademark of Apple Computer, Inc. PSPICE

is a trademark of MicroSim Corp.

Information furnished by Linear Technology Corporation

is believed to be accurate and reliable. However, Linear

Technology makes no representation that the circuits

described herein will not infringe on existing patent rights.

Linear Technology Magazine • November 1997

Acrobat is a trademark of Adobe Systems, Inc.; Windows

is a registered trademark of Microsoft Corp.; Macintosh is

a registered trademark of Apple Computer, Inc.

CD-ROM

LinearView — LinearView™ CD-ROM version 2.0 is

Linear Technology’s latest interactive CD-ROM. It al-

lows you to instantly access thousands of pages of

product and applications information, covering Linear

Technology’s complete line of high performance ana-

log products, with easy-to-use search tools.

The LinearView CD-ROM includes the complete prod-

uct speciﬁcations from Linear Technology’s Databook

library (Volumes I–V) and the complete Applications

Handbook collection (Volumes I–III). Our extensive

collection of Design Notes and the complete collection

Linear Technology

magazine are also included.

A powerful search engine built into the LinearView CD-

ROM enables you to select parts by various criteria,

such as device parameters, keywords or part numbers.

All product categories are represented: data conver-

sion, references, ampliﬁers, power products, ﬁlters

and interface circuits. Up-to-date versions of Linear

Technology’s software design tools, SwitcherCAD,

DESIGN TOOLS, continued from page 39

Micropower SwitcherCAD, FilterCAD, Noise Disk and

Spice Macromodel library, are also included. Every-

thing you need to know about Linear Technology’s

products and applications is readily accessible via

LinearView. LinearView 2.0 runs under Windows

3.1,

Windows 95 and Macintosh

System 7.0 or later.

Available at no charge.

World Wide Web Site

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technical information covering the Company’s prod-

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Other areas within the site include help, news and

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FAX: (972) 380-5138

Linear Technology Corporation

9430 Research Blvd.

Echelon IV Suite 400

Austin, TX 78759

Phone: (512) 343-3679

FAX: (512) 343-3680

Linear Technology Corporation

Houston, TX 77478

Phone: (972) 733-3071

FAX: (972) 380-5138

Linear Technology Corporation

5510 Six Forks Road, Suite 102

Raleigh, NC 27609

Phone: (919) 870-5106

FAX: (919) 870-8831

CENTRAL REGION

Linear Technology Corporation

2010 E. Algonquin Road, Suite 209

Schaumburg, IL 60173

Phone: (847) 925-0860

FAX: (847) 925-0878

Linear Technology Corporation

Kenosha, WI 53144

Phone: (414) 859-1900

FAX: (414) 859-1974

SOUTHWEST REGION

Linear Technology Corporation

21243 Ventura Blvd., Suite 227

Woodland Hills, CA 91364

Phone: (818) 703-0835

FAX: (818) 703-0517

Linear Technology Corporation

15375 Barranca Parkway, Suite A-211

Irvine, CA 92718

Phone: (714) 453-4650

FAX: (714) 453-4765

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