AVAILABLE
EVALUATION KIT AVAILABLE
Functional Diagrams
Pin Configurations appear at end of data sheet.
Functional Diagrams continued at end of data sheet.
UCSP is a trademark of Maxim Integrated Products, Inc.
For pricing, delivery, and ordering information, please contact Maxim Direct
at 1-888-629-4642, or visit Maxim’s website at www.maximintegrated.com.
General Description
The MAX16834 is a current-mode high-brightness LED
(HB LED) driver for boost, boost-buck, SEPIC, and high-
side buck topologies. In addition to driving an n-channel
power MOSFET switch controlled by the switching con-
troller, it also drives an n-channel PWM dimming switch to
achieve LED PWM dimming. The MAX16834 integrates
all the building blocks necessary to implement a fixed-fre-
quency HB LED driver with wide-range dimming control.
The MAX16834 features constant-frequency peak cur-
rent-mode control with programmable slope compensa-
tion to control the duty cycle of the PWM controller.
A dimming driver designed to drive an external n-chan-
nel MOSFET in series with the LED string provides
wide-range dimming control up to 20kHz. In addition to
PWM dimming, the MAX16834 provides analog dim-
ming using a DC input at REFI. The programmable
switching frequency (100kHz to 1MHz) allows design
optimization for efficiency and board space reduction.
A single resistor from RT/SYNC to ground sets the
switching frequency from 100kHz to 1MHz while an
external clock signal at RT/SYNC disables the internal
oscillator and allows the MAX16834 to synchronize to
an external clock. The MAX16834’s integrated high-
side current-sense amplifier eliminates the need for a
separate high-side LED current-sense amplifier in
boost-buck applications.
The MAX16834 operates over a wide supply range of
4.75V to 28V and includes a 3A sink/source gate driver
for driving a power MOSFET in high-power LED driver
applications. It can also operate at input voltages
greater than 28V in boost configuration with an external
voltage clamp. The MAX16834 is also suitable for DC-
DC converter applications such as boost or boost-
buck. Additional features include external enable/
disable input, an on-chip oscillator, fault indicator out-
put (FLT) for LED open/short or overtemperature condi-
tions, and an overvoltage protection sense input
(OVP+) for true overvoltage protection.
The MAX16834 is available in a thermally enhanced
4mm x 4mm, 20-pin TQFN-EP package and in a thermal-
ly enhanced 20-pin TSSOP-EP package and is specified
over the automotive -40°C to +125°C temperature range.
Applications
Single-String LED LCD Backlighting
Automotive Rear and Front Lighting
Projection System RGB LED Light Sources
Architectural and Decorative Lighting (MR16, M111)
Spot and Ambient Lights
DC-DC Boost/Boost-Buck Converters
Features
oWide Input Operating Voltage Range (4.75V to 28V)
oWorks for Input Voltage > 28V with External
Voltage Clamp on VIN for Boost Converter
o3000:1 PWM Dimming/Analog Dimming
oIntegrated PWM Dimming MOSFET Driver
oIntegrated High-Side Current-Sense Amplifier for
LED Current Sense in Boost-Buck Converter
o100kHz to 1MHz Programmable High-Frequency
Operation
oExternal Clock Synchronization Input
oProgrammable UVLO
oInternal 7V Low-Dropout Regulator
oFault Output (FLT) for Overvoltage, Overcurrent,
and Thermal Warning Faults
oProgrammable True Differential Overvoltage
Protection
o20-Pin TQFN-EP and TSSOP-EP Packages
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
VIN BOOST LED DRIVER
IN
ON
NDRV
CS
ANALOG
DIM
OFF
DIMOUT
SENSE+
PWMDIM
PGND
LED+
LEDs
LED-
MAX16834
REFI
Simplified Application Circuit
Ordering Information
+
Denotes a lead(Pb)-free/RoHS-compliant package.
*
EP = Exposed pad.
/V denotes an automotive qualified part.
/VY denotes an automotive qualified part in a “side wettable
flank” package.
PART TEMP RANGE PIN-PACKAGE
MAX16834ATP+
-40°C to +125°C
20 TQFN-EP*
MAX16834ATP/V+
-40°C to +125°C
20 TQFN-EP*
MAX16834AUP+
-40°C to +125°C
20 TSSOP-EP*
MAX16834AUP/V+
-40°C to +125°C
20 TSSOP-EP*
MAX16834AGP/VY+
-40°C to +125°C
20 QFN-EP*
Pin Configurations appear at end of data sheet.
MAX16834
19-4235; Rev 4; 8/11
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA= TJ= -40°C to +125°C, unless otherwise noted. Typical values are at TA= +25°C.)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
Note 1: Package thermal resistances were obtained using the method described in JEDEC specification JESD51-7, using a four-
layer board. For detailed information on package thermal considerations, refer to www.maxim-ic.com/thermal-tutorial.
IN, HV, LV to SGND................................................-0.3V to +30V
OVP+, SENSE+, DIMOUT, CLV to SGND ..............-0.3V to +30V
SENSE+ to LV........................................................-0.3V to +0.3V
HV, IN to LV ............................................................-0.3V to +30V
OVP+, CLV, DIMOUT to LV ......................................-0.3V to +6V
PGND to SGND .....................................................-0.3V to +0.3V
VCC to SGND..........................................................-0.3V to +12V
NDRV to PGND...........................................-0.3V to (VCC + 0.3V)
All Other Pins to SGND.............................................-0.3V to +6V
NDRV Continuous Current................................................±50mA
DIMOUT Continuous Current..............................................±2mA
VCC Short-Circuit Current to SGND Duration ...........................1s
Continuous Power Dissipation (TA= +70°C)
QFN, TQFN (derate 25.6mW/°C* above +70°C).........2051mW
TSSOP (derate 26.5mW/°C above +70°C) .....................2122mW
Operating Temperature Range .........................-40°C to +125°C
Junction Temperature......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
Soldering Temperature (reflow) .......................................+260°C
*
As per JEDEC51 standard (multilayer board).
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Input Voltage Range VIN 4.75 28 V
Quiescent Supply Current IQExcluding ILED 610mA
Shutdown Supply Current ISHDN VUVEN = 0V 30 60 µA
INTERNAL LINEAR REGULATOR (VCC)
Output Voltage VCC 0 ≤ ICC ≤ 50mA, 9.5V ≤ VIN ≤ 28V 6.3 7 7.7 V
Dropout Voltage VDO ICC = 35mA (Note 2) 0.65 1 V
Short-Circuit Current VCC = 0V, VIN = 12V 80 300 mA
LINEAR REGULATOR (CLV)
Output Voltage (VCLV - VLV)0 ≤ ICLV ≤ 2mA, 6V ≤ VHV ≤ 28V,
6V ≤ V(HV-LV) ≤ 22V 4.7 5 5.3 V
Dropout Voltage VDO ICLV = 2mA, 0 ≤ VLV ≤ 23.3V (Note 3) 0.5 V
Short-Circuit Current VCLV = 12V, VIN = 12V, VHV = 24V 2.2 10 mA
REFERENCE VOLTAGE (REF)
Output Voltage VREF 0 ≤ IREF ≤ 1mA, 4.75V ≤ VIN ≤ 28V 3.625 3.70 3.775 V
REF Short-Circuit Current VREF = 0V 30 mA
UNDERVOLTAGE LOCKOUT/ENABLE INPUT (UVEN)
UVEN On Threshold Voltage VUVEN_THUP 1.395 1.435 1.475 V
UVEN Threshold Voltage
Hysteresis 200 mV
Input Leakage Current ILEAK VUVEN = 0V 1 µA
PWMDIM
PWMDIM On Threshold Voltage VPWMDIM 1.395 1.435 1.475 V
PACKAGE THERMAL CHARACTERISTICS (Note 1)
Junction-to-Ambient Thermal Resistance (θJA)
QFN, TQFN ........................................................................39°C/W
TSSOP .............................................................................37.7°C/W
Junction-to-Case Thermal Resistance (θJC)
QFN, TQFN .....................................................................6°C/W
TSSOP.............................................................................2°C/W
MAX16834
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
PWMDIM Threshold Voltage
Hysteresis 200 mV
Input Leakage Current VPWMDIM = 0V 1 µA
OSCILLATOR
RRT/SYNC = 5kΩ0.9 1 1.1 MHz
Oscillator Frequency fOSC RRT/SYNC = 25kΩ180 200 220 kHz
Oscillator Frequency Range (Note 4) 100 1000 kHz
External Sync Input Clock High
Threshold (Note 4) 2 V
External Sync Input Clock Low
Threshold (Note 4) 0.4 V
External Sync Input High Pulse
Width (Note 4) 200 ns
Maximum External Sync Period 50 µs
SLOPE COMPENSATION (SC)
SC Pullup Current ISCPU VSC = 100mV 80 100 120 µA
SC Discharge Resistance RSCD VSC = 100mV 8 Ω
REFI
REFI Input Bias Current VREFI = 1V 1 µA
REFI Input Common-Mode Range (Note 4) 0 2 V
SENSE+
SENSE+ Input Bias Current (VSENSE+ - VLV) = 100mV 250 µA
HIGH-SIDE LED CURRENT-SENSE AMPLIFIER (VSENSE+ - VLV)
Input Offset Voltage VLV > 5V, (VSENSE+ - VLV) = 5mV -2.4 0 +2.4 mV
Voltage Gain AVVLV > 5V, (VSENSE+ - VLV) = 0.2V 9.7 9.9 10.1 V/V
(VSENSE+ - VLV) = 0.1V, no load 1.8 MHz
3dB Bandwidth (VSENSE+ - VLV) = 0.02V, no load 600 kHz
LOW-SIDE LED CURRENT-SENSE AMPLIFIER
Input Offset Voltage VLV < 1V, (VSENSE+ - VLV) = 0V -2 0 +2 mV
Voltage Gain AVVLV < 1V, (VSENSE+ - VLV) = 0.2V 9.7 9.9 10.1 V/V
3dB Bandwidth 600 kHz
CURRENT ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)
Transconductance gmVCOMP = 2V, VPWMDIM = 5V 400 500 600 µS
Open-Loop DC Gain AV60 dB
Input Offset Voltage -10 0 +10 mV
COMP Voltage Range VCOMP (Note 4) 0.4 2.5 V
PWM COMPARATOR
Input Offset Voltage 0.6 0.65 0.70 V
Propagation Delay tPD 50mV overdrive 40 ns
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA= TJ= -40°C to +125°C, unless otherwise noted. Typical values are at TA= +25°C.)
MAX16834
Maxim Integrated
3
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
ELECTRICAL CHARACTERISTICS (continued)
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA= TJ= -40°C to +125°C, unless otherwise noted. Typical values are at TA= +25°C.)
PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS
Minimum On-Time tON(MIN) On-time includes blanking time 100 ns
Duty Cycle (Note 4) 90 99.5 %
CURRENT PEAK LIMIT COMPARATOR
Trip Threshold Voltage 0.25 0.3 0.35 V
Propagation Delay 50mV overdrive with respect to NDRV 40 ns
OVERVOLTAGE PROTECTION INPUT (OVP+)
OVP+ On Threshold Voltage VOVP_ON 1.375 1.435 1.495 V
OVP+ Hysteresis 200 mV
OVP+ Input Leakage Current (VOVP - VLV) = 1.235V -1 +1 µA
HIGH-SIDE LED SHORT COMPARATOR
Off Threshold VCLV - VLV 4.0 4.3 4.6 V
On Threshold VCLV - VLV 4.1 4.4 4.7 V
Error Reject Blankout fOSC = 500kHz 256 µs
LOW-SIDE LED SHORT COMPARATOR
Off Threshold 0.27 0.30 0.33 V
Error Reject Blankout 5µs
HICCUP TIMER
Hiccup Time fOSC = 500kHz 8.2 ms
GATE-DRIVER OUTPUT (NDRV)
NDRV Peak Pullup Current VCC = 7V 3 A
NDRV Peak Pulldown Current VCC = 7V 3 A
p-Channel MOSFET RDSON (VCC - VNDRV) = 0.1V 1.2 1.9 Ω
n-Channel MOSFET RDSON VNDRV = 0.1V 0.9 1.7 Ω
DIMOUT
DIMOUT Peak Pullup Current (VCLV - VLV) = 5V 25 50 mA
DIMOUT Peak Pulldown Current (VCLV - VLV) = 5V 25 50 mA
p-Channel MOSFET RDSON (VCLV - VDIMOUT) = 0.1V 31 Ω
n-Channel MOSFET RDSON (VDIMOUT - VLV) = 0.1V 25 Ω
PWMDIM to DIMOUT
Propagation Delay 200 ns
FAULT FLAG (FLT)
FLT Pulldown Current VFLT = 0.2V 2 5 10 mA
FLT Leakage Current VFLT = 1.0V 1 µA
Thermal Warning On Threshold +140 °C
Thermal Warning Threshold
Hysteresis 20 °C
Note 2: Dropout voltage is defined as VIN - VCC, when VCC is 100mV below the value of VCC for VIN = 9.5V.
Note 3: Dropout is defined as VHV - VCLV, when VCLV is 100mV below the value of VCLV for VHV = 8V.
Note 4: Not production tested. Guaranteed by design.
MAX16834
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
VREF vs. TEMPERATURE
MAX16834 toc01
TEMPERATURE (°C)
VREF (V)
11095-25 -10 535 50 6520 80
3.60
3.62
3.64
3.66
3.68
3.70
3.72
3.74
-40 125
VIN = 12V
VREF vs. SUPPLY VOLTAGE
MAX16834 toc02
SUPPLY VOLTAGE (V)
VREF (V)
242016128
3.55
3.60
3.65
3.70
3.75
3.80
3.50
428
VREF vs. IREF
MAX16834 toc03
IREF (mA)
V
REF
(V)
981 2 3 5 64 7
3.6985
3.6990
3.6995
3.7000
3.7005
3.7010
3.7015
3.7020
3.6980
010
VIN = 12V
SUPPLY CURRENT
vs. SUPPLY VOLTAGE
MAX16834 toc04
SUPPLY VOLTAGE (V)
SUPPLY CURRENT (mA)
242016128
2
4
6
8
10
12
14
16
18
20
0
428
PWMDIM = 0
SUPPLY CURRENT
vs. TEMPERATURE
MAX16834 toc05
TEMPERATURE (°C)
SUPPLY CURRENT (mA)
1109565 80-10 520 35 50-25
1
2
3
4
5
6
7
8
9
10
0
-40 125
VIN = 12V
PWMDIM = 0
RT vs. SWITCHING FREQUENCY
MAX16834 toc06
SWITCHING FREQUENCY (kHz)
RT (kΩ)
10
100
1
1000100
VIN = 12V
SWITCHING FREQUENCY
vs. TEMPERATURE
MAX16834 toc07
TEMPERATURE (°C)
SWITCHING FREQUENCY (kHz)
1109565 80-10 520 35 50-25
592
593
594
595
596
597
598
599
600
601
602
603
604
605
591
590
-40 125
VIN = 12V
VCC vs. ICC
MAX16834 toc08
ICC (mA)
VCC (V)
908060 7020 30 40 5010
6.92
6.94
6.96
6.98
7.00
7.02
7.04
7.06
7.08
7.10
7.12
7.14
7.16
6.90
0 100
VIN = 12V
VCC vs. ICC
MAX16834 toc09
ICC (mA)
VCC (V)
908070605040302010
6.9
7.0
7.1
7.2
6.8
0 100
VIN = 12V
TA = +125°C
TA = +100°C
TA = +25°C
TA = -40°C
Typical Operating Characteristics
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA= +25°C, unless otherwise noted.)
MAX16834
Maxim Integrated
5
Pin Description
PIN
Q F N , T Q F N TSSOP NAME FUNCTION
1 3 OVP+
LED-String Overvoltage Protection Input. Connect a resistive voltage-divider between the
positive output, OVP+, and LV to set the overvoltage threshold. OVP+ has a 1.435V threshold
voltage with a 200mV hysteresis.
2 4 SGND Signal Ground
3 5 COMP Error-Amplifier Output. Connect an RC network from COMP to SGND for stable operation. See
the Feedback Compensation section.
4 6 REF 3.7V Reference Output Voltage. Bypass REF to SGND with a 0.1µF to 0.22µF ceramic
5 7 REFI Current Reference Input. VREFI provides a reference voltage for the current-sense amplifier to
set the LED current.
68SC
Current-Mode Slope Compensation Setting. Connect to an appropriate external capacitor from
SC to SGND to generate a ramp signal for stable operation.
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Typical Operating Characteristics (continued)
(VIN = VHV = 12V, VUVEN = 5V, VLV = VPWMDIM = VSGND, CVCC = 4.7µF, CLCV = 100nF, CREF = 100nF, RSENSE+ = 0.1Ω,
RRT = 10kΩ, TA= +25°C, unless otherwise noted.)
VCC vs. VIN
MAX16834 toc10
VIN (V)
VCC (V)
26
22
1814
10
7.02
7.04
7.06
7.08
7.10
7.12
7.14
7.16
7.18
7.20
7.00
6
TA = +125°CTA = +25°CTA = -40°C
NDRV RISE/FALL TIME
vs. CAPACITANCE
MAX16834 toc11
CAPACITANCE (nF)
NDRV RISE TIME (ns)
987654321
10
20
30
40
50
0
010
VIN = 12V
RISE TIME
FALL TIME
VCLV vs. VHV
MAX16834 toc13
VHV (V)
VCLV (V)
26
22
18
14
10
5.01
5.02
5.03
5.04
5.05
5.06
5.07
5.08
5.09
5.10
5.00
6
VIN = 12V
VCLV vs. ICLV
MAX16834 toc12
ICLV (mA)
VCLV (V)
4.54.03.0 3.51.0 1.5 2.0 2.50.5
0.50
1.00
1.50
2.00
2.50
3.00
3.50
4.00
4.50
5.00
5.50
0
05.0
VIN = 12V
MAX16834
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Pin Description (continued)
PIN
Q F N , T Q F N TSSOP NAME FUNCTION
79FLT Active-Low, Open-Drain Fault Indicator Output. See the Fault Indicator (
FLT
) section.
8 10 RT/SYNC
Resistor-Programmable Switching Frequency Setting/Sync Control Input. Connect a resistor
from RT/SYNC to SGND to set the switching frequency. Drive RT/SYNC to synchronize the
switching frequency with an external clock.
9 11 UVEN
Undervoltage-Lockout (UVLO) Threshold/Enable Input. UVEN is a dual-function adjustable
UVLO threshold input with an enable feature. Connect UVEN to VIN through a resistive voltage-
divider to program the UVLO threshold. Observe the absolute maximum value for this pin.
10 12 PWMDIM PWM Dimming Input. Connect to an external PWM signal for dimming operation.
11 13 CS Current-Sense Amplifier Positive Input. Connect a resistor from CS to PGND to set the inductor
peak current limit.
12 14 PGND Power Ground
13 15 NDRV External n-Channel Gate-Driver Output
14 16 VCC 7V Low-Dropout Voltage Regulator. Bypass to PGND with at least a 1µF low-ESR ceramic
capacitor. VCC provides power to the n-channel gate driver (NDRV).
15 17 IN Positive Power-Supply Input. Bypass to PGND with at least a 0.1µF ceramic capacitor.
16 18 HV High-Side Positive Supply Input Referred to LV. HV provides power to high-side linear regulator
17 19 CLV 5V High-Side Regulator Output. CLV provides power to the dimming MOSFET driver. Connect a
0.1µF to 1µF ceramic capacitor from CLV to LV for stable operation.
18 20 DIMOUT External Dimming MOSFET Gate Driver. DIMOUT is capable of sinking/sourcing 50mA.
19 1 LV High-Side Reference Voltage Input. Connect to SGND for boost configuration. Connect to IN for
boost-buck configuration.
20 2 SENSE+ LED Current-Sense Positive Input. Connect a bypass capacitor of at least 0.1µF between
SENSE+ and LV close to the IC.
——EP
Exposed Pad. Connect EP to a large-area contiguous copper ground plane for effective power
dissipation. Do not use as the main IC ground connection. EP must be connected to SGND.
Detailed Description
The MAX16834 is a current-mode, high-brightness LED
(HB LED) driver designed to control a single-string LED
current regulator with two external n-channel MOSFETs.
The MAX16834 integrates all the building blocks nec-
essary to implement a fixed-frequency HB LED driver
with wide-range dimming control. The MAX16834
allows implementation of different converter topologies
such as SEPIC, boost, boost-buck, or high-side buck
current regulator.
The MAX16834 features a constant-frequency, peak-cur-
rent-mode control with programmable slope compensa-
tion to control the duty cycle of the PWM controller. A
dimming driver offers a wide-range dimming control for
the external n-channel MOSFET in series with the LED
string. In addition to PWM dimming, the MAX16834
allows for analog dimming of LED current.
The MAX16834 switching frequency (100kHz to 1MHz)
is adjustable using a single resistor from RT/SYNC. The
MAX16834 disables the internal oscillator and synchro-
nizes if an external clock is applied to RT/SYNC. The
switching MOSFET driver sinks and sources up to 3A,
making it suitable for high-power MOSFETs driving in
HB LED applications, and the dimming control allows
for wide PWM dimming at frequencies up to 20kHz.
The MAX16834 is suitable for boost and boost-buck
LED drivers (Figures 2 and 3).
The MAX16834 alone operates over a wide 4.75V to
28V supply range. With a voltage clamp that limits the
IN pin voltage to less than 28V, it can operate in boost
configuration for input voltages greater than 28V.
Additional features include external enable/disable
input, an on-chip oscillator, fault indicator output (FLT)
for LED open/short or overtemperature conditions, and
an overvoltage protection circuit for true differential
overvoltage protection (Figure 1).
MAX16834
Maxim Integrated
7
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
AND
128 TOSC
ERROR
REJECT
DELAY
5µs ERROR
REJECT
DELAY
4096 TOSC
HICCUP
TIMER
FLTB
SENSE+
4.3V
VLV
VBG
VBG
VBG
VBG
0.3V
0.6V
REFERENCE
OR
OT
AND
AND AND
PWMDIM
OT
HIGH-SIDE
5V
REGULATOR
VBG
7V
LDO
RAMP
GENERATOR
PWM
COMP
OSC
BLANK
5kΩ
0.3V
AV = 9.9
UVLO
S
REF
IN
UVEN
RT/SYNC
SC
CS
REFI
SENSE+
COMP
HV
LV
PWMDIM
OVP+
VCC
NDRV
PGND
FLT
CLV
DIMOUT
TO
INTERNAL
CIRCUITRY
CURRENT-LIMIT
COMPARATOR
LED CURRENT-
SENSE AMPLIFIERS
ERROR
AMPLIFIER
LV REFERENCE
SWITCH
Q
R
NDRVB
TEMPERATURE
SENSE
VLV
VLV
FLTA
FLTB
REFHI
FLTAFLTB
VLV
VHV
VREF
VREF
LPF
VIN
VLV
REFHI
gm
NDRVB
SGND
Figure 1. Internal Block Diagram
MAX16834
Maxim Integrated
The MAX16834 is also suitable for DC-DC converter
applications such as boost or boost-buck (Figures 6
and 7). Other applications include boost LED drivers
with automotive load dump protection (Figure 4) and
high-side buck LED drivers (Figure 5).
Undervoltage Lockout/Enable
The MAX16834 features an adjustable UVLO using the
enable input (UVEN). Connect UVEN to VIN through a
resistive divider to set the UVLO threshold. The
MAX16834 is enabled when the VUVEN exceeds the
1.435V (typ) threshold. See the
Setting the UVLO
Threshold
section for more information.
UVEN also functions as an enable/disable input to the
device. Drive UVEN low to disable the output and high
to enable the output.
Reference Voltage (REF)
The MAX16834 features a 3.7V reference output, REF.
REF provides power to most of the internal circuit blocks
except for the output drivers and is capable of sourcing
1mA to external circuits. Connect a 0.1µF to 0.22µF
ceramic capacitor from REF to SGND. Connect REF to
REFI through a resistive divider to set the LED current.
Reference Input (REFI)
The output current is proportional to the voltage at
REFI. Applying an external DC voltage at REFI or using
a potentiometer from REF to SGND allows analog dim-
ming of the output current.
High-Side Reference Voltage Input (LV)
LV is a reference input. Connect LV to SGND for boost
and SEPIC topologies. Connect LV to IN for boost-buck
and high-side buck topologies.
Dimming Driver Regulator
Input Voltage (HV)
The voltage at HV provides the input voltage for the
dimming driver regulator. For boost or SEPIC topology,
connect HV either to IN or to VCC. For boost-buck, con-
nect HV to a voltage higher than IN. The voltage at HV
must not exceed 28V with respect to PGND. For the
high-side buck, connect HV to IN.
Dimming MOSFET Driver (DIMOUT)
The MAX16834 requires an external n-channel MOSFET
for PWM dimming. Connect the gate of the MOSFET to
the output of the dimming driver, DIMOUT, for normal
operation. The dimming driver is capable of sinking or
sourcing up to 50mA of current.
n-Channel MOSFET Switch Driver (NDRV)
The MAX16834 drives an external n-channel switching
MOSFET. NDRV swings between VCC and PGND.
NDRV is capable of sinking/sourcing 3A of peak current,
allowing the MAX16834 to switch MOSFETs in high-
power applications. The average current demanded
from the supply to drive the external MOSFET depends
on the total gate charge (QG) and the operating
frequency of the converter, fSW. Use the following equa-
tion to calculate the driver supply current INDRV
required for the switching MOSFET:
INDRV = QGx fSW
Pulse Dimming Inputs (PWMDIM)
The MAX16834 offers a dimming input (PWMDIM) for
pulse-width modulating the output current. PWM dim-
ming can be achieved by driving PWMDIM with a pul-
sating voltage source. When the voltage at PWMDIM is
greater than 1.435V, the PWM dimming MOSFET turns
on and when the voltage on PWMDIM is below 1.235V,
the PWM dimming MOSFET turns off.
High-Side Linear Regulator (VCLV)
The MAX16834’s 5V high-side regulator (CLV) powers
up the dimming MOSFET driver. VCLV is measured with
respect to LV and sources up to 2mA of current.
Bypass CLV to LV with a 0.1µF to 1µF low-ESR ceramic
capacitor. The maximum voltage on CLV with respect
to PGND must not exceed 28V. This limits the input volt-
age for boost-buck topology.
Low-Side Linear Regulator (VCC)
The MAX16834’s 7V low-side linear regulator (VCC) pow-
ers up the switching MOSFET driver with sourcing capa-
bility of up to 50mA. Use at least a 1µF low-ESR ceramic
capacitor from VCC to PGND for stable operation.
LED Current-Sense Input (SENSE+)
The differential voltage from SENSE+ to LV is fed to an
internal current-sense amplifier. This amplified signal is
then connected to the negative input of the transcon-
ductance error amplifier. The voltage gain factor of this
amplifier is 9.9 (typ).
Whenever VLV is greater than 5V, the input impedance
of the LED current-sense amplifier seen at the SENSE+
pin is 1kΩ±30%. In that condition, a bias current of
20µA (±30%) is pulled from SENSE+, in addition to the
current due to the 1kΩresistor. When VLV is less than
1V, the amplifier input (SENSE+ pin) is in high imped-
ance and the bias current of 20µA (±30%) is pushed
out of that pin.
Always have a bypass capacitor of at least 0.1µF value
between SENSE+ and LV and close to the IC.
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
Maxim Integrated
9
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Internal Transconductance Error Amplifier
The MAX16834 has a built-in transconductance amplifi-
er used to amplify the error signal inside the feedback
loop. The amplified current-sense signal is connected
to the negative input of the gmamplifier with the current
reference connected to REFI. The output of the op amp
is controlled by the input at PWMDIM. When the signal
at PWMDIM is high, the output of the op amp connects
to COMP; when the signal at PWMDIM is low, the out-
put of the op amp disconnects from COMP to preserve
the charge on the compensation capacitor. When the
voltage at PWMDIM goes high, the voltage on the com-
pensation capacitor forces the converter into a steady
state. COMP is connected to the negative input of the
PWM comparator with CMOS inputs, which draw very
little current from the compensation capacitor at COMP
and thus prevent discharge of the compensation
capacitor when the PWMDIM input is low.
Internal Oscillator
The internal oscillator of the MAX16834 is programma-
ble from 100kHz to 1MHz using a single resistor at
RT/SYNC. Use the following formula to calculate the
switching frequency:
where RT is the resistor from RT/SYNC to SGND.
The MAX16834 synchronizes to an external clock signal
at RT/SYNC. The application of an external clock dis-
ables the internal oscillator allowing the MAX16834 to
use the external clock for switching operation. The
internal oscillator is enabled if the external clock is
absent for more than 50µs. The synchronizing pulse
minimum width for proper synchronization is 200ns.
Switching MOSFET
Current-Sense Input (CS)
CS is part of the current-mode control loop. The switch-
ing control uses the voltage on CS, set by RCS, to termi-
nate the on pulse width of the switching cycle, thus
achieving peak current-mode control. Internal leading-
edge blanking is provided to prevent premature turn-off
of the switching MOSFET in each switching cycle.
Slope Compensation (SC)
The MAX16834 uses an internal-ramp generator for
slope compensation. The ramp signal also resets at the
beginning of each cycle and slews at the rate pro-
grammed by the external capacitor connected at SC.
The current source charging the capacitor is 100µA.
Overvoltage Protection (OVP+)
OVP+ sets the overvoltage threshold limit across the
LEDs. Use a resistive divider between output OVP+
and LV to set the overvoltage threshold limit. An internal
overvoltage protection comparator senses the differen-
tial voltage across OVP+ and LV. If the differential volt-
age is greater than 1.435V, NDRV is disabled and FLT
asserts. When the differential voltage drops by 200mV,
NDRV is enabled and FLT deasserts. The PWM dim-
ming MOSFET is still controlled by the PWMDIM input.
Fault Indicator (
FLT
)
The MAX16834 features an active-low, open-drain fault
indicator (FLT). FLT asserts when one of the following
occurs:
1) Overvoltage across the LED string
2) Short-circuit condition across the LED string, or
3) Overtemperature condition
When the output voltage drops below the overvoltage
set point minus the hysteresis, FLT deasserts. Similarly
during the short-circuit period, the fault signal
deasserts when the dimming MOSFET is on, which
happens every hiccup cycle during short circuit. During
overtemperature fault, the FLT signal is the inverse of
the PWM input.
Applications Information
Setting the UVLO Threshold
The UVLO threshold is set by resistors R1 and R2 (see
Figure 2). The MAX16834 turns on when the voltage
across R2 exceeds 1.435V, the UVLO threshold. Use
the following equation to set the desired UVLO thresh-
old:
In a typical application, use a 10kΩresistor for R2 and
then calculate R1 based on the desired UVLO threshold.
Setting the Overvoltage Threshold
The overvoltage threshold is set by resistors R4 and R9
(see Figure 2). The overvoltage circuit in the MAX16834
is activated when the voltage on OVP+ with respect to
LV exceeds 1.435V. Use the following equation to set
the desired overvoltage threshold:
VVRRR
OV =+.( )1 435 4 9 9
VVRRR
UVEN =+.( )1 435 1 2 2
f (kHz) 5000k
RT(k ) (kHz)
OSC =×
Ω
Ω
MAX16834
10
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Programming the LED Current
The LED current is programmed using the voltage on
REFI and the LED current-sense resistor R10 (see
Figure 2). The current is given by:
where VREF is 3.7V and the resistors R5, R6, and R10
are in ohms. The regulation voltage on the LED current-
sense resistor must not exceed 0.3V to prevent activa-
tion of the LED short-circuit protection circuit.
Boost Configuration
In the boost converter (Figure 2), the average inductor
current varies with the line voltage. The maximum aver-
age current occurs at the lowest line voltage. For the
boost converter, the average inductor current is equal
to the input current.
Calculate maximum duty cycle using the below equation.
where VLED is the forward voltage of the LED string in
volts, VDis the forward drop of the rectifier diode D1 in
volts (approximately 0.6V), VINMIN is the minimum input
supply voltage in volts, and VFET is the average drain to
source voltage of the MOSFET Q1 in volts when it is on.
Use an approximate value of 0.2V initially to calculate
DMAX. A more accurate value of the maximum duty
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.
Use the following equations to calculate the maximum
average inductor current ILAVG, peak-to-peak inductor
current ripple ∆IL, and the peak inductor current ILPin
amperes:
DVVV
VVV
MAX
LED D INMIN
LEDDFET
=+−
+−
IVR
RRR A
LED REF
=×
×+×
()
5
10 6 5 9 9().
VIN
LV
NDRV Q1
ON
OFF
CS
C1
PWMDIM
SGND
D1
R10
R8
MAX16834
FLT
DIMOUT
SENSE+
OVP+
CLV
COMP
PGND
C3
UVEN
HV
SC
RT/SYNC
C2
IN
L1
VCC
C5
REF
C4
R3
REFI
R5
R6
R2
R1
R9
R4
C7
R7
Q2
C6
C8
LED+
LEDs
LED-
Figure 2. Boost LED Driver
MAX16834
Maxim Integrated
11
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Allowing the peak-to-peak inductor ripple (∆IL) to be
±30% of the average inductor current:
and
The inductance value (L) of the inductor L1 in henries
(H) is calculated as:
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and ∆IL is in amperes.
Choose an inductor that has a minimum inductance
greater than the calculated value. The current rating of
the inductor should be higher than ILPat the operating
temperature.
Boost-Buck Configuration
In the boost-buck LED driver (Figure 3), the average
inductor current is equal to the input current plus the
LED current.
Calculate maximum duty cycle using the following
equation:
where VLED is the forward voltage of the LED string in
volts, VDis the forward drop of the rectifier diode D1
(approximately 0.6V) in volts, VINMIN is the minimum
input supply voltage in volts, and VFET is the average
drain to source voltage of the MOSFET Q1 in volts when
it is on. Use an approximate value of 0.2V initially to cal-
culate DMAX. A more accurate value of maximum duty
cycle can be calculated once the power MOSFET is
selected based on the maximum inductor current.
Use the below equations to calculate the maximum
average inductor current ILAVG, peak-to-peak inductor
current ripple ∆IL, and the peak inductor current ILPin
amperes:
Allowing the peak-to-peak inductor ripple ∆ILto be
±30% of the average inductor current:
The inductance value (L) of the inductor L1 in henries is
calculated as:
where fSW is the switching frequency in hertz, VINMIN
and VFET are in volts, and ∆IL is in amperes. Choose an
inductor that has a minimum inductance greater than
the calculated value.
Peak Current-Sense Resistor (R8)
The value of the switch current-sense resistor R8 for the
boost and boost-buck configurations is calculated as
follows:
where 0.25V is the minimum peak current-sense thresh-
old, ILPis the peak inductor current in amperes, and
the factor 1.25 provides a 25% margin to account for
tolerances. The worst cycle-by-cycle current limiter trig-
gers at 350mV (max). The ISAT of the inductor should
be higher than 0.35V/R8.
Output Capacitor
The function of the output capacitor is to reduce the
output ripple to acceptable levels. The ESR, ESL, and
the bulk capacitance of the output capacitor contribute
to the output ripple. In most applications, the output
ESR and ESL effects can be dramatically reduced by
using low-ESR ceramic capacitors. To reduce the ESL
and ESR effects, connect multiple ceramic capacitors
in parallel to achieve the required bulk capacitance. To
minimize audible noise generated by the ceramic
capacitors during PWM dimming, it may be necessary
to minimize the number of ceramic capacitors on the
output. In these cases an additional electrolytic or tan-
talum capacitor provides most of the bulk capacitance.
Boost and boost-buck configurations: The calcula-
tion of the output capacitance is the same for both
boost and boost-buck configurations. The output ripple
is caused by the ESR and the bulk capacitance of the
output capacitor if the ESL effect is considered negligi-
ble. For simplicity, assume that the contributions from
RILP
8025
125
=×
.
(.)
Ω
LVVD
fI
INMIN FET MAX
SW L
=−×
×
()
∆
∆
∆
IIL
IL IL I
LAVG
PAVG
L
=××
=+
03 2
2
.
IL I
D
AVG LED
MAX
=−1
DVV
VVV V
MAX LED D
LED D INMIN FET
=+
++ −
LVVD
fI
INMIN FET MAX
SW L
=−×
×
()
∆
IL IL I
PAVG
L
=+
∆
2
∆IIL
LAVG
=××03 2.
IL I
D
AVG LED
MAX
=−1
MAX16834
12
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
ESR and the bulk capacitance are equal, allowing 50%
of the ripple for the bulk capacitance. The capacitance
is given by:
where ILED is in amperes, COUT is in farads, fSW is in
hertz, and ∆VOUTRIPPLE is in volts. The remaining 50%
of allowable ripple is for the ESR of the output capaci-
tor. Based on this, the ESR of the output capacitor is
given by:
where ILPis the peak inductor current in amperes.
Use the below equation to calculate the RMS current
rating of the output capacitor:
Input Capacitor
The input filter capacitor bypasses the ripple current
drawn by the converter and reduces the amplitude of
high-frequency current conducted to the input supply.
The ESR, ESL, and the bulk capacitance of the input
capacitor contribute to the input ripple. Use a low-ESR
input capacitor that can handle the maximum input
RMS ripple current from the converter.
For the boost configuration, the input current is the
same as the inductor current. For boost-buck
I(IL (1 D )) D
IL
COUT(RMS) AVG MAX 2MAX
A
=××
+
-
(VVG ××DD
MAX MAX
)( )
21-
ESR V
IL
COUT OUTRIPPLE
P
<×
∆Ω()
()2
CID
Vf
OUT LED MAX
OUTRIPPLE SW
≥××
×
2
∆
VIN
LV
NDRV Q1
ON
OFF
CS
C1
PWMDIM
SGND
D1
R10
R8
MAX16834
HV
DIMOUT
SENSE+
OVP+
CLV
COMP
PGND
VIN
UVEN
SC
RT/SYNC
C2
IN
L1
VCC
C5
REF
C4
R3
REFI
R5
R6
R2
R1
R9
R4
C7
R7
FLT
Q2
C6
C8
LED+
LEDs
LED-
C3
Figure 3. Boost-Buck LED Driver (VLED+ < 28V)
MAX16834
Maxim Integrated
13
configuration, the input current is the inductor current
minus the LED current. But for both configurations, the
ripple current that the input filter capacitor has to sup-
ply is the same as the inductor ripple current with the
condition that the output filter capacitor should be con-
nected to ground for boost-buck configuration. This
reduces the size of the input capacitor, as the inductor
current is continuous with maximum ±30% ripple.
Neglecting the effect of LED current ripple, the calcula-
tion of the input capacitor for boost as well as boost-
buck configurations is the same.
Neglecting the effect of the ESL, the ESR, and the bulk
capacitance at the input contributes to the input voltage
ripple. For simplicity, assume that the contribution from
the ESR and the bulk capacitance is equal. This allows
50% of the ripple for the bulk capacitance. The capaci-
tance is given by:
where ∆ILis in amperes, CIN is in farads, fSW is in hertz,
and ∆VIN is in volts. The remaining 50% of allowable
ripple is for the ESR of the output capacitor. Based on
this, the ESR of the input capacitor is given by:
where ∆IL is in amperes, ESRCIN is in ohms, and ∆VIN
is in volts.
Use the below equation to calculate the RMS current
rating of the input capacitor:
Slope Compensation
Slope compensation should be added to converters
with peak current-mode control operating in continuous
conduction mode with more than 50% duty cycle to
avoid current loop instability and subharmonic oscilla-
tions. The minimum amount of slope added to the peak
inductor current to stabilize the current control loop is
half of the falling slope of the inductor.
In the MAX16834, the slope compensating ramp is
added to the current-sense signal before it is fed to the
PWM comparator. Connect a capacitor (C2 in the appli-
cation circuit) from SC to ground for slope compensa-
tion. This capacitor is charged with a 100µA current
source and discharged at the beginning of each switch-
ing cycle to generate the slope compensation ramp.
The value of the slope compensation capacitor C2 is
calculated as shown below:
Boost configuration:
where C2 is in farads, L is the inductance of the induc-
tor L1 in henries, 100µA is the pullup current from SC,
VLED and VINMIN are in volts, and R8 is the switch cur-
rent-sense resistor in ohms.
Boost-buck configuration:
where C2 is in farads, L is the inductance of the induc-
tor L1 in henries, 100µA is the pullup current from SC,
VLED is in volts, and R8 is the switch current-sense
resistor in ohms.
Selection of Power Semiconductors
Switching MOSFET
The switching MOSFET (Q1) should have a voltage rat-
ing sufficient to withstand the maximum output voltage
together with the diode drop of the rectifier diode D1
and any possible overshoot due to ringing caused by
parasitic inductances and capacitances. Use a
MOSFET with a drain-to-source voltage rating higher
than that calculated by the following equations:
Boost configuration:
where VDS is the drain-to-source voltage in volts and
VDis the forward drop of the rectifier diode D1. The fac-
tor of 1.2 provides a 20% safety margin.
Boost-buck configuration:
where VDS is the drain-to-source voltage in volts and
VDis the forward drop of the rectifier diode D1. The fac-
tor of 1.2 provides a 20% safety margin.
The continuous drain current rating of the selected
MOSFET, when the case temperature is at +70°C,
should be greater than the value calculated by the fol-
VVV V
DS LED INMAX D
=+ +
()
×12.
VVV
DS LED D
=+
()
×12.
C2 (V ) R8
-
LED
=×× ×
××
3 100 10
2
6
L
C2 (V - V ) R8
-
LED INMIN
=×× ×
××
3 100 10
2
6
L
I3
CIN(RMS) =∆IL
2
ESR V
I
CIN IN
L
<×
∆
∆2
CI
Vf
IN L
IN SW
≥××
∆
∆4
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
14
Maxim Integrated
lowing equation. The MOSFET must be mounted on a
board as per manufacturer specifications to dissipate
the heat.
The RMS current rating of the switching MOSFET Q1 is
calculated as follows for boost and boost-buck configu-
rations:
where IDRMS is the MOSFET Q1’s drain RMS current in
amperes.
The MOSFET Q1 will dissipate power due to both
switching losses as well as conduction losses. The con-
duction losses in the MOSFET is calculated as follows:
where RDSON is the on-resistance of Q1 in ohms with
an assumed junction temperature of +100°C, PCOND is
in watts, and ILAVG is in amperes.
Use the following equations to calculate the switching
losses in the MOSFET:
Boost configuration:
Boost-buck configuration:
where IGON and IGOFF are the gate currents of the
MOSFET Q1 in amperes when it is turned on and
turned off, respectively, VLED and VINMAX are in volts,
ILAVG is in amperes, fSW is in hertz, and CGD is the
gate-to-drain MOSFET capacitance in farads.
Choose a MOSFET that has a higher power rating than
that calculated by the following equation when the
MOSFET case temperature is at +70°C:
Rectifier Diode
Use a Schottky diode as the rectifier (D1) for fast
switching and to reduce power dissipation. The select-
ed Schottky diode must have a voltage rating 20%
above the maximum converter output voltage. The max-
imum converter output voltage is VLED in boost configu-
ration and VLED + VINMAX in boost-buck configuration.
The current rating of the diode should be greater than
ID in the following equation:
Dimming MOSFET
Select a dimming MOSFET (Q2) with continuous current
rating at +70°C, higher than the LED current by 30%.
The drain-to-source voltage rating of the dimming
MOSFET must be higher than VLED by 20%.
Feedback Compensation
The LED current control loop comprising of the switch-
ing converter, the LED current amplifier, and the error
amplifier should be compensated for stable control of
the LED current. The switching converter small-signal
transfer function has a right half-plane (RHP) zero for
both boost and boost-buck configurations as the induc-
tor current is in continuous conduction mode. The RHP
zero adds a 20dB/decade gain together with a 90°
phase lag, which is difficult to compensate. The easiest
way to avoid this zero is to roll off the loop gain to 0dB
at a frequency less than one-fifth of the RHP zero fre-
quency with a -20dB/decade slope.
The worst-case RHP zero frequency (fZRHP) is calculat-
ed as follows:
Boost configuration:
Boost-buck configuration:
where fZRHP is in hertz, VLED is in volts, L is the induc-
tance value of L1 in henries (H), and ILED is in amperes.
The switching converter small-signal transfer function
also has an output pole for both boost and boost-buck
configurations. The effective output impedance that
determines the output pole frequency together with the
output filter capacitance is calculated as:
fD
D
ZRHP MAX
MAX
=×
×× ×
V
2LI
LED
LED
()12
-
π
fD
ZRHP MAX
=×
××
V
2LI
LED
LED
()12
-
π
IIL D
DAVG MAX
=× ×().115-
PWP WPW
TOT COND SW
() () ()=+
PIL V V C f
I
SW AVG LED INMAX GD SW
=×+ ××
⎛
⎝
⎜⎞
⎠
⎟
×
()
2
2
1
GGIG
ON OFF
+
⎛
⎝
⎜⎞
⎠
⎟
1
PIL V C f
IG IG
SW AVG LED GD SW
ON OF
=×××
⎛
⎝
⎜⎞
⎠
⎟
×+
2
2
11
FF
⎛
⎝
⎜⎞
⎠
⎟
PILDR
COND AVG MAX DSON
=
()
××
2
ID IL D
RMS AVG MAX
=
()
×
⎛
⎝
⎜⎞
⎠
⎟×
213.
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
Maxim Integrated
15
Boost configuration:
Boost-buck configuration:
where RLED is the dynamic impedance (rate of change
of voltage with current) of the LED string at the operat-
ing current, R10 is the LED current-sense resistor in
ohms, VLED is in volts, and ILED is in amperes.
The output pole frequency for both boost and boost-
buck configurations is calculated as follows:
where fP2 is in hertz, COUT is the output filter capaci-
tance in farads, ROUT is the effective output impedance
in ohms calculated above.
Compensation components R7 and C7 perform two
functions. C7 introduces a low-frequency pole that
introduces a -20dB/decade slope into the loop gain. R7
flattens the gain of the error amplifier for frequencies
above the zero formed by R7 and C7. For compensa-
tion, this zero is placed at the output pole frequency fP2
such that it provides a -20dB/decade slope for frequen-
cies above fP2 for the complete loop gain.
The value of R7 needed to fix the total loop gain at fP2
such that the total loop gain crosses 0dB at
-20dB/decade at one-fifth of the RHP zero can be cal-
culated as follows:
where R7 is the compensation resistor in ohms, fZRHP
and fP2 are in hertz, R8 is the switch current-sense
resistor in ohms, R10 is the LED current-sense resistor
in ohms, factor 9.9 is the gain of the LED current ampli-
fier, and GMCOMP is the transconductance of the error
amplifier in Siemens.
The value of C7 can be calculated as:
where C7 is in farads, fP2 is in hertz, and R7 is in ohms.
To minimize switching frequency noise, an additional
capacitor can be added in parallel with the series com-
bination of R7 and C7. The pole from this capacitor and
R7 must be a decade higher than the loop crossover
frequency.
Short-Circuit Protection
Boost Configuration
In the boost configuration (Figure 2), if the LED string is
shorted then the excess current flowing in the LED cur-
rent-sense resistor will cause NDRV to stop switching.
The input voltage will appear on the output capacitor,
and this causes very high peak currents to flow in the
LED current-sense resistor R10 because the dimming
MOSFET (Q2) is on. Once the voltage across the LED
current-sense resistor exceeds 300mV for more than
5µs, then the dimming MOSFET Q2 turns off and stays
off for 4096 switching clock cycles. At the same time,
NDRV is also off. The MAX16834 goes into the hiccup
mode and recovers from hiccup once the short has
been removed. The power dissipation in the dimming
MOSFET (Q2) is minimized during a short across the
LED string. During the same period, FLT only goes high
when the dimming MOSFET is on.
Boost-Buck Configuration
In the case of the boost-buck configuration (Figure 3),
once an LED string short occurs then the behavior is
different. A short across the LED string causes a high
current spike due to the external capacitors at the out-
put. The regulation loop will cause NDRV to stop
switching. This causes the voltage on HV to drop if its
voltage is derived from LED+. The voltage on CLV will
drop, and this drop is detected after 128 clock cycles.
The dimming MOSFET and the switching MOSFET will
stop switching. It stays off for 4096 clock cycles, and
the cycle repeats itself. The short across the LED string
will cause the MAX16834 to go into a hiccup mode. At
the same time the FLT signal asserts itself for 4096
clock cycles every hiccup cycle. In the case where the
HV voltage is derived from a source different than
LED+, then the LED current will stay in regulation even
during a short across the LED string. In this case, FLT
does not assert itself during the short.
CRf
P
71
72
=××2π
RfR
fDR GM
ZRHP
PMAX COMP
78
51 1099
2
=×
××− × ××().
fCR
POUT OUT
2
1
=××2π
RRRV
RRID V
OUT LED LED
LED LED MAX
=+×
+×× +
()
()
10
10 LLED
RRRV
RRIV
OUT LED LED
LED LED LED
=+×
+×+
()
()
10
10
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
16
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
VIN
LV
NDRV Q1
ON
OFF
CS
C1
Q3
D2
24V
PWMDIM
SGND
D1
R10
R8
MAX16834
FLT
DIMOUT
SENSE+
OVP+
CLV
COMP
PGND
C3
UVEN
HV
SC
RT/SYNC
C2
C8
IN
L1
VCC
C5
REF
C4
R3
REFI
R5
R6
R2
R1
R9
R4
C7
R7
Q2
C6
C9
LED+
LEDs
LED-
Figure 4. Boost LED Driver with Automotive Load Dump Protection
MAX16834
Maxim Integrated
17
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
VIN
LV
NDRV Q1
ON
OFF
CS
C1
PWMDIM
SGND R10
R8
MAX16834
HV
DIMOUT
SENSE+
OVP+
CLV
COMP
PGND
VLV
VLV
UVEN
SC
RT/SYNC
C2
IN
VCC
C5
REF
C4
R3
REFI
R5
R6
R2
R1
R9
R4
C7
R7
FLT
Q2
C6
C8
LED+
LEDs
LED-
L1
VLV
D1 C3
Figure 5. High-Side Buck LED Driver
MAX16834
18
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
VIN
LV
NDRV Q1
CS
C1
PWMDIM
SGND
D1
R10
R8
MAX16834
FLT
DIMOUT
SENSE+
OVP+
CLV
COMP
PGND
VOUT
VREF
UVEN
HV
SC
RT/SYNC
C2
IN
L1
VCC
C5
REF
C4
R3
REFI
R5
R6
R2
R1
R9
C3 R4
OPTIONAL C7
C6
R7
Figure 6. Boost DC-DC Converter
MAX16834
Maxim Integrated
19
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
VIN
LV
NDRV Q1
CS
C1
PWMDIM
SGND
D1
R10
R8
MAX16834
HV
DIMOUT
SENSE+
OVP+
CLV
COMP
PGND
VOUT
VREF
VIN
UVEN
FLT
SC
RT/SYNC
C2
IN
L1
VCC
C5
REF
C4
R3
REFI
R5
R6
R2
R1
R9
C3 R4 R11
C7
C6
N.C.
R7
Figure 7. Boost-Buck DC-DC Converter
MAX16834
20
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Layout Recommendations
Typically, there are two sources of noise emission in a
switching power supply: high di/dt loops and high dv/dt
surfaces. For example, traces that carry the drain cur-
rent often form high di/dt loops. Similarly, the heatsink
of the MOSFET connected to the device drain presents
a dv/dt source; therefore, minimize the surface area of
the heatsink as much as is compatible with the MOS-
FET power dissipation or shield it. Keep all PCB traces
carrying switching currents as short as possible to mini-
mize current loops. Use ground planes for best results.
Careful PCB layout is critical to achieve low switching
losses and clean, stable operation. Use a multilayer
board whenever possible for better noise immunity and
power dissipation. Follow these guidelines for good
PCB layout:
1) Use a large contiguous copper plane under the
MAX16834 package. Ensure that all heat-dissipat-
ing components have adequate cooling.
2) Isolate the power components and high-current
path from the sensitive analog circuitry.
3) Keep the high-current paths short, especially at the
ground terminals. This practice is essential for sta-
ble, jitter-free operation. Keep switching loops short
such that:
a) The anode of D1 must be connected very close
to the drain of the MOSFET Q1.
b) The cathode of D1 must be connected very
close to COUT.
c) COUT and the current-sense resistor R8 must be
connected directly to the ground plane.
4) Connect PGND and SGND to a star-point configura-
tion.
5) Keep the power traces and load connections short.
This practice is essential for high efficiency. Use
thick copper PCBs (2oz vs.1oz) to enhance full-load
efficiency.
6) Route high-speed switching nodes away from the
sensitive analog areas. Use an internal PCB layer
for the PGND and SGND plane as an EMI shield to
keep radiated noise away from the device, feed-
back dividers, and analog bypass capacitors.
7) To prevent discharge of the compensation capaci-
tors during the off-time of the dimming cycle,
ensure that the PCB area close to these compo-
nents has extremely low leakage. Discharge of
these capacitors due to leakage results in reduced
performance of the dimming circuitry.
MAX16834
Maxim Integrated
21
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
MAX16834
QFN, TQFN
TOP VIEW
19
20
18
17
7
6
8
SGND
REF
REFI
9
OVP+
VCC
PGND
CS
IN
1
+
2
DIMOUT
45
15 14 12 11
LV
SENSE+
UVEN
RT/SYNC
FLT
SC
COMP NDRV
3
13
CLV
16
*EP
*EP = EXPOSED PAD.
10 PWMDIM
HV
20
19
18
17
16
15
14
1
2
3
4
5
6
7
DIMOUT
CLV
HV
INSGND
OVP+
SENSE+
LV
TOP VIEW
MAX16834
VCC
NDRV
PGNDREFI
REF
138CSSC
129 PWMDIMFLT
1110 UVENRT/SYNC
COMP
TSSOP
+
Pin Configurations
Chip Information
PROCESS: BiCMOS–DMOS
PACKAGE
TYPE
PACKAGE
CODE
OUTLINE
NO.
LAND
PATTERN NO.
20 TQFN T2044-3 21-0139 90-0037
20 TSSOP U20E+1 21-0108 90-0114
20 QFN G2044Y+1 21-0576 90-0360
Package Information
For the latest package outline information and land patterns
(footprints), go to www.maxim-ic.com/packages. Note that a
“+”, “#”, or “-” in the package code indicates RoHS status only.
Package drawings may show a different suffix character, but
the drawing pertains to the package regardless of RoHS status.
MAX16834
22
Maxim Integrated
High-Power LED Driver with Integrated High-Side LED
Current Sense and PWM Dimming MOSFET Driver
Revision History
REVISION
NUMBER
REVISION
DATE DESCRIPTION PAGES
CHANGED
0 8/08 Initial release —
1 2/09
Added TSSOP package and automotive version. Also updated Electrical
Characteristics, Pin Description, Detailed Description, and LED Current-
Sense Input (SENSE+) section, Pin Configuration and Package Information
1, 2, 6, 7, 8, 9, 22
2 5/09 Added automotive version of TQFN package 1
3 1/10 Added requirement for a capacitor on the SENSE+ pin 2, 3, 4, 7, 9, 11,
13, 17–20
4 8/11 Added side wettable flank package 1, 6, 7, 22
MAX16834
23
Maxim Integrated 160 Rio Robles, San Jose, CA 95134 USA 1-408-601-1000
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are implied.
Maxim reserves the right to change the circuitry and specifications without notice at any time. The parametric values (min and max limits) shown in the Electrical
Characteristics table are guaranteed. Other parametric values quoted in this data sheet are provided for guidance.
© 2011 Maxim Integrated The Maxim logo and Maxim Integrated are trademarks of Maxim Integrated Products, Inc.