2016-2017 Microchip Technology Inc. DS20005520B-page 1
MIC28511
Features
• 4.6V to 60V Operating Input Voltage Supply
• Up to 3A Output Current
• Integrated High-Side and Low-Side N-Channel
MOSFETs
• HyperLight Load® (MIC28511-1) and Hyper
Speed Control® (MIC28511-2) Architecture
• Enable Input and Power Good (PGOOD) Output
• Programmable Current-Limit and Foldback
“Hiccup” Mode Short-Circuit Protection
• Built-In 5V Regulator for Single-Supply Operation
• Adjustable 200 kHz to 680 kHz Switching
Frequency
• Fixed 5 ms Soft-Start
• Internal Compensation and Thermal Shutdown
• Thermally-Enhanced 24-Pin 3 mm x 4 mm FQFN
Package
• –40°C to +125°C Junction Temperature Range
Applications
• Industrial Power Supplies
• Distributed Supply Regulation
• Base Station Power Supplies
• Wall Transformer Regulation
• High-Voltage Single-Board Systems
General Description
The MIC28511 is a synchronous step-down switching
regulator with internal power switches capable of
providing up to 3A output current from a wide input
supply range from 4.6V to 60V. The output voltage is
adjustable down to 0.8V with a guaranteed accuracy of
±1%. A constant switching frequency can be
programmed from 200 kHz to 680 kHz. The Hyper
Speed Control® and HyperLight Load® architectures of
the MIC28511 allow for high VIN (low VOUT) operation
and ultra-fast transient response while reducing the
required output capacitance and providing very good
light-load efficiency.
The MIC28511 offers a full suite of features to ensure
protection under fault conditions. These include
undervoltage lockout to ensure proper operation under
power sag conditions, internal soft-start to reduce
inrush current, foldback current-limit, “hiccup” mode
short-circuit protection and thermal shutdown.
Package Type
MIC28511
24-Pin 3 mm x 4 mm FQFN (FL)
DL
PGND
DH
PVIN
LX
BST
PVDD
VDD
ILIM
VIN
EN
PGOOD
SW
PGND
PGND
FREQ
SW
PGND
PGND
PVIN
7
8
9101112
18
17
16
15
14
13
24 23 22 21
20
19
1
2
3
4
5
6
PVIN
PVIN
FB
AGND
25 (PVIN)
26 (PGND)
27 (SW)
60VIN, 3A Synchronous Buck Regulator
MIC28511
DS20005520B-page 2 2016-2017 Microchip Technology Inc.
Typical Application Circuit
Functional Block Diagram
MIC28511
3x4 FQFN
4.7μF
100k
V
OUT
5V (0A TO 3A)
GND
V
IN
V – 0V
100k
EN
PGND
VIN
PVDD
FREQ
VDD
FB
SW
BST
ILIM
PVIN
MIC28511
10
100k
10k
470pF
47μF
× 2
1.91k
0.1μF
6.8μH
0.1μF
2.2μF
2.2k
SW
EN
PGND
PVIN
gm
PGOOD
VDD
BST
V
REF
0.8V
POWER GOOD
COMPARATOR
PVDD
FREQ
M1
M2
R1
R2
R
INJ
C
INJ
C
OUT
C
IN
3.3V
CFF
R
LIM
R
PGOOD
D
BST
C
BST
R3
R4
C
VDD
ON
OFF
PVDD
L
VIN
VOUT
VIN
DH
DL
10,
11,
22,
23,
26
1
2
12,
21,
27
5
4, 7, 8, 9, 2 5
6
3
LX
R
BST
18
PGND
ILIM
ZCD
LSD
HSD
17 2019
LINEAR
REGULATOR UVLO
16
THERMAL
SHUTDOWN
24 FIXED TON
ESTIMATION
SOFT-START
CONTROL
LOGIC
1413
15
X90%
CURRENT
LIMIT
DETECTION
AGND FB
COMPENSATION
2016-2017 Microchip Technology Inc. DS20005520B-page 3
MIC28511
1.0 ELECTRICAL CHARACTERISTICS
Absolute Maximum Ratings †
PVIN, VIN to PGND..................................................................................................................................... –0.3V to +65V
VDD, PVDD to PGND..................................................................................................................................... –0.3V to +6V
VBST to VSW, VLX ......................................................................................................................................... –0.3V to +6V
VBST to PGND.....................................................................................................................................–0.3V to (VIN + 6V)
VSW to PGND...................................................................................................................................–0.3V to (VIN + 0.3V)
VLX, VFREQ, VILIM, VEN to AGND .....................................................................................................–0.3V to (VIN + 0.3V)
VFB, VPG to AGND ......................................................................................................................... –0.3V to (VDD + 0.3V)
PGND to AGND ........................................................................................................................................ –0.3V to +0.3V
ESD Rating(1) (HBM) .............................................................................................................................................. 1.5 kV
ESD Rating(1) (MM) ..................................................................................................................................................150V
Operating Ratings ‡
Supply Voltage (PVIN, VIN)......................................................................................................................... +4.6V to +60V
Enable Input (VEN) ..............................................................................................................................................0V to VIN
VSW, VFREQ, VILIM ...............................................................................................................................................0V to VIN
† Notice: Stresses above those listed under “Absolute Maximum Ratings” may cause permanent damage to the device.
This is a stress rating only and functional operation of the device at those or any other conditions above those indicated
in the operational sections of this specification is not intended. Exposure to maximum rating conditions for extended
periods may affect device reliability.
‡ Notice: The device is not guaranteed to function outside its operating ratings.
Note 1: Devices are ESD sensitive. Handling precautions are recommended. Human body model, 1.5 k in series
with 100 pF.
MIC28511
DS20005520B-page 4 2016-2017 Microchip Technology Inc.
TABLE 1-1: ELECTRICAL CHARACTERISTICS
Electrical Characteristics: VIN = 12V, TA = 25°C, unless noted. Bold values indicate –40°C TJ +125°C.
(Note 1).
Parameters Min. Typ. Max. Units Conditions
Power Supply Input
Input Voltage Range (PVIN,
VIN)
4.6 —60 V—
Quiescent Supply Current — 0.4 0.75 mA VFB = 1.5V (MIC28511-1)
—0.71.5 VFB = 1.5V (MIC28511-2)
Shutdown Supply Current — 0.1 10 µA SW unconnected, VEN = 0V
VDD Supply
VDD Output Voltage 4.8 5.2 5.4 VV
IN = 7V to 60V, IVDD = 10 mA
VDD UVLO Threshold 3.8 4.2 4.6 VV
DD rising
VDD UVLO Hysteresis — 400 — mV —
Load Regulation at 40 mA 0.6 2 4.0 % —
Reference
Feedback Reference Voltage 0.792 0.8 0.808 V0°C TJ +85°C (±1.0%)
0.784 0.8 0.816 –40°C TJ +125°C (±2%)
FB Bias Current — 5 500 nA VFB = 0.8V
Enable Control
EN Logic Level High 1.8 —— V—
EN Logic Level Low — — 0.6 —
EN Hysteresis — 200 — mV —
EN Bias Current — 5 40 µA VEN = 12V
Oscillator
Switching Frequency 450 680 800 kHz VFREQ = VIN
—340 — V
FREQ = 50% x VIN
Maximum Duty Cycle — 85 — % —
Minimum Duty Cycle — 0 — VFB > 0.8V
Minimum Off-Time 110 200 270 ns —
Internal MOSFET
High-Side NMOS
On-Resistance
—51 — m—
Low-Side NMOS
On-Resistance
—28 — —
Short-Circuit Protection
Current-Limit Threshold –30 –14 0 mV VFB = 0.79V
Short-Circuit Threshold –24 –7 8 VFB = 0V
Current-Limit Source Current 50 70 90 µA VFB = 0.79V
Short-Circuit Source Current 25 36 43 VFB = 0V
Note 1: Specification for packaged product only.
2016-2017 Microchip Technology Inc. DS20005520B-page 5
MIC28511
Leakage
SW, BST Leakage Current — — 50 µA —
Power Good (PGOOD)
PGOOD Threshold Voltage 85 90 95 %VOUT Sweep VFB from low to high
PGOOD Hysteresis — 6 — Sweep VFB from high to low
PGOOD Delay Time — 100 — µs Sweep VFB from low to high
PGOOD Low Voltage — 70 200 mV VFB < 90% x VNOM, IPGOOD = 1 mA
Thermal Protection
Overtemperature Shutdown — 160 — °C TJ Rising
Overtemperature Shutdown
Hysteresis
—15 — °C—
Soft-Start
Soft-Start Time — 5 — ms —
TABLE 1-1: ELECTRICAL CHARACTERISTICS (CONTINUED)
Electrical Characteristics: VIN = 12V, TA = 25°C, unless noted. Bold values indicate –40°C TJ +125°C.
(Note 1).
Parameters Min. Typ. Max. Units Conditions
Note 1: Specification for packaged product only.
MIC28511
DS20005520B-page 6 2016-2017 Microchip Technology Inc.
TEMPERATURE SPECIFICATIONS
Parameters Sym. Min. Typ. Max. Units Conditions
Temperature Ranges
Junction Operating Temperature TJ–40 — +125 °C Note 1
Storage Temperature Range TS–65 — +150 °C —
Maximum Junction Temperature TJ— — +150 °C —
Lead Temperature — — — +300 °C Soldering, 10s
Package Thermal Resistances
Thermal Resistance 3 mm x 4 mm
FQFN-24LD
JA —30 —°C/W—
Note 1: The maximum allowable power dissipation is a function of ambient temperature, the maximum allowable
junction temperature and the thermal resistance from junction to air (i.e., TA, TJ, JA). Exceeding the
maximum allowable power dissipation will cause the device operating junction temperature to exceed the
maximum +125°C rating. Sustained junction temperatures above +125°C can impact the device reliability.
2016-2017 Microchip Technology Inc. DS20005520B-page 7
MIC28511
2.0 TYPICAL PERFORMANCE CURVES
FIGURE 2-1: VIN Operating Supply
Current vs. Input Voltage (MIC28511-1).
FIGURE 2-2: VIN Shutdown Current vs.
Input Voltage.
FIGURE 2-3: VDD Voltage vs. Input
Voltage (MIC28511-1).
FIGURE 2-4: Output Voltage vs. Input
Voltage (MIC28511-1).
FIGURE 2-5: Enable Threshold vs. Input
Voltage (MIC28511-1).
FIGURE 2-6: VDD UVLO Threshold vs.
Temperature (MIC28511-1).
Note: The graphs and tables provided following this note are a statistical summary based on a limited number of
samples and are provided for informational purposes only. The performance characteristics listed herein
are not tested or guaranteed. In some graphs or tables, the data presented may be outside the specified
operating range (e.g., outside specified power supply range) and therefore outside the warranted range.
MIC28511
DS20005520B-page 8 2016-2017 Microchip Technology Inc.
FIGURE 2-7: Output Peak Current Limit
vs. Temperature (MIC28511-1).
FIGURE 2-8: Feedback Voltage vs.
Temperature (MIC28511-1).
FIGURE 2-9: En ab le T hres ho ld vs.
Temperature (MIC28511-1).
.
FIGURE 2-10: Output Voltage vs. Output
Current (MIC2 8511-1).
FIGURE 2-11: Efficiency (VIN = 12V) vs.
Output Current (MIC28511-1).
FIGURE 2-12: Efficiency (VIN = 24V) vs.
Output Current (MIC28511-1).
2016-2017 Microchip Technology Inc. DS20005520B-page 9
MIC28511
FIGURE 2-13: Efficiency (VIN = 48V) vs.
Output Current (MIC28511-1).
FIGURE 2-14: Switching Frequency vs.
Output Current (MIC28511-1).
FIGURE 2-15: IC Power Dissipation vs.
Output Current (MIC28511-1).
FIGURE 2-16: 12V Input Thermal Derating
(MIC28511-1).
FIGURE 2-17: IC Power Dissipation vs.
Output Current (MIC28511-1).
FIGURE 2-18: IC Power Dissipation vs.
Output Current (MIC28511-1).
MIC28511
DS20005520B-page 10 2016-2017 Microchip Technology Inc.
FIGURE 2-19: 24V Input Thermal Deratin g
(MIC28511-1).
FIGURE 2-20: 48V Input Thermal Deratin g
(MIC28511-1).
FIGURE 2-21: VIN Operating Supply
Current vs. Input Voltage (MIC28511-2).
FIGURE 2-22: Output Voltage vs. Input
Voltage (MIC28511-2).
FIGURE 2-23: Feedback Voltage vs.
Temperature (MIC28511-2).
FIGURE 2-24: Output Peak Current Limit
vs. Temperature (MIC28511-2).
2016-2017 Microchip Technology Inc. DS20005520B-page 11
MIC28511
FIGURE 2-25: Output Voltage vs. Output
Current (MIC28511-2).
FIGURE 2-26: Efficiency (VIN = 12V) vs.
Output Current (MIC28511-2).
FIGURE 2-27: Efficiency (VIN = 24V) vs.
Output Current (MIC28511-2).
FIGURE 2-28: Efficiency (VIN = 48V) vs.
Output Current (MIC28511-2).
FIGURE 2-29: Switch Frequency vs.
Output Current (MIC28511-2).
FIGURE 2-30: IC Power Dissipation vs.
Output Current (MIC28511-2).
MIC28511
DS20005520B-page 12 2016-2017 Microchip Technology Inc.
FIGURE 2-31: 12V Input Thermal Deratin g
(MIC28511-2).
FIGURE 2-32: IC Power Dissipation vs.
Output Current (MIC28511-2).
FIGURE 2-33: IC Power Dissipation vs.
Output Current (MIC28511-2).
FIGURE 2-34: 24V Input Thermal Derating
(MIC28511-2).
FIGURE 2-35: 48V Input Thermal Derating
(MIC28511-2).
FIGURE 2-36: Enable Turn-On Delay and
Rise Time.
Time (2.0ms/div)
IL
(2A/div)
VEN
(2V/div)
VOUT
(2V/div)
VIN = 12V
VOUT = 5V
IOUT = 3A
2016-2017 Microchip Technology Inc. DS20005520B-page 13
MIC28511
FIGURE 2-37: Start-Up with Pre-Biased
Output at 1V.
FIGURE 2-38: Start-Up with Pre-Biased
Output at 2V.
FIGURE 2-39: Enable Turn-Off Delay and
Fall Time.
FIGURE 2-40: VIN Turn -O n .
FIGURE 2-41: VIN Turn-Off.
FIGURE 2-42: Enable Turn-On/Turn-Off.
Time (2.0ms/div)
IL
(2A/div)
VEN
(2V/div)
VOUT
(2V/div)
VIN = 12V
VOUT = 5V
IOUT = 3A
Time (10ms/div)
VEN
(2V/div)
VOUT
(2V/div)
VIN = 12V
VOUT = 5V
IOUT = 3A
MIC28511
DS20005520B-page 14 2016-2017 Microchip Technology Inc.
FIGURE 2-43: VIN UVLO Thresholds.
FIGURE 2-44: Power-Up into Short-Circuit.
FIGURE 2-45: Enabled into Short.
FIGURE 2-46: Output Peak Current-Limit
Threshold.
FIGURE 2-47: Short-Circuit.
FIGURE 2-48: Output Recovery from
Short-Circuit.
Time (20ms/div)
VIN
(2V/div)
VOUT
(2V/div)
VOUT = 3.3V
IOUT = 0.5A
VEN = 5V
Time (200ms/div)
VIN
(5V/div)
VOUT
(10mV/div)
VIN = 12V
VOUT = 5V
IOUT = SHORT
IL
(2A/div)
Time (4ms/div)
IL
(2A/div)
VEN
(2V/div)
VOUT
(10mV/div)
VIN = 12V
VOUT = 5V
IOUT = SHORT
Time (40ms/div)
IL
(5A/div)
VOUT
(5V/div)
VIN
= 12V
VOUT
= 5V
Time (20μs/div)
VOUT
(2V/div)
VIN = 12V
VOUT = 5V
IL
(2A/div)
Time (200μs/div)
VOUT
(2V/div)
VIN = 12V
VOUT = 5V
IOUT = 3A TO SHORT
IL
(2A/div)
2016-2017 Microchip Technology Inc. DS20005520B-page 15
MIC28511
FIGURE 2-49: Output Recovery from
Short-Circuit (Zoomed-In Version).
FIGURE 2-50: Output Recovery from
Thermal Shutdown.
FIGURE 2-51: MIC28511-2 Switching
Waveforms, IOUT = 3A.
FIGURE 2-52: MIC28511-2 Switching
Waveforms, IOUT = 0A.
FIGURE 2-53: MIC28511-1 Switching
Waveforms, IOUT = 0A.
FIGURE 2-54: MIC28511-1 Switching
Waveforms, IOUT = 3A.
Time (100μs/div)
VOUT
(2V/div)
VIN
= 12V
VOUT
= 5V
IOUT = 3A TO SHORT
IL
(2A/div)
Time (4.0ms/div)
VOUT
(2V/div)
VIN
= 12V
VOUT
= 5V
IOUT = 1A
VSW
(5V/div)
Time (1.0μs/div)
VOUT
(10mV/div)
(AC-COUPLED)
VIN
= 12V
VOUT
= 5V
IOUT = 3A
VSW
(10V/div)
IL
(2$/div)
Time (1.0μs/div)
VOUT
(10mV/div)
(AC-COUPLED)
VIN
= 12V, VOUT
= 5V, IOUT = 0A
VSW
(10V/div)
IL
($/div)
Time (400μs/div)
VOUT
(50mV/div)
(AC-COUPLED)
VIN
= 12V, VOUT
= 5V, IOUT = 0A
VSW
(10V/div)
IL
(1$/div)
Time (1μs/div)
VOUT
(20mV/div)
(AC-COUPLED)
VIN
= 12V
VOUT
= 5V
IOUT = 3A
VSW
(10V/div)
IL
(2$/div)
MIC28511
DS20005520B-page 16 2016-2017 Microchip Technology Inc.
FIGURE 2-55: MIC28511-2 Transient
Response.
FIGURE 2-56: MIC28511-1 Transient
Response.
FIGURE 2-57: Power Good at VIN Soft
Turn-On.
FIGURE 2-58: Power Good at VIN Soft
Turn-Off.
Time (100μs/div)
VOUT
(500mV/div)
(AC-COUPLED)
VIN
= 12V
VOUT
= 5V
IOUT = 0A TO 3A
IOUT
(2A/div)
Time (100μs/div)
VOUT
(500mV/div)
(AC-COUPLED)
VIN
= 12V
VOUT
= 5V
IOUT = 0A TO 3A
IOUT
(2A/div)
Time (400ms/div)
VIN
(10V/div)
VOUT
(5V/div)
VIN = 12V
VOUT = 5V
IOUT = 0A
VPG
(5V/div)
Time (20.0ms/div)
VIN
(10V/div)
VOUT
(5V/div)
VIN
= 12V
VOUT
= 5V
IOUT
= 0A
VPG
(5V/div)
2016-2017 Microchip Technology Inc. DS20005520B-page 17
MIC28511
3.0 PIN DESCRIPTIONS
The descriptions of the pins are listed in Table 3-1.
TABLE 3-1: PIN FUNCTION TABLE
Pin Number Symbol Description
1 DL Low-Side Gate Drive. Internal low-side power MOSFET gate connection. This pin must
be left unconnected or floating.
2 PGND PGND is the return path for the low-side driver circuit. Connect to the source of low-side
MOSFETs (PGND, pins 10, 11 22, 23 and 26) through a low-impedance path.
3 DH High-Side Gate Drive. Internal high-side power MOSFET gate connection. This pin
must be left unconnected or floating.
4, 7, 8, 9, 25
(25 is ePad)
PVIN Power Input Voltage. The PVIN pins supply power to the internal power switch. Connect
all PVIN pins together and bypass locally with ceramic capacitors. The positive terminal
of the input capacitor should be placed as close as possible to the PVIN pins, the
negative terminal of the input capacitor should be placed as close as possible to the
PGND pins 10,11, 22, 23, and 26.
5 LX The LX pin is the return path for the high-side driver circuit. Connect the negative
terminal of the bootstrap capacitor directly to this pin. Also connect this pin to the SW
pins 12, 21, and 27, with a low-impedance path. The controller monitors voltages on this
and PGND for zero current detection.
6 BST Bootstrap Pin. This pin provides bootstrap supply for the high-side gate driver circuit.
Connect a 0.1 µF capacitor and an optional resistor in series from the LX (pin 5) to the
BST.
10, 11, 22, 23,
26
(26 is ePad)
PGND Power Ground. These pins are connected to the source of the low-side MOSFET. They
are the return path for the step-down regulator power stage and should be tied together.
The negative terminal of the input decoupling capacitor should be placed as close as
possible to these pins.
12, 21, 27
(27 is ePad)
SW Switch Node. The SW pins are the internal power switch outputs. These pins should be
tied together and connected to the output inductor.
13 AGND Analog Ground. The analog ground for VDD and the control circuitry. The analog ground
return path should be separate from the power ground (PGND) return path.
14 FB Feedback Input. The FB pin sets the regulated output voltage relative to the internal
reference. This pin is connected to a resistor divider from the regulated output such that
the FB pin is at 0.8V when the output is at the desired voltage.
15 PGOOD The power good output is an open drain output requiring an external pull-up resistor to
external bias. This pin is a high impedance open circuit when the voltage at FB pin is
higher than 90% of the feedback reference voltage (typically 0.8V).
16 EN Enable Input. The EN pin enables the regulator. When the pin is pulled below the
threshold, the regulator will shut down to an ultra-low current state. A precise threshold
voltage allows the pin to operate as an accurate UVLO. Do not tie EN to VDD
17 VIN Supply voltage for the internal LDO. The VIN operating voltage range is from 4.6V to
60V. A ceramic capacitor from VIN to AGND is required for decoupling. The decoupling
capacitor should be placed as close as possible to the supply pin.
18 ILIM Current Limit Setting. Connect a resistor from this pin to the SW pin node to allow for
accurate current limit sensing programming of the internal low-side power MOSFET.
19 VDD Internal +5V Linear Regulator: VDD is the internal supply bus for the IC. Connect to an
external 1 µF bypass capacitor. When VIN is <5.5V, this regulator operates in drop-out
mode. Connect VDD to VIN.
20 PVDD A 5V supply input for the low-side N-channel MOSFET driver circuit, which can be tied
to VDD externally. A 1 F ceramic capacitor from PVDD to PGND is recommended for
decoupling.
24 FREQ Switching Frequency Adjust pin. Connect this pin to VIN to operate at 680 kHz. Place a
resistor divider network from VIN to the FREQ pin to program the switching frequency.
MIC28511
DS20005520B-page 18 2016-2017 Microchip Technology Inc.
4.0 FUNCTIONAL DESCRIPTION
The MIC28511 is an adaptive on-time synchronous
buck regulator with integrated high-side and low-side
MOSFETs suitable for high-input voltage to low-output
voltage conversion applications. It is designed to
operate over a wide input voltage range (4.6V to 60V),
which is suitable for automotive and industrial
applications. The output is adjustable with an external
resistive divider. An adaptive on-time control scheme is
employed to produce a constant switching frequency in
continuous-conduction mode and reduced switching
frequency in discontinuous-operation mode, improving
light-load efficiency. Overcurrent protection is
implemented by sensing low-side MOSFET’s RDS(ON).
The device features internal soft-start, enable, UVLO,
and thermal shutdown.
4.1 Theory of Operation
As illustrated in the Functional Block Diagram, the
output voltage of the MIC28511 is sensed by the
feedback (FB) pin via voltage dividers R1 and R2, and
compared to a 0.8V reference voltage VREF at the error
comparator through a low-gain transconductance (gM)
amplifier. If the feedback voltage decreases and the
amplifier output is below 0.8V, then the error
comparator will trigger the control logic and generate
an ON-time period. The ON-time period length is
predetermined by the fixed tON estimator circuitry:
EQUATION 4-1:
At the end of the ON-time period, the internal high-side
driver turns off the high-side MOSFET and the low-side
driver turns on the low-side MOSFET. The OFF-time
period length depends upon the feedback voltage in
most cases. When the feedback voltage decreases
and the output of the gM amplifier is below 0.8V, then
the ON-time period is triggered and the OFF-time
period ends. If the OFF-time period determined by the
feedback voltage is less than the minimum OFF-time
tOFF(MIN), which is about 200 ns (typical), the MIC28511
control logic will apply the tOFF(MIN) instead. The
tOFF(MIN) is required to maintain enough energy in the
boost capacitor (CBST) to drive the high-side MOSFET.
The maximum duty cycle is obtained from
Equation 4-2.
EQUATION 4-2:
It is not recommended to use MIC28511 with an
OFF-time close to tOFF(MIN) during steady-state
operation.
The adaptive ON-time control scheme results in a
constant switching frequency in the MIC28511. The
actual ON-time and resulting switching frequency will
vary with the different rising and falling times of the
external MOSFETs. Also, the minimum tON results in a
lower switching frequency in high VIN to VOUT
applications. During load transients, the switching
frequency is changed due to the varying OFF-time.
Figure 4-1 shows the allowable range of the output
voltage versus the input voltage. The minimum output
voltage is 0.8V, which is limited by the reference
voltage. The maximum output voltage is 24V, which is
limited by the internal circuitry.
FIGURE 4-1: Allowable Output Voltage
Range vs. Input Voltage.
To illustrate the control loop operation, both the
steady-state and load transient scenarios will be
analyzed.
Figure 4-2 shows the MIC28511 control loop timing
during steady-state operation. During steady-state, the
gM amplifier senses the feedback voltage ripple, which
is proportional to the output voltage ripple and the
inductor current ripple, to trigger the ON-time period.
The ON-time is predetermined by the tON estimator.
The termination of the OFF-time is controlled by the
feedback voltage. At the valley of the feedback voltage
ripple, which occurs when VFB falls below VREF
, the
OFF period ends and the next ON-time period is
triggered through the control logic circuitry.
tON ESTIMATED
VOUT
VIN fSW
-----------------------=
Where:
VOUT Output Voltage
VIN Power Stage Input Voltage
fSW Switching Frequency
DMAX 1tOFF MIN
–fSW
=
pg
fSW = 600kHz
fSW = 400kHz
fSW = 200kHz
ALLOWABLE RANGE
0.8V (MINIMUM)
5
INPUT VOLTAGE (V)
15 25 35 45 55
OUTPUT VOLTAGE (V)
30
25
20
15
10
5
0
2016-2017 Microchip Technology Inc. DS20005520B-page 19
MIC28511
FIGURE 4-2: MIC28511 Control Loop
Timing.
Figure 4-3 shows the operation of the MIC28511 during
a load transient. The output voltage drops due to the
sudden load increase, which causes the VFB to be less
than VREF
. This will cause the error comparator to
trigger an ON-time period. At the end of the ON-time
period, a minimum OFF-time tOFF(MIN) is generated to
charge CBST because the feedback voltage is still
below VREF
. Then, the next ON-time period is triggered
due to the low feedback voltage. Therefore, the
switching frequency changes during the load transient,
but returns to the nominal fixed frequency once the
output has stabilized at the new load current level. With
the varying duty cycle and switching frequency, the
output recovery time is fast and the output voltage
deviation is small in MIC28511 converter.
FIGURE 4-3: MIC28511 Load Transient
Response.
Unlike true current-mode control, the MIC28511 uses
the output voltage ripple to trigger an ON-time period.
The output voltage ripple is proportional to the inductor
current ripple if the ESR of the output capacitor is large
enough. The MIC28511 control loop has the advantage
of eliminating the need for slope compensation.
In order to meet the stability requirements, the
MIC28511 feedback voltage ripple should be in phase
with the inductor current ripple and large enough to be
sensed by the gM amplifier and the error comparator.
The recommended feedback voltage ripple is 20 mV ~
100 mV.
If a low-ESR output capacitor is selected, then the
feedback voltage ripple may be too small to be sensed
by the gM amplifier and the error comparator. Also, if
the ESR of the output capacitor is very low, the output
voltage ripple and the feedback voltage ripple are not
necessarily in phase with the inductor current ripple. In
these cases, ripple injection is required to ensure
proper operation. Please refer to the Ripple Injection
subsection for more details about the ripple injection
technique.
4.2 Discontinuous Mode (MIC28511-1
Only)
In continuous mode, the inductor current is always
greater than zero; however, at light loads the
MIC28511-1 is able to force the inductor current to
operate in discontinuous mode. Discontinuous mode
occurs when the inductor current falls to zero, as
indicated by trace (IL) shown in Figure 4-4. During this
period, the efficiency is optimized by shutting down all
the non-essential circuits and minimizing the supply
current. The MIC28511-1 wakes up and turns on the
high-side MOSFET when the feedback voltage VFB
drops below 0.8V.
The MIC28511-1 has a zero crossing comparator that
monitors the inductor current by sensing the voltage
drop across the low-side MOSFET during its ON-time.
If the VFB > 0.8V and the inductor current goes slightly
negative, then the MIC28511-1 automatically powers
down most of the IC circuitry and goes into a low-power
mode.
Once the MIC28511-1 goes into discontinuous mode,
both DH and DL are low, which turns off the high-side
and low-side MOSFETs. The load current is supplied
by the output capacitors and VOUT drops. If the drop of
VOUT causes VFB to go below VREF
, then all the circuits
will wake up into normal continuous mode. First, the
bias currents of most circuits reduced during the
discontinuous mode are restored, and then a tON pulse
is triggered before the drivers are turned on to avoid
any possible glitches. Finally, the high-side driver is
turned on. Figure 4-4 shows the control loop timing in
discontinuous mode.
¨IL(PP)
¨VOUT(PP) = ESRC × ¨IL(PP)
OUT
¨VFB(PP) = ¨VOUT(PP) × R2
R1 + R2
TRIGGER ON-TIME IF VFB IS BELOW VREF
ESTIMATED ON-TIME
IL
IOUT
VOUT
VFB
VREF
9'+
IOUT
VOUT
VFB
9'+
VREF
TOFF(MIN)
NO LOAD
FULL LOAD
MIC28511
DS20005520B-page 20 2016-2017 Microchip Technology Inc.
FIGURE 4-4: MIC28511-1 Control Loop
Timing (Discontinuous Mode).
During discontinuous mode, the bias current of most
circuits are reduced. As a result, the total power supply
current during discontinuous mode is only about
450 A, allowing the MIC28511-1 to achieve high
efficiency in light load applications.
4.3 VDD Regulator
The MIC28511 provides a 5V regulated VDD to bias
internal circuitry for VIN ranging from 5.5V to 60V.
When VIN is less than 5.5V, VDD should be tied to VIN
pins to bypass the internal linear regulator.
4.4 Soft-Start
Soft-start reduces the power supply inrush current at
startup by controlling the output voltage rise time while
the output capacitor charges.
The MIC28511 implements an internal digital soft-start
by ramping up the 0.8V reference voltage (VREF) from
0 to 100% in about 5 ms with 9.7 mV steps. This
controls the output voltage rate of rise at turn on,
minimizing inrush current and eliminating output
voltage overshoot. Once the soft-start cycle ends, the
related circuitry is disabled to reduce current
consumption.
4.5 Current Limit
The MIC28511 uses the RDS(ON) of the internal
low-side power MOSFET to sense overcurrent
conditions. In each switching cycle, the inductor current
is sensed by monitoring the low-side MOSFET during
its ON period. The sensed voltage, V(ILIM), is compared
with the power ground (PGND) after a blanking time of
150 ns.
The voltage drop of the resistor RILIM is compared with
the low-side MOSFET voltage drop to set the
overcurrent trip level. The small capacitor connected
from the ILIM pin to PGND can be added to filter the
switching node ringing, allowing a better short limit
measurement. The time constant created by RILIM and
the filter capacitor should be much less than the
minimum off time.
The overcurrent limit can be programmed by using
Equation 4-3:
EQUATION 4-3:
The peak-to-peak inductor current ripple is calculated
with Equation 4-4.
EQUATION 4-4:
The MOSFET RDS(ON) varies 30% to 40% with
temperature; therefore, it is recommended to use the
RDS(ON) at maximum junction temperature with a 20%
margin to calculate RILIM in Equation 4-3.
In case of hard short, the current-limit threshold is
folded down to allow an indefinite hard short on the
output without any destructive effect. It is mandatory to
make sure that the inductor current used to charge the
output capacitor during soft-start is under the folded
ESTIMATED ON-TIME
IL
ZC
VFB
VREF
VD/
0
VD+
IL CROSSES 0 AND VFB > 0.8.
DISCONTINUOUS MODE STARTS
VFB < 0.8. WAKEUP FROM
DISCONTINUOUS MODE.
RILIM ICLIM 0.5+ILPP
RDS ON
VCL
+
ICL
-----------------------------------------------------------------------------------------------------=
Where:
ICLIM Desired Output Current Limit
IL(PP) Inductor Current Peak-to-Peak
Use Equation 4-4 to calculate the
inductor ripple current
RDS(ON) On-Resistance of Low-Side Power
MOSFET
VCL Current-limit threshold.
14 mV (typical).
See the Electrical Characteristics table.
ICL Current-limit source current.
70 µA (typical). See the Electrical Char-
acteristics table.
ILPP
VOUT VIN MAX
VOUT
–
VIN MAX
fSW
L
--------------------------------------------------------------------=
2016-2017 Microchip Technology Inc. DS20005520B-page 21
MIC28511
short limit; otherwise the supply will go into hiccup
mode and may not be finishing the soft-start
successfully.
When operating at higher input voltages, it is
recommended to look at the safe operating area (SOA)
curves while selecting the output current limit
threshold. SOA curves for MIC28511 are given through
Figure 2-19 and Figure 2-20 in Section 2.0, Typical
Performance Curves.
4.6 Output Current Thermal Derating
MIC28511 is packaged in 3 mm x 4 mm FQFN
package. The output current derates with temperature
when operated with higher input voltage at higher
temperature as shown in typical operating curves in
Figure 2-19 and Figure 2-20 for MIC28511-1, and
Figure 2-34 and Figure 2-35 for MIC28511-2.
4.7 Power Good (PGOOD)
The power good (PGOOD) pin is an open-drain output
that indicates logic-high when the output is nominally
90% of its steady state voltage.
4.8 MOSFET Gate Drive
The Functional Block Diagram shows a bootstrap
circuit, consisting of DBST
, CBST
, and RBST
. This circuit
supplies energy to the high-side drive circuit. Capacitor
CBST is charged, while the low-side MOSFET is on,
and the voltage on the SW pin is approximately 0V.
When the high-side MOSFET driver is turned on,
energy from CBST is used to turn the MOSFET on. As
the high-side MOSFET turns on, the voltage on the SW
pin increases to approximately VIN. Diode DBST is
reverse-biased and CBST floats high while continuing to
bias the high-side gate driver. The bias current of the
high-side driver is less than 10 mA, so a 0.1 F to 1 F
capacitor is sufficient to hold the gate voltage with
minimal droop for the power stroke (high-side
switching) cycle, i.e.
BST = 10 mA x 1.25 s/0.1 F = 125 mV. When the
low-side MOSFET is turned back on, CBST is then
recharged through the boost diode. A 30 resistor
RBST
, which is in series with the BST pin, is required to
slow down the turn-on time of the high-side N-channel
MOSFET.
MIC28511
DS20005520B-page 22 2016-2017 Microchip Technology Inc.
5.0 APPLICATION INFORMATION
5.1 Output Voltage Setting
Components
The MIC28511 requires two resistors to set the output
voltage as shown in Figure 5-1.
FIGURE 5-1: Voltage Divider
Configuration.
The output voltage is determined by Equation 5-1.
EQUATION 5-1:
A typical value of R1 used on the standard evaluation
board is 10 k. If R1 is too large, it may allow noise to
be introduced into the voltage feedback loop. If R1 is
too small in value, it will decrease the efficiency of the
buck converter, especially at light loads. Once R1 is
selected, R2 can be calculated using Equation 5-2:
EQUATION 5-2:
5.2 Setting the Switching Frequency
The MIC28511 switching frequency can be adjusted by
changing the resistor divider network from VIN.
FIGURE 5-2: Switching Frequency
Adjustment.
Equation 5-3 gives the estimated switching frequency.
EQUATION 5-3:
Figure 5-3 shows the switching frequency versus the
resistor R17 when R19 = 100 k.
FIGURE 5-3: Switching Frequency vs.
R17.
5.3 Inductor Selection
Values for inductance, peak and RMS currents are
required to select the output inductor. The input and
output voltages and the inductance value determine
the peak-to-peak inductor ripple current. Generally,
higher inductance values are used with higher input
voltages. Larger peak-to-peak ripple currents will
increase the power dissipation in the inductor and
gM AMP
VREF
FB
R1
R2
VOUT VFB 1R1
R2
-------+
=
Where:
VFB 0.8V
R2VFB R1
VOUT VFB
–
-----------------------------=
fSW f0
R17
R17 R19+
--------------------------
=
Where:
f0Switching frequency when R17 is open;
typically 680 kHz.
10 100 1000 10000
R17 (k)
V
IN
= 60V
V
IN
= 12V
R19 = 100k, I
OUT
= 1A
SW FREQ (kHz)
700
600
500
400
300
200
100
0
2016-2017 Microchip Technology Inc. DS20005520B-page 23
MIC28511
MOSFETs. Larger output ripple currents will also
require more output capacitance to smooth out the
larger ripple current. Smaller peak-to-peak ripple
currents require a larger inductance value and
therefore a larger and more expensive inductor. A good
compromise between size, loss and cost is to set the
inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is
calculated by:
EQUATION 5-4:
In continuous conduction mode, the peak inductor
current is equal to the average output current plus one
half of the peak-to-peak inductor current ripple.
EQUATION 5-5:
The RMS inductor current is used to calculate the I2R
losses in the inductor.
EQUATION 5-6:
Maximizing efficiency requires the proper selection of
core material and minimizing the winding resistance.
The high-frequency operation of the MIC28511
requires the use of ferrite materials for all but the most
cost-sensitive applications. Lower-cost iron powder
cores may be used but the increase in core loss will
reduce the efficiency of the buck converter. This is
especially noticeable at low output power. The winding
resistance decreases efficiency at the higher output
current levels.
The winding resistance must be minimized although
this usually comes at the expense of a larger inductor.
The power dissipated in the inductor is equal to the sum
of the core and copper losses. At higher output loads,
the core losses are usually insignificant and can be
ignored. At lower output currents, the core losses can
be a significant contributor. Core loss information is
usually available from the magnetics vendor. Copper
loss in the inductor is calculated by Equation 5-7:
EQUATION 5-7:
The resistance of the copper wire, DCR, increases with
the temperature. The value of the winding resistance
used should be at the operating temperature.
EQUATION 5-8:
5.4 Output Capacitor Selection
The type of the output capacitor is usually determined
by its equivalent series resistance (ESR). Voltage and
RMS current capability are also important factors in
selecting an output capacitor. Recommended capacitor
types are ceramic, tantalum, low-ESR aluminum
electrolytic, OS-CON and POSCAP. For high ESR
electrolytic capacitors, ESR is the main cause of the
output ripple. The output capacitor ESR also affects the
control loop from a stability point of view. For a low ESR
ceramic output capacitor, ripple is dominated by the
reactive impedance. The maximum value of ESR is
calculated by Equation 5-9.
EQUATION 5-9:
The total output ripple is a combination of voltage
ripples caused by the ESR and output capacitance.
The total ripple is calculated by Equation 5-10.
EQUATION 5-10:
LVOUT VIN MAX
VOUT
–
VIN MAX
ILPP
fSW
--------------------------------------------------------------------=
Where:
fSW Switching Frequency
IL(PP) The peak-to-peak inductor current
ripple; typically 20% of the maximum
output current
ILPK IOUT 0.5+ILPP
=
ILRMS IOUT MAX
2ILPP
2
12
---------------------+=
PLCU ILRMS
2DCR=
DCR HT DCR 20C10.0042+THT20C
–=
Where:
THTemperature of wire under full load
T20C Ambient temperature
DCR(20C) Room temperature winding resistance
(usually specified by the manufacturer)
ESRCOUT
VOUT PP
ILPP
---------------------------
Where:
VOUT(PP) Peak-to-Peak Output Voltage Ripple
IL(PP) Peak-to-Peak Inductor Current Ripple
VOUT PP
ILPP
COUT fSW
8
--------------------------------------
2ILPP
ESRCOUT
2
+
=
Where:
COUT Output Capacitance Value
fSW Switching Frequency
MIC28511
DS20005520B-page 24 2016-2017 Microchip Technology Inc.
As described in the Theory of Operation subsection of
the Functional Description, the MIC28511 requires at
least 20 mV peak-to-peak ripple at the FB pin for the gM
amplifier and the error comparator to operate properly.
Also, the ripple on FB pin should be in phase with the
inductor current. Therefore, the output voltage ripple
caused by the output capacitors value should be much
smaller than the ripple caused by the output capacitor
ESR. If low-ESR capacitors, such as ceramic
capacitors, are selected as the output capacitors, a
ripple injection method should be applied to provide the
enough feedback voltage ripple. Refer to the Ripple
Injection subsection for details.
The voltage rating of the capacitor should be twice the
output voltage for a tantalum and 20% greater for
aluminum electrolytic or OS-CON. The output capacitor
RMS current is calculated in Equation 5-11.
EQUATION 5-11:
The power dissipated in the output capacitor is:
EQUATION 5-12:
5.5 Input Capacitor Selection
The input capacitor for the power stage input VIN
should be selected for ripple current rating and voltage
rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the
input supply on. A tantalum input capacitor’s voltage
rating should be at least two times the maximum input
voltage to maximize reliability. Aluminum electrolytic,
OS-CON, and multilayer polymer film capacitors can
handle the higher inrush currents without voltage
de-rating. The input voltage ripple will primarily depend
on the input capacitor’s ESR. The peak input current is
equal to the peak inductor current, so:
EQUATION 5-13:
The input capacitor must be rated for the input current
ripple. The RMS value of input capacitor current is
determined at the maximum output current. Assuming
the peak-to-peak inductor current ripple is low:
EQUATION 5-14:
The power dissipated in the input capacitor is:
EQUATION 5-15:
5.6 Ripple Injection
The VFB ripple required for proper operation of the
MIC28511’s gM amplifier and error comparator is
20 mV to 100 mV. However, the output voltage ripple is
generally designed as 1% to 2% of the output voltage.
If the feedback voltage ripple is so small that the gM
amplifier and error comparator can’t sense it, then the
MIC28511 will lose control and the output voltage is not
regulated. In order to have some amount of VFB ripple,
a ripple injection method is applied for low output
voltage ripple applications.
The applications are divided into three situations
according to the amount of the feedback voltage ripple:
• Enough ripple at the feedback voltage due to the
large ESR of the output capacitors (Figure 5-4).
The converter is stable without any ripple
injection.
FIGURE 5-4: Enough Ripple at FB.
The feedback voltage ripple is:
EQUATION 5-16:
• Inadequate ripple at the feedback voltage due to
the small ESR of the output capacitors.
The output voltage ripple is fed into the FB pin
through a feed-forward capacitor, CFF in this
situation, as shown in Figure 5-5. The typical CFF
value is selected by using Equation 5-17.
ICOUT RMS
ILPP
12
------------------=
PDISS COUT
ICOUT RMS
2ESRCOUT
=
VIN
ILPK
ESRCIN
=
ICIN RMS
IOUT MAXD1D–
PDISS CINICIN RMS
2ESRCIN
=
SW
FB
R1
R2 ESR
C
OUT
MIC28511
L
VFB PP
R2
R1R2+
--------------------ESRCOUT
ILPP
=
Where:
IL(PP) Peak-to-Peak Value of the Inductor
Current Ripple
2016-2017 Microchip Technology Inc. DS20005520B-page 25
MIC28511
EQUATION 5-17:
With the feed-forward capacitor, the feedback
voltage ripple is very close to the output voltage
ripple.
EQUATION 5-18:
FIGURE 5-5: Inadequate Ripple at FB.
• Virtually no ripple at the FB pin voltage due to the
very low ESR of the output capacitors.
In this situation, the output voltage ripple is less than
20 mV. Therefore, additional ripple is injected into
the FB pin from the switching node SW via a resistor
RINJ and a capacitor CINJ, as shown in Figure 5-6.
FIGURE 5-6: Invisible Ripple at FB.
The injected ripple is calculated via:
EQUATION 5-19:
EQUATION 5-20:
In Equation 5-19 and Equation 5-20, it is assumed that
the time constant associated with CFF must be much
greater than the switching period:
EQUATION 5-21:
If the voltage divider resistors R1 and R2 are in the k
range, a CFF of 1 nF to 100 nF can easily satisfy the
large time constant requirements. Also, a 100 nF
injection capacitor CINJ is used in order to be
considered as short for a wide range of the
frequencies.
The process of sizing the ripple injection resistor and
capacitors is as follows.
•Select C
FF to feed all output ripples into the
feedback pin and make sure the large time
constant assumption is satisfied. Typical choice of
CFF is 1 nF to 100 nF if R1 and R2 are in the k
range.
•Select R
INJ according to the expected feedback
voltage ripple using Equation 5-22:
EQUATION 5-22:
The value of RINJ is calculated using Equation 5-23.
EQUATION 5-23:
•Select C
INJ as 100 nF, which could be considered
as short for a wide range of the frequencies.
R1CFF
10
fSW
--------
VFB PP
ESR ILPP
MIC28511
SW
FB R1
R2 ESR
C
OUT
CFF
L
MIC28511
SW
FB
R1
R2 ESR
C
OUT
C
FF
R
INJ
C
INJ
L
VFB PP
VIN Kdiv
D1D–1
fSW
-----------------
=
Where:
VIN Power stage input voltage
D Duty cycle
fSW Switching frequency
τ(R1//R2//RINJ) x CFF
Kdiv R1//R2
RINJ R1//R2+
-----------------------------------=
1
fSW
-----------------T
---=1«
Kdiv VFB PP
VIN
----------------------- fSW
D1D–
----------------------------
=
RINJ R1//R2
1
Kdiv
---------- 1–
=
MIC28511
DS20005520B-page 26 2016-2017 Microchip Technology Inc.
6.0 PCB LAYOUT GUIDELINES
PCB Layout is critical to achieve reliable, stable and
efficient performance. A ground plane is required to
control EMI and minimize the inductance in power,
signal and return paths.
Figure 6-1 is optimized from a small form-factor point of
view and shows the top and bottom layers of a
four-layer PCB. It is recommended to use Mid-Layer 1
as a continuous ground plane.
FIGURE 6-1: Top and Bottom Layers of a
Four-Layer Board.
The following guidelines should be followed to ensure
proper operation of the MIC28511 converter.
6.1 IC
• The analog ground pin (AGND) must be
connected directly to the ground planes. Do not
route the AGND pin to the PGND pin on the top
layer.
• Place the IC close to the point-of-load (POL).
• Use copper planes to route the input and output
power lines.
• Analog and power grounds should be kept
separate and connected at only one location.
6.2 Input Capacitor
• Place the input capacitors on the same side of the
board and as close to the PVIN and PGND pins as
possible.
• Place several vias to the ground plane close to
the input capacitor ground terminal.
• Use either X7R or X5R dielectric input capacitors.
Do not use Y5V or Z5U type capacitors.
• Do not replace the ceramic input capacitor with
any other type of capacitor. Any type of capacitor
can be placed in parallel with the input capacitor.
• If a Tantalum input capacitor is placed in parallel
with the input capacitor, it must be recommended
for switching regulator applications and the
operating voltage must be derated by 50%.
• In “Hot-Plug” applications, a Tantalum or
Electrolytic bypass capacitor must be used to limit
the overvoltage spike seen on the input supply
with power is suddenly applied.
6.3 SW Node
• Do not route any digital lines underneath or close
to the SW node.
• Keep the switch node (SW) away from the
feedback (FB) pin.
6.4 Output Capacitor
• Use a copper island to connect the output
capacitor ground terminal to the input capacitor
ground terminal.
• Phase margin will change as the output capacitor
value and ESR changes. Contact the factory if the
output capacitor is different from what is shown in
the BOM.
• The feedback trace should be separate from the
power trace and connected as close as possible
to the output capacitor. Sensing a long
high-current load trace can degrade the DC load
regulation.
6.5 Thermal Measurements
Measuring the IC’s case temperature is recommended
to ensure it is within its operating limits. Although this
might seem like a very elementary task, it is easy to get
erroneous results. The most common mistake is to use
the standard thermal couple that comes with a thermal
meter. This thermal couple wire gauge is large, typically
22 gauge, and behaves like a heat sink, resulting in a
lower case measurement.
Two methods of temperature measurement are using a
smaller thermal couple wire or an infrared
thermometer. If a thermal couple wire is used, it must
be constructed of 36 gauge wire or higher then (smaller
wire size) to minimize the wire heat-sinking effect. In
addition, the thermal couple tip must be covered in
either thermal grease or thermal glue to make sure that
the thermal couple junction is making good contact with
the case of the IC. Omega brand thermal couple
(5SC-TT-K-36-36) is adequate for most applications.
Wherever possible, an infrared thermometer is
recommended. The measurement spot size of most
infrared thermometers is too large for an accurate
reading on a small form factor ICs. However, an IR
thermometer from Optris has a 1 mm spot size, which
makes it a good choice for measuring the hottest point
on the case. An optional stand makes it easy to hold the
beam on the IC for long periods of time.
For more information about the Evaluation board layout, please contact Microchip sales.
2016-2017 Microchip Technology Inc. DS20005520B-page 27
MIC28511
7.0 PACKAGING INFORMATION
7.1 Package Marking Information
Example 24-lead FQFN (3 x 4 mm)
Legend: XX...X Customer-specific information
Y Year code (last digit of calendar year)
YY Year code (last 2 digits of calendar year)
WW Week code (week of January 1 is week ‘01’)
NNN Alphanumeric traceability code
Pb-free JEDEC® designator for Matte Tin (Sn)
*This package is Pb-free. The Pb-free JEDEC designator ( )
can be found on the outer packaging for this package.
Note: In the event the full Microchip part number cannot be marked on one line, it will
be carried over to the next line, thus limiting the number of available
characters for customer-specific information.
3
e
3
e
MIC28511
DS20005520B-page 28 2016-2017 Microchip Technology Inc.
24-Lead FQFN 3 mm x 4 mm Package Outline and Recommended Land Pattern
Note: For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
2016-2017 Microchip Technology Inc. DS20005520B-page 29
MIC28511
Note: For the most current package drawings, please see the Microchip Packaging Specification located at
http://www.microchip.com/packaging
MIC28511
DS20005520B-page 30 2016-2017 Microchip Technology Inc.
2016-2017 Microchip Technology Inc. DS20005520B-page 31
MIC28511
APPENDIX A: REVISION HISTORY
Revision B (November 2017)
The following is the list of modifications:
• Updated Absolute Maximum Ratings †.
• Updated Figure 2-22, Equation 4-3 and
Equation 5-10.
• Updated Section 4.5, Current Limit.
• Added Section 4.6, Output Current Thermal Der-
ating.
• Various typographical edits.
Revision A (May 2016)
• Converted Micrel document MIC28511 to
Microchip data sheet template DS20005520A.
• Minor text changes throughout.
MIC28511
DS20005520B-page 32 2016-2017 Microchip Technology Inc.
NOTES:
2016-2017 Microchip Technology Inc. DS20005520B-page 33
MIC28511
PRODUCT IDENTIFICATION SYSTEM
To order or obtain information, e.g., on pricing or delivery, contact your local Microchip representative or sales office.
Examples:
a) MIC28511-1YFL: 60VIN, 3A Synchronous
Buck Regulator, HyperLight
Load, –40°C to +125°C
Junction Temperature
Range, 24LD FQFN
b) MIC28511-2YFL: 60VIN, 3A Synchronous
Buck Regulator, Hyper
Speed Control, –40°C to
+125°C Junction
Temperature Range,
24LD FQFN
PART NO. XX
Package
Device
Device: MIC28511: 60VIN, 3A Synchronous Buck Regulator
Architecture: 1 = HyperLight Load
2 = Hyper Speed Control
Temperature: Y = –40°C to +125°C
Package: FL = 24-Pin 3 mm x 4 mm FQFN; Note 1
X
Temperature
X
Architecture
Note 1: FQFN is a lead-free package. Pb-Free lead
finish is Matte Tin.
MIC28511
DS20005520B-page 34 2016-2017 Microchip Technology Inc.
NOTES:
2016-2017 Microchip Technology Inc. DS20005520B-page 35
Information contained in this publication regarding device
applications and the like is provided only for your convenience
and may be superseded by updates. It is your responsibility to
ensure that your application meets with your specifications.
MICROCHIP MAKES NO REPRESENTATIONS OR
WARRANTIES OF ANY KIND WHETHER EXPRESS OR
IMPLIED, WRITTEN OR ORAL, STATUTORY OR
OTHERWISE, RELATED TO THE INFORMATION,
INCLUDING BUT NOT LIMITED TO ITS CONDITION,
QUALITY, PERFORMANCE, MERCHANTABILITY OR
FITNESS FOR PURPOSE. Microchip disclaims all liability
arising from this information and its use. Use of Microchip
devices in life support and/or safety applications is entirely at
the buyer’s risk, and the buyer agrees to defend, indemnify and
hold harmless Microchip from any and all damages, claims,
suits, or expenses resulting from such use. No licenses are
conveyed, implicitly or otherwise, under any Microchip
intellectual property rights unless otherwise stated.
Trademarks
The Microchip name and logo, the Microchip logo, AnyRate, AVR,
AVR logo, AVR Freaks, BeaconThings, BitCloud, CryptoMemory,
CryptoRF, dsPIC, FlashFlex, flexPWR, Heldo, JukeBlox, KEELOQ,
KEELOQ logo, Kleer, LANCheck, LINK MD, maXStylus,
maXTouch, MediaLB, megaAVR, MOST, MOST logo, MPLAB,
OptoLyzer, PIC, picoPower, PICSTART, PIC32 logo, Prochip
Designer, QTouch, RightTouch, SAM-BA, SpyNIC, SST, SST
Logo, SuperFlash, tinyAVR, UNI/O, and XMEGA are registered
trademarks of Microchip Technology Incorporated in the U.S.A.
and other countries.
ClockWorks, The Embedded Control Solutions Company,
EtherSynch, Hyper Speed Control, HyperLight Load, IntelliMOS,
mTouch, Precision Edge, and Quiet-Wire are registered
trademarks of Microchip Technology Incorporated in the U.S.A.
Adjacent Key Suppression, AKS, Analog-for-the-Digital Age, Any
Capacitor, AnyIn, AnyOut, BodyCom, chipKIT, chipKIT logo,
CodeGuard, CryptoAuthentication, CryptoCompanion,
CryptoController, dsPICDEM, dsPICDEM.net, Dynamic Average
Matching, DAM, ECAN, EtherGREEN, In-Circuit Serial
Programming, ICSP, Inter-Chip Connectivity, JitterBlocker,
KleerNet, KleerNet logo, Mindi, MiWi, motorBench, MPASM, MPF,
MPLAB Certified logo, MPLIB, MPLINK, MultiTRAK, NetDetach,
Omniscient Code Generation, PICDEM, PICDEM.net, PICkit,
PICtail, PureSilicon, QMatrix, RightTouch logo, REAL ICE, Ripple
Blocker, SAM-ICE, Serial Quad I/O, SMART-I.S., SQI,
SuperSwitcher, SuperSwitcher II, Total Endurance, TSHARC,
USBCheck, VariSense, ViewSpan, WiperLock, Wireless DNA, and
ZENA are trademarks of Microchip Technology Incorporated in the
U.S.A. and other countries.
SQTP is a service mark of Microchip Technology Incorporated in
the U.S.A.
Silicon Storage Technology is a registered trademark of Microchip
Technology Inc. in other countries.
GestIC is a registered trademark of Microchip Technology
Germany II GmbH & Co. KG, a subsidiary of Microchip Technology
Inc., in other countries.
All other trademarks mentioned herein are property of their
respective companies.
© 2016-2017, Microchip Technology Incorporated, All Rights
Reserved.
ISBN: 978-1-5224-2403-1
Note the following details of the code protection feature on Microchip devices:
• Microchip products meet the specification contained in their particular Microchip Data Sheet.
• Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the
intended manner and under normal conditions.
• There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our
knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data
Sheets. Most likely, the person doing so is engaged in theft of intellectual property.
• Microchip is willing to work with the customer who is concerned about the integrity of their code.
• Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not
mean that we are guaranteeing the product as “unbreakable.”
Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of our
products. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such acts
allow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.
Microchip received ISO/TS-16949:2009 certification for its worldwide
headquarters, design and wafer fabrication facilities in Chandler and
Tempe, Arizona; Gresham, Oregon and design centers in California
and India. The Company’ s quality system processes and procedures
are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping
devices, Serial EEPROMs, micro perip hera ls, n onvolat ile memory and
analog products . In add ition, Microchip’s quality system for the design
and manufacture of development systems is ISO 9001:2000 certified.
QUALITYMANAGEMENTS
YSTEM
CERTIFIEDBYDNV
== ISO/TS16949==
DS20005520B-page 36 2016-2017 Microchip Technology Inc.
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Worldwide Sales and Service
10/25/17
