LM1876
Overture
™Audio Power Amplifier Series
Dual 20W Audio Power Amplifier with Mute and Standby
Modes
General Description
The LM1876 is a stereo audio amplifier capable of delivering
typically 20W per channel of continuous average output
power into a 4Ωor 8Ωload with less than 0.1%(THD + N).
Each amplifier has an independent smooth transition fade-in/
out mute and a power conserving standby mode which can
be controlled by external logic.
The performance of the LM1876, utilizing its Self Peak In-
stantaneous Temperature (˚Ke) (SPiKe™) Protection Cir-
cuitry, places it in a class above discrete and hybrid amplifi-
ers by providing an inherently, dynamically protected Safe
Operating Area (SOA). SPiKe Protection means that these
parts are safeguarded at the output against overvoltage, un-
dervoltage, overloads, including thermal runaway and in-
stantaneous temperature peaks.
Key Specifications
jTHD+N at 1 kHz at 2 x 15W continuous average
output power into 4Ωor 8Ω: 0.1%(max)
jTHD+N at 1 kHz at continuous average
output power of 2 x 20W into 8Ω: 0.009%(typ)
jStandby current: 4.2 mA (typ)
Features
nSPiKe Protection
nMinimal amount of external components necessary
nQuiet fade-in/out mute mode
nStandby-mode
nIsolated 15-lead TO-220 package
nNon-Isolated 15-lead TO-220 package
Applications
nHigh-end stereo TVs
nComponent stereo
nCompact stereo
Typical Application
Note: Numbers in parentheses represent pinout for amplifier B.
*Optional component dependent upon specific design requirements.
Connection Diagram
SPiKe™Protection and Overture™are trademarks of National Semiconductor Corporation.
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FIGURE 1. Typical Audio Amplifier Application Circuit
Plastic Package
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Top View
Isolated Package
Order Number LM1876TF
See NS Package Number TF15B
Non-Isolated Package
Order Number LM1876T
See NS Package Number TA15A
February 1998
LM1876 Overture
™
Audio Power Amplifier Series
Dual 20W Audio Power Amplifier with Mute and Standby Modes
© 1999 National Semiconductor Corporation DS012072 www.national.com
Absolute Maximum Ratings (Notes 4, 5)
If Military/Aerospace specified devices are required,
please contact the National Semiconductor Sales Office/
Distributors for availability and specifications.
Supply Voltage |V
CC
|+|V
EE
|
(No Input) 64V
Supply Voltage |V
CC
|+|V
EE
|
(with Input) 64V
Common Mode Input Voltage (V
CC
or V
EE
) and
|V
CC
|+|V
EE
|≤54V
Differential Input Voltage 54V
Output Current Internally Limited
Power Dissipation (Note 6) 62.5W
ESD Susceptability (Note 7) 2000V
Junction Temperature (Note 8) 150˚C
Thermal Resistance
Isolated TF-Package
θ
JC
2˚C/W
Non-Isolated T-Package
θ
JC
1˚C/W
Soldering Information
TF Package (10 sec.) 260˚C
Storage Temperature −40˚C to +150˚C
Operating Ratings (Notes 4, 5)
Temperature Range
T
MIN
≤T
A
≤T
MAX
−20˚C ≤T
A
≤+85˚C
Supply Voltage |V
CC
|+|V
EE
| (Note 1) 20V to 64V
Electrical Characteristics (Notes 4, 5)
The following specifications apply for V
CC
=+22V, V
EE
=−22V with R
L
=8Ωunless otherwise specified. Limits apply for T
A
=
25˚C.
Symbol Parameter Conditions LM1876 Units
(Limits)
Typical Limit
(Note 9) (Note 10)
|V
CC
| + Power Supply Voltage GND − V
EE
≥9V 20 V (min)
|V
EE
| (Note 11) 64 V (max)
P
O
Output Power THD + N =0.1%(max),
(Note 3) (Continuous Average) f =1 kHz
|V
CC
|=|V
EE
|=22V, R
L
=8Ω20 15 W/ch (min)
|V
CC
|=|V
EE
|=20V, R
L
=4Ω(Note 13) 22 15 W/ch (min)
THD + N Total Harmonic Distortion 15 W/ch, R
L
=8Ω0.08 %
Plus Noise 15 W/ch, R
L
=4Ω,|V
CC
|=|V
EE
|=20V 0.1 %
20 Hz ≤f≤20 kHz, A
V
=26 dB
X
talk
Channel Separation f =1 kHz, V
O
=10.9 Vrms 80 dB
SR
(Note 3) Slew Rate V
IN
=1.414 Vrms, t
rise
=2 ns 18 12 V/µs (min)
I
total
Total Quiescent Power Both Amplifiers V
CM
=0V,
(Note 2) Supply Current V
O
=0V, I
O
=0mA
Standby: Off 50 80 mA (max)
Standby: On 4.2 6 mA (max)
V
OS
(Note 2) Input Offset Voltage V
CM
=0V, I
O
=0 mA 2.0 15 mV (max)
I
B
Input Bias Current V
CM
=0V, I
O
=0 mA 0.2 0.5 µA (max)
I
OS
Input Offset Current V
CM
=0V, I
O
=0 mA 0.002 0.2 µA (max)
I
O
Output Current Limit |V
CC
|=|V
EE
|=10V, t
ON
=10 ms, 3.5 2.9 Apk (min)
V
O
=0V
V
OD
Output Dropout Voltage |V
CC
–V
O
|, V
CC
=20V, I
O
=+100 mA 1.8 2.3 V (max)
(Note 2) (Note 12) |V
O
–V
EE
|, V
EE
=−20V, I
O
=−100 mA 2.5 3.2 V (max)
PSRR Power Supply Rejection Ratio V
CC
=25V to 10V, V
EE
=−25V, 115 85 dB (min)
(Note 2) V
CM
=0V, I
O
=0mA
V
CC
=25V, V
EE
=−25V to −10V 110 85 dB (min)
V
CM
=0V, I
O
=0mA
CMRR Common Mode Rejection Ratio V
CC
=35V to 10V, V
EE
=−10V to −35V, 110 80 dB (min)
(Note 2) V
CM
=10V to −10V, I
O
=0mA
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Electrical Characteristics (Notes 4, 5) (Continued)
The following specifications apply for V
CC
=+22V, V
EE
=−22V with R
L
=8Ωunless otherwise specified. Limits apply for T
A
=
25˚C.
Symbol Parameter Conditions LM1876 Units
(Limits)
Typical Limit
(Note 9) (Note 10)
A
VOL
(Note 2) Open Loop Voltage Gain R
L
=2kΩ,∆V
O
=20 V 110 90 dB (min)
GBWP Gain Bandwidth Product f
O
=100 kHz, V
IN
=50 mVrms 7.5 5 MHz (min)
e
IN
Input Noise IHF—A Weighting Filter 2.0 8 µV (max)
(Note 3) R
IN
=600Ω(Input Referred)
SNR Signal-to-Noise Ratio P
O
=1W, A—Weighted, 98 dB
Measured at 1 kHz, R
S
=25Ω
P
O
=15W, A—Weighted 108 dB
Measured at 1 kHz, R
S
=25Ω
A
M
Mute Attenuation Pin 6,11 at 2.5V 115 80 dB (min)
Standby
Pin
V
IL
Standby Low Input Voltage Not in Standby Mode 0.8 V (max)
V
IH
Standby High Input Voltage In Standby Mode 2.0 2.5 V (min)
Mute pin
V
IL
Mute Low Input Voltage Outputs Not Muted 0.8 V (max)
V
IH
Mute High Input Voltage Outputs Muted 2.0 2.5 V (min)
Note 1: Operation is guaranteed up to 64V, however, distortion may be introduced from SPiKe Protection Circuitry if proper thermal considerations are not taken into
account. Refer to the Application Information section for a complete explanation.
Note 2: DC Electrical Test; Refer to Test Circuit #1.
Note 3: AC Electrical Test; Refer to Test Circuit #2.
Note 4: All voltages are measured with respect to the GND pins (5, 10), unless otherwise specified.
Note 5: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is func-
tional, but do not guarantee specific performance limits. Electrical Characteristics state DC andAC electrical specifications under particular test conditions which guar-
antee specific performance limits. This assumes that the device is within the Operating Ratings. Specifications are not guaranteed for parameters where no limit is
given, however, the typical value is a good indication of device performance.
Note 6: For operating at case temperatures above 25˚C, the device must be derated based on a 150˚C maximum junction temperature and a thermal resistance of
θJC =2˚C/W (junction to case) for the TF package and θJC =1˚C/W for the T package. Refer to the section Determining the Correct Heat Sink in the Application In-
formation section.
Note 7: Human body model, 100 pF discharged through a 1.5 kΩresistor.
Note 8: The operating junction temperature maximum is 150˚C, however, the instantaneous Safe Operating Area temperature is 250˚C.
Note 9: Typicals are measured at 25˚C and represent the parametric norm.
Note 10: Limits are guarantees that all parts are tested in production to meet the stated values.
Note 11: VEE must have at least −9V at its pin with reference to ground in order for the under-voltage protection circuitry to be disabled. In addition, the voltage dif-
ferential between VCC and VEE must be greater than 14V.
Note 12: The output dropout voltage, VOD, is the supply voltage minus the clipping voltage. Refer to the Clipping Voltage vs. Supply Voltage graph in the Typical Per-
formance Characteristics section.
Note 13: Fora4Ωload, and with ±20V supplies, the LM1876 can deliver typically 22W of continuous average output power with less than 0.1%(THD + N). With
supplies above ±20V, the LM1876 cannot deliver more than 22W into a 4Ωdue to current limiting of the output transistors. Thus, increasing the power supply above
±20V will only increase the internal power dissipation, not the possible output power. Increased power dissipation will require a larger heat sink as explained in the
Application Information section.
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Test Circuit #1(Note 2) (DC Electrical Test Circuit)
Test Circuit #2(Note 3) (AC Electrical Test Circuit)
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Bridged Amplifier Application Circuit
Single Supply Application Circuit
Note: *Optional components dependent upon specific design requirements.
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FIGURE 2. Bridged Amplifier Application Circuit
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FIGURE 3. Single Supply Amplifier Application Circuit
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Auxiliary Amplifier Application Circuit
Equivalent Schematic (excluding active protection circuitry)
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FIGURE 4. Special Audio Amplifier Application Circuit
LM1876 (per Amp)
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External Components Description
Components Functional Description
1R
B
Prevents currents from entering the amplifier’s non-inverting input which may be passed through to the
load upon power down of the system due to the low input impedance of the circuitry when the
undervoltage circuitry is off. This phenomenon occurs when the supply voltages are below 1.5V.
2R
i
Inverting input resistance to provide AC gain in conjunction with R
f
.
3R
f
Feedback resistance to provide AC gain in conjunction with R
i
.
4C
i
(Note 14) Feedback capacitor which ensures unity gain at DC. Also creates a highpass filter with R
i
at f
C
=
1/(2πR
i
C
i
).
5C
S
Provides power supply filtering and bypassing. Refer to the Supply Bypassing application section for
proper placement and selection of bypass capacitors.
6R
V
(Note 14) Acts as a volume control by setting the input voltage level.
7R
IN
(Note 14) Sets the amplifier’s input terminals DC bias point when C
IN
is present in the circuit. Also works with C
IN
to
create a highpass filter at f
C
=1/(2πR
IN
C
IN
). Refer to
Figure 4
.
8C
IN
(Note 14) Input capacitor which blocks the input signal’s DC offsets from being passed onto the amplifier’s inputs.
9R
SN
(Note 14) Works with C
SN
to stabilize the output stage by creating a pole that reduces high frequency instabilities.
10 C
SN
(Note 14) Works with R
SN
to stabilize the output stage by creating a pole that reduces high frequency instabilities.
The pole is set at f
C
=1/(2πR
SN
C
SN
). Refer to
Figure 4
.
11 L (Note 14) Provides high impedance at high frequencies so that R may decouple a highly capacitive load and reduce
the Q of the series resonant circuit. Also provides a low impedance at low frequencies to short out R and
pass audio signals to the load. Refer to
Figure 4
.
12 R (Note 14)
13 R
A
Provides DC voltage biasing for the transistor Q1 in single supply operation.
14 C
A
Provides bias filtering for single supply operation.
15 R
INP
(Note 14) Limits the voltage difference between the amplifier’s inputs for single supply operation. Refer to the Clicks
and Pops application section for a more detailed explanation of the function of R
INP
.
16 R
BI
Provides input bias current for single supply operation. Refer to the Clicks and Pops application section
for a more detailed explanation of the function of R
BI
.
17 R
E
Establishes a fixed DC current for the transistor Q1 in single supply operation. This resistor stabilizes the
half-supply point along with C
A
.
Note 14: Optional components dependent upon specific design requirements.
Typical Performance Characteristics
THD+NvsFrequency
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THD+NvsFrequency
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THD+NvsFrequency
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Typical Performance Characteristics (Continued)
THD+Nvs
Output Power
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THD+Nvs
Output Power
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THD+Nvs
Output Power
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THD+Nvs
Output Power
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THD+Nvs
Output Power
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THD+Nvs
Output Power
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Clipping Voltage vs
Supply Voltage
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Clipping Voltage vs
Supply Voltage
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Clipping Voltage vs
Supply Voltage
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Typical Performance Characteristics (Continued)
Output Power vs
Load Resistance
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Power Dissipation vs
Output Power
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Power Dissipation vs
Output Power
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Output Power vs
Supply Voltage
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Output Mute vs
Mute Pin Voltage
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Output Mute vs
Mute Pin Voltage
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Channel Separation vs
Frequency
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Pulse Response
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Large Signal Response
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Typical Performance Characteristics (Continued)
Power Supply
Rejection Ratio
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Common-Mode
Rejection Ratio
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Open Loop
Frequency Response
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Safe Area
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SPiKe Protection
Response
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Supply Current vs
Supply Voltage
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Pulse Thermal
Resistance
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Pulse Thermal
Resistance
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Supply Current vs
Output Voltage
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Typical Performance Characteristics (Continued)
Pulse Power Limit
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Pulse Power Limit
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Supply Current vs
Case Temperature
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Supply Current (I
CC
)vs
Standby Pin Voltage
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Supply Current (I
EE
)vs
Standby Pin Voltage
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Input Bias Current vs
Case Temperature
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Application Information
MUTE MODE
By placing a logic-high voltage on the mute pins, the signal
going into the amplifiers will be muted. If the mute pins are
left floating or connected to a logic-low voltage, the amplifi-
ers will be in a non-muted state. There are two mute pins,
one for each amplifier, so that one channel can be muted
without muting the other if the application requires such a
configuration. Refer to the Typical Performance Character-
istics section for curves concerning Mute Attenuation vs
Mute Pin Voltage.
STANDBY MODE
The standby mode of the LM1876 allows the user to drasti-
cally reduce power consumption when the amplifiers are
idle. By placing a logic-high voltage on the standby pins, the
amplifiers will go into Standby Mode. In this mode, the cur-
rent drawn from the V
CC
supply is typically less than 10 µA
total for both amplifiers. The current drawn from the V
EE
sup-
ply is typically 4.2 mA. Clearly, there is a significant reduction
in idle power consumption when using the standby mode.
There are two Standby pins, so that one channel can be put
in standby mode without putting the other amplifier in
standby if the application requires such flexibility. Refer to
the Typical Performance Characteristics section for
curves showing Supply Current vs. Standby Pin Voltage for
both supplies.
UNDER-VOLTAGE PROTECTION
Upon system power-up, the under-voltage protection cir-
cuitry allows the power supplies and their corresponding ca-
pacitors to come up close to their full values before turning
on the LM1876 such that no DC output spikes occur. Upon
turn-off, the output of the LM1876 is brought to ground be-
fore the power supplies such that no transients occur at
power-down.
OVER-VOLTAGE PROTECTION
The LM1876 contains over-voltage protection circuitry that
limits the output current to approximately 3.5 Apk while also
providing voltage clamping, though not through internal
clamping diodes. The clamping effect is quite the same,
however, the output transistors are designed to work alter-
nately by sinking large current spikes.
SPiKe PROTECTION
The LM1876 is protected from instantaneous
peak-temperature stressing of the power transistor array.
The Safe Operating graph in the Typical Performance
Characteristics section shows the area of device operation
where SPiKe Protection Circuitry is not enabled. The wave-
form to the right of the SOA graph exemplifies how the dy-
namic protection will cause waveform distortion when en-
abled.
THERMAL PROTECTION
The LM1876 has a sophisticated thermal protection scheme
to prevent long-term thermal stress of the device. When the
temperature on the die reaches 165˚C, the LM1876 shuts
down. It starts operating again when the die temperature
drops to about 155˚C, but if the temperature again begins to
rise, shutdown will occur again at 165˚C. Therefore, the de-
vice is allowed to heat up to a relatively high temperature if
the fault condition is temporary, but a sustained fault will
cause the device to cycle in a Schmitt Trigger fashion be-
tween the thermal shutdown temperature limits of 165˚C and
155˚C. This greatly reduces the stress imposed on the IC by
thermal cycling, which in turn improves its reliability under
sustained fault conditions.
Since the die temperature is directly dependent upon the
heat sink used, the heat sink should be chosen such that
thermal shutdown will not be reached during normal opera-
tion. Using the best heat sink possible within the cost and
space constraints of the system will improve the long-term
reliability of any power semiconductor device, as discussed
in the Determining the Correct Heat Sink Section.
DETERMlNlNG MAXIMUM POWER DISSIPATION
Power dissipation within the integrated circuit package is a
very important parameter requiring a thorough understand-
ing if optimum power output is to be obtained. An incorrect
maximum power dissipation calculation may result in inad-
equate heat sinking causing thermal shutdown and thus lim-
iting the output power.
Equation (1) exemplifies the theoretical maximum power dis-
sipation point of each amplifier where V
CC
is the total supply
voltage. P
DMAX
=V
CC
2/2π
2
R
L
(1)
Thus by knowing the total supply voltage and rated output
load, the maximum power dissipation point can be calcu-
lated. The package dissipation is twice the number which re-
sults from equation (1) since there are two amplifiers in each
LM1876. Refer to the graphs of Power Dissipation versus
Output Power in the Typical Performance Characteristics
section which show the actual full range of power dissipation
not just the maximum theoretical point that results from
equation (1).
DETERMINING THE CORRECT HEAT SINK
The choice of a heat sink for a high-power audio amplifier is
made entirely to keep the die temperature at a level such
that the thermal protection circuitry does not operate under
normal circumstances.
The thermal resistance from the die (junction) to the outside
air (ambient) is a combination of three thermal resistances,
θ
JC
,θ
CS
, and θ
SA
. In addition, the thermal resistance, θ
JC
(junction to case), of the LM1876TF is 2˚C/W and the
LM1876T is 1˚C/W. Using Thermalloy Thermacote thermal
compound, the thermal resistance, θ
CS
(case to sink), is
about 0.2˚C/W. Since convection heat flow (power dissipa-
tion) is analogous to current flow, thermal resistance is
analogous to electrical resistance, and temperature drops
are analogous to voltage drops, the power dissipation out of
the LM1876 is equal to the following:
P
DMAX
=(T
JMAX
−T
AMB
)/θ
JA
(2)
where T
JMAX
=150˚C, T
AMB
is the system ambient tempera-
ture and θ
JA
=θ
JC
+θ
CS
+θ
SA
.
Once the maximum package power dissipation has been
calculated using equation (1), the maximum thermal resis-
tance, θ
SA
, (heat sink to ambient) in ˚C/W for a heat sink can
be calculated. This calculation is made using equation (3)
which is derived by solving for θ
SA
in equation (2).
θ
SA
=[(T
JMAX
−T
AMB
)−P
DMAX
(θ
JC
+θ
CS
)]/P
DMAX
(3)
Again it must be noted that the value of θ
SA
is dependent
upon the system designer’s amplifier requirements. If the
ambient temperature that the audio amplifier is to be working
www.national.com 12
Application Information (Continued)
under is higher than 25˚C, then the thermal resistance for the
heat sink, given all other things are equal, will need to be
smaller.
SUPPLY BYPASSING
The LM1876 has excellent power supply rejection and does
not require a regulated supply. However, to improve system
performance as well as eliminate possible oscillations, the
LM1876 should have its supply leads bypassed with
low-inductance capacitors having short leads that are lo-
cated close to the package terminals. Inadequate power
supply bypassing will manifest itself by a low frequency oscil-
lation known as “motorboating” or by high frequency insta-
bilities. These instabilities can be eliminated through multiple
bypassing utilizing a large tantalum or electrolytic capacitor
(10 µF or larger) which is used to absorb low frequency
variations and a small ceramic capacitor (0.1 µF) to prevent
any high frequency feedback through the power supply lines.
If adequate bypassing is not provided, the current in the sup-
ply leads which is a rectified component of the load current
may be fed back into internal circuitry. This signal causes
distortion at high frequencies requiring that the supplies be
bypassed at the package terminals with an electrolytic ca-
pacitor of 470 µF or more.
BRIDGED AMPLIFIER APPLICATION
The LM1876 has two operational amplifiers internally, allow-
ing for a few different amplifier configurations. One of these
configurations is referred to as “bridged mode” and involves
driving the load differentially through the LM1876’s outputs.
This configuration is shown in
Figure 2
. Bridged mode op-
eration is different from the classical single-ended amplifier
configuration where one side of its load is connected to
ground.
A bridge amplifier design has a distinct advantage over the
single-ended configuration, as it provides differential drive to
the load, thus doubling output swing for a specified supply
voltage. Consequently, theoretically four times the output
power is possible as compared to a single-ended amplifier
under the same conditions. This increase in attainable output
power assumes that the amplifier is not current limited or
clipped.
A direct consequence of the increased power delivered to
the load by a bridge amplifier is an increase in internal power
dissipation. For each operational amplifier in a bridge con-
figuration, the internal power dissipation will increase by a
factor of two over the single ended dissipation. Thus, for an
audio power amplifier such as the LM1876, which has two
operational amplifiers in one package, the package dissipa-
tion will increase by a factor of four. To calculate the
LM1876’s maximum power dissipation point for a bridged
load, multiply equation (1) by a factor of four.
This value of P
DMAX
can be used to calculate the correct size
heat sink for a bridged amplifier application. Since the inter-
nal dissipation for a given power supply and load is in-
creased by using bridged-mode, the heatsink’s θ
SA
will have
to decrease accordingly as shown by equation (3). Refer to
the section, Determining the Correct Heat Sink, for a more
detailed discussion of proper heat sinking for a given appli-
cation.
SINGLE-SUPPLY AMPLIFIER APPLICATION
The typical application of the LM1876 is a split supply ampli-
fier. But as shown in
Figure 3
, the LM1876 can also be used
in a single power supply configuration. This involves using
some external components to create a half-supply bias
which is used as the reference for the inputs and outputs.
Thus, the signal will swing around half-supply much like it
swings around ground in a split-supply application. Along
with proper circuit biasing, a few other considerations must
be accounted for to take advantage of all of the LM1876
functions.
The LM1876 possesses a mute and standby function with in-
ternal logic gates that are half-supply referenced. Thus, to
enable either the Mute or Standby function, the voltage at
these pins must be a minimum of 2.5V above half-supply. In
single-supply systems, devices such as microprocessors
and simple logic circuits used to control the mute and
standby functions, are usually referenced to ground, not
half-supply. Thus, to use these devices to control the logic
circuitry of the LM1876, a “level shifter,” like the one shown in
Figure 5
, must be employed. A level shifter is not needed in
a split-supply configuration since ground is also half-supply.
When the voltage at the Logic Input node is 0V, the 2N3904
is “off” and thus resistor R
c
pulls up mute or standby input to
the supply. This enables the mute or standby function. When
the Logic Input is 5V, the 2N3904 is “on” and consequently,
the voltage at the collector is essentially 0V. This will disable
the mute or standby function, and thus the amplifier will be in
its normal mode of operation. R
shift
, along with C
shift
, creates
an RC time constant that reduces transients when the mute
or standby functions are enabled or disabled. Additionally,
R
shift
limits the current supplied by the internal logic gates of
the LM1876 which insures device reliability. Refer to the
Mute Mode and Standby Mode sections in the Application
Information section for a more detailed description of these
functions.
CLICKS AND POPS
In the typical application of the LM1876 as a split-supply au-
dio power amplifier, the IC exhibits excellent “click” and “pop”
performance when utilizing the mute and standby modes. In
addition, the device employs Under-Voltage Protection,
which eliminates unwanted power-up and power-down tran-
sients. The basis for these functions are a stable and con-
stant half-supply potential. In a split-supply application,
ground is the stable half-supply potential. But in a
single-supply application, the half-supply needs to charge up
just like the supply rail, V
CC
. This makes the task of attaining
a clickless and popless turn-on more challenging. Any un-
even charging of the amplifier inputs will result in output
clicks and pops due to the differential input topology of the
LM1876.
DS012072-12
FIGURE 5. Level Shift Circuit
www.national.com13
Application Information (Continued)
To achieve a transient free power-up and power-down, the
voltage seen at the input terminals should be ideally the
same. Such a signal will be common-mode in nature, and
will be rejected by the LM1876. In
Figure 3
, the resistor R
INP
serves to keep the inputs at the same potential by limiting the
voltage difference possible between the two nodes. This
should significantly reduce any type of turn-on pop, due to an
uneven charging of the amplifier inputs. This charging is
based on a specific application loading and thus, the system
designer may need to adjust these values for optimal perfor-
mance.
As shown in
Figure 3
, the resistors labeled R
BI
help bias up
the LM1876 off the half-supply node at the emitter of the
2N3904. But due to the input and output coupling capacitors
in the circuit, along with the negative feedback, there are two
different values of R
BI
, namely 10 kΩand 200 kΩ. These re-
sistors bring up the inputs at the same rate resulting in a pop-
less turn-on.Adjusting these resistors values slightly may re-
duce pops resulting from power supplies that ramp
extremely quick or exhibit overshoot during system turn-on.
AUDIO POWER AMPLlFIER DESIGN
Design a 15W/8ΩAudio Amplifier
Given:
Power Output 15 Wrms
Load Impedance 8Ω
Input Level 1 Vrms(max)
Input Impedance 47 kΩ
Bandwidth 20 Hz−20 kHz
±0.25 dB
A designer must first determine the power supply require-
ments in terms of both voltage and current needed to obtain
the specified output power. V
OPEAK
can be determined from
equation (4) and I
OPEAK
from equation (5).
(4)
(5)
To determine the maximum supply voltage the following con-
ditions must be considered. Add the dropout voltage to the
peak output swing V
OPEAK
, to get the supply rail at a current
of I
OPEAK
. The regulation of the supply determines the un-
loaded voltage which is usually about 15%higher. The sup-
ply voltage will also rise 10%during high line conditions.
Therefore the maximum supply voltage is obtained from the
following equation.
Max supplies ≈±(V
OPEAK
+V
OD
) (1 + regulation) (1.1)
For 15W of output power into an 8Ωload, the required
V
OPEAK
is 15.49V. A minimum supply rail of 20.5V results
from adding V
OPEAK
and V
OD
. With regulation, the maximum
supplies are ±26V and the required I
OPEAK
is 1.94A from
equation (5). It should be noted that for a dual 15W amplifier
into an 8Ωload the I
OPEAK
drawn from the supplies is twice
1.94 Apk or 3.88 Apk. At this point it is a good idea to check
the Power Output vs Supply Voltage to ensure that the re-
quired output power is obtainable from the device while
maintaining low THD+N. In addition, the designer should
verify that with the required power supply voltage and load
impedance, that the required heatsink value θ
SA
is feasible
given system cost and size constraints. Once the heatsink
issues have been addressed, the required gain can be deter-
mined from Equation (6).
(6)
From equation 6, the minimum A
V
is: A
V
≥11.
By selecting a gain of 21, and with a feedback resistor, R
f
=
20 kΩ, the value of R
i
follows from equation (7).
R
i
=R
f
(A
V
− 1) (7)
Thus with R
i
=1kΩa non-inverting gain of 21 will result.
Since the desired input impedance was 47 kΩ, a value of 47
kΩwas selected for R
IN
. The final design step is to address
the bandwidth requirements which must be stated as a pair
of −3 dB frequency points. Five times away from a −3 dB
point is 0.17 dB down from passband response which is bet-
ter than the required ±0.25 dB specified. This fact results in
a low and high frequency pole of 4 Hz and 100 kHz respec-
tively. As stated in the External Components section, R
i
in
conjunction with C
i
create a high-pass filter.
C
i
≥1/(2π*1kΩ*4 Hz) =39.8 µF; use 39 µF.
The high frequency pole is determined by the product of the
desired high frequency pole, f
H
, and the gain, A
V
. With a
A
V
=21 and f
H
=100 kHz, the resulting GBWP is 2.1 MHz,
which is less than the guaranteed minimum GBWP of the
LM1876 of 5 MHz. This will ensure that the high frequency
response of the amplifier will be no worse than 0.17 dB down
at 20 kHz which is well within the bandwidth requirements of
the design.
www.national.com 14
Physical Dimensions inches (millimeters) unless otherwise noted
Isolated TO-220 15-Lead Package
Order Number LM1876TF
NS Package Number TF15B
www.national.com15
Physical Dimensions inches (millimeters) unless otherwise noted (Continued)
LIFE SUPPORT POLICY
NATIONAL’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT
DEVICES OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT AND GENERAL
COUNSEL OF NATIONAL SEMICONDUCTOR CORPORATION. As used herein:
1. Life support devices or systems are devices or
systems which, (a) are intended for surgical implant
into the body, or (b) support or sustain life, and
whose failure to perform when properly used in
accordance with instructions for use provided in the
labeling, can be reasonably expected to result in a
significant injury to the user.
2. A critical component is any component of a life
support device or system whose failure to perform
can be reasonably expected to cause the failure of
the life support device or system, or to affect its
safety or effectiveness.
National Semiconductor
Corporation
Americas
Tel: 1-800-272-9959
Fax: 1-800-737-7018
Email: support@nsc.com
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Europe Fax: +49 (0) 1 80-530 85 86
Email: europe.support@nsc.com
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Response Group
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Japan Ltd.
Tel: 81-3-5639-7560
Fax: 81-3-5639-7507
www.national.com
Non-Isolated TO-220 15-Lead Package
Order Number LM1876T
NS Package Number TA15A
LM1876 Overture
™
Audio Power Amplifier Series
Dual 20W Audio Power Amplifier with Mute and Standby Modes
National does not assume any responsibility for use of any circuitry described, no circuit patent licenses are implied and National reserves the right at any time without notice to change said circuitry and specifications.
