MIC2169 Micrel MIC2169 500kHz PWM Synchronous Buck Control IC General Description Features The MIC2169 is a high-efficiency, simple to use 500kHz PWM synchronous buck control IC housed in a small MSOP-10 package. The MIC2169 allows compact DC/DC solutions with a minimal external component count and cost. The MIC2169 operates using a 3V to 14.5V input, without the need for any additional bias voltage. The output voltage can be precisely regulated down to 0.8V. The adaptive all N-Channel MOSFET drive scheme allows efficiencies, over 95%, across a wide load range. The MIC2169 senses current across the high-side N-Channel MOSFET, eliminating the need for an expensive and lossy current-sense resistor. Current limit accuracy is maintained via a positive temperature coefficient that tracks the increasing RDS(ON) of the external MOSFET. Additional cost and space are saved by the internal in-rush-current limiting and digital soft-start. The MIC2169 is available in a 10-pin MSOP package, with a wide junction operating range of -40C to +125C. All support documentation can be found on Micrel's web site at: www.micrel.com. * * * * * * * * * * * * 3V to 14.5V input voltage range Adjustable output voltage down to 0.8V Up to 95% efficiency 500kHz PWM operation Adjustable current limit senses high-side N-Channel MOSFET current No external current-sense resistor Adaptive gate drive increases efficiency Fast transient response - Externally compensated Overvoltage protection protects the load in fault conditions Dual mode current limit speeds up recovery time Hiccup mode short-circuit protection Small size MSOP 10-lead package Applications * * * * * * * Point-of-load DC/DC conversion Set-top boxes Graphic cards LCD power supplies Telecom power supplies Networking power supplies Cable modems and routers Typical Application VIN = 5V SD103BWS 95 90 4.7F VIN BST CS MIC2169 HSD EFFICIENCY (%) 0.1F VDD 1k IRF7821 2.5H 3.3V VSW COMP/EN LSD 100nF 150pF 4k GND MIC2169 Efficiency 100 100F IRF7821 10k 150F x 2 FB 85 80 75 70 65 60 55 50 3.24k VIN = 5V VOUT = 3.3V 0 2 4 6 8 10 12 14 16 ILOAD (A) MIC2169 Adjustable Output 500kHz Converter Micrel, Inc. * 2180 Fortune Drive * San Jose, CA 95131 * USA * tel + 1 (408) 944-0800 * fax + 1 (408) 474-1000 * http://www.micrel.com March 2009 1 M9999-032409 MIC2169 Micrel Ordering Information Part Number Pb-Free Part Number Frequency Junction Temp. Range Package MIC2169BMM MIC2169YMM 500kHz -40C to +125C 10-lead MSOP Pin Configuration VIN 1 10 BST VDD 2 9 HSD CS 3 8 VSW COMP 4 7 LSD FB 5 6 GND 10-Pin MSOP (MM) Pin Description Pin Number Pin Name Pin Function 1 VIN Supply Voltage (Input): 3V to 14.5V. 2 VDD 5V Internal Linear Regulator (Output): VDD is the external MOSFET gate drive supply voltage and an internal supply bus for the IC. When VIN is <5V, this regulator operates in dropout mode. 3 CS Current Sense (Input): Current-limit comparator noninverting input. The current limit is sensed across the MOSFET during the ON time. The current can be set by the resistor in series with the CS pin. 4 COMP Compensation (Input): Pin for external compensation. . 5 FB 6 GND Ground (Return). 7 LSD Low-Side Drive (Output): High-current driver output for external synchronous MOSFET. 8 VSW Switch (Return): High-side MOSFET driver return. 9 HSD High-Side Drive (Output): High-current output-driver for the high-side MOSFET. When VIN is between 3.0V to 5V, 2.5V threshold MOSFETs should be used. At VIN > 5V, 5V threshold MOSFETs should be used. 10 BST Boost (Input): Provides the drive voltage for the high-side MOSFET driver. The gate-drive voltage is higher than the source voltage by VIN minus a diode drop. M9999-032409 Feedback (Input): Input to error amplifier. Regulates error amplifier to 0.8V. 2 March 2009 MIC2169 Micrel Absolute Maximum Ratings(1) Operating Ratings(2) Supply Voltage (VIN) ................................................... 15.5V Booststrapped Voltage (VBST) .................................VIN +5V Junction Temperature (TJ) ..................-40C TJ +125C Storage Temperature (TS) ........................ -65C to +150C Supply Voltage (VIN) ..................................... +3V to +14.5V Output Voltage Range .......................... 0.8V to VIN x DMAX Package Thermal Resistance JA 10-lead MSOP ............................................. 180C/W Electrical Characteristics(3) TJ = 25C, VIN = 5V; bold values indicate -40C < TJ < +125C; unless otherwise specified. Parameter Condition Min Typ Max Units Feedback Voltage Reference ( 1%) 0.792 0.8 0.808 V Feedback Voltage Reference ( 2% over temp) 0.784 0.8 0.816 V 30 100 nA Feedback Bias Current Output Voltage Line Regulation 0.03 %/V Output Voltage Load Regulation 0.5 % 0.6 % Output Voltage Total Regulation 3V VIN 14.5V; 1A IOUT 10A; (VOUT = 2.5V)(4) Oscillator Section Oscillator Frequency 450 Maximum Duty Cycle 92 Minimum On-Time(4) 500 550 kHz % 30 60 ns 1.5 3 mA 5 5.3 V Input and VDD Supply PWM Mode Supply Current VCS = VIN -0.25V; VFB = 0.7V (output switching but excluding external MOSFET gate current.) Digital Supply Voltage (VDD) VIN 6V 4.7 Error Amplifier DC Gain 70 dB Transconductance 1 ms 8.5 A Soft-Start Soft-Start Current After timeout of internal timer. See "Soft-Start" section. Current Sense CS Over Current Trip Point VCS = VIN -0.25V 160 200 240 A +1800 Temperature Coefficient ppm/C Output Fault Correction Thresholds Upper Threshold, VFB_OVT (relative to VFB) +3 % Lower Threshold, VFB_UVT (relative to VFB) -3 % Notes: 1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifications do not apply when operating the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max), the junction-to-ambient thermal resistance, JA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive die temperature, and the regulator will go into thermal shutdown. 2. Devices are ESD sensitive, handling precautions required. 3. Specification for packaged product only. 4. Guaranteed by design. March 2009 3 M9999-032409 MIC2169 Micrel Electrical Characteristics(5) Parameter Condition Min Typ Max Units Gate Drivers Rise/Fall Time Into 3000pF at VIN > 5V Output Driver Impedance Source, VIN = 5V 30 Sink, VIN = 5V 6 Source, VIN = 3V 10 Sink, VIN = 3V Driver Non-Overlap Time ns 6 10 Note 6 10 20 ns Notes: 5. Specification for packaged product only. 6. Guaranteed by design. M9999-032409 4 March 2009 MIC2169 Micrel Typical Characteristics VIN = 5V 2.0 VFB (V) 0.7985 0.5 5 10 SUPPLY VOLTAGE (V) 15 VDD Line Regulation VDD REGULATOR VOLTAGE (V) 3 2 0.796 0.794 1 0.792 -60 -30 0 30 60 90 120 150 TEMPERATURE (C) 0 FREQUENCY (kHz) 1.0 0 5 10 15 550 540 530 520 510 500 490 480 Oscillator Frequency vs. Temperature 470 0.5 0.0 -60 -30 0 30 60 90 120 150 TEMPERATURE (C) 460 450 -60 -30 0 30 60 90 120 150 TEMPERATURE (C) Current Limit Foldback Overcurrent Trip Point vs. Temperature 4 0 260 5 10 15 VDD Load Regulation 5.02 5.00 4.98 4.96 4.94 4.92 4.90 0 VIN (V) VDD Line Regulation vs. Temperature 5.0 4.5 4.0 3.5 3.0 2.5 2.0 1.5 0.7980 VIN (V) 4 VDD (V) VFB (V) 0 5 0.802 0.798 0.7995 0.7990 1.0 0.804 VDD LINE REGULATION (%) 0.8000 1.5 6 0.800 VFB Line Regulation 0.8010 0.8005 VFB vs. Temperature 0.806 PWM Mode Supply Current vs. Supply Voltage 5 10 15 20 25 LOAD CURRENT (mA) 30 Oscillator Frequency vs. Supply Voltage 1.5 FREQUENCY VARIATION (%) 2.9 2.7 2.5 2.3 2.1 1.9 1.7 1.5 1.3 1.1 0.9 0.7 0.5 -40 -20 0 20 40 60 80 100120140 TEMPERATURE (C) QUIESCENT CURRENT (mA) IDD (mA) PWM Mode Supply Current vs. Temperature 1.0 0.5 0 -0.5 -1.0 -1.5 0 5 10 15 VIN (V) 240 220 ICS ( A) VOUT (V) 3 2 1 0 0 March 2009 200 180 160 Top MOSFET = Si4800 140 RCS = 1k 120 2 4 6 ILOAD (A) 8 10 100 -60 -30 0 30 60 90 120 150 TEMPERATURE (C) 5 M9999-032409 MIC2169 Micrel Functional Diagram CIN RCS VIN CS VDD 5V LDO D1 Current Limit Comparator VDD 5V High-Side Driver 5V Bandgap Reference BG Valid SW Clamp & Startup Current Ramp Clock CBST 2 RSW Driver Logic 5V Soft-Start & Digital Delay Counter Q1 BOOST Current Limit Reference 0.8V HSD L1 1.4 1000pF VOUT COUT 5V Low-Side Driver LSD Q2 PWM Comparator Enable Error Loop 0.8V VREF +3% VREF 3% Error Amp FB Hys Comparator R3 R2 MIC2169 COMP C2 GND C1 R1 MIC2169 Block Diagram Functional Description voltage. This causes the output voltage of the error amplifier to go high. This will also increase the PWM comparator tON time of the top side MOSFET, causing the output voltage to go up and bringing VOUT back in regulation. Soft-Start The COMP pin on the MIC2169 is used for the following two functions: 1. External compensation to stabilize the voltage control loop. 2. Soft-start. For better understanding of the soft-start feature, assume VIN = 12V. The COMP pin has an internal 6.5A current source that charges the external compensation capacitor. As soon as this voltage rises to 180mV (t = Cap_COMP x 0.18V/8.5A), the MIC2169 allows the internal VDD linear regulator to power up and as soon as it crosses the undervoltage lockout of 2.6V, the chip's internal oscillator starts switching. At this point, the COMP pin current source increases to 40A and an internal 11-bit counter starts counting. This takes approximately 2ms to complete. During counting, the COMP voltage is clamped at 0.65V. After this counting cycle, the COMP current source is reduced to 8.5A and the COMP pin voltage rises from 0.65V to 0.95V, the bottom edge of the saw-tooth oscillator. This is the beginning of 0% duty cycle which increases slowly causing the output voltage to rise slowly. The MIC2169 has The MIC2169 is a voltage mode, synchronous step-down switching regulator controller designed for high power without the use of an external sense resistor. It includes an internal soft-start function (which reduces the power supply input surge current at start-up by controlling the output voltage rise time), a PWM generator, a reference voltage, two MOSFET drivers, and short-circuit current limiting circuitry to form a complete 500kHz switching regulator. Theory of Operation The MIC2169 is a voltage mode step-down regulator. The block diagram, above, illustrates the voltage control loop. The output voltage variation due, to load or line changes, will be sensed by the inverting input of the transconductance error amplifier via the feedback resistors R3, and R2 and compared to a reference voltage at the non-inverting input. This will cause a small change in the DC voltage level at the output of the error amplifier which is the input to the PWM comparator. The other input to the comparator is a 5V triangular waveform. The comparator generates a rectangular waveform whose width tON is equal to the time from the start of the clock cycle t0 until t1, the time the triangle crosses the output waveform of the error amplifier. To illustrate the control loop, assume the output voltage drops due to sudden load turn-on, this would cause the inverting input of the error amplifier which is a divided down version of VOUT to be slightly less than the reference M9999-032409 6 March 2009 MIC2169 Micrel two hysteretic comparators that are enabled when VOUT is within 3% of steady state. When the output voltage reaches 97% of programmed output voltage, then the gm error amplifier is enabled along with the hysteretic comparator. From this point onwards, the voltage control loop (gm error amplifier) is fully in control and will regulate the output voltage. Soft-start time can be calculated approximately by adding the following four time frames: t1 = Cap_COMP x 0.18V/8.5A t2 = 12 bit counter, approx 2ms t3 = Cap_COMP x 0.3V/8.5A where: Inductor Ripple Current = VOUT x 200A is the internal sink current to program the MIC2169 current limit. The MOSFET RDS(ON) varies 30% to 40% with temperature; therefore, it is recommended that a 50% margin be added to the load current (ILOAD) in the above equation to avoid false current limiting due to increased MOSFET junction temperature rise. It is also recommended to connect the RCS resistor directly to the drain of the top MOSFET Q1, and the RSW resistor to the source of Q1 to accurately sense the MOSFETs RDS(ON). To make the MIC2169 insensitive to board layout and noise generated by the switch node. For this a 1.4 resistor and a 1000pF capacitor is recommended between the switch node and ground. A 0.1F capacitor, in parallel with RCS, should be connected to filter some of the switching noise. Internal VDD Supply Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 + t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms Current Limit The MIC2169 uses the RDS(ON) of the top power MOSFET to measure output current. Since it uses the drain to source resistance of the power MOSFET, it is not very accurate. However, this scheme is adequate to protect the power supply and external components during a fault condition by cutting back the time the top MOSFET is on if the feedback voltage is greater than 0.67V. In case of a hard short when feedback voltage is less than 0.67V, the MIC2169 discharges the COMP capacitor to 0.65V, resets the digital counter and automatically shuts off the top gate drive, and the gm error amplifier and the -3% hysteretic comparators are completely disabled and the soft-start cycles restarts. This mode of operation is called the "hiccup mode" and its purpose is to protect the down stream load in case of a hard short. The circuit in Figure 1 illustrates the MIC2169 current limiting circuit. The MIC2169 controller internally generates VDD for self biasing and to provide power to the gate drives. This VDD supply is generated through a low-dropout regulator and generates 5V from VIN supply greater than 5V. For supply voltage less than 5V, the VDD linear regulator is approximately 200mV in dropout. Therefore, it is recommended to short the VDD supply to the input supply through a 5 resistor for input supplies between 2.9V to 5V. MOSFET Gate Drive The MIC2169 high-side drive circuit is designed to switch an N-Channel MOSFET. The block diagram on page 6 shows a bootstrap circuit, consisting of D1 and CBST. It supplies energy to the high-side drive circuit. Capacitor CBST is charged while the low-side MOSFET is on and the voltage on the VSW pin is approximately 0V. When the high-side MOSFET driver is turned on, energy from CBST is used to turn the MOSFET on. As the MOSFET turns on, the voltage on the VSW pin increases to approximately VIN. Diode D1 is reversed biased and CBST floats high while continuing to keep the high-side MOSFET on. When the low-side switch is turned back on, CBST is recharged through D1. The drive voltage is derived from the internal 5V VDD bias supply. The nominal low-side gate drive voltage is 5V and the nominal high-side gate drive voltage is approximately 4.5V due the voltage drop across D1. An approximate 20ns delay between the high- and low-side driver transition is used to prevent current from simultaneously flowing unimpeded through both MOSFETs. MOSFET Selection The MIC2169 controller works from input voltages of 3V to 13.2V and has an internal 5V regulator to provide power to turn the external N-Channel power MOSFETs for high- and low-side switches. For applications where VIN < 5V, the internal VDD regulator operates in dropout mode, and it is necessary that the power MOSFETs used are sub-logic level and are in full conduction mode for VGS of 2.5V. For applications when VIN > 5V; logic-level MOSFETs, whose operation is specified VIN HSD 0.1F Q1 MOSFET N 2 L1 Inductor 1.4 RCS CS LSD Q2 MOSFET N 1000pF VOUT C1 COUT 200A Figure 1. The MIC2169 Current Limiting Circuit The current limiting resistor RCS is calculated by the following equation: RCS = RDS(ON) Q1 x IL 200A IL = I LOAD + March 2009 VIN x FSWITCHING x L FSWITCHING = 500kHz Cap_COMP V t4 = OUT x 0.5 x 8.5 A VIN C2 CIN (VIN - VOUT ) Equation (1) 1 2 (Inductor Ripple Current) 7 M9999-032409 MIC2169 Micrel where: at VGS = 4.5V must be used. It is important to note the on-resistance of a MOSFET increases with rising temperature. A 75C rise in junction temperature will increase the channel resistance of the MOSFET by 50% to 75% of the resistance specified at 25C. This change in resistance must be accounted for when calculating MOSFET power dissipation and in calculating the value of current-sense (CS) resistor. Total gate charge is the charge required to turn the MOSFET on and off under specified operating conditions (VDS and VGS). The gate charge is supplied by the MIC2169 gate-drive circuit. At 500kHz switching frequency and above, the gate charge can be a significant source of power dissipation in the MIC2169. At low output load, this power dissipation is noticeable as a reduction in efficiency. The average current required to drive the high-side MOSFET is: PCONDUCTION = I SW(rms)2 x RSW PAC = PAC(off) + PAC(on) RSW = on-resistance of the MOSFET switch V D = duty cycle O VIN Making the assumption the turn-on and turn-off transition times are equal; the transition times can be approximated by: where: IG[high-side](avg) = average high-side MOSFET gate current. QG = total gate charge for the high-side MOSFET taken from manufacturer's data sheet for VGS = 5V. The low-side MOSFET is turned on and off at VDS = 0 because the freewheeling diode is conducting during this time. The switching loss for the low-side MOSFET is usually negligible. Also, the gate-drive current for the low-side MOSFET is more accurately calculated using CISS at VDS = 0 instead of gate charge. For the low-side MOSFET: PAC = (VIN +VD ) x IPK x tT x fS where: tT = switching transition time (typically 20ns to 50ns) VD = freewheeling diode drop, typically 0.5V fS it the switching frequency, nominally 500kHz The low-side MOSFET switching losses are negligible and can be ignored for these calculations. Inductor Selection Values for inductance, peak, and RMS currents are required to select the output inductor. The input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. Generally, higher inductance values are used with higher input voltages. Larger peak-to-peak ripple currents will increase the power dissipation in the inductor and MOSFETs. Larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. Smaller peak-to-peak ripple currents require a larger inductance value and therefore, a larger and more expensive inductor. A good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. The inductance value is calculated by the equation below. IG[low-side](avg) = CISS x VGS x fS Since the current from the gate drive comes from the input voltage, the power dissipated in the MIC2169, due to gate drive, is: ( ) A convenient figure of merit for switching MOSFETs is the on resistance times the total gate charge RDS(ON) x QG. Lower numbers translate into higher efficiency. Low gate-charge logic-level MOSFETs are a good choice for use with the MIC2169. Parameters that are important to MOSFET switch selection are: * Voltage rating * On-resistance * Total gate charge The voltage ratings for the top and bottom MOSFET are essentially equal to the input voltage. A safety factor of 20% should be added to the VDS(max) of the MOSFETs to account for voltage spikes due to circuit parasitics. The power dissipated in the switching transistor is the sum of the conduction losses during the on-time (PCONDUCTION) and the switching losses that occur during the period of time when the MOSFETs turn on and off (PAC). L= VOUT x(VIN (max) - VOUT ) VIN (max) x fS x 0.2 x IOUT (max) where: fS = switching frequency, 500kHz 0.2 = ratio of AC ripple current to DC output current VIN(max) = maximum input voltage The peak-to-peak inductor current (AC ripple current) is: IPP = VOUT x (VIN ( max) - VOUT ) VIN ( max) x fS x L The peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor ripple current. PSW = PCONDUCTION + PAC M9999-032409 IG where: CISS and COSS are measured at VDS = 0 IG = gate-drive current (1A for the MIC2169) The total high-side MOSFET switching loss is: IG[high-side](avg) = QG x fS PGATEDRIVE = VIN IG[high-side](avg) + IG[low-side](avg) CISS x VGS + COSS x VIN tT = 8 March 2009 MIC2169 Micrel IPP = peak-to-peak inductor ripple current IPK = IOUT (max) + 0.5 x IPP The total output ripple is a combination of the ESR output capacitance. The total ripple is calculated below: The RMS inductor current is used to calculate the I2 x R losses in the inductor. IINDUCTOR(rms) 1 IP = IOUT (max ) x 1+ 3 IOUT (max) 2 2 VOUT = IC 2 OUT(rms) IPP = 12 The power dissipated in the output capacitor is: PDISS(C OUT ) = IC OUT(rms)2 x RESR(C OUT ) Input Capacitor Selection The input capacitor should be selected for ripple current rating and voltage rating. Tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. To maximize reliability, tantalum input capacitor voltage rating should be at least two times the maximum input voltage. Aluminum electrolytic, OS-CON, and multilayer polymer film capacitors can handle the higher inrush currents without voltage derating. The input voltage ripple will primarily depends upon the input capacitor's ESR. The peak input current is equal to the peak inductor current, so: PINDUCTORCu = IINDUCTOR(rms)2 x R WINDING The resistance of the copper wire, RWINDING, increases with temperature. The value of the winding resistance used should be at the operating temperature: ) R WINDING(hot) = R WINDING(20C) x 1 + 0.0042 x (THOT - T20C ) where: THOT = temperature of the wire under operating load T20C = ambient temperature RWINDING(20C) is room temperature winding resistance (usually specified by the manufacturer) Output Capacitor Selection The output capacitor values are usually determined capacitors ESR (equivalent series resistance). Voltage and RMS current capability are two other important factors to consider when selecting the output capacitor. Recommended capacitors are tantalum, low-ESR aluminum electrolytics, and POSCAPS. The output capacitor's ESR is usually the main cause of output ripple. The output capacitor ESR also affects the overall voltage feedback loop from stability point of view. See: "Feedback Loop Compensation" section for more information. The maximum value of ESR is calculated: RESR ) where: D = duty cycle COUT = output capacitance value fS = switching frequency The voltage rating of capacitor should be twice the voltage for a tantalum and 20% greater for an aluminum electrolytic. The output capacitor RMS current is calculated below: Maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. The high frequency operation of the MIC2169 requires the use of ferrite materials for all but the most cost sensitive applications. Lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. This is especially noticeable at low output power. The winding resistance decreases efficiency at the higher output current levels. The winding resistance must be minimized although this usually comes at the expense of a larger inductor. The power dissipated in the inductor is equal to the sum of the core and copper losses. At higher output loads, the core losses are usually insignificant and can be ignored. At lower output currents, the core losses can be a significant contributor. Core loss information is usually available from the magnetics vendor. Copper loss in the inductor is calculated by the equation below: ( ( IPP x (1 - D) + IPP x RESR COUT x fS VIN = IINDUCTOR(peak) x RESR(C IN ) The input capacitor must be rated for the input current ripple. The RMS value of input capacitor current is determined at the maximum output current. Assuming the peak-to-peak inductor ripple current is low: ICIN (rms) IOUT (max ) x D x (1- D) The power dissipated in the input capacitor is: PDISS(C IN ) = IC IN (rms) 2 x RESR(C IN ) Voltage Setting Components The MIC2169 requires two resistors to set the output voltage as shown in Figure 2. VOUT IPP where: VOUT = peak-to-peak output voltage ripple March 2009 9 M9999-032409 MIC2169 Micrel lost in the diode is proportional to the forward voltage drop of the diode. As the high-side MOSFET starts to turn on, the body diode becomes a short circuit for the reverse recovery period, dissipating additional power. The diode recovery and the circuit inductance will cause ringing during the high-side MOSFET turn-on. An external Schottky diode conducts at a lower forward voltage preventing the body diode in the MOSFET from turning on. The lower forward voltage drop dissipates less power than the body diode. The lack of a reverse recovery mechanism in a Schottky diode causes less ringing and less power loss. Depending upon the circuit components and operating conditions, an external Schottky diode will give a 1/2% to 1% improvement in efficiency. Feedback Loop Compensation The MIC2169 controller comes with an internal transconductance error amplifier used for compensating the voltage feedback loop by placing a capacitor (C1) in series with a resistor (R1) and another capacitor C2 in parallel from the COMP pin-to-ground. See "Functional Block Diagram." Power Stage The power stage of a voltage mode controller has an inductor, L1, with its winding resistance (DCR) connected to the output capacitor, COUT, with its electrical series resistance (ESR) as shown in Figure 3. The transfer function G(s), for such a system is: R1 Error Amp FB 7 R2 VREF 0.8V MIC2169 [adj.] Figure 2. Voltage-Divider Configuration Where: VREF for the MIC2169 is typically 0.8V The output voltage is determined by the equation: R1 VO = VREF x 1 + R2 A typical value of R1 can be between 3k and 10k. If R1 is too large, it may allow noise to be introduced into the voltage feedback loop. If R1 is too small, in value, it will decrease the efficiency of the power supply, especially at light loads. Once R1 is selected, R2 can be calculated using: R2 = VREF x R1 VO - VREF L External Schottky Diode An external freewheeling diode is used to keep the inductor current flow continuous while both MOSFETs are turned off. This dead time prevents current from flowing unimpeded through both MOSFETs and is typically 15ns. The diode conducts twice during each switching cycle. Although the average current through this diode is small, the diode must be able to handle the peak current. DCR VO ESR COUT Figure 3. The Output LC Filter in a Voltage Mode Buck Converter ID(avg) = IOUT x 2 x 15ns x fS (1+ ESR x s x C) G(s) = 2 DCR x s x C + s x L x C + 1+ ESR x s x C The reverse voltage requirement of the diode is: VDIODE(rrm) = VIN Plotting this transfer function with the following assumed values (L=2 H, DCR=0.009, COUT=1000F, ESR=0.025) gives lot of insight as to why one needs to compensate the loop by adding resistor and capacitors on the COMP pin. Figures 4 and 5 show the gain curve and phase curve for the above transfer function. The power dissipated by the Schottky diode is: PDIODE = ID(avg) x VF where: VF = forward voltage at the peak diode current The external Schottky diode, D1, is not necessary for circuit operation since the low-side MOSFET contains a parasitic body diode. The external diode will improve efficiency and decrease high frequency noise. If the MOSFET body diode is used, it must be rated to handle the peak and average current. The body diode has a relatively slow reverse recovery time and a relatively high forward voltage drop. The power 30 30 GAIN 7.5 -15 -37.5 -80 -80 100 100 3 1.10 4 1 .10 f 5 1 .10 6 1 .10 1000000 Figure 4. The Gain Curve for G(s) M9999-032409 10 March 2009 MIC2169 Micrel gm Error Amplifier It is undesirable to have high error amplifier gain at high frequencies because high frequency noise spikes would be picked up and transmitted at large amplitude to the output, thus, gain should be permitted to fall off at high frequencies. At low frequency, it is desireable to have high open-loop gain to attenuate the power line ripple. Thus, the error amplifier gain should be allowed to increase rapidly at low frequencies. The transfer function with R1, C1, and C2 for the internal gm error amplifier can be approximated by the following equation: 1 + R1 x S x C1 Error Amplifier(z) = gm x s x C1 + C2 1+ R1x C1x C2 x S ( ) C1 + C2 Figure 5. Phase Curve for G(s) It can be seen from the transfer function G(s) and the gain curve that the output inductor and capacitor create a two pole system with a break frequency at: fLC = The above equation can be simplified by assuming C2<<C1, 1 2 x L x COUT 1 + R1 x S x C1 Error Amplifier(z) = gm x s x (C1)(1+ R1x C2 x S) Therefore, fLC = 3.6kHz From the above transfer function, one can see that R1 and C1 introduce a zero and R1 and C2 a pole at the following frequencies: Fzero= 1/2 x R1 x C1 Fpole = 1/2 x C2 x R1 Fpole@origin = 1/2 x C1 Figures 7 and 8 show the gain and phase curves for the above transfer function with R1 = 9.3k, C1 = 1000pF, C2 = 100pF, and gm = .005-1. It can be seen that at 50kHz, the error amplifier exhibits approximately 45 of phase margin. By looking at the phase curve, it can be seen that the output capacitor ESR (0.025) cancels one of the two poles (LCOUT) system by introducing a zero at: fZERO = 1 2 x x ESR x COUT Therefore, FZERO = 6.36kHz. From the point of view of compensating the voltage loop, it is recommended to use higher ESR output capacitors since they provide a 90 phase gain in the power path. For comparison purposes, Figure 6 shows the same phase curve with an ESR value of 0.002. Figure 7. Error Amplifier Gain Curve Figure 6. The Phase Curve with ESR = 0.002 It can be seen from Figure 5 that at 50kHz, the phase is approximately -90 versus Figure 6 where the number is -150. This means that the transconductance error amplifier has to provide a phase boost of about 45 to achieve a closed-loop phase margin of 45 at a crossover frequency of 50kHz for Figure 4, versus 105 for Figure 6. The simple RC and C2 compensation scheme allows a maximum error amplifier phase boost of about 90. Therefore, it is easier to stabilize the MIC2169 voltage control loop by using high ESR value output capacitors. March 2009 11 M9999-032409 MIC2169 Micrel 100 71.607 OPEN LOOP GAIN MARGIN 50 0 42.933 50 100 100 3 1.10 4 6 5 1 .10 f 1 .10 1 .10 1000000 Figure 9. Open-Loop Gain Margin Figure 8. Error Amplifier Phase Curve Total Open-Loop Response The open-loop response for the MIC2169 controller is easily obtained by adding the power path and the error amplifier gains together, since they already are in Log scale. It is desirable to have the gain curve intersect zero dB at tens of kilohertz, this is commonly called crossover frequency; the phase margin at crossover frequency should be at least 45. Phase margins of 30 or less cause the power supply to have substantial ringing when subjected to transients, and have little tolerance for component or environmental variations. Figures 9 and 10 show the open-loop gain and phase margin. It can be seen from Figure 9 that the gain curve intersects the 0dB at approximately 50kHz, and from Figure 10, that at 50kHz, the phase shows approximately 50 of margin. OPEN LOOP PHASE MARGIN 250 269.097 300 350 360 10 10 100 3 1.10 4 1 .10 f 5 1 .10 6 1 .10 1000000 Figure 10. Open-Loop Phase Margin M9999-032409 12 March 2009 MIC2169 Micrel Design Example Layout and Checklist: 1. Connect the current limiting (R2) resistor directly to the drain of top MOSFET Q3. 2. Use a 5 resistor from the input supply to the VIN pin on the MIC2169. Also, place a 1F ceramic capacitor from this pin to GND, preferably not thru a via. 3. The feedback resistors R3 and R4/R5/R6 should be placed close to the FB pin. The top side of R3 should connect directly to the output node. Run this trace away from the switch node (junction of Q3, Q2, and L1). The bottom side of R3 should connect to the GND pin on the MIC2169. 4. The compensation resistor and capacitors should be placed right next to the COMP pin and the other side should connect directly to the GND pin on the MIC2169 rather than going to the plane. 5. Add a 1.4 resistor and a 1000pF capacitor from the switch node to ground pin. See page 7, Current Limiting section for more detail. 6. Add place holders for gate resistors on the top and bottom MOSFET gate drives. If necessary, gate resistors of 10 or less should be used. J1 +Vin 5-12V 7. Low gate charge MOSFETs should be used to maximize efficiency, such as Si4800, Si4804BDY, IRF7821, IRF8910, FDS6680A and FDS6912A, etc. 8. Compensation component GND, feedback resistor ground, chip ground should all run together and connect to the output capacitor ground. See demo board layout, top layer. 9. The 10F ceramic capacitor should be placed between the drain of the top MOSFET and the source of the bottom MOSFET. 10. The 10F ceramic capacitor should be placed right on the VDD pin without any vias. 11. The source of the bottom MOSFET should connect directly to the input capacitor GND with a thick trace. The output capacitor and the input capacitor should connect directly to the GND plane. 12. Place a 0.01F to 0.1F ceramic capacitor in parallel with the CS resistor to filter any switching noise. +VIN Cin=AVX TPSD686M020R0070 R2 470 ohm C4 10uF/6V D1 SD103BWS L1 Cout=AVX TPSD337M006R0045 CDRH127 / LD-1R0-MC 1.0uH 2 1 1 4R02 Ohm 8 7 6 5 MIC2169 C14 DIN C9 Open R1 R7 100K R8 4.02K R13 4 Q2 IRF7821 RES 1 2 3 FB C7 330uF R12 47 J4 1 + Vout C8 Open R3 10K R14 Open D2 C15 100pF C12 0.1uF/25V C11 Open R4 3.16k 5 R5 R6 4.64K 11.3K C 3.3V B 2.5V A 1.5V 1 2 4 6 GND 1 3 5 6 J3 1 0 Ohm 2 1 C10 0.1uF LSD 7 COMP/EN GND 3 4 Q1 2N7002E J2 SHDN C6 330uF/6.3V + 2 R10 1 RES + 2 VSW 8 R11 2 U1 9 1N5819HW HSD 1 Q3 IRF7821 1 2 3 BST 2 C5 0.1uF/25V 4 10 1 C16 0.1uF 8 7 6 5 1 Vin C13 1uF/16V 2 R9 10 1 C1 10uF/16V 2 2 + 2 C3 68uF 20V Vdd + 3 C2 68uF/20V CS 1 1 1 JP2 HEADER 3X2 J5 1 GND MIC2169BMM Evaluation Board Schematic March 2009 13 M9999-032409 MIC2169 Micrel MIC2169BMM Bill of Materials Item Part Number Manufacturer Description U1 MIC2169-YMM Micrel, Inc. Buck controller 1 Q2, Q3 IRF7821-TR IR 30V, N channel HEXFET , Power MOSFET 2 SI4390DY Vishay OR 0 D1 SD103BWS Vishay 30V , Schottky Diode 1 D2 1N5819HW Diodes Inc. 40V , Schottky Diode 1 SL04 Vishay OR 0 CMMSH1-40 Central Semi OR 0 CDRH127LDNP-1R0NC Sumida 1.0uH, 10A inductor 1 HC5-1R0 Cooper Electronic OR 0 L1 Qty. SER1360-1R0 Coilcraft OR 0 C1 C3225X7R1C106M TDK 10uF/16V, X7R Ceramic cap. 1 C2 , C3. TPSD686M020R0070 AVX 68uF, 20V Tantalum 2 594D686X0020D2T Vishay/Sprague OR 0 C2012X5R0J106M TDK 10uF/6.3V, 0805 Ceramic cap. 1 CM21X5R106M06AT AVX OR 0 3 C4 C5, C10 , C12 VJ1206Y104KXXAT Vishay Victramon 0.1uF/25V Ceramic cap. C6, C7 TPSD337M006R0045 AVX 330uF, 6.3V, Tantalum 2 C8 594D337X06R3D2T Vishay/Sprague Open 0 Vishay Dale open 0 C2012X7R1C105K TDK 1uF/16V, 0805 Ceramic cap. 1 GRM21BR71C105KA01B. muRata OR 0 VJ1206S105KXJAT Vishay Victramon OR 0 DIN 0 C15 VJ0603A102KXXAT Vishay Victramon 1000pF /25V, 0603 , NPO 1 C16 VJ0603Y104KXXAT Vishay Victramon 0.1uF/25V Ceramic cap. 1 R2 CRCW06034700JRT1 Vishay 470 Ohm , 0603, 1/16W, 5%. 1 R3 CRCW08051002FRT1 Vishay 10K / 0805 1/10W, 1% 1 R4 CRCW08053161FRT1 Vishay 3.16K /0805, 1/10W , 1% 1 R5 CRCW08054641FRT1 Vishay 4.64K /0805, 1/10W , 1% 1 C9 ,C11. C13 C14 R6 CRCW08051132FRT1 Vishay 11.3K / 0805, 1/10W, 1% 1 R8 CRCW06034021FRT1 Vishay 4.02K ,0603,1/16W, 1% 1 R9, CRCW12065R00FRT1 Vishay 5 ohm , 1/8W , 1206 , 1% 2 R10 CRCW12062R00FRT1 Vishay 2 Ohm , 1/8 W , 1206 , 1% 1 R12 CRCW12061R40FRT1 Vishay 1.4 Ohm , 1/8 W , 1206 , 1% 1 R14 J1, J3, J4, J5 2551-2-00-01-00-00-07-0 Notes: 1. Micrel.Inc 2. Vishay corp 3. Diodes. Inc 4. Sumida 5. TDK 6. muRata 7. AVX 8. International Rectifier 9. Fairchild Semiconductor 10. Cooper Electronic 11. Coilcraft 12. Central Semi March 2009 MilMax Open 0 Turret Pins 4 408-944-0800 206-452-5664 805-446-4800 408-321-9660 847-803-6100 800-831-9172 843-448-9411 847-803-6100 207-775-8100 561-752-5000 1-800-322-2645 631-435-1110 14 M9999-032409 MIC2169 Micrel Package Information 10-Pin MSOP (MM) MICREL, INC. TEL 2180 FORTUNE DRIVE + 1 (408) 944-0800 FAX SAN JOSE, CA 95131 + 1 (408) 944-0970 WEB USA http://www.micrel.com The information furnished by Micrel in this datasheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer. Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A Purchaser's use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser's own risk and Purchaser agrees to fully indemnify Micrel for any damages resulting from such use or sale. (c) 2005 Micrel, Incorporated. March 2009 15 M9999-032409 Mouser Electronics Authorized Distributor Click to View Pricing, Inventory, Delivery & Lifecycle Information: Micrel: MIC2169BYMM TR MIC2169BYMM MIC2169YMM MIC2169YMM TR MIC2169BYMM-TR MIC2169YMM-TR