February 1993 - Linear Technology

FEBRUARY 1993 VOLUME III NUMBER 1

New LTC1148/LTC1149

Switching Regulators

Maximize Efficiency

from Milliamps to Amps

IN THIS ISSUE . . .

COVER ARTICLE

New Switching Regulators

Maximize Efficiency ....... 1

Milton Wilcox and Randy Flatness

Editor's Page .................. 2

Richard Markell

DESIGN FEATURES

New Isolators Replace

“Optos”........................... 3

James Herr

High Speed, Precision,

Single-Supply Op Amps .. 5

William H. Gross

5V/3V, 12-Bit ADC Perfor–

mance

Comparison ......... 8

William Rempfer

DESIGN IDEAS

A Simple, Efficient Laser

Power Supply ................. 13

Jim Williams

Switching Regulator

Provides

15V from an

8V – 40V Input ................ 14

Brian Huffman

A Twelve-Bit, Micropower

Battery-Current Monitor . 16

Sammy Lum

200mA Output,

1.5V-to-5V Converter ....... 17

Jim Williams

LT1158 H-Bridge uses

Ground-Referenced Current

Sensing for Protection .... 18

Peter Schwartz

Multi-Output 3w Power

Supply Uses 2 AA Cells ... 20

Steve Pietkiewicz

LTC1157 Switch for 3.3V

PC Card Power ................ 22

Tim Skovmand

New Device Cameos ........ 23

by Milton Wilcox and

Randy Flatness

Continued on page 10

Introduction

The LTC1148 and LTC1149 are the

first stepdown switching regulators

that provide extremely high operating

efficiencies (typically greater than 90%)

over the entire load-current range de-

manded by the next generation of so-

phisticated notebook computers,

cellular phones, and handheld instru-

ments. The LTC1148/LTC1149 extends

battery life by providing efficiencies,

when these devices are in sleep or

standby modes, that are nearly as

high as those for full-power operation.

Furthermore, losses are reduced to

the point where no heat sinking is

required.

The extremely wide operating range

is illustrated by the typical efficiency

curve of Figure 1. The LTC1148 and

LTC1149 accomplish this feat by auto-

matically and smoothly changing from

synchronously switched current-mode

operation at high output currents to

Burst Mode

operation at low output

currents. Members of the LTC1148/

LTC1149 family can operate from in-

put voltages as low as 4V (LTC1148) to

as high as 48V (LTC1149). They are all

capable of 100% duty cycles for very-

low-dropout operation, and all have

built-in current limiting. Line and load

transient response is excellent under

a variety of conditions, including when

making the transition from Burst Mode

operation to full-current operation.

The small size and high efficiency of

the LTC1148/LTC1149 family make

DC-to-DC conversion feasible in the

most restricted spaces of today’s por-

table electronics. Take, for example,

the problem of locally dropping 5V

to 3.3V on a logic board. In many

cases the dissipation of a linear regu-

lator, even for this modest voltage

drop, is unacceptable, because there

is simply no way to remove the heat

from the enclosed space. A linear

regulator delivering 1A in this ap-

plication would dissipate over 1.7W.

Figure 1. Thanks to Burst Mode

operation,

the LTC1148-5 is capable of greater than 90%

efficiency from 20mA to 2A of output current

LOAD CURRENT (A)

20mA

EFFICIENCY (%)

100

0.2A 2A

1148_1.eps

LTC1148-5

= 10V

LINEAR TECHNOLOG



LINEAR TECHNOLOG



LINEAR TECHNOLOG



Linear Technology Magazine • February 1993

DESIGN FEATURES

LTC Switching Regulators Break the 90%

Efficiency Barrier—The LTC Technology

Machine Marches On

This issue of Linear Technology

proudly spotlights our LTC1148/1149

switching regulators, which break the

elusive 90% efficiency barrier. Not only

do we pass the barrier but we burst

through, without stopping, across the

entire load range from milliamps to

amps. Our lead article introduces the

LTC1148/LTC1149 synchronous

switching regulators, which provide

very high efficiencies at all current

levels. In addition to the 94%-efficient

5 volt to 3.3 volt converter mentioned

in the article, we have achieved the

following efficiencies in the lab since

the article was written:

• An LTC1148 converter using 6-to-

12 volt input with a 5 volt output at

4 amps output current. This

converter achieves greater than

90% efficiency.

• An LTC1148 circuit that converts

6 volts to 5 volts at 1/2 amp,

achieving 96% efficiency.

• An LTC1149 24 volt to 5 volt

converter with output current of >1

amp, achieving 90% efficiency.

• A high-current circuit using the

LT1158 (featured in the February

1992 LT Magazine), which converts

5 volts to 3.3 volts. This circuit’s

efficiency is 90% at 10 amps and

88% at 15 amps.

Efficiency, by itself, is no panacea,

but higher and higher efficiencies im-

ply less and less heat in your enclo-

sure. This may not be a problem for a

remote data-collection system in the

Canadian North, but it certainly is a

problem in notebook and laptop com-

puters. (If you require additional

information on these leading-edge de-

velopments in power-supply efficien-

cies, do not hesitate to call the

Applications Group at the LTC fac-

tory.)

Another design feature highlights

signal isolation applications and cir-

cuitry using a new isolation device

from LTC. This device, the LTC1145/

LTC1146 uses a special integrated-

circuit lead frame to form the isolation

capacitors. The device uses two dice

(one driver and one receiver) coupled

FAE Cameo: Tom Mosteller

through the isolation capacitors on the

opposite ends of the lead frame. The

article also describes how the design

provides sufficient high-voltage isola-

tion for even the most critical require-

ments with high transient immunity.

High speed, precision operational

amplifiers are also featured in this

issue, as are new 12-bit ADCs. Six new

high-speed op amps—their circuit to-

pologies, performance specifications,

and some application areas—are dis-

cussed at length in another Design

Feature. Analog-to-Digital converters

are featured in an article describing the

new 3.3 and 5 volt, 12-bit converters

from LTC.

Also in this issue, I am pleased to

note, is a large collection of Design

Ideas. These range from a helium-neon

laser power supply, to a multi-output

power supply from two AA batteries, to

an H-bridge driver circuit for a 15-amp

DC motor. Circuitry for PCMCIA card

power conditioning and 12-bit battery

current monitoring are also featured in

the Design Idea area. Good reading!

LTC has eighteen Field Application

Engineers (or FAEs) spread through-

out the world to assist customers in the

design and selection of circuits avail-

able from LTC. Each FAE has a sys-

tem-design background and has a

working knowledge of the world from a

system designer’s point of view. All of

our FAEs are available by phone and,

in certain situations, in person, to help

with the design of your circuitry. This

space will profile one FAE per issue.

Tom Mosteller works out of LTC’s

Northeast Office. He covers the states

of Pennsylvania, Maryland, West Vir-

ginia, Virginia, and Delaware, and parts

of New York and New Jersey. Tom’s

by Richard Markell

expertise is in the areas of signal pro-

cessing and switching power supply

design. He also has extensive experi-

ence in process control. “My greatest

challenge in the signal processing field,”

writes Tom, “was an ECG-monitor front

end that amplified millivolts while ig-

noring kilovolts. This monitor had an

isolated power supply, low-level am-

plifiers, slew-rate limiters, switched-

capacitor filters, a pressure-bridge

exciter and amplifiers, and A-to-D

converters.”

Tom also enjoys writing. He wrote

most of Application Note 50 on a por-

table computer while on a balcony

overlooking the beach in Lewes,

Delaware (while he was on vacation,

no less!). Tom’s byline has also ap-

peared in PC Magazine, Circuit Cellar

Ink, and other publications.

Tom has been married to his wife

Rose for eleven years. They have one

son, Kyle, who is seven years old and in

the second grade. Tom enjoys playing

computer games and writing stories

with Kyle on the computer. (Who en-

joys computer games?) He also enjoys

audio, YMCA Indian Guides, camping

and fishing with Kyle, and helping with

Rose’s software business. Tom can be

reached through LTC’s Northeast Re-

gion Sales Office as listed on the back

of this magazine.

EDITOR'S PAGE

Linear Technology Magazine • February 1993

DESIGN FEATURES

The LTC1145 and LTC1146 are a

new generation of signal isolators. Pre-

viously, signal isolation was accom-

plished by means of opto-isolators.

Light from an LED was detected across

a physical isolation barrier by either a

photo diode or transistor and con-

verted to an electrical signal. Isolation

levels up to thousands of volts were

easily achieved.

Attempts have been made to provide

signal isolation on a single silicon die.

Problems arose due to reliability con-

straints of damage from ESD or over-

voltage. A new technique, using a

capacitive lead frame, overcomes the

problems associated with single-pack-

age signal isolation. Further, this tech-

nique is suitable for use in thin

surface-mount packages—a solution

not available with opto-isolators. The

data rates are 200kbps for the LTC1145

and 20kbps for the LTC1146. Both

parts can sustain over 1000V across

their isolation barriers.

Circuit Design

The isolator IC includes two dice

(see Figure 1). The first is a driver

designed to drive two isolation capaci-

tors. The second contains a receiver

and filter which detect the drive signal

and provide a digital output. The driver

is powered from the input signal. This

allows interchangeability with opto-

isolators, since no auxiliary power

source is needed on the input side. The

driver contains a high-frequency oscil-

lator whose differential outputs are

capacitively coupled to the receiver.

The receiver detects the presence or

absence of the driver’s oscillations and

outputs a logic 1 or 0, as appropriate.

The driver consists of a bias circuit,

a Schmitt trigger, and a differential

output oscillator to drive the isolation

capacitors. The Schmitt trigger pre-

vents the oscillator from driving the

capacitors until the input rises to a

sufficient level (3V for the LTC1145 or

2V for the LTC1146) to provide a clean

square-wave output. Two versions of

the drivers are available, one with a

5MHz oscillator frequency (LTC1145),

and one with a 400kHz oscillator fre-

quency (LTC1146). The 5MHz unit pro-

vides reliable data communication at

200kbps whereas the 400kHz unit can

operate at 20kbps. Speed has its price:

the LTC1145 consumes 600µA while

the LTC1146 requires only 60µA of

input current. These power levels are

significantly below LED opto-isolators.

The receiver is designed to detect

differential drive from the capacitors. A

differential input comparator with

300mV of hysteresis detects the in-

coming signal while rejecting common-

mode noise. The output of the

comparator triggers a one-shot circuit

with a period longer than that of the

drive frequency. The retriggerable one-

shot has a constant output as long as

the input signal exists. When an input

signal is absent for one time-out pe-

riod, the one-shot goes low.

Also connected to the one-shot is a

two-bit counter that works as a digital

filter. The counter is clocked from a

second oscillator running at around

10MHz. The output of the counter

requires four oscillator clock periods to

change state and is gated by the output

of the one-shot. When an input signal

triggers the one-shot, a logical 1 is

presented to the data input of the filter.

Four clock periods later the output of

the filter goes high. If the signal is

absent for four clock cycles, the filter

output returns low. This digital filter

eliminates possible erratic operation

from spikes or temporary overloads of

the input amplifier. The noise or inter-

ference signal must exist for four clock

periods of the digital filter before an

output-stage change can occur.

The LTC1145’s digital filter can be

clocked externally to lower its

effective bandwidth. The LTC1146’s

New Isolators Replace “Optos” by James Herr

DATA INPUT

GND1 GND2

EDQ

1-SHOT

OSCILLATOR

IN OUT

FILTER

CD IN

CLOCK

DETECTOR

1-SHOT

OUTPUT

DATA

OUTPUT

EXT OSC

TTL

BUFFER

–

OSCILLATOR TTL

BUFFER

DRIVER RECEIVER

ISOLATION

BARRIER

1145_1.eps

Figure 1. Two-chip design provides isolation barrier

Linear Technology Magazine • February 1993

DESIGN FEATURES

digital-filter bandwidth

can be reduced by connect-

ing a capacitor at pin 8.

Lower filter bandwidths

are desirable to eliminate

longer noise bursts. Experi-

mental work using this

isolator and filter has

shown no corrupted data

over a wide range of com-

mon-mode inputs.

Isolation Capacitors

A specialized lead frame

is needed to form the isolation capaci-

tors. This lead frame can be manufac-

tured with the same high-volume

techniques used for conventional in-

tegrated-circuit lead frames, and at

approximately the same cost. The two

dice (driver and receiver) are placed at

the opposite ends of the lead frame

and coupled through the isolation

capacitors. The capacitance between

the input and output is on the order

of 1pF. This provides sufficient isola-

tion in even the most critical of applica-

tions and is suitable for handling high

voltages with high dV/dt.

Each 1pF capacitor is formed by

three parallel metal fingers spaced

about 20 mils apart. The capacitors’

opto-isolator replacement.

One possible application is

an isolated RS232 receiver.

The D

pin of LTC1145 is

driven by an RS232 signal

through a 5.1kΩ resistor

(Figure 3). The D

OUT

pin of

the LTC1145 presents

isolated, TTL-compatible

output signals. The GND2

pin of the LTC1145 is con-

nected to the same ground

potential as the receiving

end of the link. The isolator

can accommodate differences of up to

1kV between GND1 and GND2.

Another application is an isolated,

thermocouple-sensed temperature-to-

frequency converter (see Figure 4). The

output of I

produces a 0kHz–1kHz

pulse train in response to a 0°C to

100°C temperature excursion (see LTC

Application Note 45 for the details).

The pulses from I

drive the D

pin of

LTC1146. The GND1 pin is connected

to the same ground potential as I

. The

OUT

pin of LTC1146 presents isolated,

TTL-compatible output signals. The

circuit consumes only 460µA maxi-

mum, allowing it to operate from a 9V

battery.

Figure 4. Isolated temperature-to-frequency converter

118

OSC

GND2

GND1

OUT

ISOLATION

BARRIER

1145_3.eps

RS232

5.1kΩ

+5V

TTL OUT

118

OSC

GND2

GND1

OUT

ISOLATION

BARRIER

1145_2.eps

metal fingers and bonding posts re-

place the 5 center pins on each side of

the 18 pin package, as illustrated in

Figure 2. The dielectric for the capaci-

tors is the plastic package molding

compound. The material has a high

dielectric constant and a high break-

down voltage.

Applications

The LTC1145/LTC1146 can be used

in a wide range of applications where

voltage transients, differential ground

potentials, or high noise may be en-

countered, such as isolated serial data

interfaces, isolated analog-to-digital

converters for process control, iso-

lated FET drivers, and low-power

–

1145_4.eps

–

A1

LTC1049

–

LT1025

+6V

GND R 6.81k*

1.5k

100°C

TRIM

C3

0.47µF

100k

"A"

CONTROL

AMPLIFIER

0.02µF

390pF

†

+6V

Q2

2N3906

10k

C1

100pF

Q1

2N3904

C4

300pF

100k OUTPUT

0-100°C = 0-1kHz

VF➞

LTC201

CHARGE

PUMP

240k +6V

6.8µFLT1004-1.2

TYPE K

THERMOCOUPLE

†



= 74C14

= POLYSTYRENE

FOR GENERAL PURPOSE 4mV FULL SCALE V F DELETE

THERMOCOUPLE/LT1025 PAIR AND DRIVE POINT "A."

➞

*IRC/TRW-MTR-5/+120PPM

LTC1146

OUT

GND2

GND1

Figure 3. Isolated, low-power RS232

Figure 2. LTC1145/

LTC1146 pin description

Linear Technology Magazine • February 1993

DESIGN FEATURES

High Speed Comes to Precision,

Single-Supply Op Amps by William H. Gross

of many transistors and require care-

ful thermal layout; therefore precision

op amps require much larger dice than

simple commodity op amps. For this

reason precision quad op amps have

not been available in the narrow SO

package; they are simply too wide to fit.

To address this problem, LTC devel-

oped a special lead frame for the stan-

dard narrow SO16 mold. This new lead

frame maximizes the area available for

the die, but has an internal length-to-

width ratio of 3 to 1. If one die were used

to make a quad op amp for this pack-

age, it would also have to have a length-

to-width ratio (aspect ratio) of 3 to 1.

The introduction of LTC’s 12-bit A-

to-D converters that operate on 3.3V

supplies, such as the LTC1282,

LTC1287, and LTC1289, emphasizes

the need for low voltage, 12-bit-accu-

rate op amps. For converters operating

with a 2.5V reference, we would like an

op amp whose output swings from

ground to 2.5V. The new LTC A-to-D

converters include the sample-and-hold

on the same chip; therefore an op amp

is needed that settles cleanly after the

input hold cap is connected. For the

160kHz-sampling LTC1282, the am-

plifier must deliver several milliamps of

output current and settle in less than

1µs. Since the LSB in a 12-bit system

using a 2.5V reference is only 610µV,

the settling must have no thermal tails.

Process Technology

The LT1211 family of amplifiers are

manufactured on LTC’s complemen-

tary bipolar process. The low drift and

noise requirements of the amplifier,

combined with the low open-loop out-

put impedance required to drive sam-

pling A-to-Ds precludes using a CMOS

process. PNP transistors are used in

the input stage so the common-mode

range includes the negative supply. To

make a fast amplifier we need fast

PNPs, hence we could not use our

standard bipolar process with lateral

PNPs. The complementary process has

36V NPNs and PNPs that both have

cutoff frequencies of 600MHz. Both

transistors in this process are opti-

mized for high gain, excellent match-

ing, and low noise.

Circuit Topology

The circuit topology of the LT1211 is

conventional, with some high speed

innovations. The amplifier consists of

two gain stages. The first is a differen-

tial-voltage to single-ended-current

(transconductance) stage whose out-

put drives a high-input-impedance,

inverting voltage-gain stage configured

High aspect ratios cause manufactur-

ing problems that increase costs. To

solve this problem, we die-attach two

dual op amp dice to the lead frame and

bond them out with the standard pin-

out. This gives us an effective aspect

ratio of 3 to 1 without having to actually

make a die with that aspect ratio.

Performance Goals

There is a trend to lower supply

voltages for analog signal processing.

Engineers can no longer depend on

having ±15V or ±12V supplies avail-

able. Often, only a single +5V supply is

available, and the trend is to +3.3V.

Our design goal was to make these

single-supply op amps operate well

with supply voltages as low as 2.5V.

The LT1211–LT1216 Family

LTC proudly introduces a new fam-

ily of dual and quad single-supply pre-

cision op amps. These new amplifiers

are fast, with slew rates of up to 50V/µs

and gain-bandwidth products to

28MHz. All members of the family are

unity-gain stable and operate on any

single supply between 2.5V and 36V as

well as on split supplies from ±2V to

±18V. The design of an op amp involves

a trade off between slew rate and sup-

ply current consumption. Rather than

force a major compromise, three new

amplifiers were developed. Each of the

three amplifier types is offered as both

a dual and a quad; hence there are six

part numbers.

The LT1211 dual and the LT1212

quad amplifiers draw the least quies-

cent current in the family, only 1.3mA

per amplifier. The LT1211/LT1212

op amps have a 14MHz gain-band-

width product and a peak slew rate of

7V/µs. They settle to 0.01% of a 10V

step in 2.2µs.

The LT1213 dual and the LT1214

quad amplifiers have twice the gain-

bandwidth product of the LT1211/

LT1212 amps at 28MHz. The peak slew

rate of the LT1213/LT1214 is 12V/µs

and the supply current goes up to 3mA

per amplifier. They settle to 0.01% of a

10V step in 1.2µs.

The LT1215 dual and the LT1216

quad amplifiers are the fastest in the

family, with a slew rate of 50V/µs. The

gain-bandwidth product is 23MHz, and

the quiescent supply current is 5mA

per amplifier. They settle to 0.01% of a

10V step in 480ns.

The dual op amps are available with

the industry standard pin-out in either

8-pin SO or 8-pin mini-DIP packages.

The quad op amps have the standard

pin-out in the 14 pin DIP and the same

pin-out, with two no-connect pins on

one end, in the narrow (150mil) SO16

package. LTC is the first to offer a pre-

cision quad op amp in the narrow SO

package. Precision amplifiers consist

There is a trend

to lower supply voltages for

analog signal processing.

Engineers can no longer

depend on having

15V or

12V supplies available.

Often, only a single

supply is available, and the

trend is to

3.3V

Linear Technology Magazine • February 1993

DESIGN FEATURES

as a Miller integrator. An emitter fol-

lower buffers the output of the second

stage and sources current into the

load. The current in this follower is

monitored with a second loop that

provides the output sink current.

One advantage of general-purpose

op amps is that they can take large

differential inputs without excessive

input current flowing. This is because

they use lateral PNPs with emitter-

base breakdown voltages greater than

36V in the input stage. Conventional

NPNs and fast PNPs have low emitter-

base breakdown voltages and require

clamps across the inputs to protect

them. To make the LT1211 as easy to

use as possible, we use lateral PNP

transistors in the input stage. Refer-

ring to the simplified schematic (Fig-

ure 1), the input PNP emitter followers

(Q1 and Q2) are lateral transistors

with high breakdown. The differential-

amplifier PNPs (Q3 and Q4) that con-

vert the differential voltage to a current

are fast transistors. Because Q1 and

Q2 drive the high impedance bases of

Q3 and Q4, the base-emitter capaci-

tance of Q1 and Q2 couples the input

signal to the faster PNPs (Q3 and Q4)

even above the cutoff frequency of the

lateral transistors.

An active load, Q5 through Q9,

maximizes the gain of the first stage for

low noise and low offset-voltage drift.

The base current of Q8 matches the

base current of Q10 for low drift. The

capacitor C

introduces a pole and a

zero in the open-loop gain by rolling off

half the input-stage gain above 1MHz.

This reduces the unity-gain frequency

to half the gain-bandwidth product

and therefore increases the gain and

phase margin.

The output current from the first

stage drives the second stage, consist-

ing of Q10, Q11, and Q12. Q10 and

Q11 are emitter followers to increase

the input impedance of this second

stage. Q12 operates in a common-

emitter configuration with a current-

source load for maximum voltage gain.

The capacitor C

turns the gain stage

into a Miller integrator. For good phase

margin in the amplifier, the integrator

must work well at the amplifier unity-

gain frequency. Since Q10 and Q11

operate at fairly low currents, they

generate significant phase shift that

limits the accuracy of the integrator. To

improve the frequency response of this

stage, we add C

and R

to feed-for-

ward the signal around Q10 and Q11.

The output stage buffers the second

stage with emitter follower Q15 and a

current-sink circuit. In order to sink

output current and swing all the way

to the negative supply, an NPN tran-

sistor, Q16, must drive the output.

Q14’s collector current is one tenth

that of follower Q15’s and is sub-

tracted from Q13’s emitter current.

Then Q13’s collector current is com-

pared with current source I8; the ex-

cess drives Q16. When the current in

Q15 drops, more current drives Q16

and the amplifier sinks current. Ca-

pacitor C

stabilizes this feedback loop,

which includes common-base tran-

sistor Q13. Because the gain of the

loop is quite large, Q15 never turns off

and the open loop output impedance

stays low for overall amplifier stability.

First order, the overall DC gain of

the op amp is the input stage transcon-

ductance times the current gain of

Q10, Q11, Q12, and Q15, times the

load resistor. The transconductance

of the LT1211 first stage is 500nA/mV;

the gain of the transistors is about 100

each; with a 500Ω load the calculated

gain is twenty-five million (25V per

µV). The actual gain is about 2.5 mil-

lion, due to second-order effects like

Early voltage. The IC layout is opti-

mized to eliminate thermal feedback

that would reduce gain. The layout

also optimizes channel-to-channel

separation, which is typically 100nV/

V (140dB).

Performance

Table 1 describes the typical AC per-

formance of each of these amplifiers.

Table 2 summarizes the DC electri-

cal performance of the low cost grades

of these new amplifiers. The dual op

amps also have selections for improved

offset voltage and drift in the DIP

packages.

Applications

Instead of describing several known

op amp circuits, I will now discuss

some general applications. The LT1211

family of amplifiers are the optimum

solution whenever a combination of

speed and accuracy is needed on low

supply voltage.

–IN

Q3 Q4

+IN

Q10 C

Q11

Q12



Q14

BIAS

–

OUT

1211_1.eps

Q15

Q13

Q1

LQ2

Q16

Figure 1. LT1211 simplified schematic

Linear Technology Magazine • February 1993

DESIGN FEATURES

With multimedia becoming more

important, the need for CD-quality

audio amplifiers that operate on single

+5V supplies is growing. The LT1211

can handle 1V

rms

at a non-inverting

gain of one while operating on single

+5V supply. The LT1211 delivers dis-

tortion-free signals over the full 20Hz-

to-20kHz range; THD is less than

0.001% for all signals up to 1V

rms

Referenced to 1V, the signal-to-noise

ratio is over 110dB and the supply

current is only 2.6mA for both stereo

channels.

The LT1216 is an ideal op amp for

precision anti-aliasing filters. The

20MHz gain bandwidth supports high-

Q filters up to 1MHz and the high slew

rate results in a power bandwidth of

2.5MHz at 2.5V

P–P

. The quad is a natu-

ral for state-variable filters where three

or four op amps are needed for each

pole pair. The low offset-voltage drift of

these amplifiers ensures that self-cali-

brating systems are accurate between

calibrations.

The LTC1196 is an 8-bit A-to-D

converter that operates on a 3.3V sup-

ply with an external 2.5V reference.

This converter is fast; it samples the

too bad considering that there are

4085 codes that are OK. When using

the LTC1289, (another 3.3V, 12-bit A-

to-D converter) these codes can be

recovered because an external refer-

ence is used. The external reference

can be positioned a little above ground

to shift the full 4096 codes into the

output range of the LT1211.

Summary

The LT1211 family brings a level of

precision and speed to low voltage sys-

tems that was previously unavailable.

Operation is guaranteed over the full

military temperature range with single

supplies as low as 2.5V. Input offset

voltage, input bias current, open-loop

gain, and CMRR are comparable with

the best op amps available today. The

three amplifiers span a seven-to-one

range of slew rate with a four-to-one

range of supply current. The open-loop

output impedance is low and these

amps settle to microvolts in fractions of

a microsecond, making them ideal for

data-acquisition systems. These am-

plifiers are available in small surface-

mount packages, including for the first

time the narrow SO16 for the quads.

input at a 450kHz rate. The input

range is from 10mV±5mV to 2.5V±5mV.

The LT1211/LT1213/LT1215 output

swing is guaranteed to be such that

only code “0000 0000” is missing when

the devices are operated on the same

3.3V single supply.

The LTC1282 is a 12-bit A-to-D

converter with an internal reference

that operates on a single 3.3V supply.

The nominal input range is from 0V to

2.5V. Since the LT1211 can only swing

to within 5mV of ground, the lowest

nine codes cannot be used. This is not

The LT1211 Family

of amplifiers are the

optimum solution

whenever a combination

of speed and accuracy

is needed on low

supply voltage

DC Parameters LT1211/12 LT1213/14 LT1215/16

Max offset voltage 275µV 275µV 450µV

Max offset voltage drift 6µV/°C6µV/°C10µV/°C

Max input offset current 30nA 40nA 120nA

Max input bias current 125nA 200nA 600nA

Min input voltage range

VS = 3.3V 0V – 1.8V 0V – 1.8V 0V – 1.3V

VS = 5.0V 0V – 3.5V 0V – 3.5V 0V – 3.0V

Min CMRR 86dB 86dB 86dB

Min output voltage swing

VS = 3.3V 0.01V – 2.5V 0.01V – 2.5V 0.01V – 2.5V

VS = 5.0V 0.01V – 4.2V 0.01V – 4.2V 0.01V – 4.2V

Min open loop gain

VS = 3.3V or 5.0V 250V/mV 250V/mV 150V/mV

VS = ±15V 1200V/mV 1200V/mV 1000V/mV

Min channel separation 128dB 128dB 130dB

Min output current 20mA 30mA 30mA

Max supply current per amp 1.8mA 3.8mA 6.6mA

Operating supply voltage 2.5V – 36V 2.5V – 36V 2.5V – 36V

AC Parameters LT1211/12 LT1213/14 LT1215/16

Slew rate, VS = ±15V 7V/µs 12V/µs 50V/µs

Slew rate, VS = +5V 4V/µs 8V/µs 30V/µs

Settling time, 2V to 0.01% 800ns 500ns 250ns

Settling time, 10V to 0.01% 2.2µs 1.2µs 480ns

Gain bandwidth product 14MHz 28MHz 23MHz

Unity gain cross frequency 7MHz 13MHz 12MHz

Phase margin 55°45°45°

Noise voltage, 0.1 to 10Hz 125nVP-P 125nVP-P 400nVP-P

Spot noise voltage, 10Hz 12.5nV√Hz 10nV√Hz 15nV√Hz

Spot noise voltage, 1kHz 12nV√Hz 10nV√Hz 12.5nV√Hz

Table 1. Typical AC performance Table 2. Guaranteed DC performance of low-cost grades

Linear Technology Magazine • February 1993

DESIGN FEATURES

Five and Three Volt, 12-Bit ADC

Performance Comparison

Four new sampling A/D converters

from Linear Technology, the LTC1273,

LTC1275, LTC1276 and LTC1282,

stand out above the crowd. These new

5V and 3V 12-bit ADCs offer the best

speed/power performance available

today (see Figure 1). They also provide

precision references, internally

trimmed clocks, and fast sample-and-

holds. With additional features such

as single-supply operation and high-

impedance analog inputs, they reduce

system complexity and cost. This ar-

ticle will describe the new ADCs and

discuss the performance and power

trade-offs that should be considered

in selecting a 5V or 3V A/D converter.

Complete ADCs Provide

Lowest Power, Highest Speed

on Single or Dual Supplies

The LTC1273, LTC1275, LTC1276,

and LTC1282 provide complete A/D

solutions at previously impossible

speed/power levels. As shown in Table

1, the LTC1273, LTC1275, and

LTC1276 all have the same 300kHz

maximum sampling rate and 75mW

typical power dissipation. The LTC1273

digitizes 0-to-5V inputs from a single

5V rail. The LTC1275 and LTC1276

operate on ±5V rails and digitize ±2.5V

and ±5V inputs, respectively.

by William Rempfer

Power Input Sample S/(N + D) typ PDISS

Device Supplies Range Rate at fINPUT (typ)

LTC1273 +5V 0–5V 300kHz 70dB at 100kHz 75mW

LTC1275/6 ±5V ±2.5V/±5V 300kHz 70dB at 100kHz 75mW

LTC1282 +3V 0–2.5V 140kHz 68dB at 70kHz 12mW

or ±3V or ±1.25V 140kHz

5V ADCs Sample at

300kHz on 75mW of Power

The LTC1273, LTC1275, and

LTC1276 have excellent DC specs,

including ±1/2LSB linearity and

25ppm/°C full-scale drift. In addition,

they have excellent dynamic perfor-

mance. As Figure 2 shows, the ADCs

typically provide 72dB of Signal to

Noise plus Distortion (11.7 effective

bits) at the maximum sample rate of

300kHz. The S/(N + D) ratio is over

70dB (11.3 effective bits) for input

frequencies up to 100kHz.

Table 1. Four new complete ADCs offer high speed and low power on single or dual 5V or 3V

supplies

SAMPLE RATE/POWER RATIO (kHz/mW)

1273_1.eps

010 100 1000 10000

SAMPLE RATE (kHz)

AD7880

AD7870

MAX163/4/7

AD1674 AD678

ADS7800 AD7886

AD1671

CS5412

★

LTC1273/5/6 ON 5V OR ±5V

LTC1282 ON 3V OR ±3V

CS5012A

LTC1292

Figure 1. The LTC 1273/5/6 and the LTC1282 have up to

45 times higher speed/power ratios than competitive ADCs

This 300kHz sample rate and dy-

namic performance comes at a power

level that is more stingy than that of

any other ADC in this speed range.

Figure 1 shows a graph of speed/

power ratios for the competitive ADCs.

The speed/power ratio is defined as

the maximum sample rate in kHz di-

vided by the typical power dissipation

The LTC1282 samples at 140kHz

and typically dissipates only 12mW

from either 3V or ±3V supplies. It

digitizes 0–2.5V inputs from a single

3V supply or ±1.25V inputs from ±3V

supplies.

A complete ADC system is provided

by the on-chip sample-and-holds, pre-

cision references, and internally

trimmed clocks. The high-impedance

analog inputs are easy to drive and

can be multiplexed without buffer am-

plifiers. A single 5V or 3V power supply

is all that is needed to digitize unipolar

inputs. (Bipolar inputs require ±5V or

±3V supplies but the negative supply

draws only microamperes of current).

But most significant are the speed/

power ratios, which are higher than

those of any other ADC in this speed

range.

(kHz)

ENOBS (EFFECTIVE NUMBER OF BITS)

10 100

1273_3.eps

LTC1273

= 5V

= 300kHz

LTC1282

= 3V

= 140kHz

S/(N + D) (dB)

Figure 2. The 300kHz LT1273 gives 70dB

S/(N + D) with 100kHz inputs, the

140kHz LTC1282 gives 68dB at Nyquist

Linear Technology Magazine • February 1993

DESIGN FEATURES

in mW. The 4.0kHz/mW of the

LTC1273, LTC1275, and LTC1276 is

better than the best competitive ADC.

Table 2 shows a competitive analy-

sis of currently available ADCs. The

LTC1273, LTC1275, and LTC1276 of-

fer advantages over the rest in every

area, including performance, function,

and power.

Even More Power Savings:

3V ADC Samples at 140kHz

on 12mW

The low-power, 3V LTC1282 pro-

vides even more impressive speed/

power performance. As fast and dy-

namically accurate as many power hun-

gry, dual- and triple-supply ADCs, this

complete 3V or ±3V sampling ADC pro-

vides extremely good performance on

only 12mW of power. DC specs include

±1/2LSB maximum linearity and the

internal reference provides 25ppm

maximum full-scale drift. Figure 2

shows 11.4 effective bits at a 140kHz

sample rate with 11.0 effective bits at

the Nyquist frequency of 70kHz. The

speed/power ratio, as shown in Figure

1, is an outstanding 11.7kHz/mW.

The LTC1282 is ideal for 3V systems

but will also find uses in 5V designs

where the lowest possible power con-

sumption is required. It interfaces eas-

ily to 3V logic but can also talk well to

5V systems. The LTC1282 can receive

5V CMOS levels directly and its 0-to-

3V outputs can meet 5V TTL levels and

connect directly to 5V systems.

Performance Comparison

Table 3 compares the performance

of the new ADCs to another recent,

low-cost product, the AD1674. The 5V

LTC1273 offers three times the speed

at one fifth the power, and the 3V

device goes even further. The table

shows that using the 3V LTC1282

gives even greater savings in power

than the LTC1273, with only modest

reductions in speed, accuracy, and

noise. The power dissipation has been

reduced six times with only a 50%

reduction in speed. Linearity and drift

don’t degrade at all in going to the 3V

device. The noise of the LTC1282 is

slightly higher, due to the reduced

input span and the lower operating

current, but the converter still gives

more than 70dB typical S/(N + D).

Compared to the AD1674, the LTC1282

offers 40% higher sampling rate and

30 times lower power.

Conclusion

These new 5V and 3V ADCs offer

the best speed/power performance

available today. They also provide pre-

cision references, internally trimmed

clocks, and fast sample-and-holds.

With additional features such as single-

supply operation and high-impedance

analog inputs, they reduce system com-

plexity and cost. For performance,

power, and cost, these new ADCs must

be considered for new designs.

LTC1273/5 AD1674 AD7870/5/6 AD7800 AD678 ADS7880 MAX163/4/7

Internal reference ✔✔✔ ✔✔✔

Internal clock (no crystal req’d) ✔✔✔✔✔✔

S/(N + D) at fIN=100kHz (typ) 70dB 68dB — — 64dB 69dB 69dB

High-impedance analog input ✔✔✔

Unipolar and bipolar inputs ✔✔ ✔

≥300kHz sample rate ✔✔

Single-supply operation LTC1273 ✔

Low power (<150mW) ✔

3V upgrade path ✔

Table 2. Competitive analysis of current

ADCs

LTC1273 LTC1282 AD1674

Parameter on +5V on +3V or ±3V on +5V or ±15V

Power dissipation (typ) 75mW 12mW 385mW

Sample rate 300kHz 140kHz 100kHz

Conversion time (max) 2.7µs6µs10µs

INL (max) ±1/2LSB ±1/2LSB ±1/2LSB

Typical ENOBs 11.7 11.4 11.5

Linear input bandwidth

(ENOBs >11 bits) 125kHz 70kHz 100kHz

Table 3. Performance comparison

Linear Technology Magazine • February 1993

DESIGN FEATURES

SHUTDOWN

N DRIVE

SENSE

–

SENSE

P DRIVE

LTC1148-3.3

S GND P GND

1µF

0V = NORMAL

>1.5V = SHUTDOWN

3300pF

1k 470pF

L1

50µHR



50mΩ

Si9430 100µF



IRLR024

1N5817 330µF

OUT



3.3V/2A

1000pF

1148_2.eps

=

L1 = SL-1R050J, KRL BANTRY (603) 668-3210

CTX50-2-MP, COILTRONICS (305) 781-8900

Figure 3. The LTC1148/LTC1149 architecture features constant-offtime current-mode operation, synchronous switch-

ing,

and automatic transition to Burst Mode

operation

Continued from page 1

1148_3.eps

LOW

DROPOUT

10V REG

LTC1149

ONLY

–

P DRIVE

N DRIVE

–

1-SHOT

SENSE

–

OUT

1.25V

REF

OFF

able, fixed 3.3V, and fixed 5V versions

and is available in both DIP and SOIC

(narrow) surface-mount packages.

High Performance

with High Efficiency

The LTC1148/LTC1149 synchro-

nous switching regulator controllers

use the constant off-time, current-

The LTC1148 5V to 3.3V converter

shown in Figure 2 has 94% efficiency

at 1A output. In other words, the

LTC1148 dissipates only 200mW while

delivering 3.3W to the load.

Table 1 gives an overview of the

different members of the LTC1148/

LTC1149 family and several of their

applications. Each device has adjust-

mode architecture shown in Figure 3.

Current-mode operation was judged

to be mandatory for its well known

advantages of clean start-up, accu-

rate current limit, and excellent line

and load regulation. The constant off-

time architecture adds to this list sim-

plicity (neither an oscillator nor ramp

compensation is required), inherent

100% duty cycle in dropout condi-

tions, and constant inductor-ripple

current. Because the off-time is con-

stant, the operating frequency changes

with input voltage. For example, in an

LTC1149-5 application, the frequency

will increase 50% when V

is doubled

from 10V to 20V.

To maximize the operating efficiency

over a wide current range, loss-reduc-

ing circuit techniques must be carefully

applied. For example, synchronous

switching (replacing the diode in a

stepdown regulator with a switched

transistor) may buy several percentage

points in efficiency at high output cur-

rents, but will cost many more if allowed

to continue at low output currents. This

is the principal reason why the

Figure 2. This LTC1148 5V-to-3V converter circuit achieves 94% efficiency at 1A output

current

Linear Technology Magazine • February 1993

DESIGN FEATURES



OUT

10mA

EFFICIENCY/LOSS (%)

100

30mA

1148_4.eps

100mA 300mA 3A1A

GATE CHARGE

LTC1148

Figure 4. Burst Mode

operation limits gate

charge and DC supply current at low output

currents; synchronous switching limits losses

at high output currents

LTC1148 LTC1148-3.3 LTC1148-5 LTC1149 LTC1149-3.3 LTC1149-5

Continuous input

voltage ≤ 48V ✔✔✔

Continuous input

voltage ≤ 12V ✔✔✔

Low dropout 5V ✔✔

Adjustable/

multiple output ✔✔

5V to 3.3V/

4-cell to 3.3V ✔

LTC1148 and LTC1149 change to

Burst Mode

operation as the output

current drops.

The continuous mode operation is as

follows: The external P-channel

MOSFET switch is turned on at the end

of the off-time and turned off when the

inductor current has ramped up to the

current-comparator threshold. During

the off-time, the N-channel MOSFET

synchronous switch is turned on to re-

duce the dissipation that would other-

wise occur in the diode. At the end of

the constant off-time, the P-channel

MOSFET is again turned on and the

cycle repeats. Adaptive anti-shoot-

through circuitry prevents simulta-

neous conduction of the two MOSFETs

regardless of their relative sizes, and

gate resistors are neither required nor

recommended.

Burst Mode

Operation—the

Low-Current Efficiency Saver

The LTC1148/LTC1149 burst oper-

ating mode is automatically invoked

when the current required by the load

is less than the minimum current sup-

plied by continuous operation. In Burst

Mode

operation the output voltage is

regulated via a hysteretic comparator

which, when tripped, shuts down both

MOSFET drivers and much of the con-

trol circuitry to conserve DC supply cur-

rent. From the time the comparator

trips, until the lower comparator

threshold is reached, the load current

is supplied entirely by the charge stored

Superb Start-up Manners

When starting a load from ground or

recovering from a short circuit, the

LTC1148 and LTC1149 offer superb

control of inductor current, with no

voltage overshoot when the regulated

voltage is reached. This is accomplished

by making the off-time proportional to

the output voltage as the output capaci-

tor charges. When V

OUT

= 0, the off-time

is lengthened to retain tight control of

the peak inductor current at the very

low duty cycles required. As the output

voltage increases, the off-time is pro-

gressively shortened until it reaches the

normal operating point. The result is

the typical start-up behavior shown in

Figure 5.

Line regulation can be just as vital

as load regulation in battery-operated

systems. This is because plugging in a

wall adapter often increases the regu-

lator input voltage to over double the

battery voltage. The LTC1148/

LTC1149 can handle these large steps

with no effect on the output voltage.

in the output capacitor. When the out-

put capacitor discharges to the lower

threshold, the main loop again turns on

briefly at a low current level to recharge

the capacitor. This cycle repeats at a

progressively slower rate as the output

current is reduced.

Table 1. LTC1148/LTC1149 features

Figure 5. When starting up, the LTC1149/

LTC1149 supply a constant current until the

regulated voltage is reached. Note the lack of

overshoot

The LTC1148/LTC1149

switching regulators not only

beat both linear regulators

and other switching

regulators in efficiency,

but also offer superior

dropout performance

Figure 4 shows how the efficiency

losses in a typical LTC1148 regulator

are apportioned as a result of the

action of Burst Mode

operation. The

gate-charge loss deserves special

attention, since it is responsible for

much of the efficiency lost in the mid-

current region. Each time a MOSFET

gate is switched from low to high to low

again, a packet of charge, dQ, moves

from V

to ground. The resulting

dQ/dt is a current out of V

that is typi-

cally much larger than the DC supply

current. If Burst Mode

operation was

not employed at low output currents,

the gate-charge loss alone would be

enough to push efficiency down to un-

acceptable levels. With Burst Mode

operation, the DC supply current

represents the lone (and unavoidable)

loss component, which continues to be-

come a higher percentage as output

current is reduced.

1148_5.eps

OUT



1V/DIV

IND



1A/DIV

Linear Technology Magazine • February 1993

DESIGN FEATURES

SHUTDOWN

N DRIVE

SENSE

–

SENSE

P DRIVE

LTC1148-5

S GND P GND

1µF

0V = NORMAL

>1.5V = SHUTDOWN

3300pF

1k 470pF

Si9430 100µF



5.2V-12V

Si9410

1N5817 330µF

OUT



5V/2A

1000pF

1148_6.eps

L1

62µHR



50mΩ

=

L1 = SL-1R050J, KRL BANTRY (603) 668-3210

CTX62-2-MP, COILTRONICS (305) 781-8900

Figure 6. The LTC1148/LTC1149 architecture provides 100% duty cycle, allowing very low

dropout operation. This LTC1148-5 circuit supports a 1A load with the input voltage only

200mV above the output

Very Low Dropout Operation

The LTC1148/LTC1149 switching

regulators not only beat both linear

regulators and other switching regula-

tors in efficiency, but also offer supe-

rior dropout performance. As the input

voltage on the LTC1148 or LTC1149

drops, the loop extends the on-time for

the switch (remember, the off-time is

constant), thereby keeping the induc-

tor ripple current constant. When V

–

OUT

drops below approximately 1.5V,

OFF

is reduced to compensate for the

decreasing frequency. Ultimately, the

LTC1148 and LTC1149 regulators drop

out smoothly, with the P-channel

MOSFET switch turning on at 100%

duty cycle or DC.

With the switch turned on at a 100%

duty cycle, the dropout voltage is lim-

ited only by the load current multiplied

by the total DC resistance of the

MOSFET, the inductor, and the cur-

rent-sense resistor. For example, the

low dropout 5V regulator shown in

Figure 6 has a total resistance in drop-

out of less than 200mΩ. This gives it a

dropout voltage of less than 200mV at

1A output current. Furthermore, when

the regulator is in a dropout condition,

the ground current is limited to the DC

supply current (1.5mA for the LTC1148

and 2.5mA for the LTC1149) indepen-

dent of load current.

High Voltage Capability

he LTC1149 versions offer an op-

erating-voltage capability far higher

than those found in other BiCMOS-

based technologies. This is because

the LTC1149 is actually a hybrid of a

BiCMOS controller chip and a 60V

bipolar companion chip that adds the

low dropout regulator and P-channel

drive level-shift circuits shown in Fig-

ure 3. The resulting combination,

available in a narrow 16-lead SOIC

package, extends LTC1148-like per-

formance to input voltages as high as

48V (60V absolute maximum). This

allows the LTC1149 to be used in such

applications as automotive and dis-

tributed power, as well as portable

equipment operating off high-voltage

AC adapters.

Figure 7. The LTC1149 extends high efficiency operation to high input voltages. This DC-to-DC

converter achieves 87% efficiency in dropping 48V to 5V

the dead-time between the conduction

of the two power MOSFETs and pro-

vides the highest possible operating

efficiency in continuous mode.

Conclusion

The LTC1148/LTC1149 family of

synchronous, stepdown switching

regulators offers breakthroughs in

the areas of low-current operating

efficiency, high-current operating ef-

ficiency, very low dropout, and wide

input voltage compliance. This per-

formance will be vital to extending

battery life in the next generation of

portable electronics.

SHUTDOWN2

N GATE

SENSE

–

SENSE

P GATE

LTC1149-5

S GND P,R GNDS

0V = NORMAL

>2V = SHUTDOWN

3300pF

1k C



680pF

IRFR9024

100µF

100V



48V

IRFR024

MBR380

150µF

OS-CON

OUT



5V/2.5A

1000pF

1148_7.eps

P DRIVE

0.047µF

1N4148

SHUTDOWN1

1µF

0.068µF

CAP

L1

75µHR



39mΩ

=

L1 = SL-1R039J, KRL BANTRY (603) 668-3210

CTX75-2-MP, COILTRONICS (305) 781-8900

A 48V to 5V DC-to-DC converter

that achieves 87% efficiency at 1A load

current is shown in Figure 7. In this

application the synchronous switch

plays a vital role, since the P-channel

switch duty cycle is only a little more

than 10%, meaning that the N-chan-

nel synchronous switch conducts

nearly 90% of the time in continuous

mode. Fortunately, low R

DS(ON)

N-chan-

nel MOSFETs that reduce losses to the

point that heat sinking is not required,

even during continuous short-circuit

operation, are readily available. The

small Schottky diode shown across the

N-channel MOSFET conducts only in

Linear Technology Magazine • February 1993

DESIGN IDEAS

Helium-neon lasers, used for a vari-

ety of tasks, are difficult loads for a

power supply. They typically need al-

most 10 kilovolts to start conduction,

although they require about 1500 volts

to maintain conduction at their speci-

fied operating currents. Powering a

laser usually involves some form of

start-up circuitry to generate the initial

breakdown voltage and a separate sup-

ply for sustaining conduction. Figure

1’s circuit considerably simplifies driv-

ing the laser. The start-up and sus-

taining functions have been combined

into a single, closed-loop current source

with over 10 kilovolts of compliance.

When power is applied, the laser does

not conduct and the voltage across the

190Ω resistor is zero. The LT1170

switching regulator FB pin sees no

feedback voltage, and its switch pin

A Simple, Efficient Laser Power Supply

by Jim Williams

loop.

The LT1170 adjusts its pulse-

width drive to L2 to maintain the FB pin

at 1.23 volts, regardless of changes in

operating conditions. Hence, the laser

sees constant current drive, in this

case 6.5mA. Other currents are obtain-

able by varying the 190Ω value. The

IN4002 diode string clamps excessive

voltages when laser conduction first

begins, protecting the LT1170. The 10µF

capacitor at the V

pin frequency com-

pensates the loop and the MUR405

maintains L1’s current flow when the

LT1170 V

pin is not conducting. The

circuit will start and run the laser over a

9–35 volt input range with an electrical

efficiency of about 75%.

References

1. Williams, Jim. “Illumination Circuitry for Liquid

Crystal Displays” Linear Technology Application

Note 49, August 1992.

) provides full duty cycle pulse-

width modulation to L2. Current flows

from L1’s center tap through Q1 and

Q2 into L2 and the LT1170. This cur-

rent flow causes Q1 and Q2 to switch,

alternately driving L1. The 0.47µF ca-

pacitor resonates with L1, providing a

boosted, sine-wave drive. L1 provides

substantial step-up, causing about

3500 volts to appear at its secondary.

The capacitors and diodes associated

with L1’s secondary form a voltage

tripler, producing over 10 kilovolts

across the laser. The laser breaks down

and current begins to flow through it.

The 47kΩ resistor limits current and

isolates the laser’s load characteristic.

The current flow causes a voltage to

appear across the 190Ω resistor. A

filtered version of this voltage appears

at the LT1170 FB pin, closing a control

LASER

190Ω

1N4002

(ALL)

0.1µF

10k

10µF

GND

2.2µF

IN

9V TO 35V

150Ω

MUR405 L2

150µH

LT1170

5413

11 HV DIODES

1800pF

10kV

0.01

5kV

1800pF

10kV

47k

2.2µF

0.47µF

HV DIODES =

0.47µF =

Q1, Q2 =

L1 =

L2 =

LASER =

SEMTECH-FM-50

WIMA 3X 0.15µF TYPE MKP-20

ZETEX ZTX-849

COILTRONICS CTX02-11128

COILTRONICS CTX150-3-52, COILTRONICS (305) 781-8900

HUGHES 3121H-P

10k

Q1 Q2

LASER_1.eps

Figure 1. Laser power supply

Linear Technology Magazine • February 1993

DESIGN IDEAS

Many systems require that ±15V

supplies for analog circuitry be derived

from an input voltage that may be

above or below the 15V output. The

split-supply requirement is usually ful-

filled by a switcher with a multiple-

secondary transformer, or by multiple

switchers. An alternative approach,

shown in Figure 1, uses an LT1074

switching regulator IC, two inductors,

and a “flying” capacitor to generate a

dual-output supply that accepts a wide

range of input voltages. This solution is

particularly noteworthy because it uses

only one switching regulator IC and

does not require a transformer. Induc-

tors are preferred over transformers

because they are readily available and

more economical.

The operating waveforms for the

circuit are shown in Figure 2. During

the switching cycle, the LT1074’s V

pin swings between the input voltage

) and the negative output voltage

(–V

OUT

). (The ability of the LT1074’s V

pin to swing below ground is unusual—

most other 5-pin-buck switching regu-

lator ICs cannot do this.) Trace A shows

the waveform of the V

pin voltage,

and Trace B is the current flowing

through the power switch.

While the LT1074 power switch is

on, current flows from the input volt-

age source through the switch, through

capacitor C2 and inductor L1 (Trace

C), and into the load. A portion of the

switch current also flows into inductor

L2 (Trace D). This current is used to

recharge C2 and C4 during the switch-

off time to a potential equal to the

positive output voltage (+V

OUT

). The

current waveforms for both inductors

occur on top of a DC level.

The waveforms are virtually identical

because the inductors have identical

values, and because the same voltage

potentials are applied across them dur-

ing the switching cycles.

Switching Regulator Provides ±15V

Output from an 8V – 40V Input,

without a Transformer by Brian Huffman

Figure 1. Schematic diagram for ±15V ver-

sion

1074_1.eps

L2

50µH

VR1

LT1074

GND

C2

470µF, 25V

D1

MUR410

C6

0.01µF

R4

20k

R5

20k

C7

0.01µF

C1

1000µF

50V

L1

50µH

R2

7.50k

R3

1.30k

C3

470µF

25V

C4

470µF

25V

–V

OUT



–15V, 0.5A

D2

MUR410

R1

3.3k

C5

0.01µF



8V-40V

OUT



+15V, 0.5A

2.21V* (1 + R2/R3) (EQ. 1)

–V

OUT



NICHICON UPL1H102MRH

NICHICON UPL1E471MPH

MOTOROLA MUR410

COILTRONICS CTX50-2-52 (305) 781-8900

OUT

= 

OUT

= 

C1 = 

C2, C3, C4 = 

D1, D2 = 

L1, L2 =

When the switch turns off, the cur-

rent in L1 and L2 begins to ramp

downward, causing the voltages across

them to reverse polarity and forcing the

voltage at the V

pin below ground.

The V

pin voltage falls until diodes

D1 (Trace E) and D2 (Trace F) are

forward biased. During this interval,

the voltage on the V

pin is equal to a

diode drop below the negative output

voltage (–V

OUT

). L2’s current then cir-

culates between both D1 and D2, charg-

ing C2 and C4. The energy stored in L1

is used to replace the energy lost by C2

and C4 during the switch-on time.

Trace G is capacitor C2’s current wave-

form. Capacitor C4’s current waveform

(Trace F) is the same as diode D2’s

current less the DC component. As-

suming that the forward voltage drops

of diodes D1 and D2 are equal, the

negative output voltage (–V

OUT

) will be

equal to the positive output voltage

(+V

OUT

). After the switch turns on again,

the cycle is repeated.

Figure 2. LT1074 switching waveforms

A = 20V/DIV

B = 2A/DIV

, I

C = 1A/DIV

, I

D = 1A/DIV

E = 1A/DIV

, I

F = 1A/DIV

, I

G = 1A/DIV

5µs/DIV

Linear Technology Magazine • February 1993

DESIGN IDEAS

Figure 3 shows the excellent regula-

tion of the negative output voltage for

various output currents. The negative

output voltage tracks the positive sup-

ply (+V

OUT

) within 200mV for load varia-

tions from 50mA to 500mA. Negative

output load current should not exceed

the positive output load by more than a

factor of 4; the imbalance causes loop

instabilities. For common load condi-

tions the two output voltages track

each other perfectly.

Another advantage of this circuit is

that inductor L1 acts as both an energy

storage element and as a smoothing

filter for the positive output (+V

OUT

The output ripple voltage has a trian-

gular waveshape whose amplitude is

determined by the inductor ripple cur-

rent (see Trace C of Figure 2) and the

ESR (Effective Series Resistance) of the

output capacitor (C3). This type of

ripple is usually small, so a post filter is

not necessary.

Figure 4 shows the efficiency for a

0.5A common load at various input

voltages. The two main loss elements

are the output diodes (D1 and D2) and

the LT1074 power switch. At low input

voltages, the efficiency drops because

the switch’s saturation voltage becomes

a higher percentage of the available

input supply.

The output voltage is controlled by

the LT1074 internal error amplifier.

This error amplifier compares a frac-

tion of the output voltage, via the R2–

R3 divider network shown in Figure 1,

with an internal 2.21V reference volt-

age and then varies the duty cycle until

the two values are equal. The RC net-

work (R1 and C5 in Figure 1) connected

to the V

pin along with the R4/R5 and

C6/C7 network provides sufficient com-

pensation to stabilize the control loop.

Equation 1 can be used to determine

the output voltage.

Figure 5 shows the circuit's –5V

load-regulation characteristics, and

Figure 6 shows its efficiency.

Refer to the schematic diagram in

Figure 7 for modified component val-

ues to provide ±5V at 1 Amp.

–VOUT (V)

15.3

1074_3.eps

0.5

–IOUT (A)

+IOUT = 0.5A

+IOUT = –IOUT

0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45

15.2

15.1

15.0

14.9

14.8

14.7

14.6

EFFICIENCY (%)

1074_4.eps

INPUT VOLTAGE (V)

5 253035

50 10 15 20

1074_1a.eps

L2

50µH

VR1

LT1074

GND

C2

680µF, 16V

D1

MBR360

C6

0.01µF

R4

20k

R5

20k

C7

0.01µF

C1

1000µF

50V

L1

50µH

R2

2.80k

R3

2.21k

C3

680µF,

16V

C4

680µF, 16V

–V

OUT

–5V, 1A

D2

MBR360

R1

C5

0.033µF



8V-40V

OUT



+5V, 1A

2.21V* (1 + R2/R3) (EQ. 1)

–V

OUT



NICHICON UPL1H102MRH

NICHICON UPL1C681MPH

MOTOROLA MBR360

COILTRONICS CTX50-2-52 (305) 781-8900

OUT

= 

OUT

= 

C1 = 

C2, C3, C4 = 

D1, D2 = 

L1, L2 =

Figure 7. Schematic diagram for ±5V version

Figure 5. –5V output dropout characteristics Figure 6. ±5V efficiency characteristics with 1A

common load

Figure 3. –15V output dropout characteristics Figure 4. ±15V efficiency characteristics with

0.5V common load

INPUT VOLTAGE (V)

EFFICIENCY (%)

15 40

1074_6.eps

510 20 3525 30

–I

OUT

(A)

4.8

–V

OUT

(V)

5.0

5.1

5.3

5.4

5.6

5.7

0.2 0.5 0.7 1.0

1074_5.eps

0.1 0.3 0.4 0.6 0.8 0.9

4.9

5.2

5.5

OUT

= –I

OUT

= 1A

Linear Technology Magazine • February 1993

DESIGN IDEAS

A Twelve-Bit, Micropower

Battery-Current Monitor

Introduction

The LTC1297 forms the core of the

micropower battery-current monitor

shown in Figure 1. This 12-bit data-

acquisition system features an auto-

matic power shutdown that is activated

after each conversion. In shutdown

the supply current is reduced to 6µA,

typically. As shown in Figure 2, the

average power-supply current of the

LTC1297 varies from milliamperes to

a few microamperes as the sampling

frequency is reduced. This circuit

draws only 190µA from a 6V to 12V

battery when the sampling frequency

is less than 10 samples per second.

Wake-up time is limited by that re-

quired by the LTC1297 (5.5µs). For

long periods of inactivity, the circuit’s

supply current can be further reduced

to 20µA by using the shutdown feature

on the LT1121. More wake-up time is

required when using this mode of shut-

down. It is usually determined by the

amount of capacitance in the circuit

by Sammy Lum

SAMPLE

(kHz)

AVERAGE I

(µA)

100

1000

10000

0.001 1 10 100

1297_2.eps

10.01 0.1

and the available charging current

from the regulator.

The Battery-Current Monitor

The battery voltage of 6V to 12V is

regulated down to 5V by the LT1121

micropower regulator. A sense resistor

of 0.05Ω is placed in series with the

battery to convert the battery current

to a voltage. Full scale is designed for

2A, giving a resolution of 0.5mA with

the 12-bit ADC. The LTC1047 ampli-

fies the voltage across the sense resis-

tor by 25 V/V. This goes through an RC

lowpass filter before being fed into the

input of the LTC1297. The RC filter

serves two functions. First, it helps

band-limit the input noise to the ADC.

Second, the capacitor helps the

LTC1047 recover from transients due

to the switching input capacitor of the

LTC1297. The LT1004 provides the

full-scale reference for the ADC. A low

battery detection circuit has been cre-

ated by using the other half of the

LTC1047 as a comparator. Its trip point

has been set to 5V plus the dropout

voltage of the LT1121. Because data is

transmitted serially to and from the

microprocessor or microcontroller, this

current-monitor circuit can be located

close to the battery.

Figure 1. A micropower battery-current monitor using the LTC1297 12-bit data acquisition system

–

+IN

–IN

GND

VCC

CLK

DOUT

VREF

LOAD

510kΩ6V-12V

0.05Ω

2A FULL

SCALE

30.1kΩ

750kΩ

0.1µF

10Ω

1/2

LTC1047

1µF

0.33µF

1N4148

+5V

GND

OUT

SHDN

LT1121

LTC1297

22µF

TANT

0.1µF

LT1004-2.5

20MΩ

LO BATTERY

TO AND

FROM µP

–

1/2

LTC1047

1297_1.eps

470kΩ

100kΩ

Figure 2. Power-supply current versus

sampling frequency for the LTC1297

Linear Technology Magazine • February 1993

DESIGN IDEAS

Some 1.5V powered systems, such

as two-way survival radios, remote,

transducer-fed data-acquisition sys-

tems, and the like, require much more

power than stand-alone IC regulators

can provide. Figure 1’s design supplies

a 5V output with 200mA capacity.

The circuit is essentially a flyback

regulator. The LT1170 switching

regulator’s low saturation losses and

ease of use permit high power opera-

tion and design simplicity. Unfortu-

nately this device has a 3V minimum

supply requirement. Bootstrapping its

supply pin from the 5V output is pos-

sible, but requires some form of start-

up mechanism. The 1.5V powered

LT1073 switching regulator forms a

start-up loop. When power is applied,

the LT1073 starts, causing its V

to periodically pull current through L1.

L1 responds with high-voltage flyback

pulses. These pulses are rectified and

200mA Output, 1.5V-to-5V Converter

by Jim Williams

stored in the 500µF capacitor, produc-

ing the circuit’s DC output. The out-

put-divider string is set up so the

LT1073 turns off when the circuit’s

output crosses about 4.5V. Under these

conditions the LT1073 can no longer

drive L1, but the LT1170 can. When

the start-up circuit turns off, the

LT1170 V

pin has adequate supply

voltage and it can operate. There is

some overlap between start-up loop

turn-off and LT1170 turn-on, but this

has no detrimental effect. The start-up

loop must function over a wide range of

loads and battery voltages. Start-up

currents are about 1 ampere, necessi-

tating attention to the LT1073’s satu-

ration and drive characteristics. The

worst case is a nearly depleted battery

and heavy output loading. Figure 2

plots input/output characteristics for

the circuit. Note that the circuit will

start into all loads with V

BATT

= 1.2V.

Start-up is possible down to 1.0V at

reduced loads. Once the circuit has

started, the plot shows it will drive full

200mA loads down to V

BATT

= 0.6V (a

very dead battery). Figure 3 graphs

efficiency at two supply voltages over a

range of output currents. Performance

is attractive, although at lower cur-

rents circuit quiescent power degrades

efficiency. Fixed junction saturation

losses are responsible for lower overall

efficiency at the lower supply voltage.

References

1. Williams, Jim and Brian Huffman. “Some Thoughts

on DC–DC converters”, pages 13–17, “1.5V to 5V

Converters.” Linear Technology Corporation Appli-

cation Note 29, October 1988.

OUTPUT CURRENT (mA)

EFFICIENCY (%)

100

40 100 180

CELL_3.eps

20 60 80 120 140 200160

= 1.2V

= 1.5V

OUT

= 5.0V

OUTPUT CURRENT (mA)

MINIMUM INPUT VOLTAGE TO MAINTAIN VOUT = 5.0V

0.2

1.0

1.4

40 100 180 220

CELL_2.eps

20 60 80 120 140 200160

1.2

0.8

0.6

0.4

START

RUN

SW1

SW2

500µF

1N5823

LT1073

L1 =

* = COILTRONICS CTX20-5-52, COILTRONICS (305) 781-8900

1% METAL FILM RESISTOR

CELL_1.eps

22µF

6.8µF

GND

LT1170

GND

220µF

L1

20µH

1.5V

OUT

3.74k*

1k*

240Ω*

Figure 1. 200mA output, 1.5V-to-5V converter Figure 3. Efficiency versus operating point

for Figure 1

Figure 2. Input/output data for Figure 1

Linear Technology Magazine • February 1993

DESIGN IDEAS

The LT1158 half-bridge motor driver

incorporates a number of powerful

protection features. Some of these,

such as its adaptive gate drive, are

dedicated in function. Others are open

to a variety of uses, depending upon

application requirements. The circuit

shown in Figure 1 takes advantage of

the wide common-mode input range of

the LT1158’s FAULT comparator to

perform ground-referenced current

sensing in an H-bridge motor driver.

By using ground-referenced sensing,

protection can easily be provided

against overloaded, stalled, or shorted

motors. For overloads and stalls, the

circuit becomes a constant-current

chopper, regulating the motor’s arma-

ture current to a preset maximum

value. For shorted loads, the circuit

protects itself by operating at a very

low duty cycle until the short is cleared.

Setting Up For Ground-

Referenced Sensing

The circuit of Figure 1 is essentially

a straightforward LT1158 H-bridge, of

the “sign/magnitude” type. (See the

LT1158 data sheet for a description of

component functions.) In many

LT1158 applications, a current-sense

resistor is placed in each upper

MOSFET source lead. This circuit, how-

ever, senses the IR drop across one

resistor (R1) common to the sources of

both lower MOSFETs. In Figure 1,

U1’s FAULT output activates the con-

stant-current protection mode (for

overloads and stalls), and U2’s FAULT

output indicates a shorted load. Hence,

given a maximum continuous motor

current of 15A, R1’s value is easily

determined: V

SENSE(+)

of U1 must ex-

ceed V

SENSE(–)

by the LT1158’s internal

threshold of 110mV in order for FAULT

to go low. 15A x R1 = 0.110V, so R1 =

(0.110V/15A) ≅7.5mΩ. The FAULT pin

of U2 should go low when I

is 24A, so

a 1.6:1 voltage divider is added at U2’s

SENSE(+) input. R2, R3, C1, and C2

filter any switching spikes which ap-

pear across R1.

Closing the Loop on Overloads

If the motor is overloaded or stalled,

its back EMF will drop, causing the

armature current to increase at a rate

determined primarily by the motor’s

inductance. Without protection, this

current could rise to a value limited

only by supply voltage and circuit re-

sistance. The necessary protection is

provided via the feedback loop formed

by U1’s FAULT output, U3A, U4B, and

U4D. When I

exceeds 15A, the FAULT

pin of U1 conducts, triggering the 40µs

monostable U3A. The “Q” output of

U3A in turn forces the outputs of U4B

and U4D to a logic low state, turning off

Q1 or Q3, and turning on both Q2 and

Q4. For the time during which U3A’s

“Q” output is high, the motor current

decays through the path formed by the

motor’s resistance, plus the “on” resis-

tance of Q2 and Q4 in series. In this

application, turning both lower MOS-

FETs on is preferable to forcing all four

MOSFETs off, as it provides a low-

resistance recirculation path for the

motor current. This reduces motor and

supply ripple currents, as well as MOS-

FET dissipation. At the end of U3A’s

40µs timeout, the H-bridge turns on

again. If the overload still exists, the

current quickly builds up to the U1

“FAULT” trip point again, and the 40ms

timeout repeats. This feedback loop

holds the motor current approximately

constant at 15A for any combination of

supply voltage and duty cycle that

would otherwise cause an excess cur-

rent condition. When the motors cur-

rent draw falls below 15A, the circuit

resumes normal operation.

LT1158 H-Bridge uses Ground-

Referenced Current Sensing

for System Protection by Peter Schwartz

Opening the Loop on Shorts

In the event of a short across the

motor terminals, the current through

the H-bridge rises faster than the U1/

U3A loop can regulate it. This could

easily exceed the safe operating area

limits of the MOSFETs. The solution is

simple: when the FAULT comparator

of U2 detects that I

≥ 24A, monostable

U3B is triggered. The “Q’” output of

U3B will then hold the ENABLE line of

the two LT1158s “low” for 10ms, re-

sulting in a rapid shutdown and a very

low duty cycle. After the 10ms shut-

down interval, U3B’s “Q’” output will

return high, and the bridge will be re-

enabled. If the motor remains shorted,

U3B is triggered again, causing an-

other 10ms shutdown. When the short

is cleared, circuit operation returns to

that described above.

A Final Note

As a class, sign/magnitude H-bridge

systems are susceptible to MOSFET

and/or motor damage if the motor

velocity is accelerated rapidly, or the

state of the DIRECTION line is

switched while the motor is rotating.

This is especially true if the motor/

load system has high inertia. The cir-

cuit of Figure 1 is designed to provide

protection under these conditions: the

motor may be commanded to acceler-

ate and to change direction with no

precautions. For the case of decelera-

tion, however, it’s generally best to use

a controlled velocity profile. If a spe-

cific application requires the ability to

operate with no restrictions upon the

rate of change of duty cycle, there are

straightforward modifications to Fig-

ure 1 which allow this. Please contact

the factory for more information.

Linear Technology Magazine • February 1993

DESIGN IDEAS

Figure 1. H-bridge motor driver with ground referenced current sensing



FAULT

ENABLE

INPUT

BIAS

GND

SENSE

–

SENSE

B GATE FB

B GATE DR

T SOURCE

T GATE FB

BOOST

BOOST DR

T GATE DR

FAULT

ENABLE

INPUT

BIAS

GND 7

SENSE

–

SENSE

B GATE FB

B GATE DR

T SOURCE

T GATE FB

BOOST

BOOST DR

T GATE DR

270

100

C2

0.01µF

160

R3

100

R2

100

R1

0.0075

39k +5V

REXT/CEXT

CEXT

CLR

+5V

74HC221

4.7k

U4A

74HC02

1N4148

4.7k

PWM (“MAGNITUDE”)

DIRECTION (“SIGN”)

U4B

74HC02 4

U4C

74HC02

910

12 13

U4D

74HC02

47k

+5V

10µF

0.01µF

LT1158

220k +5V

REXT/CEXT

CEXT

CLR

+5V

74HC221

Q2

IRFZ44**

Q1

IRFZ44**

Q4

IRFZ44**

Q3

IRFZ44**

+24V

470µF* 470µF*

DC MOTOR

(15A CONT)

MBR170

1N4148

MBR170

0.1µF

1N4148

0.1µF10µF

0.047µF

10k***

47k

+5V

0.01µF

LT1158

Q1-Q4 MOUNTED ON HEAT SINK

LOW ESR CAPACITORS (SPRAGUE 673D, ETC.)

DIODE SHOWN IS THE MOSFET’S INTEGRAL 

DRAIN-BODY DIODE.

PULLDOWN FOR “ENABLE” LINE IN CASE

+5V IS NOT PRESENT.



*

**

***

1000pF

U3A

TWISTED-PAIR TWISTED-PAIR

U3B

C1

0.01µF

Linear Technology Magazine • February 1993

DESIGN IDEAS

Multi-Output Three-Watt Power Supply

Operates from Two AA Cells Steve Pietkiewicz

Portable, battery-operated micropro-

cessor systems often have power sup-

ply requirements beyond what existing

low-voltage IC switching regulators can

deliver. Also, a multiplicity of voltages

are usually required to supply sub-

systems such as main logic, flash

memory VPP supply, LCD contrast,

and modem. Previous approaches to

this problem used a separate DC–DC

converter circuit for each output, in-

creasing system cost and complexity.

The approach described in this article

combines the classic multi-output fly-

back topology with an LT1110 mi-

cropower, low-voltage DC–DC

converter. The negative-to-positive to-

pology provides for operation from a

1.8V to 7.5V input, a key provision for

systems that must operate with either

an AC adaptor or two AA cells. Mi-

cropower circuitry reduces quiescent

current to 400µA no-load. The circuit

can provide 5V at up to 400mA, 12V at

60mA, +28V at 2mA and –5V at 50mA

from an input voltage as low as 2.0V.

The LT1110 micropower, Burst

Mode

DC–DC converter IC functions

as the controller in the circuit (Figure

1). The LT1110 toggles its SW1 pin

when the voltage at its FB pin drops

below 220mV. The power device in the

circuit is Q2, a Zetex ZTX-849. This

remarkable device, which comes in a

small TO-92-type package, can handle

collector currents exceeding 6A with

guaranteed beta of 100. Saturation

voltage of the device with a forced beta

of 50 is 250mV at 5A I

. Base drive for

Q2 is provided by Q1, whose drive is

supplied by the SW1 pin of the LT1110.

Q2 is turned off by Q3, whose base is

AC coupled to the SW1 pin by the

2200pF capacitor. Q3, a 2N2369, is a

very fast device; it pulls the base charge

out of Q2 in 50ns. Q2 is kept off by the

2200Ω base-emitter resistor R4. The

primary winding of the trans–former,

L1, functions as the regulated (main)

secondary winding during the flyback

phase. The voltage across L1 is forced

to 5V during this phase. Hence, other

indirectly regulated voltages can be

achieved by the use of secondary wind-

Figure 1. Multi-output power supply delivers +5V at 400mA, +12V at 60mA, –5V at 50mA and +28V at 2mA from 2AA cells

2 AA

CELLS

C1

150µF

OS-CON

R6

432k

R7

21.5k

Q6

2N3906

GND SW2

SW1

IC1

LT1110

R9

4700

R10

150

Q5

2N3904

Q4

2N3904

R8

100

R1

220

Q1

2N4403

R2

220

Q2

ZETEX

ZTX-849

R3

C2

2200pF

R11

2200

D1

1N4148

R4

2200

C6

4.7µF

D5

1N5818

D4

1N4148

D3

1N5818

D2

1N5821

C3

47µF

C4

47µF

C5

33µF

+5V/

400mA

+12V/

60mA

+28V/

2mA

–5V/

50mA

T1*

R5**

20mΩ

TDK EPC-15 CORE, PC40 MATERIAL, 6 MIL GAP OR

FERROXCUBE EFD 15 CORE, 3C85 MATERIAL, 6 MIL GAP

L1 = 6T #20

L2 = 7T #32

L3 = 16T #38

L4 = 6T #34

DALE LVR-3

*T1 =

**R5 =

C1, C3, C4, C5 = SANYO OS-CON

ZETEX: (516) 543-7100

 = HEAVY, HIGH CURRENT TRACES

1110_1.eps

Q3

2N2369

L1 L2

Linear Technology Magazine • February 1993

DESIGN IDEAS

ngs with appropriate turns ratios.

The 12V output does not posses 5%

regulation from zero to full load, but a

micropower, linear, low-dropout regu-

lator such as the LT1121 can be used

to achieve the desired voltage regula-

tion. Negative outputs can be gener-

ated merely by reversing the phasing

of additional secondary windings, as

is done with L4 to obtain –5V output.

Feedback is accomplished by the level-

shift network comprising Q6 and R6.

Q6’s collector is fed into R7, closing

the loop.

Switch-current sensing and con-

trol is essential when throwing lots of

amperes around. Variations in V

and t

due to manufacturing spread

can result in large peak current

changes if sensing and control are not

implemented. Many Burst Mode

regulators contain no provision for

current sensing, but the LT1110 is an

exception. The LT1110 switch will

turn off when the voltage at the I

reaches 600mV less than the voltage

at the V

pin. A 600mV shunt would

reduce system efficiency severely in a

2.0V input converter, so a pre-bias

voltage drop is developed by current

source Q4–Q5 flowing through resis-

tor R8. Approximately 480mV is de-

veloped across R8, reducing the drop

across sense resistor R5 to 120mV.

This voltage drop represents 6% of

the 2V input, causing some loss of

efficiency, but the current sense func-

tion allows operation with inputs as

high as 7.5V.

Bypass capacitor C1 should be

placed close to the DC–DC converter

circuitry. The low-ESR OS-CON type

should be used. An inexpensive, high-

ESR unit can result in poor efficiency.

The main 5V output capacitors should

also be OS-CON types. The peak cur-

rent into these units is over 4A.

Skimping on output capacitors can

result in costly field failures. High

peak currents also necessitate care-

ful printed-circuit layout. The high-

current paths (highlighted in Figure

1) should be made extra-wide and as

short as possible.

Efficiency for the circuit is approxi-

mately 70% over an input range of 2

to 3.2V and .5W to 2W total output

power. The circuit will supply a 5V,

100µA “sleep” mode load for over 3

months from a pair of alkaline AA

cells (Figure 2). A 5V, 1mA load lasts

28 days with alkalines. However,

NiCad cells are recommended for the

power source, as the relatively high

internal impedance of the alkaline

cells deliver only 8% more operating

time than a pair of 600mAHr NiCad

cells when delivering 5V at 200mA

load, as detailed in Figure 3.

TIME (MINUTES)

1.0

BATTERY/OUTPUT VOLTAGE (V)

2.0

2.5

3.5

4.0

5.0

5.5

15 30 45 60

1110_.3eps

5 10 2025 3540 5055

1.5

3.0

4.5

NiCAD

ALKALINE

BATTERY

OUTPUT

DAYS

BATTERY/OUTPUT VOLTAGE (V)

60 120

1110_2.eps

20 40 80 100

OUTPUT

LOAD

= 100µA

BATTERY

Figure 2. Sleep mode lifetime, 5V output, 100µA load current Figure 3. Battery lifetime for I

LOAD

= 200mA

Linear Technology Magazine • February 1993

DESIGN IDEAS

Computers designed to accept PC

cards—plug-in modules specified by

the Personal Computer Memory Card

International Association (PCMCIA)—

have special hardware features to ac-

commodate these pocket-sized cards.

PCMCIA-compliant cards require power

management electronics which con-

form to the height restrictions of the

three standard configurations: 3.3mm,

5mm and 10.5mm. These height limi-

tations dramatically reduce the avail-

able options for power management on

the card itself. For example, high-effi-

ciency switching regulators to convert

the incoming 5V down to 3.3V for the

on-card 3.3V logic require relatively

large magnetics and filter capacitors,

which are not always available in pack-

aging thin enough to meet the tight

height requirements.

One possible approach to the prob-

lem of supplying power to a 3.3V PC

LTC1157 Switch

for 3.3V PC Card Power by Tim Skovmand

card is to switch the input supply

voltage from 5V to 3.3V after the card

has been inserted and the attribute

ROM has informed the computer of

the card’s voltage and current

requirements. The switching regula-

tor, housed in the computer, switches

the power supplied to the connector

from 5V to 3.3V.

A window comparator and ultra-low

drop switch on the PC card, Q1 in

Figure 1, closes after the supply volt-

age drops from 5V to 3.3V, ensuring

that the sensitive 3.3V logic on the card

is never powered by more than 3.6V or

less than 2.4V. A second switch, Q2, is

provided on the card to interrupt power

to 3.3V loads that can be idled when

not in use.

The built-in charge pumps in the

LTC1157 drive the gates of the low

DS(ON)

N-channel MOSFETs to 8.7V

when powered from a 3.3V supply.

(P-channels cannot be used at 3.3V

because they do not have guaranteed

DS(ON)

with V

< 3.3V.) The LT1017

and the LTC1157 are both micropower

and are supplied by a filter, R5 and C2,

which holds the supply pins high long

enough to ensure that the MOSFET

gates are fully discharged immediately

after the card is disconnected from the

power supply. A large bleed resistor,

R6, further ensures that the high-im-

pedance gate of Q1 is not inadvertently

charged-up when the card is removed

or when it is stored.

All of the components shown in

Figure 1 are available in thin, surface-

mount packaging and occupy a very

small amount of surface area. Further,

the power dissipation is extremely low

because the LTC1157 and LT1017 are

micropower and the MOSFET switches

are very low R

DS(ON)

SENSITIVE

3.3V

LOGIC

SENSITIVE

3.3V

LOGIC

Q1

MTD3055EL

R6

5.1M

Q2

MTD3055EL

3.3V

R1

150k

R4

100k C2

10µF

6.3V

R2

49.9k

R3

100k

1% LT1004-1.2

C1

0.1µF

SW ON/OFF

FROM µP

–

ATTRIBUTE

ROM

R5

510

LTC1157

GND

IN2

IN1

–

1/2

LT1017

1/2

LT1017

1157_1.eps

Figure 1. 3.3V PCMCIA card power switching

Linear Technology Magazine • February 1993

DESIGN IDEAS

For further information on the

above or any other devices men-

tioned in this issue of Linear Tech-

nology, use the reader service card

or call the LTC literature-service

number: (800) 637-5545. Ask for

the pertinent data sheets and ap-

plication notes.

Information furnished by Linear Technology Cor-

poration is believed to be accurate and reliable.

However, no responsibility is assumed for its use.

Linear Technology makes no representation that

the circuits described herein will not infringe on

existing patent rights.

New Device Cameos

LT1116 12ns, Single-Supply,

Ground-Sensing Comparator

The LT1116 is a high-speed (12ns)

comparator capable of sensing signals

down to ground while operating from a

single +5V supply rail. The comparator

can also operate from split ±5V supply

rails, where the input common-mode

range extends from 2 volts below the

positive rail to the negative supply rail.

The LT1116 is pin-compatible with

the industry standard LT1016. Like

the LT1016, the LT1116 is stable

through its output-transition region,

which makes it easy to use over a wide

range of operating conditions. The

device’s complementary output stages

provide active drive in both directions

for increased speed when driving TTL

logic or passive loads. The LT1116 has

a latch pin for synchronizing or retain-

ing data. Latch setup and hold times

are typically 2ns—commensurate with

the device’s propagation delay.

The LT1116’s tight offset (1mV typi-

cal) and high gain specifications

(3000V/V typical) make it an ideal

choice for high-speed applications such

as zero-crossing detectors, triggers,

sampling circuits, A/D converters, cur-

rent sensing for switching regulators,

and line receivers for data communica-

tion. Linear Technology’s Application

Note 13 describes practical design tech-

niques for high-speed comparators.

The LT1116 is available in 8-lead

SOIC and 8-pin mini-DIP packages.

The LTC1255 Dual, 24V

High-Side MOSFET Driver

The LTC1255 dual, 24V high-side

gate driver is designed to drive two

standard N-channel power MOSFETs

in a high-side switch configuration.

The LTC1255 contains two indepen-

dent, on-chip charge pumps so that

less expensive, lower R

DS(ON)

N-chan-

nel MOSFETs can be used in place of P-

channel switches. The charge pumps

require no external components and

have been designed to be very efficient.

All of the circuitry to drive, control,

and protect the power MOSFET and

load is provided by the LTC1255. The

input is compatible with both TTL and

CMOS logic families and the standby

current with the input switched off is

only 12 microamps from a 12V supply.

The quiescent current rises to 240

microamps with the switch turned on

and the charge pump producing 24V

from a 12V supply.

The MOSFET and load are protected

by a sense circuit that trips when an

over-current condition is detected at

the drain end of the power MOSFET. A

built-in 10-microsecond delay ensures

that the LTC1255 protection circuitry

is not false-triggered by transient load

or power supply conditions. A longer

RC delay can be added externally to

accommodate loads with large tran-

sient start-up current requirements,

such as lamps or DC motors.

The 9–24V operating range of the

LTC1255 makes it the ideal choice for

many automotive and industrial

applications, as well as 8–12 cell note-

book-computer battery switching ap-

plications. The LTC1255 is available in

both 8-lead SO and 8-lead DIP packag-

ing and is rated over both the indus-

trial and commercial operating

temperature ranges.

LT1331, LT1341, and

LT1342 RS232 Transceivers

Three new RS232 transceivers ex-

pand LTC’s line of interface circuits for

PC-compatible applications. Each of

these transceivers contains three driv-

ers and five receivers to support the

serial interface requirements of per-

sonal computers. All three feature

±10kV ESD protection on the RS232

line pins, operate to 120kbaud, and

contain on-chip charge-pump circuitry

to allow operation from standard logic

power supplies. Pin-outs are compat-

ible with the LT1137, and each trans-

ceiver has low-power SHUTDOWN and

DRIVER DISABLE operating modes

to allow optimization of system

power consumption based upon signal

requirements.

The LT1341 features one low-power

receiver that remains active while the

circuit is in SHUTDOWN mode. Draw-

ing only 60µA of power, the keep-alive

receiver may be used to monitor a data

line to control system wake-up. This is

especially useful in battery-operated

systems.

The LT1342 matches LT1137A per-

formance with the addition of 3V logic-

interface capability. The circuit is ideal

for systems with both 3V and 5V power

supplies. Power consumption is 12mA

from the 5V supply and 0.1mA from

the 3V power supply. In SHUTDOWN,

power consumption drops to near zero.

The LT1331 provides more flexible

power management features for mixed

5V/3V systems, and delivers RS562

level outputs when used in 3V-only

systems. Two power supply pins, V

and V

, are used to power the circuit.

Either may be used at 3V or 5V. The V

supply powers the charge pump and

driver circuits and may be turned

off in SHUTDOWN. With V

= 5V, full

RS232 output levels are supported.

With V

= 3.3V, outputs are at RS562

levels. The receivers are powered from

supply V

. V

current drain is 3mA with

all receivers active or 60µA in SHUT-

DOWN with one receiver active.

All three circuits are available in 28-

pin DIP, SOIC, and SSOP packages.

NEW DEVICE CAMEOS

Linear Technology Magazine • February 1993

DESIGN IDEAS

DESIGN TOOLS

LINEAR TECHNOLOGY CORPORATION

1630 McCarthy Boulevard

Milpitas, CA 95035-7487

(408) 432-1900

Literature Department (800) 637-5545

Applications on Disk

NOISE DISK

This IBM-PC (or compatible) progam allows the user to

calculate circuit noise using LTC op amps, determine the

best LTC op amp for a low noise application, display the

noise data for LTC op amps, calculate resistor noise, and

calculate noise using specs for any op amp.

Available at no charge.

SPICE MACROMODEL DISK

This IBM-PC (or compatible) high density diskette contains

the library of LTC op amp SPICE macromodels. The

models can be used with any version of SPICE for general

analog circuit simulations. The diskette also contains work-

ing circuit examples using the models, and a demonstration

copy of PSPICE

by MicroSim.

Available at no charge.

Technical Books

1990 Linear Databook — This 1,440 page collection

of data sheets covers op amps, voltage regulators,

references, comparators, filters, PWMs, data conversion

and interface products (bipolar and CMOS), in both com-

mercial and military grades. The catalog features well over

300 devices.

$10.00

1992 Linear Databook Supplement — This 1248 page

supplement to the

1990 Linear Databook

is a collection of

all products introduced since then. The catalog contains full

data sheets for over 140 devices. The

1992 Linear Databook

Supplement

is a companion to the

1990 Linear Databook

which should not be discarded.

$10.00

Linear Applications Handbook — 928 pages full of

application ideas covered in depth by 40 Application Notes

and 33 Design Notes. This catalog covers a broad range of

“real world” linear circuitry. In addition to detailed, systems-

oriented circuits, this handbook contains broad tutorial

content together with liberal use of schematics and scope

photography. A special feature in this edition includes a 22-

page section on SPICE macromodels.

$20.00

Monolithic Filter Handbook — This 232 page book comes

with a disk which runs on PCs. Together, the book and disk

assist in the selection, design and implementation of the

right switched capacitor filter circuit. The disk contains

standard filter responses as well as a custom mode. The

handbook contains over 20 data sheets, Design Notes and

Application Notes.

$40.00

SwitcherCAD Handbook — This 144 page manual, in-

cluding disk, guides the user through SwitcherCAD – a

powerful PC software tool which aids in the design and

optimization of switching regulators. The program can cut

days off the design cycle by selecting topologies, calculat-

ing operating points and specifying component values and

manufacturer's part numbers.

$20.00

FRANCE

Linear Technology S.A.R.L.

Immeuble “Le Quartz”

58 Chemin de la Justice

92290 Chatenay Mallabry

France

Phone: 33-1-46316161

FAX: 33-1-46314613

GERMANY

Linear Technology GMBH

Untere Hauptstr. 9

D-8057 Eching

Germany

Phone: 49-89-3194710

FAX: 49-89-3194821

JAPAN

Linear Technology KK

5F YZ Building

4-4-12 Iidabashi Chiyoda-Ku

Tokyo, 102 Japan

Phone: 81-3-3237-7891

FAX: 81-3-3237-8010

KOREA

Linear Technology Korea Branch

Namsong Building, #505

Itaewon-Dong 260-199

Yongsan-Ku, Seoul

Korea

Phone: 82-2-792-1617

FAX: 82-2-792-1619

SINGAPORE

Linear Technology Pte. Ltd.

101 Boon Keng Road

#02-15 Kallang Ind. Estates

Singapore 1233

Phone: 65-293-5322

FAX: 65-292-0398

TAIWAN

Linear Technology Corporation

Rm. 801, No. 46, Sec. 2

Chung Shan N. Rd.

Taipei, Taiwan, R.O.C.

Phone: 886-2-521-7575

FAX: 886-2-562-2285

UNITED KINGDOM

Linear Technology (UK) Ltd.

The Coliseum, Riverside Way

Camberley, Surrey GU15 3YL

United Kingdom

Phone: 011-44-276-677676

FAX: 011-44-276-64851

International

Sales Offices

CENTRAL REGION

Linear Technology Corporation

Chesapeake Square

229 Mitchell Court, Suite A-25

Addison, IL 60101

Phone: (708) 620-6910

FAX: (708) 620-6977

NORTHEAST REGION

Linear Technology Corporation

One Oxford Valley

2300 E. Lincoln Hwy., Suite 306

Langhorne, PA 19047

Phone: (215) 757-8578

FAX: (215) 757-5631

NORTHWEST REGION

Linear Technology Corporation

782 Sycamore Dr.

Milpitas, CA 95035

Phone: (408) 428-2050

FAX: (408) 432-6331

SOUTHEAST REGION

Linear Technology Corporation

17060 Dallas Parkway

Suite 208

Dallas, TX 75248

Phone: (214) 733-3071

FAX: (214) 380-5138

SOUTHWEST REGION

Linear Technology Corporation

22141 Ventura Boulevard

Suite 206

Woodland Hills, CA 91364

Phone: (818) 703-0835

FAX: (818) 703-0517

U.S. Area

Sales Offices

World Headquarters

Linear Technology Corporation

1630 McCarthy Boulevard

Milpitas, CA 95035-7487

Phone: (408) 432-1900

FAX: (408) 434-0507