Device Operating
Temperature Range Package
EASY SWITCHER
1.0 A STEP–DOWN
VOLTAGE REGULATOR
ORDERING INFORMATION
LM2575T–**
LM2575TV–** TJ = –40° to +125°C
Straight Lead
Vertical Mount
DEVICE TYPE/NOMINAL OUTPUT VOLTAGE
LM2575–3.3
LM2575–5
LM2575–12
LM2575–15
LM2575–Adj
3.3 V
5.0 V
12 V
15 V
1.23 V to 37 V
D2T SUFFIX
PLASTIC PACKAGE
CASE 936A
(D2PAK)
1
5
Order this document by LM2575/D
T SUFFIX
PLASTIC PACKAGE
CASE 314D
TV SUFFIX
PLASTIC PACKAGE
CASE 314B
Pin 1. Vin
2. Output
3. Ground
4. Feedback
5. ON/OFF
LM2575D2T–** Surface Mount
1
5
1
5
Heatsink surface (shown as terminal 6 in case outline
drawing) is connected to Pin 3.
Heatsink surface
connected to Pin 3.
** = Voltage Option, ie. 3.3, 5.0, 12, 15 V and
** =\Adjustable Output.
SEMICONDUCTOR
TECHNICAL DATA
1
MOTOROLA ANALOG IC DEVICE DATA
The LM2575 series of regulators are monolithic integrated circuits ideally
suited for easy and convenient design of a step–down switching regulator
(buck converter). All circuits of this series are capable of driving a 1.0 A load
with excellent line and load regulation. These devices are available in fixed
output voltages of 3.3 V, 5.0 V, 12 V, 15 V, and an adjustable output version.
These regulators were designed to minimize the number of external
components to simplify the power supply design. Standard series of
inductors optimised for use with the LM2575 are offered by several dif ferent
inductor manufacturers.
Since the LM2575 converter is a switch–mode power supply , its ef ficiency
is significantly higher in comparison with popular three–terminal linear
regulators, especially with higher input voltages. In many cases, the power
dissipated by the LM2575 regulator is so low, that no heatsink is required or
its size could be reduced dramatically.
The LM2575 features include a guaranteed ±4% tolerance on output
voltage within specified input voltages and output load conditions, and ±10%
on the oscillator frequency (±2% over 0°C to 125°C). External shutdown is
included, featuring 80 µA typical standby current. The output switch includes
cycle–by–cycle current limiting, as well as thermal shutdown for full
protection under fault conditions.
Features
•3.3 V, 5.0 V, 12 V, 15 V, and Adjustable Output Versions
•Adjustable Version Output Voltage Range of 1.23 V to 37 V ±4%
Maximum Over Line and Load Conditions
•Guaranteed 1.0 A Output Current
•Wide Input Voltage Range: 4.75 V to 40 V
•Requires Only 4 External Components
•52 kHz Fixed Frequency Internal Oscillator
•TTL Shutdown Capability, Low Power Standby Mode
•High Efficiency
•Uses Readily Available Standard Inductors
•Thermal Shutdown and Current Limit Protection
Applications
•Simple and High–Efficiency Step–Down (Buck) Regulators
•Efficient Pre–Regulator for Linear Regulators
•On–Card Switching Regulators
•Positive to Negative Converters (Buck–Boost)
•Negative Step–Up Converters
•Power Supply for Battery Chargers
Motorola, Inc. 1999 Rev 2, 07/1999
LM2575
2MOTOROLA ANALOG IC DEVICE DATA
Figure 1. Block Diagram and Typical Application
7.0 V – 40 V
Unregulated
DC Input
L1
330
µ
H
Gnd
+Vin
1
Cin
100
µ
F3ON
/OFF5
Output
2
Feedback
4
D1
1N5819 Cout
330
µ
F
Typical Application (Fixed Output Voltage Versions)
Representative Block Diagram and Typical Application
Unregulated
DC Input +Vin
1
Cout
Feedback
4
Cin
L1
D1
R2
R1
1.0 k Output
2
Gnd
3
ON/OFF
5
Reset
Latch
Thermal
Shutdown
52 kHz
Oscillator
1.235 V
Band–Gap
Reference
Freq
Shift
18 kHz
Comparator
Fixed Gain
Error Amplifier
Current
Limit
Driver
1.0 Amp
Switch
ON/OFF
3.1 V Internal
Regulator
Regulated
Output
Vout
Load
Output
Voltage V ersions
3.3 V
5.0 V
12 V
15 V
R2
(
Ω
)
1.7 k
3.1 k
8.84 k
11.3 k
For adjustable version
R1 = open, R2 = 0
Ω
LM2575
5.0 V Regulated
Output 1.0 A Load
This device contains 162 active transistors.
ABSOLUTE MAXIMUM RATINGS (Absolute Maximum Ratings indicate limits beyond
which damage to the device may occur.)
Rating Symbol Value Unit
Maximum Supply Voltage Vin 45 V
ON/OFF Pin Input Voltage – –0.3 V ≤ V ≤ +Vin V
Output V oltage to Ground (Steady–State) – –1.0 V
Power Dissipation
Case 314B and 314D (TO–220, 5–Lead) PDInternally Limited W
Thermal Resistance, Junction–to–Ambient RθJA 65 °C/W
Thermal Resistance, Junction–to–Case RθJC 5.0 °C/W
Case 936A (D2PAK) PDInternally Limited W
Thermal Resistance, Junction–to–Ambient
(Figure 34) RθJA 70 °C/W
Thermal Resistance, Junction–to–Case RθJC 5.0 °C/W
Storage Temperature Range Tstg –65 to +150 °C
Minimum ESD Rating (Human Body Model: C
= 100 pF, R = 1.5 kΩ)– 3.0 kV
Lead Temperature (Soldering, 10 s) – 260 °C
Maximum Junction Temperature TJ150 °C
NOTE: ESD data available upon request.
LM2575
3
MOTOROLA ANALOG IC DEVICE DATA
OPERATING RATINGS (Operating Ratings indicate conditions for which the device is
intended to be functional, but do not guarantee specific performance limits. For guaranteed
specifications and test conditions, see the Electrical Characteristics.)
Rating Symbol Value Unit
Operating Junction Temperature Range TJ–40 to +125 °C
Supply Voltage Vin 40 V
SYSTEM PARAMETERS ([Note 1] Test Circuit Figure 14)
ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V for
the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the operating
junction temperature range that applies [Note 2], unless otherwise noted.)
Characteristics Symbol Min Typ Max Unit
LM2575–3.3 ([Note 1] Test Circuit Figure 14)
Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C) Vout 3.234 3.3 3.366 V
Output Voltage (4.75 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V
TJ = 25°C 3.168 3.3 3.432
TJ = –40 to +125°C 3.135 – 3.465
Efficiency (Vin = 12 V, ILoad = 1.0 A) η– 75 – %
LM2575–5 ([Note 1] Test Circuit Figure 14)
Output Voltage (Vin = 12 V, ILoad = 0.2 A, TJ = 25°C) Vout 4.9 5.0 5.1 V
Output Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V
TJ = 25°C 4.8 5.0 5.2
TJ = –40 to +125°C 4.75 – 5.25
Efficiency (Vin = 12 V, ILoad = 1.0 A) η– 77 – %
LM2575–12 ([Note 1] Test Circuit Figure 14)
Output Voltage (Vin = 25 V, ILoad = 0.2 A, TJ = 25°C) Vout 11.76 12 12.24 V
Output Voltage (15 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V
TJ = 25°C 11.52 12 12.48
TJ = –40 to +125°C 11.4 – 12.6
Efficiency (Vin = 15V, ILoad = 1.0 A) η– 88 – %
LM2575–15 ([Note 1] Test Circuit Figure 14)
Output Voltage (Vin = 30 V, ILoad = 0.2 A, TJ = 25°C) Vout 14.7 15 15.3 V
Output Voltage (18 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A) Vout V
TJ = 25°C 14.4 15 15.6
TJ = –40 to +125°C 14.25 – 15.75
Efficiency (Vin = 18 V, ILoad = 1.0 A) η– 88 – %
LM2575 ADJUSTABLE VERSION ([Note 1] Test Circuit Figure 14)
Feedback Voltage (Vin = 12 V, ILoad = 0.2 A, Vout = 5.0 V, TJ = 25°C) VFB 1.217 1.23 1.243 V
Feedback Voltage (8.0 V ≤ Vin ≤ 40 V, 0.2 A ≤ ILoad ≤ 1.0 A, Vout = 5.0 V) VFB V
TJ = 25°C 1.193 1.23 1.267
TJ = –40 to +125°C 1.18 – 1.28
Efficiency (Vin = 12 V, ILoad = 1.0 A, Vout = 5.0 V) η– 77 – %
NOTES: 1.External components such as the catch diode, inductor, input and output capacitors can affect switching regulator system performance. When the
LM2575 is used as shown in the
Figure 14
test circuit, system performance will be as shown in system parameters section
.
2.Tested junction temperature range for the LM2575: Tlow = –40°C Thigh = +125°C
LM2575
4MOTOROLA ANALOG IC DEVICE DATA
DEVICE PARAMETERS
ELECTRICAL CHARACTERISTICS (Unless otherwise specified, Vin = 12 V for the 3.3 V, 5.0 V, and Adjustable version, Vin = 25 V for
the 12 V version, and Vin = 30 V for the 15 V version. ILoad = 200 mA. For typical values TJ = 25°C, for min/max values TJ is the operating
junction temperature range that applies [Note 2], unless otherwise noted.)
Characteristics Symbol Min Typ Max Unit
ALL OUTPUT VOLTAGE VERSIONS
Feedback Bias Current (Vout = 5.0 V [Adjustable V ersion Only]) IbnA
TJ = 25°C – 25 100
TJ = –40 to +125°C – – 200
Oscillator Frequency [Note 3] fosc kHz
TJ = 25°C – 52 –
TJ = 0 to +125°C 47 – 58
TJ = –40 to +125°C 42 – 63
Saturation Voltage (Iout = 1.0 A [Note 4]) Vsat V
TJ = 25°C – 1.0 1.2
TJ = –40 to +125°C – – 1.3
Max Duty Cycle (“on”) [Note 5] DC 94 98 – %
Current Limit (Peak Current [Notes 4 and 3]) ICL A
TJ = 25°C 1.7 2.3 3.0
TJ = –40 to +125°C 1.4 – 3.2
Output Leakage Current [Notes 6 and 7], TJ = 25°C ILmA
Output = 0 V – 0.8 2.0
Output = –1.0 V – 6.0 20
Quiescent Current [Note 6] IQmA
TJ = 25°C – 5.0 9.0
TJ = –40 to +125°C – – 11
Standby Quiescent Current (ON/OFF Pin = 5.0 V (“off”)) Istby µA
TJ = 25°C – 80 200
TJ = –40 to +125°C – – 400
ON/OFF Pin Logic Input Level (Test Circuit Figure 14) V
Vout = 0 V VIH
TJ = 25°C 2.2 1.4 –
TJ = –40 to +125°C 2.4 – –
Vout = Nominal Output Voltage VIL
TJ = 25°C – 1.2 1.0
TJ = –40 to +125°C – – 0.8
ON/OFF Pin Input Current (Test Circuit Figure 14) µA
ON/OFF Pin = 5.0 V (“off”), TJ = 25°C IIH –15 30
ON/OFF Pin = 0 V (“on”), TJ = 25°C IIL – 0 5.0
NOTES: 3.The oscillator frequency reduces to approximately 18 kHz in the event of an output short or an overload which causes the regulated output voltage to
drop approximately 40% from the nominal output voltage. This self protection feature lowers the average dissipation of the IC by lowering the
minimum duty cycle from 5% down to approximately 2%.
4.Output (Pin 2) sourcing current. No diode, inductor or capacitor connected to output pin.
5.Feedback (Pin 4) removed from output and connected to 0 V.
6.Feedback (Pin 4) removed from output and connected to +12 V for the Adjustable, 3.3 V, and 5.0 V versions, and +25 V for the 12 V and 15 V
versions, to force the output transistor “off”.
7.Vin = 40 V.
LM2575
5
MOTOROLA ANALOG IC DEVICE DATA
TYPICAL PERFORMANCE CHARACTERISTICS (Circuit of Figure 14)
V
out, OUTPUT VOLTAGE CHANGE (%)
0
20
–50
3.0
0
–50
2.0
0
1.2
–50
IQ, QUIESCENT CURRENT (mA)
Vin, INPUT VOLTAGE (V)
IO, OUTPUT CURRENT (A)
TJ, JUNCTION TEMPERATURE (
°
C)
Vin, INPUT VOLTAGE (V)
INPUT–OUTPUT DIFFERENTIAL (V)
TJ, JUNCTION TEMPERATURE (
°
C)
Vsat, SATURATION VOLTAGE (V)
SWITCH CURRENT (A)
V
out, OUTPUT VOLTAGE CHANGE (%)
Figure 2. Normalized Output Voltage
TJ, JUNCTION TEMPERATURE (
°
C)
Figure 3. Line Regulation
Vin = 20 V
ILoad = 200 mA
Normalized at
TJ = 25
°
C
Figure 4. Switch Saturation Voltage Figure 5. Current Limit
Figure 6. Dropout Voltage Figure 7. Quiescent Current
ILoad = 200 mA
TJ = 25
°
C
3.3 V, 5.0 V and Adj
12 V and 15 V
25
°
C
Vin = 25 V
Vout = 5.0 V
Measured at
Ground Pin
TJ = 25
°
C
ILoad = 200 mA
ILoad = 1.0 A
∆
Vout = 5%
Rind = 0.2
Ω
125
°
C
–40
°
C
5.0–25 100 201525 257550 3530 40100 125
0.8
0.4
0.4
0
0
–0.2
–0.4
0.6
0.2
1.00.6
0.2
–0.2
–0.6
2.5
1.5
0.5
0
2.0
1.0
14
10
6.0
4.0
18
12
8.0
16
1.1
0.9
0.7
0.5
1.0
0.8
0.6
1.2
0.8
0.4
1.0
0.6
1.8
1.4
1.6
0.4 –250.1 00.2 250.3 500.4 750.5 1000.6 1250.7
5.0–25 100 1525 2050 2575 30100 35125
0.8 0.9 1.0
40
ILoad = 200 mA
ILoad = 1.0 A
LM2575
6MOTOROLA ANALOG IC DEVICE DATA
OUTPUT
VOLTAGE
(PIN 2)
OUTPUT
CURRENT
(PIN 2)
INDUCTOR
OUTPUT
RIPPLE
VOLTAGE
Vout, OUTPUT VOLTAGE Istby, STANDBY QUIESCENT CURRENT ( A)
µ
100
–50
–50
10 V
–50
0
100
µ
s/DIV
IFB, FEEDBACK PIN CURRENT (nA)
TJ, JUNCTION TEMPERATURE (
°
C)
TJ, JUNCTION TEMPERATURE (
°
C)
5.0
µ
s/DIV
NORMALIZED FREQUENCY (%)
TJ, JUNCTION TEMPERATURE (
°
C)
Istby, STANDBY QUIESCENT CURRENT ( A)
µ
Figure 8. Standby Quiescent Current
Vin, INPUT VOLTAGE (V)
Figure 9. Standby Quiescent Current
Figure 10. Oscillator Frequency Figure 11. Feedback Pin Current
Figure 12. Switching Waveforms Figure 13. Load Transient Response
Vin = 12 V
VON/OFF = 5.0 V
TJ = 25
°
C
–1001.0 A
1.0
40
0
2.0
0.5
20
1.0 A
0
120
0
0
100
0.5 A
–2.0
100
–40
80
–4.0
60
40
20 mV
–8.0
20
0
–10
0
00
40
80
120
60
20
–6.0
/DIV
ILoad, LOAD CURRENT (A)
–20
–25
–25
–25
5.0
0
0
0
10
25
25
25
15
50
50
50
20
75
75
75
25
100
100
100
30
125
125
125
4035
Vin = 12 V
Normalized at 25
°
CAdjustable
Version Only
CHANGE (mV)
CURRENT
LM2575
7
MOTOROLA ANALOG IC DEVICE DATA
Figure 14. Typical Test Circuit
D1
1N5819
L1
330
µ
H
Output
2
4
Feedback
Cout
330
µ
F
/16 V
Cin
100
µ
F/50 V
LM2575–5
1
53ON
/OFFGnd
Vin
Load
Vout
Regulated
Output
Vin
Unregulated
DC Input
8.0 V – 40 V
D1
1N5819
L1
330
µ
H
Output
2
4
Feedback
Cout
330
µ
F
/16 V
Cin
100
µ
F/50 V
LM2575
Adjustable
1
53ON
/OFFGnd
Vin
Load
Vout
Regulated
Output
Unregulated
DC Input
8.0 V – 40 V
5.0 Output Voltage Versions
Adjustable Output Voltage Versions
Vout
+
Vref
ǒ
1
)
R2
R1
Ǔ
R2
+
R1
ǒ
Vout
Vref –1
Ǔ
Where Vref = 1.23 V, R1
between 1.0 k
Ω
and 5.0 k
Ω
R2
R1
+
–
+
–
PCB LAYOUT GUIDELINES
As in any switching regulator, the layout of the printed
circuit board is very important. Rapidly switching currents
associated with wiring inductance, stray capacitance and
parasitic inductance of the printed circuit board traces can
generate voltage transients which can generate
electromagnetic interferences (EMI) and affect the desired
operation. As indicated in the Figure 14, to minimize
inductance and ground loops, the length of the leads
indicated by heavy lines should be kept as short as possible.
For best results, single–point grounding (as indicated) or
ground plane construction should be used.
On the other hand, the PCB area connected to the Pin 2
(emitter of the internal switch) of the LM2575 should be kept
to a minimum in order to minimize coupling to sensitive
circuitry.
Another sensitive part of the circuit is the feedback. It is
important to keep the sensitive feedback wiring short. To
assure this, physically locate the programming resistors near
to the regulator, when using the adjustable version of the
LM2575 regulator.
LM2575
8MOTOROLA ANALOG IC DEVICE DATA
PIN FUNCTION DESCRIPTION
Pin Symbol Description (Refer to Figure 1)
1 Vin This pin is the positive input supply for the LM2575 step–down switching regulator. In order to minimize
voltage transients and to supply the switching currents needed by the regulator, a suitable input bypass
capacitor must be present (Cin in
Figure 1).
2 Output This is the emitter of the internal switch. The saturation voltage Vsat of this output switch is typically 1.0 V.
It should be kept in mind that the PCB area connected to this pin should be kept to a minimum in order to
minimize coupling to sensitive circuitry.
3 Gnd Circuit ground pin. See the information about the printed circuit board layout.
4 Feedback This pin senses regulated output voltage to complete the feedback loop. The signal is divided by the
internal resistor divider network R2, R1 and applied to the non–inverting input of the internal error amplifier .
In the Adjustable version of the LM2575 switching regulator this pin is the direct input of the error amplifier
and the resistor network R2, R1 is connected externally to allow programming of the output voltage.
5 ON/OFF It allows the switching regulator circuit to be shut down using logic level signals, thus dropping the total
input supply current to approximately 80 µA. The input threshold voltage is typically 1.4 V. Applying a
voltage above this value (up to +Vin) shuts the regulator off. If the voltage applied to this pin is lower than
1.4 V or if this pin is connected to ground, the regulator will be in the “on” condition.
DESIGN PROCEDURE
Buck Converter Basics
The LM2575 is a “Buck” or Step–Down Converter which is
the most elementary forward–mode converter. Its basic
schematic can be seen in
Figure 15.
The operation of this regulator topology has two distinct
time periods. The first one occurs when the series switch is
on, the input voltage is connected to the input of the inductor .
The output of the inductor is the output voltage, and the
rectifier (or catch diode) is reverse biased. During this period,
since there is a constant voltage source connected across
the inductor, the inductor current begins to linearly ramp
upwards, as described by the following equation:
IL(on)
+ǒ
Vin –V
out
Ǔ
ton
L
During this “on” period, energy is stored within the core
material in the form of magnetic flux. If the inductor is properly
designed, there is sufficient energy stored to carry the
requirements of the load during the “off” period.
Figure 15. Basic Buck Converter
D1
Vin
Vout
RLoad
L
Cout
Power
Switch
The next period is the “off” period of the power switch.
When the power switch turns off, the voltage across the
inductor reverses its polarity and is clamped at one diode
voltage drop below ground by catch dioded. Current now
flows through the catch diode thus maintaining the load
current loop. This removes the stored energy from the
inductor. The inductor current during this time is:
IL(off)
+ǒ
Vout –V
D
Ǔ
t
off
L
This period ends when the power switch is once again
turned on. Regulation of the converter is accomplished by
varying the duty cycle of the power switch. It is possible to
describe the duty cycle as follows:
d
+
ton
T, where T is the period of switching.
For the buck converter with ideal components, the duty
cycle can also be described as:
d
+
Vout
Vin
Figure 16 shows the buck converter idealized waveforms
of the catch diode voltage and the inductor current.
Power
Switch
Figure 16. Buck Converter Idealized Waveforms
Power
Switch
Off
Power
Switch
Off
Power
Switch
On
Power
Switch
On
Von(SW)
VD(FWD)
Time
Time
ILoad(AV)
Imin
Ipk
Diode Diode Power
Switch
Diode VoltageInductor Current
LM2575
9
MOTOROLA ANALOG IC DEVICE DATA
Procedure
(Fixed Output Voltage Version) In order to simplify the switching regulator design, a step–by–step
design procedure and example is provided.
Procedure Example
Given Parameters:
Vout = Regulated Output Voltage (3.3 V, 5.0 V, 12 V or 15 V)
Vin(max) = Maximum DC Input Voltage
ILoad(max) = Maximum Load Current
Given Parameters:
Vout = 5.0 V
Vin(max) = 20 V
ILoad(max) = 0.8 A
1. Controller IC Selection
According to the required input voltage, output voltage and
current, select the appropriate type of the controller IC output
voltage version.
1. Controller IC Selection
According to the required input voltage, output voltage, current
polarity and current value, use the LM2575–5 controller IC
2. Input Capacitor Selection (Cin)
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
input pin +Vin and ground pin Gnd. This capacitor should be
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
2. Input Capacitor Selection (Cin)
A 47 µF, 25 V aluminium electrolytic capacitor located near to
the input and ground pins provides sufficient bypassing.
3. Catch Diode Selection (D1)
A.Since the diode maximum peak current exceeds the
regulator maximum load current the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design the diode should have a
current rating equal to the maximum current limit of the
LM2575 to be able to withstand a continuous output short
B.The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
3. Catch Diode Selection (D1)
A.For this example the current rating of the diode is 1.0 A.
B.Use a 30 V 1N5818 Schottky diode, or any of the suggested
fast recovery diodes shown in the Table 4.
4. Inductor Selection (L1)
A.According to the required working conditions, select the
correct inductor value using the selection guide from
Figures 17 to 21
.
B.From the appropriate inductor selection guide, identify the
inductance region intersected by the Maximum Input
Voltage line and the Maximum Load Current line. Each
region is identified by an inductance value and an inductor
code.
C.Select an appropriate inductor from the several different
manufacturers part numbers listed in Table 1 or Table 2.
When using Table 2 for selecting the right inductor the
designer must realize that the inductor current rating must
be higher than the maximum peak current flowing through
the inductor. This maximum peak current can be calculated
as follows
:
where ton is the “on” time of the power switch and
For additional information about the inductor, see the
inductor section in the “External Components” section of
this data sheet.
Ip(max)
+
ILoad(max)
)ǒ
Vin–Vout
Ǔ
ton
2L
ton
+
Vout
Vin x1
fosc
4. Inductor Selection (L1)
A.Use the inductor selection guide shown in Figures 17 to 21.
B.From the selection guide, the inductance area intersected
by the 20 V line and 0.8 A line is L330.
C.Inductor value required is 330 µH. From the Table 1 or
Table 2
,
choose an inductor from any of the listed
manufacturers.
LM2575
10 MOTOROLA ANALOG IC DEVICE DATA
Procedure
(Fixed Output Voltage Version) (continued)In order to simplify the switching regulator design, a step–by–step
design procedure and example is provided.
Procedure Example
5. Output Capacitor Selection (Cout)
A.Since the LM2575 is a forward–mode switching regulator
with voltage mode control, its open loop 2–pole–2–zero
frequency characteristic has the dominant pole–pair
determined by the output capacitor and inductor values. For
stable operation and an acceptable ripple voltage,
(approximately 1% of the output voltage) a value between
100 µF and 470 µF is recommended.
B.Due to the fact that the higher voltage electrolytic capacitors
generally have lower ESR (Equivalent Series Resistance)
numbers, the output capacitor’s voltage rating should be at
least 1.5 times greater than the output voltage. For a 5.0 V
regulator, a rating at least 8V is appropriate, and a 10 V or
16 V rating is recommended.
5. Output Capacitor Selection (Cout)
A.Cout = 100 µF to 470 µF standard aluminium electrolytic.
B.Capacitor voltage rating = 16 V.
Procedure (Adjustable Output Version:
LM2575–Adj)
Procedure Example
Given Parameters:
Vout = Regulated Output V oltage
Vin(max) = Maximum DC Input Voltage
ILoad(max) = Maximum Load Current
Given Parameters:
Vout = 8.0 V
Vin(max) = 12 V
ILoad(max) = 1.0 A
1. Programming Output Voltage
To select the right programming resistor R1 and R2 value (see
Figure 14) use the following formula:
Resistor R1 can be between 1.0 k and 5.0 kΩ. (For best
temperature coefficient and stability with time, use 1% metal
film resistors).
Vout
+
Vref
ǒ
1
)
R2
R1
Ǔ
R2
+
R1
ǒ
Vout
Vref –1
Ǔ
where V ref = 1.23 V
1. Programming Output Voltage (selecting R1 and R2)
Select R1 and R2:
R2 = 9.91 kΩ, choose a 9.88 k metal film resistor.
R2
+
R1
ǒ
Vout
Vref
*
1
Ǔ+
1.8 k
ǒ
8.0 V
1.23 V
*
1
Ǔ
Vout
+
1.23
ǒ
1
)
R2
R1
Ǔ
Select R1 = 1.8 kΩ
2. Input Capacitor Selection (Cin)
To prevent large voltage transients from appearing at the input
and for stable operation of the converter, an aluminium or
tantalum electrolytic bypass capacitor is needed between the
input pin +Vin and ground pin Gnd This capacitor should be
located close to the IC using short leads. This capacitor should
have a low ESR (Equivalent Series Resistance) value.
For additional information see input capacitor section in the
“External Components” section of this data sheet.
2. Input Capacitor Selection (Cin)
A 100 µF aluminium electrolytic capacitor located near the
input and ground pin provides sufficient bypassing.
3. Catch Diode Selection (D1)
A.Since the diode maximum peak current exceeds the
regulator maximum load current the catch diode current
rating must be at least 1.2 times greater than the maximum
load current. For a robust design, the diode should have a
current rating equal to the maximum current limit of the
LM2575 to be able to withstand a continuous output short.
B.The reverse voltage rating of the diode should be at least
1.25 times the maximum input voltage.
3. Catch Diode Selection (D1)
A.For this example, a 3.0 A current rating is adequate.
B.Use a 20 V 1N5820 or MBR320 Schottky diode or any
suggested fast recovery diode in the Table 4.
LM2575
11
MOTOROLA ANALOG IC DEVICE DATA
Procedure (Adjustable Output Version:
LM2575–Adj) (continued)
Procedure Example
4. Inductor Selection (L1)
A.Use the following formula to calculate the inductor V olt x
microsecond [V x µs] constant:
B.Match the calculated E x T value with the corresponding
number on the vertical axis of the Inductor Value Selection
Guide shown in Figure 21. This E x T constant is a measure
of the energy handling capability of an inductor and is
dependent upon the type of core, the core area, the number
of turns, and the duty cycle.
C.Next step is to identify the inductance region intersected by
the E x T value and the maximum load current value on the
horizontal axis shown in Figure 21.
D.From the inductor code, identify the inductor value. Then
select an appropriate inductor from the Table 1 or Table 2.
The inductor chosen must be rated for a switching
frequency of 52 kHz and for a current rating of 1.15 x
IIoad.
The inductor current rating can also be determined by
calculating the inductor peak current
:
where ton is the “on” time of the power switch and
For additional information about the inductor, see the
inductor section in the “External Components” section of
this data sheet.
ExT
+
ǒ
V
in –V
out
Ǔ
Vout
Von x106
F[Hz] [V x
m
s]
Ip(max)
+
ILoad(max)
)ǒ
Vin –V
out
Ǔ
ton
2L
ton
+
Vout
Vin x1
fosc
4. Inductor Selection (L1)
A.Calculate E x T [V x µs] constant:
B.E x T = 51 [V x µs]
C.ILoad(max) = 1.0 A
Inductance Region = L220
D.Proper inductor value = 220 µH
Choose the inductor from the Table 1 or Table 2.
ExT
+
(
12 – 8.0)x8.0
12 x1000
52
+
51 [V x
m
s]
5. Output Capacitor Selection (Cout)
A.Since the LM2575 is a forward–mode switching regulator
with voltage mode control, its open loop 2–pole–2–zero
frequency characteristic has the dominant pole–pair
determined by the output capacitor and inductor values.
For stable operation, the capacitor must satisfy the
following requirement:
B.Capacitor values between 10 µF and 2000 µF will satisfy
the loop requirements for stable operation. To achieve an
acceptable output ripple voltage and transient response, the
output capacitor may need to be several times larger than
the above formula yields.
C.Due to the fact that the higher voltage electrolytic capacitors
generally have lower ESR (Equivalent Series Resistance)
numbers, the output capacitor’s voltage rating should be at
least 1.5 times greater than the output voltage. For a 5.0 V
regulator, a rating of at least 8V is appropriate, and a 10 V
or 16 V rating is recommended.
Cout
w
7.785 Vin(max)
Vout xL[µH] [µF]
5. Output Capacitor Selection (Cout)
A.
To achieve an acceptable ripple voltage, select
Cout = 100 µF electrolytic capacitor.
Cout
w
7.785 12
8.220
+
53 µF
LM2575
12 MOTOROLA ANALOG IC DEVICE DATA
INDUCTOR VALUE SELECTION GUIDE
Vin, MAXIMUM INPUT VOLTAGE (V)
Vin, MAXIMUM INPUT VOLTAGE (V) Vin, MAXIMUM INPUT VOLT AGE (V)
IL, MAXIMUM LOAD CURRENT (A) IL, MAXIMUM LOAD CURRENT (A)
IL, MAXIMUM LOAD CURRENT (A)
0.2
60
0.2
60
0.2
200
0.2
60
0.2
60
ET, VOLTAGE TIME (V s)
µ
IL, MAXIMUM LOAD CURRENT (A)
Vin, MAXIMUM INPUT VOLT AGE (V)
Figure 17. LM2575–3.3
IL, MAXIMUM LOAD CURRENT (A)
Figure 18. LM2575–5.0
Figure 19. LM2575–12 Figure 20. LM2575–15
Figure 21. LM2575–Adj
NOTE: This Inductor Value Selection Guide is applicable for continuous mode only .
H1500H1000
L680 L470
L330
L150
H1000
L100
L680
L470
L330
L220
L150
H1500 H1000
H1500 H1000
H680
H470
H680 H680
H2200 H2200
H2200 H1500 H1000
H470
H470
L330 L220 L150
L680
L680
L680 L470
L470
L470
L100
L220
L220
L330 L330
40
40
150
20
35
25
125
40
15
30
20
100
30
10
25
15
80
25
8.0
22
12
70
20
7.0
20
10
60
18
6.0
19
9.0
50
17
18
8.0
40
16
17
7.0
20
14
5.0
0.3
0.3
0.3
0.3
0.3
0.4
0.4
0.4
0.4
0.4
0.5
0.5
0.5
0.5
0.5
0.6
0.6
0.6
0.6
0.6
0.7
0.7
0.7
0.7 0.8
0.8
0.8
0.8
0.8
0.9
0.9
0.9
0.9 1.0
1.0
1.0
1.0
1.0
15
30
L220
LM2575
13
MOTOROLA ANALOG IC DEVICE DATA
Table 1. Inductor Selection Guide
Inductor
Code Inductor
Value Pulse Eng Renco AIE Tech 39
L100 100 µH PE–92108 RL2444 415–0930 77 308 BV
L150 150 µH PE–53113 RL1954 415–0953 77 358 BV
L220 220 µH PE–52626 RL1953 415–0922 77 408 BV
L330 330 µH PE–52627 RL1952 415–0926 77 458 BV
L470 470 µH PE–53114 RL1951 415–0927 –
L680 680 µH PE–52629 RL1950 415–0928 77 508 BV
H150 150 µH PE–53115 RL2445 415–0936 77 368 BV
H220 220 µH PE–53116 RL2446 430–0636 77 410 BV
H330 330 µH PE–53117 RL2447 430–0635 77 460 BV
H470 470 µH PE–53118 RL1961 430–0634 –
H680 680 µH PE–53119 RL1960 415–0935 77 510 BV
H1000 1000 µH PE–53120 RL1959 415–0934 77 558 BV
H1500 1500 µH PE–53121 RL1958 415–0933 –
H2200 2200 µH PE–53122 RL2448 415–0945 77 610 BV
Table 2. Inductor Selection Guide
Inductance Current Schott Renco Pulse Engineering Coilcraft
(µH) (A) THT SMT THT SMT THT SMT SMT
68
0.32 67143940 67144310 RL–1284–68–43 RL1500–68 PE–53804 PE–53804–S DO1608–68
68
0.58 67143990 67144360 RL–5470–6 RL1500–68 PE–53812 PE–53812–S DO3308–683
68
0.99 67144070 67144450 RL–5471–5 RL1500–68 PE–53821 PE–53821–S DO3316–683
1.78 67144140 67144520 RL–5471–5 – PE–53830 PE–53830–S DO5022P–683
100
0.48 67143980 67144350 RL–5470–5 RL1500–100 PE–53811 PE–53811–S DO3308–104
100 0.82 67144060 67144440 RL–5471–4 RL1500–100 PE–53820 PE–53820–S DO3316–104
1.47 67144130 67144510 RL–5471–4 – PE–53829 PE–53829–S DO5022P–104
150
0.39 – 67144340 RL–5470–4 RL1500–150 PE–53810 PE–53810–S DO3308–154
150 0.66 67144050 67144430 RL–5471–3 RL1500–150 PE–53819 PE–53819–S DO3316–154
1.20 67144120 67144500 RL–5471–3 – PE–53828 PE–53828–S DO5022P–154
220
0.32 67143960 67144330 RL–5470–3 RL1500–220 PE–53809 PE–53809–S DO3308–224
220 0.55 67144040 67144420 RL–5471–2 RL1500–220 PE–53818 PE–53818–S DO3316–224
1.00 67144110 67144490 RL–5471–2 – PE–53827 PE–53827–S DO5022P–224
330
0.42 67144030 67144410 RL–5471–1 RL1500–330 PE–53817 PE–53817–S DO3316–334
330
0.80 67144100 67144480 RL–5471–1 – PE–53826 PE–53826–S DO5022P–334
NOTE: Table 1 and Table 2 of this Indicator Selection Guide shows some examples of different manufacturer products suitable for design with the LM2575.
LM2575
14 MOTOROLA ANALOG IC DEVICE DATA
Table 3. Example of Several Inductor Manufacturers Phone/Fax Numbers
Pulse Engineering Inc. Phone
Fax + 1–619–674–8100
+ 1–619–674–8262
Pulse Engineering Inc. Europe Phone
Fax + 353 93 24 107
+ 353 93 24 459
Renco Electronics Inc. Phone
Fax + 1–516–645–5828
+ 1–516–586–5562
AIE Magnetics Phone
Fax + 1–813–347–2181
Coilcraft Inc. Phone
Fax + 1–708–322–2645
+ 1–708–639–1469
Coilcraft Inc., Europe Phone
Fax + 44 1236 730 595
+ 44 1236 730 627
Tech 39 Phone
Fax + 33 8425 2626
+ 33 8425 2610
Schott Corp. Phone
Fax + 1–612–475–1 173
+ 1–612–475–1786
Table 4. Diode Selection Guide gives an overview about both surface–mount and through–hole diodes for an
effective design. Device listed in bold are available from Motorola.
V
Schottky Ultra–Fast Recovery
V
1.0 A 3.0 A 1.0 A 3.0 A
VRSMT THT SMT THT SMT THT SMT THT
20 V SK12 1N5817
SR102 SK32
MBRD320 1N5820
MBR320
SR302
30 V MBRS130LT3
SK13 1N5818
SR103
11DQ03
SK33
MBRD330 1N5821
MBR330
SR303
31DQ03
MURS120T3 MUR120
11DF1
HER102
MURS320T3
40 V MBRS140T3
SK14
10BQ040
10MQ040
1N5819
SR104
11DQ04
MBRS340T3
MBRD340
30WQ04
SK34
1N5822
MBR340
SR304
31DQ04
10BF10 MURD320 MUR320
30WF10
MUR420
50 V MBRS150
10BQ050 MBR150
SR105
11DQ05
MBRD350
SK35
30WQ05
MBR350
SR305
11DQ05
31DF1
HER302
LM2575
15
MOTOROLA ANALOG IC DEVICE DATA
EXTERNAL COMPONENTS
Input Capacitor (Cin)
The Input Capacitor Should Have a Low ESR
For stable operation of the switch mode converter a low
ESR (Equivalent Series Resistance) aluminium or solid
tantalum bypass capacitor is needed between the input pin
and the ground pin to prevent large voltage transients from
appearing at the input. It must be located near the regulator
and use short leads. With most electrolytic capacitors, the
capacitance value decreases and the ESR increases with
lower temperatures. For reliable operation in temperatures
below –25°C larger values of the input capacitor may be
needed. Also paralleling a ceramic or solid tantalum
capacitor will increase the regulator stability at cold
temperatures.
RMS Current Rating of Cin
The important parameter of the input capacitor is the RMS
current rating. Capacitors that are physically large and have
large surface area will typically have higher RMS current
ratings. For a given capacitor value, a higher voltage
electrolytic capacitor will be physically larger than a lower
voltage capacitor, and thus be able to dissipate more heat to
the surrounding air, and therefore will have a higher RMS
current rating. The consequence of operating an electrolytic
capacitor above the RMS current rating is a shortened
operating life. In order to assure maximum capacitor
operating lifetime, the capacitor’s RMS ripple current rating
should be: Irms > 1.2 x d x ILoad
where d is the duty cycle, for a buck regulator
d
+
ton
T
+
Vout
Vin
and d
+
ton
T
+
|Vout|
|Vout|
)
Vin for a buck
*
boost regulator.
Output Capacitor (Cout)
For low output ripple voltage and good stability, low ESR
output capacitors are recommended. An output capacitor has
two main functions: it filters the output and provides regulator
loop stability. The ESR of the output capacitor and the
peak–to–peak value of the inductor ripple current are the
main factors contributing to the output ripple voltage
value.Standard aluminium electrolytics could be adequate for
some applications but for quality design low ESR types are
recommended.
An aluminium electrolytic capacitor’s ESR value is related
to many factors such as the capacitance value, the voltage
rating, the physical size and the type of construction. In most
cases, the higher voltage electrolytic capacitors have lower
ESR value. Often capacitors with much higher voltage
ratings may be needed to provide low ESR values that are
required for low output ripple voltage.
The Output Capacitor Requires an ESR Value
That Has an Upper and Lower Limit
As mentioned above, a low ESR value is needed for low
output ripple voltage, typically 1% to 2% of the output voltage.
But if the selected capacitor’s ESR is extremely low (below
0.05 Ω), there is a possibility of an unstable feedback loop,
resulting in oscillation at the output. This situation can occur
when a tantalum capacitor, that can have a very low ESR, is
used as the only output capacitor.
At Low Temperatures, Put in Parallel Aluminium
Electrolytic Capacitors with Tantalum Capacitors
Electrolytic capacitors are not recommended for
temperatures below –25°C. The ESR rises dramatically at
cold temperatures and typically rises 3 times at –25°C and as
much as 10 times at –40°C. Solid tantalum capacitors have
much better ESR spec at cold temperatures and are
recommended for temperatures below –25°C. They can be
also used in parallel with aluminium electrolytics. The value
of the tantalum capacitor should be about 10% or 20% of the
total capacitance. The output capacitor should have at least
50% higher RMS ripple current rating at 52 kHz than the
peak–to–peak inductor ripple current.
Catch Diode
Locate the Catch Diode Close to the LM2575
The LM2575 is a step–down buck converter; it requires a
fast diode to provide a return path for the inductor current
when the switch turns off. This diode must be located close to
the LM2575 using short leads and short printed circuit traces
to avoid EMI problems.
Use a Schottky or a Soft Switching
Ultra–Fast Recovery Diode
Since the rectifier diodes are very significant source of
losses within switching power supplies, choosing the rectifier
that best fits into the converter design is an important
process. Schottky diodes provide the best performance
because of their fast switching speed and low forward
voltage drop.
They provide the best efficiency especially in low output
voltage applications (5.0 V and lower). Another choice could
be Fast–Recovery, or Ultra–Fast Recovery diodes. It has to
be noted, that some types of these diodes with an abrupt
turnoff characteristic may cause instability or EMI troubles.
A fast–recovery diode with soft recovery characteristics
can better fulfill a quality, low noise design requirements.
Table 4 provides a list of suitable diodes for the LM2575
regulator. Standard 50/60 Hz rectifier diodes such as the
1N4001 series or 1N5400 series are NOT suitable.
Inductor
The magnetic components are the cornerstone of all
switching power supply designs. The style of the core and
the winding technique used in the magnetic component’s
design has a great influence on the reliability of the overall
power supply.
Using an improper or poorly designed inductor can cause
high voltage spikes generated by the rate of transitions in
current within the switching power supply, and the possibility
of core saturation can arise during an abnormal operational
mode. V oltage spikes can cause the semiconductors to enter
avalanche breakdown and the part can instantly fail if enough
energy is applied. It can also cause significant RFI (Radio
Frequency Interference) and EMI (Electro–Magnetic
Interference) problems.
Continuous and Discontinuous Mode of Operation
The LM2575 step–down converter can operate in both the
continuous and the discontinuous modes of operation. The
regulator works in the continuous mode when loads are
relatively heavy, the current flows through the inductor
continuously and never falls to zero. Under light load
LM2575
16 MOTOROLA ANALOG IC DEVICE DATA
conditions, the circuit will be forced to the discontinuous
mode when inductor current falls to zero for certain period of
time (see Figure 22 and Figure 23). Each mode has
distinctively different operating characteristics, which can
affect the regulator performance and requirements. In many
cases the preferred mode of operation is the continuous
mode. It offers greater output power, lower peak currents in
the switch, inductor and diode, and can have a lower output
ripple voltage. On the other hand it does require larger
inductor values to keep the inductor current flowing
continuously, especially at low output load currents and/or
high input voltages.
To simplify the inductor selection process, an inductor
selection guide for the LM2575 regulator was added to this
data sheet (Figures 17
through
21). This guide assumes that
the regulator is operating in the continuous mode, and
selects an inductor that will allow a peak–to–peak inductor
ripple current to be a certain percentage of the maximum
design load current. This percentage is allowed to change as
different design load currents are selected. For light loads
(less than approximately 200 mA) it may be desirable to
operate the regulator in the discontinuous mode, because
the inductor value and size can be kept relatively low.
Consequently, the percentage of inductor peak–to–peak
current increases. This discontinuous mode of operation is
perfectly acceptable for this type of switching converter. Any
buck regulator will be forced to enter discontinuous mode if
the load current is light enough.
Figure 22. Continuous Mode Switching
Current Waveforms
POWER SWITCH
1.0
0
0
CURRENT (A)
HORTIZONTAL TIME BASE: 5.0
µ
s/DIV
1.0
INDUCTOR
CURRENT (A)
Selecting the Right Inductor Style
Some important considerations when selecting a core
type are core material, cost, the output power of the power
supply, the physical volume the inductor must fit within, and
the amount of EMI (Electro–Magnetic Interference) shielding
that the core must provide. The inductor selection guide
covers different styles of inductors, such as pot core, E–core,
toroid and bobbin core, as well as different core materials
such as ferrites and powdered iron from different
manufacturers.
For high quality design regulators the toroid core seems to
be the best choice. Since the magnetic flux is completely
contained within the core, it generates less EMI, reducing
noise problems in sensitive circuits. The least expensive is
the bobbin core type, which consists of wire wound on a
ferrite rod core. This type of inductor generates more EMI
due to the fact that its core is open, and the magnetic flux is
not completely contained within the core.
When multiple switching regulators are located on the
same printed circuit board, open core magnetics can cause
interference between two or more of the regulator circuits,
especially at high currents due to mutual coupling. A toroid,
pot core or E–core (closed magnetic structure) should be
used in such applications.
Do Not Operate an Inductor Beyond its
Maximum Rated Current
Exceeding an inductor’s maximum current rating may
cause the inductor to overheat because of the copper wire
losses, or the core may saturate. Core saturation occurs
when the flux density is too high and consequently the cross
sectional area of the core can no longer support additional
lines of magnetic flux.
This causes the permeability of the core to drop, the
inductance value decreases rapidly and the inductor begins
to look mainly resistive. It has only the dc resistance of the
winding. This can cause the switch current to rise very rapidly
and force the LM2575 internal switch into cycle–by–cycle
current limit, thus reducing the dc output load current. This
can also result in overheating of the inductor and/or the
LM2575. Different inductor types have different saturation
characteristics, and this should be kept in mind when
selecting an inductor.
Figure 23. Discontinuous Mode Switching
Current Waveforms
0.1
0.1
0
0
HORTIZONTAL TIME BASE: 5.0
µ
s/DIV
POWER SWITCH
CURRENT (A)
INDUCTOR
CURRENT (A)
LM2575
17
MOTOROLA ANALOG IC DEVICE DATA
GENERAL RECOMMENDATIONS
Output Voltage Ripple and Transients
Source of the Output Ripple
Since the LM2575 is a switch mode power supply
regulator, its output voltage, if left unfiltered, will contain a
sawtooth ripple voltage at the switching frequency. The
output ripple voltage value ranges from 0.5% to 3% of the
output voltage. It is caused mainly by the inductor sawtooth
ripple current multiplied by the ESR of the output capacitor.
Short Voltage Spikes and How to Reduce Them
The regulator output voltage may also contain short
voltage spikes at the peaks of the sawtooth waveform (see
Figure 24). These voltage spikes are present because of the
fast switching action of the output switch, and the parasitic
inductance of the output filter capacitor. There are some
other important factors such as wiring inductance, stray
capacitance, as well as the scope probe used to evaluate
these transients, all these contribute to the amplitude of these
spikes. To minimise these voltage spikes, low inductance
capacitors should be used, and their lead lengths must be
kept short. The importance of quality printed circuit board
layout design should also be highlighted.
Figure 24. Output Ripple Voltage Waveforms
HORTIZONTAL TIME BASE: 10
µ
s/DIV
UNFILITERED
OUTPUT
VOLTAGE
VERTICAL
RESOLUTION:
20 mV/DIV
FILITERED
OUTPUT
VOLTAGE
Voltage spikes caused by switching action of the output
switch and the parasitic inductance of the output capacitor
Minimizing the Output Ripple
In order to minimise the output ripple voltage it is possible
to enlarge the inductance value of the inductor L1 and/or to
use a larger value output capacitor . There is also another way
to smooth the output by means of an additional LC filter
(20 µH, 100 µF), that can be added to the output (see
Figure 33) to further reduce the amount of output ripple and
transients. With such a filter it is possible to reduce the
output ripple voltage transients 10 times or more. Figure 24
shows the difference between filtered and unfiltered output
waveforms of the regulator shown in Figure 33.
The upper waveform is from the normal unfiltered output of
the converter, while the lower waveform shows the output
ripple voltage filtered by an additional LC filter.
Heatsinking and Thermal Considerations
The Through–Hole Package TO–220
The LM2575 is available in two packages, a 5–pin
TO–220(T, TV) and a 5–pin surface mount D2PAK(D2T).
There are many applications that require no heatsink to keep
the LM2575 junction temperature within the allowed
operating range. The TO–220 package can be used without
a heatsink for ambient temperatures up to approximately
50°C (depending on the output voltage and load current).
Higher ambient temperatures require some heatsinking,
either to the printed circuit (PC) board or an external heatsink.
The Surface Mount Package D2PAK
and its Heatsinking
The other type of package, the surface mount D2PAK, is
designed to be soldered to the copper on the PC board. The
copper and the board are the heatsink for this package and
the other heat producing components, such as the catch
diode and inductor. The PC board copper area that the
package is soldered to should be at least 0.4 in2 (or
100 mm2) and ideally should have 2 or more square inches
(1300 mm2) of 0.0028 inch copper. Additional increasing of
copper area beyond approximately 3.0 in2 (2000 mm2) will
not improve heat dissipation significantly. If further thermal
improvements are needed, double sided or multilayer PC
boards with large copper areas should be considered.
Thermal Analysis and Design
The following procedure must be performed to determine
whether or not a heatsink will be required. First determine:
1. PD(max) maximum regulator power dissipation in the
application.
2. TA(max) maximum ambient temperature in the
application.
3. TJ(max) maximum allowed junction temperature
(125°C for the LM2575). For a conservative
design, the maximum junction temperature
should not exceed 110°C to assure safe
operation. For every additional 10°C
temperature rise that the junction must
withstand, the estimated operating lifetime of
the component is halved.
4. RθJC package thermal resistance junction–case.
5. RθJA package thermal resistance junction–ambient.
(Refer to Absolute Maximum Ratings in this data sheet or
RθJC and RθJA values).
The following formula is to calculate the total power
dissipated by the LM2575:
PD = (Vin x IQ) + d x ILoad x Vsat
where d is the duty cycle and for buck converter
d
+
ton
T
+
VO
Vin,
IQ(quiescent current) and Vsat can be found in the
LM2575 data sheet,
Vin is minimum input voltage applied,
VOis the regulator output voltage,
ILoadis the load current.
The dynamic switching losses during turn–on and turn–of f
can be neglected if proper type catch diode is used.
Packages Not on a Heatsink (Free–Standing)
For a free–standing application when no heatsink is used,
the junction temperature can be determined by the following
expression: TJ = (RθJA) (PD) + TA
where (RθJA)(PD) represents the junction temperature rise
caused by the dissipated power and TA is the maximum
ambient temperature.
LM2575
18 MOTOROLA ANALOG IC DEVICE DATA
Packages on a Heatsink
If the actual operating junction temperature is greater than
the selected safe operating junction temperature determined
in step 3, than a heatsink is required. The junction
temperature will be calculated as follows:
TJ = PD (RθJA + RθCS + RθSA) + TA
where RθJC is the thermal resistance junction–case,
RθCS is the thermal resistance case–heatsink,
RθSA is the thermal resistance heatsink–ambient.
If the actual operating temperature is greater than the
selected safe operating junction temperature, then a larger
heatsink is required.
Some Aspects That can Influence Thermal Design
It should be noted that the package thermal resistance and
the junction temperature rise numbers are all approximate,
and there are many factors that will affect these numbers,
such as PC board size, shape, thickness, physical position,
location, board temperature, as well as whether the
surrounding air is moving or still.
Other factors are trace width, total printed circuit copper
area, copper thickness, single– or double–sided, multilayer
board, the amount of solder on the board or even colour of
the traces.
The size, quantity and spacing of other components on
the board can also influence its effectiveness to dissipate
the heat.
Figure 25. Inverting Buck–Boost Regulator Using the
LM2575–12 Develops –12 V @ 0.35 A
D1
1N5819
L1
100
µ
H
Output
2
4
Feedback
Unregulated
DC Input
12 V to 25 V
Cin
100
µ
F
/50 V
1
53ON
/OFFGnd
+Vin
Regulated
Output
–12 V @ 0.35 A
Cout
1800
µ
F
/16 V
LM2575–12
ADDITIONAL APPLICATIONS
Inverting Regulator
An inverting buck–boost regulator using the LM2575–12 is
shown in Figure 25. This circuit converts a positive input
voltage to a negative output voltage with a common ground
by bootstrapping the regulators ground to the negative output
voltage. By grounding the feedback pin, the regulator senses
the inverted output voltage and regulates it.
In this example the LM2575–12 is used to generate a
–12 V output. The maximum input voltage in this case
cannot exceed +28 V because the maximum voltage
appearing across the regulator is the absolute sum of the
input and output voltages and this must be limited to a
maximum of 40 V.
This circuit configuration is able to deliver approximately
0.35 A to the output when the input voltage is 12 V or higher.
At lighter loads the minimum input voltage required drops to
approximately 4.7 V, because the buck–boost regulator
topology can produce an output voltage that, in its absolute
value, is either greater or less than the input voltage.
Since the switch currents in this buck–boost configuration
are higher than in the standard buck converter topology, the
available output current is lower.
This type of buck–boost inverting regulator can also
require a larger amount of startup input current, even for light
loads. This may overload an input power source with a
current limit less than 1.5 A.
Such an amount of input startup current is needed for at
least 2.0 ms or more. The actual time depends on the output
voltage and size of the output capacitor.
Because of the relatively high startup currents required by
this inverting regulator topology, the use of a delayed startup
or an undervoltage lockout circuit is recommended.
Using a delayed startup arrangement, the input capacitor
can charge up to a higher voltage before the switch–mode
regulator begins to operate.
The high input current needed for startup is now partially
supplied by the input capacitor Cin.
Design Recommendations:
The inverting regulator operates in a different manner than
the buck converter and so a different design procedure has to
be used to select the inductor L1 or the output capacitor Cout.
The output capacitor values must be larger than is
normally required for buck converter designs. Low input
voltages or high output currents require a large value output
capacitor (in the range of thousands of µF).
The recommended range of inductor values for the
inverting converter design is between 68 µH and 220 µH. To
select an inductor with an appropriate current rating, the
inductor peak current has to be calculated.
The following formula is used to obtain the peak inductor
current:
where ton
+
|VO|
Vin
)
|VO|x1
fosc, and fosc
+
52 kHz.
Ipeak
[
ILoad (Vin
)
|VO|)
Vin
)
Vin xt
on
2L1
Under normal continuous inductor current operating
conditions, the worst case occurs when Vin is minimal.
Note that the voltage appearing across the regulator is the
absolute sum of the input and output voltage, and must not
exceed 40 V.
LM2575
19
MOTOROLA ANALOG IC DEVICE DATA
Figure 26. Inverting Buck–Boost
Regulator with Delayed Startup
D1
1N5819
L1
100
µ
H
Output
2
4
Feedback
Unregulated
DC Input
12 V to 25 V
Cin
100
µ
F
/50 V
1
35ON
/OFF Gnd
+Vin
Regulated
Output
–12 V @ 0.35 A
Cout
1800
µ
F
/16 V
LM2575–12
C1
0.1
µ
F
R1
47 k R2
47 k
It has been already mentioned above, that in some
situations, the delayed startup or the undervoltage lockout
features could be very useful. A delayed startup circuit
applied to a buck–boost converter is shown in Figure 26.
Figure 32 in the “Undervoltage Lockout” section describes an
undervoltage lockout feature for the same converter
topology.
Figure 27. Inverting Buck–Boost Regulator Shut Down
Circuit Using an Optocoupler
LM2575–XX
1
35 GndON/OFF
+Vin
R2
47 k
Cin
100
µ
F
NOTE: This picture does not show the complete circuit.
R1
47 k
R3
470
Shutdown
Input
MOC8101
–Vout
Off
On
5.0 V
0
+Vin
With the inverting configuration, the use of the ON/OFF pin
requires some level shifting techniques. This is caused by the
fact, that the ground pin of the converter IC is no longer at
ground. Now, the ON/OFF pin threshold voltage (1.4 V
approximately) has to be related to the negative output
voltage level. There are many different possible shut down
methods, two of them are shown in Figures 27
and
28.
Figure 28. Inverting Buck–Boost Regulator Shut Down
Circuit Using a PNP Transistor
NOTE: This picture does not show the complete circuit.
R2
5.6 k
Q1
2N3906
LM2575–XX
1
35 GndON/OFF
R1
12 k –Vout
+Vin
Shutdown
Input
Off
On
+V
0
+Vin
Cin
100
µ
F
Negative Boost Regulator
This example is a variation of the buck–boost topology and
is called a negative boost regulator. This regulator
experiences relatively high switch current, especially at low
input voltages. The internal switch current limiting results in
lower output load current capability.
The circuit in Figure 29 shows the negative boost
configuration. The input voltage in this application ranges
from –5.0 V to –12 V and provides a regulated –12 V output.
If the input voltage is greater than –12 V, the output will rise
above –12 V accordingly, but will not damage the regulator.
Figure 29. Negative Boost Regulator
1N5817
150
µ
H
Output
2
4
Feedback
Regulated
Output
Vout = –12 V
Load Current from
200 mA for Vin = –5.2 V
to 500 mA for Vin = –7.0 V
Unregulated
DC Input
–Vin = –5.0 V to –12 V
L1
D1
Cout
1000
µ
F
/16 V
Cin
100
µ
F
/50 V
LM2575–12
1
53 ON/OFFGnd
+Vin
LM2575
20 MOTOROLA ANALOG IC DEVICE DATA
Design Recommendations:
The same design rules as for the previous inverting
buck–boost converter can be applied. The output capacitor
Cout must be chosen larger than would be required for a
standard buck converter. Low input voltages or high output
currents require a large value output capacitor (in the range
of thousands of µF). The recommended range of inductor
values for the negative boost regulator is the same as for
inverting converter design.
Another important point is that these negative boost
converters cannot provide current limiting load protection in
the event of a short in the output so some other means, such
as a fuse, may be necessary to provide the load protection.
Delayed Startup
There are some applications, like the inverting regulator
already mentioned above, which require a higher amount of
startup current. In such cases, if the input power source is
limited, this delayed startup feature becomes very useful.
To provide a time delay between the time the input voltage
is applied and the time when the output voltage comes up,
the circuit in Figure 30 can be used. As the input voltage is
applied, the capacitor C1 charges up, and the voltage across
the resistor R2 falls down. When the voltage on the ON/OFF
pin falls below the threshold value 1.4 V, the regulator starts
up. Resistor R1 is included to limit the maximum voltage
applied to the ON/OFF pin, reduces the power supply noise
sensitivity , and also limits the capacitor C1 discharge current,
but its use is not mandatory.
When a high 50 Hz or 60 Hz (100 Hz or 120 Hz
respectively) ripple voltage exists, a long delay time can
cause some problems by coupling the ripple into the ON/OFF
pin, the regulator could be switched periodically on and off
with the line (or double) frequency.
Figure 30. Delayed Startup Circuitry
R1
47 k
LM2575–XX
1
35 GndON/OFF
R2
47 k
+Vin
+Vin
C1
0.1
µ
F
Cin
100
µ
F
NOTE: This picture does not show the complete circuit.
Undervoltage Lockout
Some applications require the regulator to remain off until
the input voltage reaches a certain threshold level. Figure 31
shows an undervoltage lockout circuit applied to a buck
regulator. A version of this circuit for buck–boost converter is
shown in Figure 32. Resistor R3 pulls the ON/OFF pin high
and keeps the regulator off until the input voltage reaches a
predetermined threshold level, which is determined by the
following expression:
Vth
[
VZ1
)ǒ
1
)
R2
R1
Ǔ
VBE (Q1)
Figure 31. Undervoltage Lockout Circuit for
Buck Converter
R2
10 k
Z1
1N5242B
R1
10 k
Q1
2N3904
R3
47 k
Vth
≈
13 V
Cin
100
µ
F
LM2575–5.0
1
35 GndON/OFF
+Vin
+Vin
NOTE: This picture does not show the complete circuit.
Figure 32. Undervoltage Lockout Circuit for
Buck–Boost Converter
R2
15 k
Z1
1N5242B
R1
15 k
Q1
2N3904
R3
68 k
Vth
≈
13 V
Cin
100
µ
F
LM2575–5.0
1
35 GndON/OFF
+Vin
+Vin
Vout = –5.0 V
NOTE: This picture does not show the complete circuit.
Adjustable Output, Low–Ripple Power Supply
A 1.0 A output current capability power supply that
features an adjustable output voltage is shown in Figure 33.
This regulator delivers 1.0 A into 1.2 V to 35 V output. The
input voltage ranges from roughly 8.0 V to 40 V. In order to
achieve a 10 or more times reduction of output ripple, an
additional L–C filter is included in this circuit.
LM2575
21
MOTOROLA ANALOG IC DEVICE DATA
Figure 33. Adjustable Power Supply with Low Ripple Voltage
D1
1N5819
L1
150
µ
H
Output
2
4
Feedback
R2
50 k
R1
1.1 k
L2
20
µ
HRegulated
Output Voltage
1.2 V to 35 V @1.0 A
Optional Output
Ripple Filter
Unregulated
DC Input
+
Cout
2200
µ
F
C1
100
µ
F
Cin
100
µ
F
/50 V
LM2575–Adj
1
53ON
/OFFGnd
+Vin
R , THERMAL RESISTANCE
JA
θ
JUNCTION-T O-AIR ( C/W)
°
30
40
50
60
70
80
1.0
1.5
2.0
2.5
3.0
3.5
010203025155.0
L, LENGTH OF COPPER (mm)
Minimum
Size Pad
2.0 oz. Copper
L
L
ÎÎÎÎ
ÎÎÎÎ
ÎÎÎÎ
Free Air
Mounted
Vertically
PD, MAXIMUM POWER DISSIPATION (W)
Figure 34. D2PAK Thermal Resistance and Maximum
Power Dissipation versus P.C.B. Copper Length
PD(max) for TA = 50
°
C
R
θ
JA
LM2575
22 MOTOROLA ANALOG IC DEVICE DATA
THE LM2575–5.0 STEP–DOWN VOLTAGE REGULATOR WITH 5.0 V @ 1.0 A OUTPUT POWER CAPABILITY.
TYPICAL APPLICATION WITH THROUGH–HOLE PC BOARD LAYOUT
DC–DC Converter
Figure 35. Schematic Diagram of the LM2575–5.0 Step–Down Converter
Figure 36. Printed Circuit Board
Component Side Figure 37. Printed Circuit Board
Copper Side
D1
1N5819
L1
330
µ
H
Output
2
4
Feedback
Unregulated
DC Input
+Vin = +7.0 V to +40 V
Cout
330
µ
F
/16 V
C1
100
µ
F
/50 V
LM2575–5.0
1
53ON
/OFFGnd
+Vin
J1
Regulated Output
+Vout1 = 5.0 V @ 1.0 A
Gndin Gndout
C1 – 100 µF, 50 V, Aluminium Electrolytic
C2 – 330 µF, 16 V, Aluminium Electrolytic
D1 – 1.0 A, 40 V, Schottky Rectifier, 1N5819
L1 – 330 µH, Tech 39: 77 458 BV, Toroid Core, Through–Hole, Pin 3 = Start, Pin 7 = Finish
NOTE: Not to scale. NOTE: Not to scale.
+Vout1
+Vin
Gndin Gndout
C1
L1 C2
D1 J1
U1 LM2575
LM2575
23
MOTOROLA ANALOG IC DEVICE DATA
THE LM2575–ADJ STEP–DOWN VOLTAGE REGULATOR WITH 8.0 V @ 1.0 A OUTPUT POWER
CAPABILITY. TYPICAL APPLICATION WITH THROUGH–HOLE PC BOARD LAYOUT
C1 – 100 µF, 50 V, Aluminium Electrolytic
C2 – 330 µF, 16 V, Aluminium Electrolytic
C3 – 100 µF, 16 V, Aluminium Electrolytic
D1 – 1.0 A, 40 V, Schottky Rectifier, 1N5819
L1 – 330 µH, Tech 39: 77 458 BV, Toroid Core, Through–Hole, Pin 3 = Start, Pin 7 = Finish
L2 – 25 µH, TDK: SFT52501, Toroid Core, Through–Hole
R1 – 1.8 k
R2 – 10 k
Figure 38. Schematic Diagram of the 8.0 V @ 1.0 V Step–Down Converter Using the LM2575–Adj
(An additional LC filter is included to achieve low output ripple voltage)
Figure 39. PC Board Component Side Figure 40. PC Board Copper Side
Vref = 1.23 V
R1 is between 1.0 k and 5.0 k
D1
1N5819
L1
330
µ
H
Output
2R2
10 k
R1
1.8 k
L2
25
µ
HRegulated
Output Filtered
Vout2 = 8.0 V @1.0 A
Unregulated
DC Input
C2
330
µ
F
/16 V
C3
100
µ
F
/16 V
C1
100
µ
F
/50 V
LM2575–Adj
1
53ON
/OFFGnd
+Vin
+Vin = +10 V to + 40 V
4 Feedback
Regulated
Output Unfiltered
Vout1 = 8.0 V @1.0 A
Vout
+
Vref
)ǒ
1
)
R2
R1
Ǔ
+Vout1
+Vin
Gndin
C1
L1 C2
D1 J1
U1 LM2575
L2
C3
+Vout2
R2 R1
Gndout
MOTOROLA
NOTE: Not to scale. NOTE: Not to scale.
References
•National Semiconductor LM2575 Data Sheet and Application Note
•National Semiconductor LM2595 Data Sheet and Application Note
•Marty Brown “Pratical Switching Power Supply Design”, Academic Press, Inc., San Diego 1990
•Ray Ridley “High Frequency Magnetics Design”, Ridley Engineering, Inc. 1995
LM2575
24 MOTOROLA ANALOG IC DEVICE DATA
T SUFFIX
PLASTIC PACKAGE
CASE 314D–03
ISSUE D
TV SUFFIX
PLASTIC PACKAGE
CASE 314B–05
ISSUE J
OUTLINE DIMENSIONS
–Q–
12345
U
K
DG
S
A
B
5 PL
J
H
L
EC
M
Q
M
0.356 (0.014) T
SEATING
PLANE
–T–
DIM MIN MAX MIN MAX
MILLIMETERSINCHES
A0.572 0.613 14.529 15.570
B0.390 0.415 9.906 10.541
C0.170 0.180 4.318 4.572
D0.025 0.038 0.635 0.965
E0.048 0.055 1.219 1.397
G0.067 BSC 1.702 BSC
H0.087 0.112 2.210 2.845
J0.015 0.025 0.381 0.635
K1.020 1.065 25.908 27.051
L0.320 0.365 8.128 9.271
Q0.140 0.153 3.556 3.886
U0.105 0.117 2.667 2.972
S0.543 0.582 13.792 14.783
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 10.92 (0.043) MAXIMUM.
V
Q
KF
UA
B
G
–P–
M
0.10 (0.254) P M
T
5X J
M
0.24 (0.610) T
OPTIONAL
CHAMFER
SLW
E
C
H
N
–T–
SEATING
PLANE
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: INCH.
3. DIMENSION D DOES NOT INCLUDE
INTERCONNECT BAR (DAMBAR) PROTRUSION.
DIMENSION D INCLUDING PROTRUSION SHALL
NOT EXCEED 0.043 (1.092) MAXIMUM.
DIM MIN MAX MIN MAX
MILLIMETERSINCHES
A0.572 0.613 14.529 15.570
B0.390 0.415 9.906 10.541
C0.170 0.180 4.318 4.572
D0.025 0.038 0.635 0.965
E0.048 0.055 1.219 1.397
F0.850 0.935 21.590 23.749
G0.067 BSC 1.702 BSC
H0.166 BSC 4.216 BSC
J0.015 0.025 0.381 0.635
K0.900 1.100 22.860 27.940
L0.320 0.365 8.128 9.271
N0.320 BSC 8.128 BSC
Q0.140 0.153 3.556 3.886
S––– 0.620 ––– 15.748
U0.468 0.505 11.888 12.827
V––– 0.735 ––– 18.669
W0.090 0.110 2.286 2.794
5X D
LM2575
25
MOTOROLA ANALOG IC DEVICE DATA
D2T SUFFIX
PLASTIC PACKAGE
CASE 936A–02
(D2PAK)
ISSUE A
OUTLINE DIMENSIONS
5 REF
A
123
K
B
S
H
D
G
C
E
ML
P
N
R
V
U
TERMINAL 6
NOTES:
1 DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2 CONTROLLING DIMENSION: INCH.
3 TAB CONTOUR OPTIONAL WITHIN DIMENSIONS
A AND K.
4 DIMENSIONS U AND V ESTABLISH A MINIMUM
MOUNTING SURFACE FOR TERMINAL 6.
5 DIMENSIONS A AND B DO NOT INCLUDE MOLD
FLASH OR GATE PROTRUSIONS. MOLD FLASH
AND GATE PROTRUSIONS NOT TO EXCEED
0.025 (0.635) MAXIMUM.
DIM
AMIN MAX MIN MAX
MILLIMETERS
0.386 0.403 9.804 10.236
INCHES
B0.356 0.368 9.042 9.347
C0.170 0.180 4.318 4.572
D0.026 0.036 0.660 0.914
E0.045 0.055 1.143 1.397
G0.067 BSC 1.702 BSC
H0.539 0.579 13.691 14.707
K0.050 REF 1.270 REF
L0.000 0.010 0.000 0.254
M0.088 0.102 2.235 2.591
N0.018 0.026 0.457 0.660
P0.058 0.078 1.473 1.981
R5 REF
S0.116 REF 2.946 REF
U0.200 MIN 5.080 MIN
V0.250 MIN 6.350 MIN
__
45
M
0.010 (0.254) T
–T–
OPTIONAL
CHAMFER
LM2575
26 MOTOROLA ANALOG IC DEVICE DATA
NOTES
LM2575
27
MOTOROLA ANALOG IC DEVICE DATA
NOTES
LM2575
28 MOTOROLA ANALOG IC DEVICE DATA
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the suitability of its products for any particular purpose, nor does Motorola assume any liability arising out of the application or use of any product or circuit, and
specifically disclaims any and all liability, including without limitation consequential or incidental damages. “Typical” parameters which may be provided in Motorola
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