MIC3223
High Power Boost LED Driver with
Integrated FET
Micrel Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel +1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
January 2010 M9999-011510-A
General Description
The MIC3223 is a constant current boost LED driver
capable of driving a series string of high power LEDs. The
MIC3223 can be used in general lighting, bulb replacement,
garden pathway lighting and other solid state illumination
applications.
The MIC3223 is a peak current mode control PWM boost
regulator and the 4.5V and 20V operating input voltage
range allows multiple applications from a 5V or a 12V bus.
The MIC3223 implements a fixed internal 1MHz switching
frequency to allow for a reduction in the design footprint
size. Power consumption has been minimized through the
implementation of a 200mV feedback voltage that provides
an accuracy of ±5%. The MIC3223 can be dimmed through
the use of a PWM signal and features an enable pin for a
low power shutdown state.
The MIC3223 is a very robust LED driver and offers the
following protection features: over voltage protection (OVP),
thermal shutdown, switch current limiting and under voltage
lockout (UVLO).
The MIC3223 is offered in a low profile exposed pad 16-pin
TSSOP package.
Data sheets and support documentation can be found on
Micrel’s web site at: www.micrel.com.
Features
• 4.5V to 20V supply voltage
• 200mV feedback voltage with an accuracy of ±5%
• Step-up output voltage (boost) conversion up to 37V
• 1MHz switching frequency
• 100m/3.5A internal power FET switch
• LEDs can be dimmed using a PWM signal
• User settable LED current (through external resistor)
• Externally programmable soft-start
• Protection features that include:
– Output over-voltage protection (OVP)
– Under-voltage lockout (UVLO)
– Over temperature protection
• Junction temperature range: -40°C to +125°C
• Available in a exposed pad 16-pin TSSOP package
Applications
• Architectural lighting
• Industrial lighting
• Signage
• Landscape lighting (garden/pathway)
• Under cabinet lighting
• MR-16 bulbs
_______________________________________________________________________________________________________________________
Typical Application
Micrel, Inc. MIC3223
January 2010 2 M9999-011510-A
Ordering Information
Part Number Junction Temp. Range Package Lead Finish
MIC3223YTSE –40° to +125°C 16-pin ePad TSSOP PB- free
Pin Configuration
16-Pin ePad TSSOP (TSE)
Pin Description
Pin Number Pin Name Pin Function
1 EN Enable (Input): Logic high enables and logic low disables operation.
2 SS
Soft Start (Input resistance of 30k). Connect a capacitor to GND for soft-start. Clamp the
pin to a known voltage to control the internal reference voltage and hence the output
current.
3 COMP Compensation Pin (Input): Add external R and C-to-GND to stabilize the converter.
4 FB Negative Input to Error Amp
5 OVP
Connect to the centre tap of an external resistor divider, the top of which is tied to Vout
and bottom-to-ground.
6 PGND Power Ground
7,8,9,10 SW Switch Node (Input): Internal NMOS switch Drain Pin
11 VIN Input Supply
12 DRVVDD
For 4.5V < VIN < 6V, connect DRVVDD to VIN. DRVVDD is the input voltage supply for
the converter’s internal power FET gate driver. For VIN > 6V, connect this pin to VDD.
13 VDD
For 4.5V < VIN < 6V, this pin becomes the input voltage supply for the converter’s internal
circuit. For VIN > 6V, this pin is an output of the internal 5.5V regulator that supplies
internal circuits. User must add 10µF decoupling capacitor from VDD-to-AGND.
14 DIM_IN PWM input to control LED dimming.
15 DIM_OUT Output driver to drive external FET for LED dimming.
16 AGND Analog Ground
17 EP Connect to Power Ground
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January 2010 3 M9999-011510-A
Absolute Maximum Ratings(1)
Supply Voltage (VIN).....................................................+22V
Switch Voltage (VSW)..................................... -0.3V to +42V
Regulated Voltage (VDD) ............................... -0.3V to +6.5V
Dimming In Voltage (VDIM_IN) ...............-0.3V to (VDD + 0.3V)
Dimming Out Voltage (VDIM_OUT)..........-0.3V to (VDD + 0.3V)
Soft-Start Voltage (VSS).......................-0.3V to (VDD + 0.3V)
Enable Voltage (VEN)............................-0.3V to (VIN + 0.3V)
Feedback Voltage (VFB) ......................-0.3V to (VDD + 0.3V)
Switch Current (ISW) ..................................Internally Limited
Comp Voltage (VCOMP).......................-0.3V to (+VDD + 0.3V)
FET Driver Supply (VDRVVDD) ......................... -0.3V to +6.5V
PGND to AGND ............................................ -0.3V to +0.3V
Over Voltage Protection (VOVP) ...........-0.3V to (VDD + 0.3V)
Peak Reflow Temperature (soldering, 10-20sec.) ..... 260°C
Storage Temperature (TS)..........................-65°C to +150°C
ESD Rating(3)................................................................+2kV
Operating Ratings(2)
Supply Voltage (VIN)...................................... +4.5V to +20V
Switch Voltage (VSW)....................................................+37V
Junction Temperature (TJ) .........................-40°C to +125°C
Junction Thermal Resistance
ePad TSSOP-16L (θJA)...................................36.5°C/W
Electrical Characteristics(4)
VIN = VEN = 12V; L = 22µH, CIN =4.7µF, COUT =2x4.7µF; TA = 25°C, BOLD values indicate –40°C TJ +125°C, unless otherwise noted.
Symbol Parameter Condition Min Typ Max Units
VIN Voltage Supply Range 4.5
20 V
VUVLO Under Voltage Lockout Monitoring for VDD 3 3.7 4.4 V
VOVP Over Voltage Protection 1.216 1.28 1.344 V
IVIN Quiescent Current VFB=250mV 2.1 5 mA
ISD Shutdown Current VEN =0V 10 µA
Room Temperature 190 200 210 mV
VFB Feedback Voltage Over Temperature 184 216 mV
IFB Feedback Input Current VFB=200mV -450 nA
VDD Internal Voltage Regulator 5.3 V
DMAX Maximum Duty Cycle 85 90 95 %
V
DD Line Regulation VLED=18V, VIN=8V to 16V, ILED=350mA 0.5 %
ISW Switch Current Limit 3.5 9 10.5 A
RSW Switch RDSON plus RCS 100 m
ISW Switch Leakage Current VEN=0, VSW=37V 0.01 10 µA
Turn On 1.5 V
VEN Enable Threshold Turn Off 0.4 V
IEN Enable Pin Current 20 40 µA
VDIM_TH_H DIM_IN Threshold High Logic High 1.5 V
VDIM_TH_L DIM_IN Threshold Low Logic Low 0.4 V
Hys DIM_IN Hysteresis 500 mV
IDIM_IN DIM_IN Pin Current VDIM_IN = 5V 1 µA
TDR Dim Delay (Rising) DIM_IN Rising 40 ns
TDF Dim Delay (Falling DIM_IN Falling 30 ns
Micrel, Inc. MIC3223
January 2010 4 M9999-011510-A
Symbol Parameter Condition Min Typ Max Units
0.7 1.3 µs
DIM MIN Minimum Dimming Pulse
DIM_IN =1µs CDIM_OUT = 1.25nF
DIM_OUT measured from 4V rising to 2.5
falling 0.5 1.5 µs
RDO DIM_OUT Resistance High DIM_OUT pull up resistance
IDIM_OUT = +2mA 70
RDO DIM_OUT Resistance Low Dim Out pull down resistance
IDIM_OUT = -2mA 40
FSW Oscillator Frequency 0.7 1 1.3 MHz
RSS Soft Start Resistance 30 46 62 k
Temperature rising 165 °C
TSD Over Temperature Threshold
Shutdown Hysteresis 10 °C
Notes
1. Exceeding the absolute maximum rating may damage the device.
2. The device is not guaranteed to function outside its operating rating.
3. Devices are ESD sensitive. Handling precautions recommended. Human body model, 1.5k in series with 100pF.
4. Specification for packaged product only.
Test Circuit
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January 2010 5 M9999-011510-A
Typical Characteristics
Feedback Voltage
v s. In put Voltage
0.190
0.192
0.194
0.196
0.198
0.200
0.202
0.204
0.206
0.208
0.210
4 9 14 19
INPUT VOLTAGE (V)
REFERENCE VOTLAGE (V)
VOUT = 30V
IOUT =0.36A
Switching Frequency
v s. Inpu t Voltage
0.9
1.0
1.0
1.1
1.1
1.2
491419
INPUT VOLTAGE (V)
SWITCHING FREQUENCY (MHz)
T = 25°C
V
DD
= V
IN
V
IN
= 4.5V to 6V
V
OUT
= 30V
I
OUT
= 0.36A
R
SW_NODE
vs.
Temperature
0.10
0.11
0.12
0.13
0.14
0.15
0.16
0.17
0.18
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (°C)
RSW_NODE (Ω)
V
IN
= 12V
V
OUT
= 36V
I
SW
= 1.3A
Switching Frequency
vs. Temperature
0.80
0.85
0.90
0.95
1.00
1.05
1.10
1.15
1.20
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (°C)
SWITCHING FREQUENCY (MHz)
V
IN
= 12V
V
OUT
= 26V
I
OUT
= 0.36A
Efficiency
vs. Output Current
80
82
84
86
88
90
92
94
96
0 0.5 1 1.5
OUTPUT CURRENT (A)
EFFICIENCY (%)
V
OUT
= 25V
8V
10V
12V
Efficiency
vs. Output Current
80
82
84
86
88
90
92
94
96
00.511.5
OUTPUT CURRENT (A)
EFFICIENCY (%)
14V
16V
V
OUT
= 25V
Efficiency
vs. Output Current
80
82
84
86
88
90
92
94
96
98
0 0.5 1 1.5
OUTPUT CURRENT (A)
EFFICIENCY (%)
18V
20V
V
OUT
= 25V
Efficiency
vs. Input Voltage
80
82
84
86
88
90
92
94
96
98
5101520
INPUT VOLTAGE (V)
EFFICIENCY (%)
V
OUT
= 25V
I
OUT
= 0.5A
T = 25°C
VDD Voltage
v s. Input Voltage
5.00
5.05
5.10
5.15
5.20
5.25
5.30
5.35
5.40
5.45
5.50
5101520
INPUT VOLT AGE (V)
VDD VOLTAGE (V)
T = 25°C
V
OUT
= 25V
I
OUT
= 0.5A
Current Limit
v s. Input Volta ge
7.0
7.5
8.0
8.5
9.0
9.5
4 9 14 19
INPUT VOLT AGE (V)
CURRENT LIMIT (A)
T = 25°C
V
IN
= 4.5V to 6V
Feedback Voltage
vs. Temperature
0.200
0.202
0.204
0.206
0.208
0.210
0.212
0.214
0.216
0.218
0.220
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (°C)
FEEDBACK VOLTAGE (V)
V
IN
= 12V
V
OUT
= 26V
I
OUT
= 0.36A
Current Limit
vs. Temperature
6.0
6.5
7.0
7.5
8.0
8.5
9.0
9.5
10.0
10.5
11.0
-40 -20 0 20 40 60 80 100 120
TEMPERATURE (°C)
CURRENT LIMIT (A)
V
IN
= 12V
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January 2010 6 M9999-011510-A
Typical Characteristics (continued)
Efficiency
vs. Output Current
80
82
84
86
88
90
92
94
96
0 0.5 1 1.5
OUTPUT CURRENT (A)
EFFICIENCY (%)
V
OUT
= 25V
10V
Efficiency
vs. Output Current
80
82
84
86
88
90
92
94
96
0 0.5 1 1.5
OUTPUT CURRENT (A)
EFFICIENCY (%)
V
OUT
= 25V
12V
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January 2010 7 M9999-011510-A
Functional Characteristics
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January 2010 8 M9999-011510-A
Functional Characteristics (continued)
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Functional Diagram
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January 2010 10 M9999-011510-A
Functional Description
A constant current output converter is the preferred
method for driving LEDs. Small variations in current
have a minimal effect on the light output, whereas small
variations in voltage have a significant impact on light
output. The MIC3223 LED driver is specifically designed
to operate as a constant current LED Driver.
The MIC3223 is designed to operate as a boost
converter, where the output voltage is greater than the
input voltage. This configuration allows for the design of
driving multiple LEDs in series to help maintain color and
brightness. The MIC3223 can also be configured as a
SEPIC converter, where the output voltage can be either
above or below the input voltage.
The MIC3223 has an input voltage range, from 4.5V and
20V, to address a diverse range of applications. In
addition, the LED current can be programmed to a wide
range of values through the use of an external resistor.
This provides design flexibility in adjusting the current for
a particular application need.
The MIC3223 features a low impedance gate driver
capable of switching large MOSFETs. This low
impedance provides higher operating efficiency.
The MIC3223 can control the brightness of the LEDs via
its PWM dimming capability. Applying a PWM signal (up
to 20kHz) to the DIM_IN pin allows for control of the
brightness of the LEDs.
The MIC3223 boost converter employs peak current
mode control. Peak current mode control offers
advantages over voltage mode control in the following
manner. Current mode control can achieve a superior
line transient performance compared to voltage mode
control and is easier to compensate than voltage mode
control, thus allowing for a less complex control loop
stability design. Page 9 of this datasheet shows the
functional block diagram.
Boost Converter operation
The boost converter is a peak current mode pulse width
modulation (PWM) converter and operates as follows. A
flip-flop (FF) is set on the leading edge of the clock
cycle. When the FF is set, a gate driver drives the power
FET on. Current flows from VIN through the inductor (L)
and through the power switch and also through the
current sense resistor to PGND. The voltage across the
current sense resistor is added to a slope compensation
ramp (needed for stability). The sum of the current sense
voltage and the slope compensation voltages (called
VCS) is fed into the positive terminal of the PWM
comparator. The other input to the PWM comparator is
the error amp output (called VEA). The error amp’s
negative input is the feedback voltage (VFB). VFB is the
voltage across RADJ (R5). In this way the output LED
current is regulated. If VFB drops, VEA increases and
therefore the power FET remains on longer so that VCS
can increase to the level of VEA. The reverse occurs
when VFB increases.
PWM Dimming
This control process just described occurs during each
DIM_IN pulse and when ever DIM_IN is high. When
DIM_IN is low, the boost converter will no longer switch
and the output voltage will drop. For high dimming ratios
use an external PWM Dimming switch as shown in the
Typical Application. When the dim pulse is on the
external switch is on and circuit operates in the closed
loop control mode as described. When the DIM_IN is low
the boost converter does not switch and the external
switch is open and no LED current can flow and the
output voltage does not droop. When DIM_IN goes high
the external switch is driven on and LED current flows.
The output voltage remains the same (about the same)
during each on and off DIM_IN pulse.
PWM Dimming can also be used in the Test Circuit in
applications that do not require high dimming ratios. In
the Test Circuit, the load is not removed from the output
voltage between DIM_IN pulses and will therefore drain
the output capacitors. The voltage that the output will
discharge to is determined by the sum of the VF (forward
voltage drops of the LEDs). When VOUT can no longer
forward bias the LEDs, then the LED current will stop
and the output capacitors will stop discharging. During
the next DIM_IN pulse VOUT has to charge back up
before the full LED current will flow. For applications that
do not require high dimming ratios.
Micrel, Inc. MIC3223
January 2010 11 M9999-011510-A
Application Information
Constant Output Current Converter
The MIC3223 is a peak current mode boost converter
designed to drive high power LEDs with a constant
current output. The MIC3223 operates with an input
voltage range from 4.5V to 20V. In the boost
configuration, the output can be set from VIN up to 37V.
The peak current mode control architecture of the
MIC3223 provides the advantages of superior line
transient response as well as an easier to design
compensation.
The MIC3223 LED driver features a built-in soft start
circuitry in order to prevent start-up surges. Other
protection features include:
• Current Limit (ILIMIT) – Current sensing for over
current and overload protection
• Over Voltage Protection (OVP) – output over
voltage protection to prevent operation above a
safe upper limit
• Under Voltage Lockout (UVLO) – UVLO designed
to prevent operation below a safe lower limit
Setting the LED Current
The current through the LED string is set via the value
chosen for the current sense resistor RADJ which is R5 in
the schematic of the Typical Application. This value can
be calculated using Equation 1:
Eq. (1)
ADJ
R
0.2V
ILED =
Another important parameter to be aware of in the boost
converter design is the ripple current. The amount of
ripple current through the LED string is equal to the
output ripple voltage divided by the LED AC resistance
(RLED – provided by the LED manufacturer) plus the
current sense resistor RADJ. The amount of allowable
ripple through the LED string is dependent upon the
application and is left to the designer’s discretion. The
equation is shown in Equation 2.
Eq. (2) )R(R
V
I
ADJLED
OUT
LED
RIPPLE
+
≈
Where
SWOUT
LED
OUT FC
DI
VRIPPLE ×
×
=
Reference Voltage
The voltage feedback loop the MIC3223 uses an internal
voltage of 200mV with an accuracy of ±5%. The
feedback voltage is the voltage drop across the current
sense resistor as shown in the Typical Application.
When in regulation the voltage at VFB will equal 200mV.
Output Over Voltage Protection (OVP)
The MIC3223 provides an OVP circuitry in order to
protect the system from an overvoltage fault condition.
This OVP threshold can be programmed through the
use of external resistors (R3 and R4 in the Typical
Application). A reference value of 1.245V is used for the
OVP. Equation 3 can be used to calculate the resistor
value for R9 to set the OVP point. Normally use 100k for
R3.
Eq. (3) 1/1.245)(V
R3
R4
OVP −
=
VDD
An internal linear regulator is used to provide the
necessary internal bias voltages. When VIN is 6V or
below connect the VDD pin to VIN. Use a 10µF ceramic
bypass capacitor.
DRVVDD
An internal linear regulator is used to provide the
necessary internal bias voltages to the gate driver that
drives the external FET. When VIN is above 6V connect
DRVVDD to VDD.
When VIN is 6V or below connect the DRVVDD pin to
VIN. Use a bypass capacitor, 10µF ceramic capacitor.
UVLO
Internal under voltage lock out (UVLO) prevents the part
from being used below a safe VIN voltage. The UVLO is
3.7V. Operation below 4.5V is not recommended.
Soft Start
Soft start is employed to lessen the inrush currents
during turn on. At turn on the following occurs;
1. After about 1.5ms CSS will start to rise in a
exponential manner according to;
⎟
⎟
⎟
⎠
⎞
⎜
⎜
⎜
⎝
⎛
−= ×
−
)C(37k
t
SS SS
e10.2V
2. According to the block diagram, VSS is the ref
node of the error amp. PWM switching start
when VSS begins to rise.
3. When the CSS is fully charged, 0.2V will be at the
error amp reference and steady state operation
begins.
4. Design for soft-start time using the above
equation.
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January 2010 12 M9999-011510-A
Figure 1. Soft start
LED Dimming
The MIC3223 LED driver can control the brightness of
the LED string via the use of pulse width modulated
(PWM) dimming. An input signal from DC up to 20kHz
can be applied to the DIM_IN pin (see Typical
Application) to pulse the LED string ON and OFF. It is
recommended to use PWM dimming signals above
120Hz to avoid any recognizable flicker by the human
eye. PWM dimming is the preferred way to dim an LED
in order to prevent color/wavelength shifting. Color
wavelength shifting will occur with analog dimming. By
employing PWM Dimming the output current level
remains constant during each DIM_IN pulse. The boost
converter switches only when DIM_IN is high. Between
DIM_IN pulses the output capacitors will slowly
discharge. The higher the DIM_IN frequency the less the
output capacitors will discharge.
PWM Dimming Limits
The minimum pulse width of the DIM_IN is determined
by the DIM_IN frequency and the L and C used in the
boost stage output filter. At low DIM_IN frequencies
lower dimming ratios can be achieved.
PWMD
PERIOD
ELED_ON_TIM
Dim_ratio =
Figure 2. DIM_IN Dimming Ratio
If high dimming ratios are required, a lower Dimming
frequency is required. During each DIM_IN pulse the
inductor current has to ramp up to it steady state value in
order for the programmed LED current to flow. The
smaller the inductance value the faster this time is and a
narrower DIM_IN pulse can be achieved. But smaller
inductance means higher ripple current.
Figure 3. PWM Dimming 20%
Figure 3 shows that switching occurs only during DIM_IN
on pulses. When DIM_IN is low the boost converter
stops switching and the external LED is turned off. The
LED current flows only when DIM_IN is high. Figure 3
shows that the compensation pin (VCOMP) does not
discharge between DIM_IN pulses. Therefore, when the
DIM_IN pulse starts again the converter resumes
operation at the same VCOMP voltage. This eliminates the
need for the comp pin to charge up during each DIM_IN
pulse and allows for high Dimming ratios.
Figure 4. PWM Dimming 10% and ILED 100Hz
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Figure 5. PWM Dimming 20% and ILED 1kHz
In Figure 4 is at 100Hz dimming frequency and Figure 5
is 1kHz dimming frequency. The time it takes for the
LED current to reach it full value is longer with a lower
Dimming frequency. The reason is the output capacitors
slowly discharge between dimming pulses.
Figure 6. PWM Dimming 20% and ILED 1kHz
Figure 6 shows the output voltage VOUT discharge
between DIM_IN pulses. The amount of discharge is
dependent on the time between DIM_IN pulses.
Figure 7. 5µs DIM_IN Pulse
Figure 7 shows the minimum DIM_IN pulse at these
operating conditions before the ILED current starts to drop
due to low VOUT. The converter is ON (switching) only
during a DIM_IN pulse.
Figure 7 shows that at this DIM_IN pulse width the
converter is ON (switching) long enough to generate the
necessary VOUT to forward bias the LED string at the
programmed current level. Therefore this condition will
result in the desired ILED.
Figure 8. 2.5µs DIM_IN Pulse
Figure 8 shows that at this DIM_IN pulse width the
converter in not ON (switching) long enough to generate
the necessary VOUT to forward bias the LED string at the
programmed current level. As a result the LED current
drops. Therefore, this condition will not result in the
desired ILED.
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Design Procedure for a LED Driver
Symbol Parameter Min Nom Max Units
Input
VIN Input Voltage 8 12 14 V
IIN Input Current 2 A
Output
LEDs Number of LEDs 5 6 7
VF Forward Voltage of LED 3.2 3.5 4.0 V
VOUT Output Voltage 16 21 28 V
ILED LED Current 0.33 0.35 0.37 A
IPP Required I Ripple 40 mA
Pout Output Power 10.36 W
DIM_IN PWM Dimming 0 100 %
OVP Output Over Voltage Protection 30 V
System
FSW Switching Frequency 1 MHz
eff Efficiency 80 %
VDIODE Forward drop of schottky diode 0.5 V
Table 1. Design example parameters
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January 2010 15 M9999-011510-A
Design Example
In this example, we will be designing a boost LED driver
operating off a 12V input. This design has been created to
drive 6 LEDs at 350mA with a ripple of about 20%. We are
designing for 80% efficiency at a switching frequency of
1MHz.
Select RADJ
Having chosen the LED drive current to be 350mA in this
example, the current can be set by choosing the RADJ
resistor from Equation 1:
0.57
0.35A
0.2V
RADJ ==
Use the next lowest standard value 0.56.
I
LED = 0.36A
The power dissipation in this resistor is:
71mWRILEDP ADJ
2
RADJ =×=
Use a resistor rated at quarter watt or higher.
Operating Duty Cycle
The operating duty cycle can be calculated using Equation
four provided below:
Eq. (4)
()
DIODEOUT
DIODEINOUT
VV
VVV
D+
+−
=
VDIODE is the Vf of the output diode D1 in the Typical
Application. It is recommended to use a schottky diode
because it has a lower Vf than a junction diode.
These can be calculated for the nominal (typical) operating
conditions, but should also be understood for the minimum
and maximum system conditions as listed below.
(
)
DIODEOUT(nom)
DIODEIN(nom)OUT(nom)
VV
VVV
Dnom +
+−
=
(
)
DIODEOUT(max)
DIODEIN(min)OUT(max)
VV
VVV
Dmax +
+−
=
(
)
DIODEOUT(min)
DIODEIN(max)OUT(min)
VV
VVV
Dmin +
+−
=
()
44.0
5.021
5.01221
Dnom =
+
−−
=
()
21- 12 + 0.5
D= =0.44
nom 21+ 0.5
Therefore Dnom = 44%, Dmax = 72% and Dmin = 15%.
Inductor Selection
First calculate the RMS input current (nominal, min and
max) for the system given the operating conditions listed in
the design example table. The minimum value of the RMS
input current is necessary to ensure proper operation.
Using Equation 5, the following values have been
calculated:
(RMS)
IN(min)
OUT(max)OUT(max)
)IN_RMS(max 1.54A
Veff
IV
I=
×
×
=
Eq (5) (RMS)
IN(nom)
OUT(nom)OUT(nom)
)IN_RMS(nom 0.74A
Veff
IV
I=
×
×
=
(RMS)
IN(max)
OUT(min)OUT(min)
)IN_RMS(min 0.46A
Veff
IV
I=
×
×
=
IOUT is the same as ILED.
Selecting the inductor current (peak-to-peak), IL_PP, to be
between 20% to 50% of IIN_RMS(nom), in this case 40%, we
obtain:
I
IN_PP(nom) = 0.4 × IIN_RMS(nom) = 0.4 × 0.74 = 0.30AP-P
It can be difficult to find large inductor values with high
saturation currents in a surface mount package. Due to
this, the percentage of the ripple current may be limited by
the available inductor. It is recommended to operate in the
continuous conduction mode. The selection of L described
here is for continuous conduction mode.
Eq. (6) V×D
IN
L=I×F
IN_PP SW
Using the nominal values, we get:
12V × 0.44
L= =18H
0.3A ×1MHz
Select the next higher standard inductor value of 22µH.
Going back and calculating the actual ripple current gives:
PP
SW
maxIN(min)
IN_PP(max) 0.26A
1MHzH22
0.728V
FL
DV
I=
×
×
=
×
×
=
μ
The average input current is different than the RMS input
current because of the ripple current. If the ripple current is
low, then the average input current nearly equals the RMS
input current. In the case where the average input current
is different than the RMS, equation 7 shows the following:
Eq. (7)
()
12
)(I
II
2
IN_PP
2
)IN_RMS(max)IN_AVE(max −=
A54.1
12
(0.24)
)54.1(I
2
2
)IN_AVE(max ≈−=
The Maximum Peak input current IL_PK can found using
Equation 8:
Eq. (8) IL_PK(max) = IIN_AVE(max) + 0.5 ×IL_PP(max) = 1.67A
The saturation current (ISAT) at the highest operating
temperature of the inductor must be rated higher than this.
The power dissipated in the inductor is:
Micrel, Inc. MIC3223
January 2010 16 M9999-011510-A
Eq. (9) PINDUCTOR = IIN_RMS(max)
2 × DCR
A Coilcraft # MSS1260-223ML is used in this example. Its
DCR is 52m, ISAT =2.7A
P
INDUCTOR = 1.542 × 52 m = 0.123W
Output Capacitor
In this LED driver application, the ILED ripple current is a
more important factor when compared to that of the output
ripple voltage (although the two are directly related). To
find the COUT for a required ILED ripple use the following
calculation:
For an output ripple ILED(ripple) = 20ma
Eq. (10)
SWLED_totalADJ)LED(ripple
nomLED(nom)
OUT F)R(RI
DI
C×+×
×
=
Find the equivalent ac resistance RLED_ac from the
datasheet of the LED. This is the inverse slope of the ILED
vs. Vf curve i.e.:
Eq. (11) LED
V
Rf
LED_ac =
In this example use RLED_ac = 0.6 for each LED.
If the LEDs are connected in series, multiply RLED_ac = 0.6
by the total number of LEDs. In this example of six LEDs,
we obtain the following:
R
LED_total Rdynamic = 6 × 0.6 = 3.6
Eq. (12)
F1.9
F)R(RI
DI
C
SWLED_totalADJ)LED(ripple
nomLED(nom)
OUT
μ
=
×+×
×
=
Use 2.2µF or higher.
There is a trade off between the output ripple and the
rising edge of the DIM_IN pulse. This is because between
PWM dimming pulses, the converter stops pulsing and
COUT will start to discharge. The amount that COUT will
discharge depends on the time between PWM Dimming
pluses. At the next DIM_IN pulse, COUT has to be charged
up to the full output voltage VOUT before the desired LED
current flows.
Input Capacitor
The input capacitor is shown in the Typical Application.
For superior performance, ceramic capacitors should be
used because of their low equivalent series resistance
(ESR). The input capacitor CIN ripple current is equal to the
ripple in the inductor. The ripple voltage across the input
capacitor, CIN is the ESR of CIN times the inductor ripple.
The input capacitor will also bypass the EMI generated by
the converter as well as any voltage spikes generated by
the inductance of the input line. For a required VIN(ripple):
Eq. (13)
F0.75
1MHz50mV8
(0.3A)
FV
I
C
SWIN(ripple)
IN_PP
IN
μ
=
××
=
×
=
This is the minimum value that should be used. To protect
the IC from inductive spikes or any overshoot, a larger
value of input capacitance may be required.
Use 2.2µF or higher as a good safe min.
Rectifier Diode Selection
A schottky diode is best used here because of the lower
forward voltage and the low reverse recovery time. The
voltage stress on the diode is the max VOUT and therefore
a diode with a higher rating than max VOUT should be used.
An 80% de-rating is recommended here as well.
Eq. (14) IDIODE(max) = IOUT(max) = 0.36A
Since IIN_AVE(max) occurs when D is at a maximum.
Eq. (15) PDIODE(max) VDIODE × IDIODE_(max)
A SK35B is used in this example, it’s VDIODE is 0.5V
P
DIODE(max) 0.5V × 0.36A = 0.18W
MIC3223 Power Losses
To find the power losses in the MIC3223:
There is about 6mA input from VIN into the VDD pin.
The internal power switch has an RDSON of about 170m
at.
P
MIC3223 = VIN × 6mA + PwrFET
Eq. (16) PwrFET = IFET_RMS(max)
2 × Rds_on_@100°
+ VOUT(max) × IIN_AVE(max) × tsw × Fsw
R
ds_on_@100° 160m
tsw 30ns is the internal Power FET ON an OFF
transition time.
1.3A
12
I
IDI
2
L_PP
2
)IN_AVE(maxSWRMS(max) =
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛+=
PwrFET = 1.3A2 × 160m + 28V × 1.54A × 30ns
× 1MHz = 1.6W
P
MIC3223 = 8 × 6mA + 1.77W = 1.66W
Snubber
A snubber is a damping resistor in series with a DC
blocking capacitor in parallel with the power switch (same
as across the flyback diode because VOUT is an ac
ground). When the power switch turns off, the drain to
source capacitance and parasitic inductance will cause a
high frequency ringing at the switch node. A snubber
circuit as shown in the application schematic may be
required if ringing is present at the switch node. A critically
damped circuit at the switch node is where R equals the
characteristic impedance of the switch node.
Micrel, Inc. MIC3223
January 2010 17 M9999-011510-A
Eq.(17)
ds
Lparisitic
R=
snubber C
The explanation of the method to find the best R snubber
is beyond the scope of this data sheet.
Use Rsnubber = 2, ½ watt and Csnubber = 470pf to 1000pf.
The power dissipation in the Rsnubber is:
Rsnubber = Csnubber × VOUT
2 × FSW
Psnubber = 470pF × 28V2 × 1MHz = 0.4W
Power Loss in the L 0.123 W
Power Loss in the sckottky diode 0.2 W
Psnubber 0.4 W
MIC3223 Power Loss 1.66 W
Total Losses 2.4W
Efficiency 80%
Table 2. Major Power Losses
Table 2 showing the Power losses in the Design Example.
OVP - Over Voltage Protection
Set OVP higher than the maximum output voltage by at
least one Volt. To find the resistor divider values for OVP
use equation 18 and set the OVP = 30V and ROVP_H =
100k:
Eq. (18) 4.33k
1.24530
1.245100k
ROVP_L =
−
×
=
Compensation
Figure 9. Current Mode Loop Diagram
Current mode control simplifies the compensation. In
current mode, the complex poles created by the output L
and C are reduced to a single pole. The explanation for
this is beyond the scope of this datasheet, but it’s
generally thought to be because the inductor becomes a
constant current source and can’t act to change phase.
From the small signal block diagram the loop transfer
function is:
Figure 10. Simplified Control Loop
Eq. (19) T(s) = Gea(s) × Gvc(s) × H(s)
Where
For a LED driver
dynamicADJ
ADJ
RR
R
H(s) +
= and
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛+=
comp
compOmea sC
1
R||Zg(s)G
Eq. (20)
()
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛+
+
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛−
⎟
⎠
⎞
⎜
⎝
⎛
⎟
⎠
⎞
⎜
⎝
⎛
=
=
2
CsR
1
RsC1
RD'
sL
1
2
RD'
Ri
1
(s)V
(s)V
(s)G
OUTdynamic
ESROUT
dynamic
2
OP
CONTROL
OUT
VC
Where
LED
OUT
OP I
V
R= Is the DC operating point of the converter.
Rdymanic is the ac load the converter sees. When the load
on the converter is a string of LEDs, Rdymanic is the series
sum of the RLED(ac) of each LED.
RLED_total is usually between 0.1 to 1 per LED. It can be
calculated from the slope of ILED vs. Vf plot of the LED.
Ri = Ai × Rcs = 0.86
Ai = 114 and Rcs 7.5m; are internal to the ic.
The equation for Gvc(s) is theoretical and should give a
good idea of where the poles and zeros are located.
Eq.(20) shows that L2
RD'
f
L
RD'
sdynamic
2
RHPZ
dynamic
2
π
=→=
is a RHP Zero. The loop bandwidth should be about 1/5 to
1/10 of the frequency of RHPZ to ensure stability. From
Equation (20) it is shown that there is only the single pole.
OUTdynamic
pole
OUTdynamic CR2
1
f
CR
1
s
π
=→= and a Zero
due to the ESR of the output capacitor.
OUTESR
ESR
OUTESR CR2
1
f
CR
1
s
π
=→=
Micrel, Inc. MIC3223
January 2010 18 M9999-011510-A
This greatly simplifies the compensation.
One needs only to get a bode plot of the transfer function
of the control to output Gvc(s) with a network analyzer
and/or calculate it. From the bode plot find what the gain of
Gvc(s) is at 10
R
fHPZ
=. Next design the error amp gain
Gea(s) so the loop gain at the cross over frequency T(fco) is
0 db where 10
R
fco HPZ
= or less.
Error Amp
Figure 11. Internal Error Amp and External Compensation
The error amp is a gm type and the gain Gea(s) is
Eq. (21) ⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛
⎟
⎟
⎠
⎞
⎜
⎜
⎝
⎛+=
comp
compOmea sC
1
R||Zg(s)G
V
0.8mA
gm= and Zo = 1.2M.
The zero is 100
R
10
f
pC2R
1
fHPZ
co
compcom
zero === .
Set the at the mid band where Gea(fco) = gm × Rcomp. At
fzero × 10 the phase boost is near its maximum.
Figure 12. Error Amp Transfer Function
Error Amp Gain and Phase
-80
-60
-40
-20
0
20
40
60
1.E+02 1.E+03 1.E+04 1.E+05 1.E+06
FREQUENCY (Hz)
GAIN (dB) / PHASE (°)
Gain
Phase
Micrel, Inc. MIC3223
January 2010 19 M9999-011510-A
Other Applications
Figure 13. MIC3223 Typical Application without External PWM Dimming Switch
Audio noise
Audio noise from the output capacitors may exits in a
standard boost LED converter. The physical dimensions
of ceramic capacitors change with the voltage applied to
them. During PWM Dimming, the output capacitors in
standard converters are subjected to fast voltage and
current transients that may cause the output capacitors
to oscillate at the PWM Dimming frequency. This is one
reason users may want PWM dimming frequencies
above the audio range.
PCB Layout
1. All typologies of DC-to-DC converters have a
Reverse Recovery Current (RRC) of the flyback
or (freewheeling) diode. Even a Schottky diode,
which is advertised as having zero RRC, it really
is not zero. The RRC of the freewheeling diode
in a boost converter is even greater than in the
Buck converter. This is because the output
voltage is higher than the input voltage and the
diode has to charge up to –VOUT during each on-
time pulse and then discharge to Vf during the
off-time.
2. Even though the RRC is very short (tens of
nanoseconds) the peak currents are high (multiple
amperes). These fast current spikes generate EMI
(electromagnetic interference). The amount of RRC
is related to the die size and internal capacitance of
the diode. It is important not to oversize (i.e. not
more than the usual de rating) the diode because
the RRC will be needlessly higher. Example: If a 2A
diode is needed do not use a higher current rated
diode because the RRC will be needlessly higher. If
a 25V diode is needed do not use a 100V etc.
3. The high RRC causes a voltage drop on the ground
trace of the PCB and if the converter control IC is
referenced to this voltage drop, the output regulation
will suffer.
4. For good output regulation, it is important to connect
the IC’s reference to the same point as the output
capacitors to avoid the voltage drop caused by RRC.
This is also called a star connection or single point
grounding.
5. Feedback trace: The high impedance traces of the
FB should be short.
Micrel, Inc. MIC3223
January 2010 20 M9999-011510-A
Evaluation Board Schematic
37V Max 1A LED Driver
Micrel, Inc. MIC3223
January 2010 21 M9999-011510-A
Bill of Materials
Item Part Number Manufacturer Description Qty
GRM319R61E475KA12D muRata(1)
C3216X7R1E475M TDK(2)
C1
12063D475KAT2A AVX(3)
Ceramic Capacitor, 4.7µF, 25V, Size 1206, X7R 1
C2 GRM188R71C273KA01D muRata Ceramic Capacitor, 0.027µF, 6.3V, Size 0603, X7R 1
GRM188R60J106ME47D muRata
C1608X5R0J106K TDK
C3, C7
08056D106MAT2A AVX
Ceramic Capacitor, 10µF, 6.3V, Size 0603, X7R 2
12105C475KAZ2A AVX
C4, C6 GRM32ER71H475KA88L muRata
Ceramic Capacitor, 4.7µF, 50V, Size 1210, X7R 2
GRM188R71C473KA01D muRata
C5 0603YC473K4T2A AVX
Ceramic Capacitor, 0.047µF, 6.3V, Size 0603, X7R 1
C8 GRM188R72A102KA37D muRata Ceramic Capacitor, 1000pF, 100V Size 0603, X7R
D1 SK35B MCC(4) Schottky Diode, 3A, 50V (SMB) 1
L1 MSD1260-223ML-LD Coilcraft(6) Inductor, 22µH, 5A 1
R1, R3 CRCW0603100KFKEA Vishay Dale(4) Resistor, 100k, 1%, Size 0603 2
R2 CRCW0603549RFKEA Vishay Dale Resistor, 549, 1%, Size 0603 1
R4 CRCW06033K24FKEA Vishay Dale Resistor, 3.24k, 1%, Size 0603 1
R5 CRCW1206R560FKEA Vishay Dale Resistor, 0.56, 1%, 1/2W, Size 1206
(for .35A LED current Change for different ILED) 1
R6 RMC 1/4 2 1% R Stackpole Electronics,
Inc.(7) Resistor, 2, 1%, 1/2W, Size 1210 1
Si2318DS Vishay Siliconix(4)
Q1 AM2340N Analog Power(8) N-Channel 40V MOSFET 1
U1 MIC3223 Micrel, Inc.(9) High Power Boost LED Driver with Integrated FET 1
Notes:
1. Murata: www.murata.com.
2. TDK: www.tdk.com.
3. AVX: www.avx.com.
4. Vishay: www.vishay.com.
5. Internacional Rectifier: www.ift.com.
6. Coilcraft: www.coilcraft.com
7. Stackpole Electronics, Inc.: www.
8. Analog Power: www.analogpowerinc.com
8. Micrel, Inc.: www.micrel.com.
Micrel, Inc. MIC3223
January 2010 22 M9999-011510-A
PCB Layout Recommendations
Top Layer
Bottom Layer
Micrel, Inc. MIC3223
January 2010 23 M9999-011510-A
Package Information
16-Pin ePad TSSOP (TSE)
Micrel, Inc. MIC3223
January 2010 24 M9999-011510-A
Recommended Land Pattern
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL +1 (408) 944-0800 FAX +1 (408) 474-1000 WEB http://www.micrel.com
The information furnished by Micrel in this data sheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its
use. Micrel reserves the right to change circuitry and specifications at any time without notification to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product
can reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant
into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significant injury to the user. A
Purchaser’s use or sale of Micrel Products for use in life support appliances, devices or systems is a Purchaser’s own risk and Purchaser agrees to fully
indemnify Micrel for any damages resulting from such use or sale.
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