Semiconductor Components Industries, LLC, 2001
April, 2001 – Rev. 2 1Publication Order Number:
UC3842A/D
UC3842A, UC3843A,
UC2842A, UC2843A
High Performance
Current Mode Controllers
The UC3842A, UC3843A series of high performance fixed
frequency current mode controllers are specifically designed for
off–line and dc–to–dc converter applications offering the designer a
cost effective solution with minimal external components. These
integrated circuits feature a trimmed oscillator for precise duty cycle
control, a temperature compensated reference, high gain error
amplifier, current sensing comparator, and a high current totem pole
output ideally suited for driving a power MOSFET.
Also included are protective features consisting of input and
reference undervoltage lockouts each with hysteresis, cycle–by–cycle
current limiting, programmable output deadtime, and a latch for single
pulse metering.
These devices are available in an 8–pin dual–in–line plastic package
as well as the 14–pin plastic surface mount (SO–14). The SO–14
package has separate power and ground pins for the totem pole output
stage.
The UCX842A has UYLO thresholds of 16 V (on) and 10 V (off),
ideally suited for off–line converters. The UCX843A is tailored for
lower voltage applications having UVLO thresholds of 8.5 V (on) and
7.6 V (off).
•Trimmed Oscillator Discharge Current for Precise Duty Cycle
Control
•Current Mode Operation to 500 kHz
•Automatic Feed Forward Compensation
•Latching PWM for Cycle–By–Cycle Current Limiting
•Internally Trimmed Reference with Undervoltage Lockout
•High Current Totem Pole Output
•Undervoltage Lockout with Hysteresis
•Low Startup and Operating Current
•Direct Interface with ON Semiconductor SENSEFET Products
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14 SO–14
D SUFFIX
CASE 751A
1
See detailed ordering and shipping information in the package
dimensions section on page 15 of this data sheet.
ORDERING INFORMATION
See general marking information in the device marking
section on page 16 of this data sheet.
DEVICE MARKING INFORMATION
1
8
PDIP–8
N SUFFIX
CASE 626
PIN CONNECTIONS
(Top View)
Vref
(Top View)
Compensation
Voltage Feedback
Current Sense
RT/CT
Vref
VCC
Output
Gnd
1
2
3
45
6
7
8
Compensation
NC
Voltage Feedback
NC
Current Sense
NC
RT/CT
NC
VCC
VC
Output
Gnd
Power Ground
1
2
3
4
5
6
7
9
8
10
11
12
13
14
UC3842A, UC3843A, UC2842A, UC2843A
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2
Figure 1. Simplified Block Diagram
5.0V
Reference
Latching
PWM
VCC
Undervoltage
Lockout
Oscillator
Error
Amplifier
7(12)
VC
7(11)
Output
6(10)
Power
Ground
5(8)
3(5)
Current
Sense
Input
Vref
8(14)
4(7)
2(3)
1(1)
Gnd 5(9)
RTCT
Voltage
Feedback
Input
R
R
+
-
Vref
Undervoltage
Lockout
Output
Compensation
Pin numbers in parenthesis are for the D suffix SO-14 package.
VCC
MAXIMUM RATINGS
Rating Symbol Value Unit
Bias and Driver Voltages (Zero Series Impedance, see also Total Device spec) VCC, VC30 V
Total Power Supply and Zener Current (ICC + IZ) 30 mA
Output Current, Source or Sink (Note 1.) IO1.0 A
Output Energy (Capacitive Load per Cycle) W 5.0 µJ
Current Sense and Voltage Feedback Inputs Vin – 0.3 to + 5.5 V
Error Amp Output Sink Current IO10 mA
Power Dissipation and Thermal Characteristics
D Suffix, Plastic Package
Maximum Power Dissipation @ TA = 25°C
Thermal Resistance, Junction–to–Air
N Suffix, Plastic Package
Maximum Power Dissipation @ TA = 25°C
Thermal Resistance, Junction–to–Air
PD
RθJA
PD
RθJA
862
145
1.25
100
mW
°C/W
W
°C/W
Operating Junction Temperature TJ+ 150 °C
Operating Ambient Temperature
UC3842A, UC3843A
UC2842A, UC2843A
TA0 to + 70
– 25 to + 85
°C
Storage Temperature Range Tstg – 65 to + 150 °C
1. Maximum Package power dissipation limits must be observed.
UC3842A, UC3843A, UC2842A, UC2843A
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3
ELECTRICAL CHARACTERISTICS (VCC = 15 V, [Note 2.], RT = 10 k, CT = 3.3 nF, TA = Tlow to Thigh [Note 3.],
unless otherwise noted.)
UC284XA UC384XA
Characteristics Symbol Min Typ Max Min Typ Max Unit
REFERENCE SECTION
Reference Output Voltage (IO = 1.0 mA, TJ = 25°C) Vref 4.95 5.0 5.05 4.9 5.0 5.1 V
Line Regulation (VCC = 12 V to 25 V) Regline – 2.0 20 – 2.0 20 mV
Load Regulation (IO = 1.0 mA to 20 mA) Regload – 3.0 25 – 3.0 25 mV
Temperature Stability TS– 0.2 – – 0.2 – mV/°C
Total Output Variation over Line, Load, Temperature Vref 4.9 – 5.1 4.82 – 5.18 V
Output Noise Voltage (f = 10 Hz to 10 kHz,
TJ = 25°C) Vn– 50 – – 50 – µV
Long Term Stability (TA = 125°C for 1000 Hours) S – 5.0 – – 5.0 – mV
Output Short Circuit Current ISC – 30 – 85 – 180 – 30 – 85 – 180 mA
OSCILLATOR SECTION
Frequency
TJ = 25°C
TA = Tlow to Thigh
fosc 47
46 52
–57
60 47
46 52
–57
60
kHz
Frequency Change with Voltage (VCC = 12 V to 25 V) ∆fosc/∆V– 0.2 1.0 – 0.2 1.0 %
Frequency Change with Temperature
TA = Tlow to Thigh ∆fosc/∆T– 5.0 – – 5.0 – %
Oscillator Voltage Swing (Peak–to–Peak) Vosc – 1.6 – – 1.6 – V
Discharge Current (Vosc = 2.0 V)
TJ = 25°C
TA = Tlow to Thigh
Idischg 7.5
7.2 8.4
–9.3
9.5 7.5
7.2 8.4
–9.3
9.5
mA
ERROR AMPLIFIER SECTION
Voltage Feedback Input (VO = 2.5 V) VFB 2.45 2.5 2.55 2.42 2.5 2.58 V
Input Bias Current (VFB = 2.7 V) IIB – –0.1 –1.0 – –0.1 –2.0 µA
Open Loop Voltage Gain (VO = 2.0 V to 4.0 V) AVOL 65 90 – 65 90 – dB
Unity Gain Bandwidth (TJ = 25°C) BW 0.7 1.0 – 0.7 1.0 – MHz
Power Supply Rejection Ratio (VCC = 12 V to 25 V) PSRR 60 70 – 60 70 – dB
Output Current
Sink (VO = 1.1 V, VFB = 2.7 V)
Source (VO = 5.0 V, VFB = 2.3 V) ISink
ISource 2.0
–0.5 12
–1.0 –
–2.0
–0.5 12
–1.0 –
–
mA
Output Voltage Swing
High State (RL = 15 k to ground, VFB = 2.3 V)
Low State (RL = 15 k to Vref, VFB = 2.7 V) VOH
VOL 5.0
–6.2
0.8 –
1.1 5.0
–6.2
0.8 –
1.1
V
2. Adjust VCC above the Startup threshold before setting to 15 V.
3. Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
Tlow =0°C for UC3842A, UC3843A Thigh = +70°C for UC3842A, UC3843A
–25°C for UC2842A, UC2843A +85°C for UC2842A, UC2843A
UC3842A, UC3843A, UC2842A, UC2843A
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4
ELECTRICAL CHARACTERISTICS (VCC = 15 V, [Note 4.], RT = 10 k, CT = 3.3 nF, TA = Tlow to Thigh [Note 5.],
unless otherwise noted.)
UC284XA UC384XA
Characteristics Symbol Min Typ Max Min Typ Max Unit
CURRENT SENSE SECTION
Current Sense Input Voltage Gain (Notes 6. & 7.) AV2.85 3.0 3.15 2.85 3.0 3.15 V/V
Maximum Current Sense Input Threshold (Note 6.) Vth 0.9 1.0 1.1 0.9 1.0 1.1 V
Power Supply Rejection Ratio
VCC = 12 to 25 V (Note 6.) PSRR – 70 – – 70 – dB
Input Bias Current IIB – –2.0 –10 – –2.0 –10 µA
Propagation Delay (Current Sense Input to Output) tPLH(in/out) – 150 300 – 150 300 ns
OUTPUT SECTION
Output Voltage
Low State (ISink = 20 mA)
Low State (ISink = 200 mA)
High State (ISink = 20 mA)
High State (ISink = 200 mA)
VOL
VOH
–
–
13
12
0.1
1.6
13.5
13.4
0.4
2.2
–
–
–
–
13
12
0.1
1.6
13.5
13.4
0.4
2.2
–
–
V
Output Voltage with UVLO Activated
VCC = 6.0 V, ISink = 1.0 mA VOL(UVLO) – 0.1 1.1 – 0.1 1.1 V
Output Voltage Rise Time (CL = 1.0 nF, TJ = 25°C) tr– 50 150 – 50 150 ns
Output Voltage Fall Time (CL = 1.0 nF, TJ = 25°C) tf– 50 150 – 50 150 ns
UNDERVOLTAGE LOCKOUT SECTION
Startup Threshold
UCX842A
UCX843A
Vth 15
7.8 16
8.4 17
9.0 14.5
7.8 16
8.4 17.5
9.0
V
Minimum Operating Voltage After Turn–On
UCX842A
UCX843A
VCC(min) 9.0
7.0 10
7.6 11
8.2 8.5
7.0 10
7.6 11.5
8.2
V
PWM SECTION
Duty Cycle
Maximum
Minimum DCmax
DCmin 94
–96
––
094
–96
––
0
%
TOTAL DEVICE
Power Supply Current (Note 4.)
Startup:
(VCC = 6.5 V for UCX843A,
(VCC = 14 V for UCX842A) Operating
ICC
–
–0.5
12 1.0
17 –
–0.5
12 1.0
17
mA
Power Supply Zener Voltage (ICC = 25 mA) VZ30 36 – 30 36 – V
4. Adjust VCC above the Startup threshold before setting to 15 V.
5. Low duty cycle pulse techniques are used during test to maintain junction temperature as close to ambient as possible.
Tlow =0°C for UC3842A, UC3843A Thigh = +70°C for UC3842A, UC3843A
–25°C for UC2842A, UC2843A +85°C for UC2842A, UC2843A
6. This parameter is measured at the latch trip point with VFB = 0 V.
7. Comparator gain is defined as: AV∆V Output Compensation
∆V Current Sense Input
UC3842A, UC3843A, UC2842A, UC2843A
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5
RTΩ, TIMING RESISTOR (k )
Figure 2. Timing Resistor versus
Oscillator Frequency Figure 3. Output Deadtime versus
Oscillator Frequency
Figure 4. Oscillator Discharge Current
versus Temperature Figure 5. Maximum Output Duty Cycle
versus Timing Resistor
Figure 6. Error Amp Small Signal
Transient Response Figure 7. Error Amp Large Signal
Transient Response
0.5 µs/DIV
20 mV/DIV
VCC = 15 V
AV = -1.0
TA = 25°C
10 k 20 k 50 k 100 k 200 k 500 k 1.0 M
fOSC, OSCILLATOR FREQUENCY (Hz)
VCC = 15 V
TA = 25°C
10 k 20 k 50 k 100 k 200 k 500 k 1.0 M
fOSC, OSCILLATOR FREQUENCY (Hz)
% DT, PERCENT OUTPUT DEADTIME
VCC = 15 V
TA = 25°C
-55 -25 0 25 50 75 100 125
TA, AMBIENT TEMPERATURE (°C)
, DISCHARGE CURRENT (mA)
dischg
I
VCC = 15 V
VOSC = 2.0 V
RT, TIMING RESISTOR (Ω)
800 1.0 k 2.0 k 3.0 k 4.0 k 6.0 k 8.0 k
, MAXIMUM OUTPUT DUTY CYCLE (%)
max
D
VCC = 15 V
CT = 3.3 nF
TA = 25°C
Idischg = 9.5 mA
Idischg = 7.2 mA
2.55 V
2.5 V
2.45 V
VCC = 15 V
AV = -1.0
TA = 25°C
0.1 µs/DIV
200 mV/DIV
2.5 V
3.0 V
2.0 V
80
50
20
8.0
5.0
2.0
0.8
100
50
20
10
5.0
2.0
1.0
9.0
8.5
8.0
7.5
7.0
100
90
80
70
60
50
40
UC3842A, UC3843A, UC2842A, UC2843A
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6
Figure 8. Error Amp Open Loop Gain and
Phase versus Frequency Figure 9. Current Sense Input Threshold
versus Error Amp Output Voltage
Figure 10. Reference Voltage Change
versus Source Current Figure 11. Reference Short Circuit Current
versus Temperature
Figure 12. Reference Load Regulation Figure 13. Reference Line Regulation
∆, OUTPUT VOLTAGE CHANGE (2.0 mV/DIV)
O
2.0 ms/DIV
V
∆, OUTPUT VOLTAGE CHANGE (2.0 mV/DIV)
O
2.0 ms/DIV
V
VCC = 12 V to 25 V
TA = 25°C
∆, REFERENCE VOLTAGE CHANGE (mV)
ref
0 20 40 60 80 100 120
Iref, REFERENCE SOURCE CURRENT (mA)
V
VCC = 15 V
TA = 55°C
TA = 125°C
, REFERENCE SHORT CIRCUIT CURRENT (mA)
SC
-55 -25 0 25 50 75 100 125
TA, AMBIENT TEMPERATURE (°C)
VCC = 15 V
RL ≤ 0.1 Ω
I
VCC = 15 V
IO = 1.0 mA to 20 mA
TA = 25°C
0
-4.0
-8.0
-12
-16
-20
-24
110
90
70
50
TA = 25°C
-20
AVOL , OPEN LOOP VOLTAGE GAIN (dB)
10 M10
f, FREQUENCY (Hz)
Gain
Phase
VCC = 15 V
VO = 2.0 V to 4.0 V
RL = 100 K
TA = 25°C
0
30
60
90
120
150
180
100 1.0 k 10 k 100 k 1.0 M
0
20
40
60
80
100
, EXCESS PHASE (DEGREES)
φ
0
VO, ERROR AMP OUTPUT VOLTAGE (V)
0
, CURRENT SENSE INPUT THRESHOLD (V
)
Vth
0.2
0.4
0.6
0.8
1.0
1.2
2.0 4.0 6.0 8.0
VCC = 15 V
TA = 25°C
TA = -55°C
TA = 125°C
UC3842A, UC3843A, UC2842A, UC2843A
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Figure 14. Output Saturation Voltage
versus Load Current Figure 15. Output Waveform
Figure 16. Output Cross Conduction Figure 17. Supply Current versus
Supply Voltage
50 ns/DIV
VCC = 15 V
CL = 1.0 nF
TA = 25°C
100 ns/DIV
VCC = 30 V
CL = 15 pF
TA = 25°C
, SUPPLY CURRENT
100 mA/DIV 20 V/DIV
I, OUTPUT VOLTAGEV
CC O
8006004002000
IO, OUTPUT LOAD CURRENT (mA)
, OUTPUT SATURATION VOLTAGE (V)
sat
V
VCC
TA = 25°C
TA = -55°C
Gnd
TA = 25°C
Source Saturation
(Load to Ground)
TA = -55°C
VCC = 15 V
80 µs Pulsed Load
120 Hz Rate
010203040
, SUPPLY CURRENT (mA)
CC
VCC , SUPPLY VOLTAGE
I
RT = 10 k
CT = 3.3 nF
VFB = 0 V
ISense = 0 V
TA = 25°C
UCX843A
UCX842A
90%
10%
0
1.0
2.0
3.0
-2.0
-1.0
0
25
20
15
10
5
0
Sink Saturation
(Load to VCC)
UC3842A, UC3843A, UC2842A, UC2843A
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8
+
-
Sink Only
Positive True Logic
=
RS
+
Internal
Bias
Reference
Regulator
Oscillator
S
R
Q
-
Vref
UVLO
3.6V
36V
VCC 7(12)
Q1
Vin
VCC
VC
7(11)
6(10)
5(8)
3(5)
+
1.0mA
Error
Amplifier
1(1)
2(3)
4(7)
8(14)
5(9)Gnd
Output
Compensation
Voltage Feedback
Input
RT
CT
Vref
-
-
PWM
Latch
Current Sense
Comparator
R
R
Power Ground
Current Sense Input
2R
R1.0V
Pin numbers in parenthesis are for the D suffix SO-14 package.
QT
+
-
+
+
-
+
-
+
VCC
UVLO
Output
2.5V
Figure 18. Representative Block Diagram
Output/
Compensation
Current Sense
Input
Latch
``Reset'' Input
Output
Capacitor CT
Latch
``Set'' Input
Large RT/Small CTSmall RT/Large CT
Figure 19. Timing Diagram
UC3842A, UC3843A, UC2842A, UC2843A
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OPERATING DESCRIPTION
The UC3842A, UC3843A series are high performance,
fixed frequency, current mode controllers. They are
specifically designed for Off–Line and dc–to–dc converter
applications offering the designer a cost effective solution
with minimal external components. A representative block
diagram is shown in Figure 18.
Oscillator
The oscillator frequency is programmed by the values
selected for the timing components RT and C T. Capacitor CT
is charged from the 5.0 V reference through resistor RT to
approximately 2.8 V and discharged to 1.2 V by an internal
current sink. During the discharge of CT, the oscillator
generates and internal blanking pulse that holds the center
input of the NOR gate high. This causes the Output to be in
a low state, thus producing a controlled amount of output
deadtime. Figure 2 shows RT versus Oscillator Frequency
and Figure 3, Output Deadtime versus Frequency, both for
given values of CT. Note that many values of R T and CT will
give the same oscillator frequency but only one combination
will yield a specific output deadtime at a given frequency.
The oscillator thresholds are temperature compensated, and
the discharge current is trimmed and guaranteed to within
±10% at TJ = 25°C. These internal circuit refinements
minimize variations of oscillator frequency and maximum
output duty cycle. The results are shown in Figures 4 and 5.
In many noise sensitive applications it may be desirable to
frequency–lock the converter to an external system clock.
This can be accomplished by applying a clock signal to the
circuit shown in Figure 21. For reliable locking, the
free–running oscillator frequency should be set about 10%
less than the clock frequency. A method for multi unit
synchronization is shown in Figure 22. By tailoring the
clock waveform, accurate Output duty cycle clamping can
be achieved.
Error Amplifier
A fully compensated Error Amplifier with access to the
inverting input and output is provided. It features a typical
dc voltage gain of 90 dB, and a unity gain bandwidth of
1.0 MHz with 57 degrees of phase margin (Figure 8). The
noninverting input is internally biased at 2.5 V and is not
pinned out. The converter output voltage is typically divided
down and monitored by the inverting input. The maximum
input bias current is –2.0 µA which can cause an output
voltage error that is equal to the product of the input bias
current and the equivalent input divider source resistance.
The Error Amp Output (Pin 1) is provide for external loop
compensation (Figure 31). The output voltage is offset by
two diode drops (≈ 1.4 V) and divided by three before it
connects to the inverting input of the Current Sense
Comparator. This guarantees that no drive pulses appear at
the Output (Pin 6) when Pin 1 is at its lowest state (VOL).
This occurs when the power supply is operating and the load
is removed, or at the beginning of a soft–start interval
(Figures 24, 25). The Error Amp minimum feedback
resistance is limited by the amplifier’s source current
(0.5 mA) and the required output voltage (VOH) to reach the
comparator’s 1.0 V clamp level:
Rf(min) ≈3.0 (1.0 V) + 1.4 V
0.5 mA = 8800 Ω
Current Sense Comparator and PWM Latch
The UC3842A, UC3843A operate as a current mode
controller, whereby output switch conduction is initiated by
the oscillator and terminated when the peak inductor current
reaches the threshold level established by the Error
Amplifier Output/Compensation (Pin1). Thus the error
signal controls the peak inductor current on a
cycle–by–cycle basis. The current Sense Comparator PWM
Latch configuration used ensures that only a single pulse
appears at the Output during any given oscillator cycle. The
inductor current is converted to a voltage by inserting the
ground referenced sense resistor RS in series with the source
of output switch Q1. This voltage is monitored by the
Current Sense Input (Pin 3) and compared a level derived
from the Error Amp Output. The peak inductor current under
normal operating conditions is controlled by the voltage at
pin 1 where:
Ipk = V(Pin 1) – 1.4 V
3 RS
Abnormal operating conditions occur when the power
supply output is overloaded or if output voltage sensing is
lost. Under these conditions, the Current Sense Comparator
threshold will be internally clamped to 1.0 V. Therefore the
maximum peak switch current is:
Ipk(max) = 1.0 V
RS
When designing a high power switching regulator it
becomes desirable to reduce the internal clamp voltage in
order to keep the power dissipation of RS to a reasonable
level. A simple method to adjust this voltage is shown in
Figure 23. The two external diodes are used to compensate
the internal diodes yielding a constant clamp voltage over
temperature. Erratic operation due to noise pickup can result
if there is an excessive reduction of the Ipk(max) clamp
voltage.
A narrow spike on the leading edge of the current
waveform can usually be observed and may cause the power
supply to exhibit an instability when the output is lightly
loaded. This spike is due to the power transformer
interwinding capacitance and output rectifier recovery time.
The addition of an RC filter on the Current Sense Input with
a time constant that approximates the spike duration will
usually eliminate the instability; refer to Figure 27.
UC3842A, UC3843A, UC2842A, UC2843A
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PIN FUNCTION DESCRIPTION
Pin
8–Pin 14–Pin Function Description
1 1 Compensation This pin is Error Amplifier output and is made available for loop compensation.
2 3 Voltage
Feedback This is the inverting input of the Error Amplifier. It is normally connected to the switching
power supply output through a resistor divider.
3 5 Current Sense A voltage proportional to inductor current is connected to this input. The PWM uses this
information to terminate the output switch conduction.
4 7 RT/CTThe Oscillator frequency and maximum Output duty cycle are programmed by connecting
resistor RT to Vref and capacitor CT to ground. Operation to 500 kHz is possible.
5 – Gnd This pin is the combined control circuitry and power ground (8–pin package only).
6 10 Output This output directly drives the gate of a power MOSFET. Peak currents up to 1.0 A are
sourced and sunk by this pin.
7 12 VCC This pin is the positive supply of the control IC.
8 14 Vref This is the reference output. It provides charging current for capacitor CT through
resistor RT.
– 8 Power Ground This pin is a separate power ground return (14–pin package only) that is connected back
to the power source. It is used to reduce the effects of switching transient noise on the
control circuitry.
–11 VCThe Output high state (VOH) is set by the voltage applied to this pin (14–pin package only).
With a separate power source connection, it can reduce the effects of switching transient
noise on the control circuitry.
– 9 Gnd This pin is the control circuitry ground return (14–pin package only) and is connected back to
the power source ground.
– 2,4,6,13 NC No connection (14–pin package only). These pins are not internally connected.
Undervoltage Lockout
Two undervoltage lockout comparators have been
incorporated to guarantee that the IC is fully functional
before the output stage is enabled. The positive power
supply terminal (VCC) and the reference output (Vref) are
each monitored by separate comparators. Each has built–in
hysteresis to prevent erratic output behavior as their
respective thresholds are crossed. The VCC comparator
upper and lower thresholds are 16 V/10 V for the UCX842A,
and 8.4 V/7.6 V for the UCX843A. The Vref comparator
upper and lower thresholds are 3.6V/3.4 V. The large
hysteresis and low startup current of the UCX842A makes
it ideally suited in off–line converter applications where
efficient bootstrap startup techniques are required
(Figure 34). The UCX843A is intended for lower voltage dc
to dc converter applications. A 36 V zener is connected as
a shunt regulator form VCC to ground. Its purpose is to
protect the IC from excessive voltage that can occur during
system startup. The minimum operating voltage for the
UCX842A is 11 V and 8.2 V for the UCX843A.
Output
These devices contain a single totem pole output stage that
was specifically designed for direct drive of power
MOSFETs. It is capable of up to ±1.0 A peak drive current
and has a typical rise and fall time of 50 ns with a 1.0 nF load.
Additional internal circuitry has been added to keep the
Output in a sinking mode whenever an undervoltage lockout
is active. This characteristic eliminates the need for an
external pull–down resistor.
The SO–14 surface mount package provides separate pins
for VC (output supply) and Power Ground. Proper
implementation will significantly reduce the level of
switching transient noise imposed on the control circuitry.
This becomes particularly useful when reducing the Ipk(max)
clamp level. The separate VC supply input allows the
designer added flexibility in tailoring the drive voltage
independent of VCC. A zener clamp is typically connected
to this input when driving power MOSFETs in systems
where VCC is greater than 20 V. Figure 26 shows proper
power and control ground connections in a current sensing
power MOSFET application.
Reference
The 5.0 V bandgap reference is trimmed to ±1.0%
tolerance a t T J = 25°C on the UC284XA, and ± 2.0% on the
UC384XA. Its primary purpose is to supply charging current
to the oscillator timing capacitor. The reference has short
circuit protection and is capable of providing in excess of
20 mA for powering additional control system circuitry.
UC3842A, UC3843A, UC2842A, UC2843A
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11
DESIGN CONSIDERATIONS
Do not attempt to construct the converter on
wire–wrap o r plug–in pr ototype boards. High Frequency
circuit layout techniques are imperative to prevent
pulsewidth jitter. This is usually caused by excessive noise
pick–up imposed on the Current Sense o r Voltage Feedback
inputs. Noise immunity can be improved by lowering circuit
impedances a t these points. The printed circuit layout should
contain a ground plane with low–current signal and
high–current switch and output grounds returning on
separate paths back to the input filter capacitor. Ceramic
bypass capacitors (0.1 µF) connected directly to VCC, VC,
and Vref may be required depending upon circuit layout.
This provides a low impedance path for filtering the high
frequency noise. All high current loops should be kept as
short as possible using heavy copper runs to minimize
radiated EMI. The Error Amp compensation circuitry and
the converter output voltage divider should be located close
to the IC and as far as possible from the power switch and
other noise generating components.
Current mode converters can exhibit subharmonic
oscillations when operating a t a duty cycle greater than 50%
with continuous inductor current. This instability is
independent of the regulators closed–loop characteristics
and is caused by the simultaneous operating conditions of
fixed frequency and peak current detecting. Figure 20A
shows the phenomenon graphically. At t0, switch
conduction begins, causing the inductor current to rise at a
slope of m1. This slope is a function of the input voltage
divided by the inductance. At t1, the Current Sense Input
reaches the threshold established by the control voltage.
This causes the switch to turn off and the current to decay at
a slope of m2 until the next oscillator cycle. The unstable
condition can be shown if a pertubation is added to the
control voltage, resulting in a small ∆I (dashed line). With
a fixed oscillator period, the current decay time is reduced,
and the minimum current at switch turn–on (t2) is increased
by ∆I + ∆I m2/m1. The minimum current at the next cycle
(t3) decreases to (∆I + ∆I m2/m1) (m2/m1). This pertubation
is multiplied by m2.m1 on each succeeding cycle, alternately
increasing and decreasing the inductor current at switch
turn–on. Several oscillator cycles may be required before
the inductor current reaches zero causing the process to
commence again. If m2/m1 is greater than 1, the converter
will be unstable. Figure 20B shows that by adding an
artificial ramp that is synchronized with the PWM clock to
the control voltage, the ∆I pertubation will decrease to zero
on succeeding cycles. This compensation ramp (m3) must
have a slope equal to or slightly greater than m2/2 for
stability. With m2/2 slope compensation, the average
inductor current follows the control voltage yielding true
current mode operation. The compensating ramp can be
derived from the oscillator and added to either the Voltage
Feedback or Current Sense inputs (Figure 33).
Figure 20. Continuous Current Waveforms
(A)
(B)
t0t1t2t3
t4t5t6
Control Voltage
∆Im1
m2
m3
m1 m2
Oscillator Period
Oscillator Period
Control Voltage
∆I
Inductor
Current ∆ I + ∆Im2
m1
m2
m1
∆I + ∆Im2
m1
Inductor
Current
UC3842A, UC3843A, UC2842A, UC2843A
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12
Figure 21. External Clock Synchronization Figure 22. External Duty Cycle Clamp and
Multi Unit Synchronization
Figure 23. Adjustable Reduction of Clamp Level Figure 24. Soft–Start Circuit
Figure 25. Adjustable Buffered Reduction of
Clamp Level with Soft–Start Figure 26. Current Sensing Power MOSFET
Virtually lossless current sensing can be achieved with the implementation of a
SENSEFET power switch. For proper operation during over current conditions, a
reduction of the Ipk(max) clamp level must be implemented. Refer to Figures 23 and 25.
The diode clamp is required if the Sync amplitude is large enough to
cause the bottom side of CT to go more than 300 mV below ground.
External
Sync
Input
47
5(9)
R
R
Bias
Osc
Vref
RT
8(14)
4(7)
2(3)
1(1)
0.01 CT
2R
R
EA
+
-
+
5(9)
R
R
Bias
Osc
8(14)
4(7)
2(3)
1(1)
2R
R
EA
+
-
+
7
5.0k
3
8
6
5
1
C
R
S
MC1455
2
RA
+
-
+
-
4
Q
5.0k
5.0k
RB
To
Additional
UCX84XA's
f = 1.44
(RA + 2RB)C Dmax = RB
RA + 2RB
5(9)
R
R
Bias
Osc
8(14)
4(7)
2(3)
1(1)
2R
R
EA
+
-
+
Q1
RS
3(5)
5(8)
1.0V
-
R
SQ
Comp/Latch
5.0Vref
VClamp
Vin
VCC
7(11)
6(10)
-
+
+
-
+
-+
7(12)
+
-
R1 R2
R2
VClamp =1.67
+ 1
+ 0.33 x 10 - 3I
pk(max) =VClamp
RS
Where: 0 ≤ VClamp ≤ 1.0 V
R2
R1
1.0mA
R1
R1 + R25(9)
R
R
Bias
Osc
8(14)
4(7)
2(3)
1(1)
2R
R
EA
+
-
+
1.0V
-
R
SQ
5.0Vref
-
+
+
-
+
C
tSoft-Start 3600C in µF
1.0M
1.0mA
5(9)
R
R
Bias
Osc
8(14)
4(7)
2(3)
1(1)
2R
R
EA
+
-
+
Q1
RS
3(5)
5(8)
1.0V
-
R
SQ
Comp/Latch
5.0Vref
VClamp
Vin
VCC
7(11)
6(10)
-
+
+
-
+
-+
7(12)
+
-
MPSA63
R1
R2
C
tSoftstart = - In 1 - VCR1 R2
C
R2
VClamp =1.67
+ 1
Ipk(max) =VClamp
RS Where: 0 ≤ VClamp ≤ 1.0 V
1.0mA
R1
3VClamp R1 + R2
RS
1/4 W
(5)
(8)
-
R
SQ
Comp/Latch
5.0Vref
Vin
VCC
(11)
(10)
-
+
+
-
+
-+
(12)
+
-
Power Ground
To Input Source
Return
VPin 5 =
If: SENSEFET = MTP10N10M
RS = 200
Then: Vpin 5 = 0.075 Ipk
SENSEFET
RS Ipk rDS(on)
M
G
D
S
K
Control CIrcuitry
Ground:
To Pin (9)
rDM(on) + RS
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13
Figure 27. Current Waveform Spike Suppression Figure 28. MOSFET Parasitic Oscillations
Figure 29. Bipolar Transistor Drive Figure 30. Isolated MOSFET Drive
Figure 31. Latched Shutdown Figure 32. Error Amplifier Compensation
The totem-pole output can furnish negative base current for enhanced
transistor turn-off, with the addition of capacitor C1.
Error Amp compensation circuit for stabilizing any current-mode topology except
for boost and flyback converters operating with continuous inductor current.
Error Amp compensation circuit for stabilizing current-mode boost and flyback
topologies operating with continuous inductor current.
The MCR101 SCR must be selected for a holding of less than 0.5 mA at TA(min).
The simple two transistor circuit can be used in place of the SCR as shown. All
resistors are 10 k.
Series gate resistor Rg will damp any high frequency parasitic oscillations
caused by the MOSFET input capacitance and any series wiring inductance
in the gate-source circuit.
The addition of the RC filter will eliminate instability caused by the leading
edge spike on the current waveform.
Q1
RS
3(5)
5(8)
-
R
SQ
Comp/Latch
5.0Vref
Vin
VCC
7(11)
6(10)
-
+
+
-
+
-+
7(12)
+
-
R
C
Q1
RS
3(5)
5(8)
-
R
SQ
Comp/Latch
5.0Vref
Vin
VCC
7(11)
6(10)
-
+
+
-
+
-+
7(12)
+
-
Rg
Q1
RS
3(5)
5(8)
Vin
6(1)
C1
IB
+
0
-
Base
Charge
Removal
ÉÉ
É
É
É
É
ÉÉ
Q1
3(5)
5(8)
-
R
SQ
Comp/Latch
5.0Vref
Vin
VCC
7(11)
6(1)
-
+
+
-
+
-+
7(12)
+
-
Np
R
CRSNS
Isolation
Boundary
VGS Waveforms
+
0
-
+
0
-
Ipk = V(pin 1) - 1.4
3 RS
NP
NS
50% DC 25% DC
5(9)
R
R
Bias
Osc
8(14)
4(7)
2(3)
1(1)
2R
R
EA
+
-
+
1.0mA
2N
3903
2N
3905
MCR
101
5(9)
2(3)
1(1)
2R
R
EA
+
-
+
1.0mA
CIRf
Ri
Rd
From VO
2.5V
5(9)
2(3)
1(1)
2R
R
EA
+
-
+
1.0mA
Cp
CIRf
From VO
Rp
Rd
Ri
2.5V
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14
Figure 33. Slope Compensation
Figure 34. 27 Watt Off–Line Flyback Regulator
The buffered oscillator ramp can be resistively summed with either the voltage feedback or current sense inputs to provide slope compensation.
Ri
Rd
-3.0 m
m
1.0V
Vin
VCC
RS
3(5)
5(8)
6(10)
7(11)
7(12)
+
-
+
-
5.0Vref
Bias
Osc
1.0mA
+
2R
R
R
R
R
SQ
CfRf
EA
1(1)
2(3)
4(7)
RT
8(14)
MPS3904
RSlope
From VO
5(9)
CT
Comp/Latch
-m
-
+
T1 - Primary: 45 Turns # 26 AWG
T1 - Secondary ± 12 V: 9 Turns # 30 AWG
T1 - (2 strands) Bifiliar Wound
T1 - Secondary 5.0 V: 4 Turns (six strands)
T1 - #26 Hexfiliar Wound
T1 - Secondary Feedback: 10 Turns #30 AWG
T1 - (2 strands) Bifiliar Wound
T1 - Core: Ferroxcube EC35-3C8
T1 - Bobbin: Ferroxcube EC35PCB1
T1 - Gap ≈ 0.01" for a primary inductance of 1.0 mH
L1 - 15 µH at 5.0 A, Coilcraft Z7156.
L2, L3 - 25 µH at 1.0 A, Coilcraft Z7157.
Comp/Latch
S
RQ
1N4935 1N4935
5.0Vref
Bias
Osc
++ 47
100
EA
+
+
7(12)
L1
5.0V/4.0A
2200 1000 +
MUR110
MBR1635
1000
1000 10
++
+L2
5.0V RTN
12V/0.3A
1N4937
L3
MUR110
±12V RTN
-12V/0.3A
T1
1.0k
470pF
3(5)
5(8)
6(10)
7(11)
22Ω
1N4937
2.7k
3300pF
4.7k
56k
250
+
115Va
c
4.7ΩMDA
202
68
5(9)
+
1(1)
2(3)
4(7)
10k
0.01
4700pF
18k
4.7k
MTP
4N50
8(14)
10
+
+
680pF
0.5Ω
150k
100pF
+
-
+
-
+
--
+
+
-
+
--
+
Test Conditions Results
Line Regulation: 5.0 V
± 12 V Vin = 95 Vac to 130 Vac ∆ = 50 mV or ± 0.5%
∆ = 24 mV or ± 0.1%
Load Regulation: 5.0 V
± 12 V Vin = 115 Vac, Iout = 1.0 A to 4.0 A
Vin = 115 Vac, Iout = 100 mA to 300 mA ∆ = 300 mV or ± 3.0%
∆ = 60 mV or ± 0.25%
Output Ripple: 5.0 V
± 12 V Vin = 115 Vac 40 mVpp
80 mVpp
Efficiency Vin = 115 Vac 70%
All outputs are at nominal load currents, unless otherwise noted.
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15
ORDERING INFORMATION
Device Operating
Temperature Range Package Shipping
UC3842AD SO–14 55 Units/Rail
UC3843AD SO–14 55 Units/Rail
UC3843AD2 SO–14 55 Units/Rail
UC3843AD2R2 SO–14 2500 Tape & Reel
UC3842ADR2 SO–14 2500 Tape & Reel
UC3843ADR2 TA = 0° to +70°CSO–14 2500 Tape & Reel
UC3842AN
A
PDIP–8 50 Units/Rail
UC3843AN PDIP–8 50 Units/Rail
UC3842AN1 PDIP–8 50 Units/Rail
UC3842AN2 PDIP–8 50 Units/Rail
UC3843AN2 PDIP–8 50 Units/Rail
UC2842AD SO–14 55 Units/Rail
UC2843AD SO–14 55 Units/Rail
UC2843AD2 SO–14 55 Units/Rail
UC2843AD2R2 SO–14 2500 Tape & Reel
UC2842ADR2 TA = –25° to +85°CSO–14 2500 Tape & Reel
UC2843ADR2
A
SO–14 2500 Tape & Reel
UC2842AN PDIP–8 50 Units/Rail
UC2843AN PDIP–8 50 Units/Rail
UC2843AN2 PDIP–8 50 Units/Rail
UC3842A, UC3843A, UC2842A, UC2843A
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16
x = 2 or 3
A = Assembly Location
WL = Wafer Lot
YY, Y = Year
WW = Work Week
SO–14
D SUFFIX
CASE 751A
MARKING DIAGRAMS
UC384xAD
AWLYWW
14
1
UC384xAN
FAWL
YYWW
UC3843AD2
AWLYWW
14
1
PDIP–8
N SUFFIX
CASE 626
UC284xAD
AWLYWW
14
1
UC284xAN
AWL
YYWW
UC2843AD2
AWLYWW
14
1
1
8
1
8
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17
PACKAGE DIMENSIONS
PDIP–8
N SUFFIX
CASE 626–05
ISSUE L
NOTES:
1. DIMENSION L TO CENTER OF LEAD WHEN
FORMED PARALLEL.
2. PACKAGE CONTOUR OPTIONAL (ROUND OR
SQUARE CORNERS).
3. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
14
58
F
NOTE 2 –A–
–B–
–T–
SEATING
PLANE
H
J
GDK
N
C
L
M
M
A
M
0.13 (0.005) B M
T
DIM MIN MAX MIN MAX
INCHESMILLIMETERS
A9.40 10.16 0.370 0.400
B6.10 6.60 0.240 0.260
C3.94 4.45 0.155 0.175
D0.38 0.51 0.015 0.020
F1.02 1.78 0.040 0.070
G2.54 BSC 0.100 BSC
H0.76 1.27 0.030 0.050
J0.20 0.30 0.008 0.012
K2.92 3.43 0.115 0.135
L7.62 BSC 0.300 BSC
M--- 10 --- 10
N0.76 1.01 0.030 0.040
SO–14
D SUFFIX
CASE 751A–03
ISSUE F
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.127 (0.005) TOTAL
IN EXCESS OF THE D DIMENSION AT
MAXIMUM MATERIAL CONDITION.
–A–
–B–
G
P7 PL
14 8
71 M
0.25 (0.010) B M
S
B
M
0.25 (0.010) A S
T
–T–
F
RX 45
SEATING
PLANE D14 PL K
C
J
M
DIM MIN MAX MIN MAX
INCHESMILLIMETERS
A8.55 8.75 0.337 0.344
B3.80 4.00 0.150 0.157
C1.35 1.75 0.054 0.068
D0.35 0.49 0.014 0.019
F0.40 1.25 0.016 0.049
G1.27 BSC 0.050 BSC
J0.19 0.25 0.008 0.009
K0.10 0.25 0.004 0.009
M0 7 0 7
P5.80 6.20 0.228 0.244
R0.25 0.50 0.010 0.019
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18
Notes
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19
Notes
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20
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