LM2743MTC/NOPB - Texas Instruments

BOOT

ISEN

PGND

VCC

PWGD

FREQ

SS/TRACK

SGND

EAO

PGND

VCC = 3.3V VIN = 3.3V

VOUT = 1.2V@4A

RFB2

CC2 RC1

RCS

CSS

RFADJ

RCC

CCC

D1CBOOT

Q1 CIN1,2

CC1 RFB1

CO1,2

LM2743

RC2 CC3

RPULL-UP +

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LM2743

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LM2743 Low Voltage N-Channel MOSFET Synchronous Buck Regulator Controller

1 Features 3 Description

The LM2743 is a high-speed synchronous buck

1• Power Stage Input Voltage from 1 V to 16 V regulator controller with an accurate feedback voltage

• Control Stage Input Voltage from 3 V to 6 V accuracy of ±2%. It can provide simple down

• Output Voltage Adjustable down to 0.6 V conversion to output voltages as low as 0.6V. Though

the control sections of the IC are rated for 3 to 6V,

• Power Good Flag and Shutdown the driver sections are designed to accept input

• Output Over-Voltage and Under-Voltage Detection supply rails as high as 16V. The use of adaptive non-

• ±2% Feedback Voltage Accuracy Over overlapping MOSFET gate drivers helps avoid

Temperature potential shoot-through problems while maintaining

high efficiency. The IC is designed for the more cost-

• Low-Side Adjustable Current Sensing effective option of driving only N-channel MOSFETs

• Adjustable Soft-Start in both the high-side and low-side positions. It senses

• Tracking and Sequencing with Shutdown and the low-side switch voltage drop for providing a

Soft-Start Pins simple, adjustable current limit.

• Switching Frequency from 50 kHz to 1 MHz The fixed-frequency voltage-mode PWM control

• TSSOP-14 Package architecture is adjustable from 50 kHz to 1 MHz with

one external resistor. This wide range of switching

2 Applications frequency gives the power supply designer the

flexibility to make better tradeoffs between

• 3.3V Buck Regulation component size, cost and efficiency.

• Cable Modem, DSL and ADSL Features include soft-start, input under-voltage

• Laser Jet and Ink Jet Printers lockout (UVLO) and Power Good (based on both

• Low Voltage Power Modules under-voltage and over-voltage detection). In

addition, the shutdown pin of the IC can be used for

• DSP, ASIC, Core, and I/O providing startup delay, and the soft-start pin can be

used for implementing precise tracking, for the

purpose of sequencing with respect to an external

rail.

Device Information(1)

PART NUMBER PACKAGE BODY SIZE (NOM)

LM2743 TSSOP (14) 4.40 mm × 5.00 mm

(1) For all available packages, see the orderable addendum at

the end of the data sheet.

Typical Application Diagram

An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Table of Contents

7.3 Feature Description................................................. 10

1 Features.................................................................. 17.4 Device Functional Modes........................................ 18

2 Applications ........................................................... 18 Application and Implementation ........................ 19

3 Description............................................................. 18.1 Application Information............................................ 19

4 Revision History..................................................... 28.2 Typical Applications ................................................ 19

5 Pin Configuration and Functions......................... 39 Power Supply Recommendations...................... 33

6 Specifications......................................................... 410 Layout................................................................... 33

6.1 Absolute Maximum Ratings ...................................... 410.1 Layout Guidelines ................................................. 33

6.2 ESD Ratings ............................................................ 410.2 Layout Example .................................................... 33

6.3 Recommended Operating Conditions....................... 411 Device and Documentation Support................. 34

6.4 Thermal Information.................................................. 411.1 Community Resources.......................................... 34

6.5 Electrical Characteristics........................................... 511.2 Trademarks........................................................... 34

6.6 Typical Characteristics.............................................. 711.3 Electrostatic Discharge Caution............................ 34

7 Detailed Description............................................ 10 11.4 Glossary................................................................ 34

7.1 Overview................................................................. 10 12 Mechanical, Packaging, and Orderable

7.2 Functional Block Diagram....................................... 10 Information........................................................... 34

4 Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Revision G (March 2013) to Revision H Page

• Added ESD Ratings table, Feature Description section, Device Functional Modes,Application and Implementation

section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and

Mechanical, Packaging, and Orderable Information section. ................................................................................................ 1

Changes from Revision F (March 2013) to Revision G Page

• Changed layout of National Data Sheet to TI format ........................................................................................................... 32

Product Folder Links: LM2743

LM2743

BOOT

ISEN

PGND

VCC

PWGD

FREQ

SS/TRACK

SGND

EAO

PGND

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

5 Pin Configuration and Functions

PW Package

14-Pin TSSOP

Top View

Pin Functions

PIN DESCRIPTION

NAME NO.

BOOT 1 Bootstrap pin. This is the supply rail for the gate drivers. When the high-side MOSFET turns on, the

voltage on this pin should be at least one gate threshold above the regulator input voltage VIN to

properly turn on the MOSFET. See MOSFET Gate Drivers for more details on how to select

MOSFETs.

LG 2 Low-gate drive pin. This is the gate drive for the low-side N-channel MOSFET. This signal is

interlocked with the high-side gate drive HG (Pin 14), so as to avoid shoot-through.

PGND 3 Power ground. This is also the ground for the low-side MOSFET driver. Both the pins must be

connected together on the PCB and form a ground plane, which is usually also the system ground.

SGND 4 Signal ground. It should be connected appropriately to the ground plane with due regard to good layout

practices in switching power regulator circuits.

VCC 5 Supply rail for the control sections of the IC.

PWGD 6 Power Good pin. This is an open drain output, which is typically meant to be connected to VCC or any

other low voltage source through a pull-up resistor. Choose the pull-up resistor so that the current

going into this pin is kept below 1 mA. For most applications a recommended value for the pull-up

resistor is 100 kΩ. The voltage on this pin is thus pulled low under output under-voltage or over-

voltage fault conditions and also under input UVLO.

ISEN 7 Current limit threshold setting pin. This sources a fixed 40 µA current. A resistor of appropriate value

should be connected between this pin and the drain of the low-side MOSFET (switch node).

EAO 8 Output of the error amplifier. The voltage level on this pin is compared with an internally generated

ramp signal to determine the duty cycle. This pin is necessary for compensating the control loop.

SS/TRACK 9 Soft-start and tracking pin. This pin is internally connected to the non-inverting input of the error

amplifier during soft-start, and in fact any time the SS/TRACK pin voltage happens to be below the

internal reference voltage. For the basic soft-start function, a capacitor of minimum value 1 nF is

connected from this pin to ground. To track the rising ramp of another power supply’s output, connect

a resistor divider from the output of that supply to this pin as described in Application and

Implementation.

FB 10 Feedback pin. This is the inverting input of the error amplifier, which is used for sensing the output

voltage and compensating the control loop.

FREQ 11 Frequency adjust pin. The switching frequency is set by connecting a resistor of suitable value

between this pin and ground. The equation for calculating the exact value is provided in Application

and Implementation, but some typical values (rounded up to the nearest standard values) are 324 kΩ

for 100 kHz, 97.6 kΩfor 300 kHz, 56.2 kΩfor 500 kHz, 24.9 kΩfor 1 MHz.

SD 12 IC shutdown pin. Pull this pin to VCC to ensure the IC is enabled. Connect to ground to disable the IC.

Under shutdown, both high-side and low-side drives are off. This pin also features a precision

threshold for power supply sequencing purposes, as well as a low threshold to ensure minimal

quiescent current.

HG 14 High-gate drive pin. This is the gate drive for the high-side N-channel MOSFET. This signal is

interlocked with LG (Pin 2) to avoid shoot-through.

Product Folder Links: LM2743

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6 Specifications

6.1 Absolute Maximum Ratings

MIN MAX UNIT

VCC –0.3 7 V

BOOT voltage –0.3 21 V

ISEN –0.3 9.5 V

All other pins –0.3 VCC + 0.3 V

TJJunction temperature 150 °C

Lead temperature (soldering, 10 s) 260 °C

Soldering information Infrared or convection (20 s) 235 °C

Tstg Storage temperature –65 150 °C

(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(2) If Military/Aerospace specified devices are required, please contact the Texas Instruments Sales Office/ Distributors for availability and

specifications.

6.2 ESD Ratings VALUE UNIT

V(ESD) Electrostatic discharge(1) Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(2) 2000 V

(1) The human body model is a 100 pF capacitor discharged through a 1.5-kΩresistor into each pin.

(2) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.

6.3 Recommended Operating Conditions MIN NOM MAX UNIT

VCC Supply voltage range 3 6 V

TJJunction temperature –40 125 °C

6.4 Thermal Information LM2743

THERMAL METRIC(1) TSSOP (PW) UNIT

14 PINS

RθJA Junction-to-ambient thermal resistance 107.9 °C/W

RθJC(top) Junction-to-case (top) thermal resistance 33.7 °C/W

RθJB Junction-to-board thermal resistance 50.7 °C/W

ψJT Junction-to-top characterization parameter 1.8 °C/W

ψJB Junction-to-board characterization parameter 50 °C/W

(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application

report, SPRA953.

Product Folder Links: LM2743

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

6.5 Electrical Characteristics

Typical limits are for TJ= 25°C only, represent the most likely parametric norm at TJ= 25°C, and are provided for reference

purposes only; minimum and maximum limits apply over the junction temperature range of –40°C to 125°C. Unless otherwise

specified, VCC = 3.3 V. Data sheet minimum and maximum specification limits are specified by design, test, or statistical

analysis. (See (1))PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

VFB FB Pin Voltage VCC = 3 V to 6 V 0.588 0.6 0.612 V

UVLO Thresholds Rising 2.76

VON V

Falling 2.42

VCC = 3.3 V, VSD = 3.3 V 1 1.5 2.1

Fsw = 600 kHz

Operating VCC Current mA

IQ_VCC VCC = 5V, VSD = 3.3 V 1 1.7 2.1

Fsw = 600 kHz

Shutdown VCC Current VCC = 3.3 V, VSD = 0 V 110 185 µA

tPWGD1 PWGD Pin Response Time VFB Rising 6 µs

tPWGD2 PWGD Pin Response Time VFB Falling 6 µs

ISS-ON SS Pin Source Current VSS = 0 V 7 10 14 µA

SS Pin Sink Current During Over VSS = 2.5 V

ISS-OC 90 µA

Current

ISEN-TH ISEN Pin Source Current Trip Point 25 40 55 µA

ERROR AMPLIFIER

Error Amplifier Unity Gain

GBW 9 MHz

Bandwidth

G Error Amplifier DC Gain 106 dB

SR Error Amplifier Slew Rate 3.2 V/µs

EAO Pin Current Sourcing and VEAO = 1.5, FB = 0.55 V 2.6

IEAO mA

Sinking Capability VEAO = 1.5, FB = 0.65 V 9.2

Error Amplifier Output Voltage Minimum 1 V

VEA Maximum 2 V

GATE DRIVE

IQ-BOOT BOOT Pin Quiescent Current VBOOT = 12 V, VSD = 0 18 90 µA

High-Side MOSFET Driver Pull-Up

RHG_UP VBOOT = 5 V at 350 mA Sourcing 3 Ω

ON resistance

High-Side MOSFET Driver Pull-

RHG_DN HG = 5 V at 350 mA Sourcing 2 Ω

Down ON resistance

Low-Side MOSFET Driver Pull-Up

RLG_UP VBOOT = 5 V at 350 mA Sourcing 3 Ω

ON resistance

Low-Side MOSFET Driver Pull-

RLG_DN LG = 5 V at 350 mA Sourcing 2 Ω

Down ON resistance

OSCILLATOR

RFADJ = 702.1 kΩ50

RFADJ = 98.74 kΩ300

fSW PWM Frequency kHz

RFADJ = 45.74 kΩ475 600 725

RFADJ = 24.91 kΩ1000

Max High-Side Duty Cycle fSW = 300 kHz 80%

D fSW = 600 kHz 76%

fSW = 1 MHz 73%

LOGIC INPUTS AND OUTPUTS

Standby High Trip Point VFB = 0.575 V, VBOOT = 3.3 V, VSD

VSTBY-IH 1.1 V

Rising

Standby Low Trip Point VFB = 0.575 V, VBOOT = 3.3 V, VSD

VSTBY-IL 0.232 V

Falling

VSD-IH SD Pin Logic High Trip Point VSD Rising 1.3 V

(1) The power MOSFETs can run on a separate 1-V to 16-V rail (Input voltage, VIN). Practical lower limit of VIN depends on selection of the

external MOSFET.

Product Folder Links: LM2743

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Electrical Characteristics (continued)

Typical limits are for TJ= 25°C only, represent the most likely parametric norm at TJ= 25°C, and are provided for reference

purposes only; minimum and maximum limits apply over the junction temperature range of –40°C to 125°C. Unless otherwise

specified, VCC = 3.3 V. Data sheet minimum and maximum specification limits are specified by design, test, or statistical

analysis. (See (1))PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

VSD-IL SD Pin Logic Low Trip Point VSD Falling 0.8 V

VPWGD-TH-LO PWGD Pin Trip Points FB Falling 0.408 0.434 0.457 V

VPWGD-TH-HI PWGD Pin Trip Points FB Rising 0.677 0.710 0.742 V

FB Falling 60

VPWGD-HYS PWGD Hysteresis mV

FB Rising 90

Product Folder Links: LM2743

TEMPERATURE (oC)

-40 -25 -10 5 80 95 110125

20 35 50 65

BOOT PIN CURRENT (mA)

5.9

6.1

6.2

6.3

6.4

6.5

6.6

6.7

TEMPERATURE (oC)

-40 -25 -10 5 80 95 110125

20 35 50 65

9.3

9.4

9.5

9.6

9.7

9.8

9.9

BOOT PIN CURRENT (mA)

EFFICIENCY (%)

100

VIN = 12V

VIN = 5V

0.0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0

OUTPUT CURRENT (A)

0.05 0.25 0.45 0.65 0.85

FREQUENCY (MHz)

1.05

VCC OPERATING CURRENT

PLUS BOOT CURRENT (mA)

0.0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0

OUTPUT CURRENT (A)

100

EFFICIENCY (%)

VIN = 3.3V

VIN = 12V

VIN = 5V

EFFICIENCY (%)

100 VIN = 3.3V

VIN = 12V

VIN = 5V

0.0 0.4 0.8 1.2 1.6 2.0 2.4 2.8 3.2 3.6 4.0

OUTPUT CURRENT (A)

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

6.6 Typical Characteristics

VCC = 3.3 V fSW = 300 kHz VCC = 3.3 V fSW = 300 kHz

Figure 1. Efficiency VOUT = 1.2 V Figure 2. Efficiency VOUT = 2.5 V

FDS6898A FET TA= 25°C

VCC = 5 V fSW = 300 kHz

Figure 4. VCC Operating Current Plus BOOT Current vs

Figure 3. Efficiency VOUT = 3.3 V Frequency

fSW = 300 kHz FDS6898A FET No-Load

Figure 6. BOOT Pin Current vs Temperature for BOOT

Figure 5. BOOT Pin Current vs Temperature for BOOT Voltage = 5 V

Voltage = 3.3 V

Product Folder Links: LM2743

2V/div

1V/div

5V/div

100 ns/DIV

2V/div

1V/div

5V/div

100 ns/DIV

0 0.4 0.8 1.2 1.6 2 2.4 2.8 3.2 3.6 4

1.200

1.201

1.202

1.203

1.204

1.205

1.206

1.207

1.208

1.209

1.210

OUTPUT VOLTAGE (V)

OUTPUT CURRENT (A)

TEMPERATURE (oC)

-55 -35 -15 85 105 125

5 25 45 65

FREQUENCY (kHz)

520

540

560

580

600

620

640

TEMPERATURE (oC)

-0.6

-0.5

-0.4

-0.3

-0.2

-0.1

0.1

0.2

INTERNAL REFERENCE

VOLTAGE ('%)

-40 -25 -10 5 80 95 110125

20 35 50 65

TEMPERATURE (oC)

-40 -25 -10 5 80 95 110125

20 35 50 65

BOOT PIN CURRENT (mA)

23.9

24.1

24.2

24.3

24.4

24.5

24.6

24.7

24.8

24.9

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

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Typical Characteristics (continued)

fSW = 300 kHz FDS6898A FET No-Load

Figure 7. BOOT Pin Current vs Temperature for BOOT Figure 8. Internal Reference Voltage vs Temperature

Voltage = 12 V

Figure 9. Frequency vs Temperature Figure 10. Output Voltage vs Output Current

VCC = 3.3 V VIN = 5 V VOUT = 1.2 V

VCC = 3.3 V VIN = 5 V VOUT = 1.2 V IOUT = 4 A CSS = 12 nF fSW = 300 kHz

IOUT = 4 A CSS = 12 nF fSW = 300 kHz

Figure 12. Switch Waveforms (HG Falling)

Figure 11. Switch Waveforms (HG Rising)

Product Folder Links: LM2743

100 mV/div

100 Ps/DIV

5V/div

VOUT

VIN

100mV/div

100 Ps/DIV

5V/div

VIN

VOUT

20mV/div

100 Ps/DIV

2A/div

VOUT

IOUT

20 mV/div

40 Ps/DIV

2A/div

VOUT

IOUT

50 mV/div

40 Ps/DIV

2A/div

VOUT

IOUT

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Typical Characteristics (continued)

IOUT = 0 A to 4 A VCC = 3.3 V VIN = 5 V

VCC = 3.3 V VIN = 5V VOUT = 1.2V VOUT = 1.2 V CSS = 12nF fSW = 300 kHz

CSS = 12 nF fSW = 300 kHz

Figure 14. Load Transient Response

Figure 13. Start-Up (No-Load)

IOUT = 4 A to 0 A VCC = 3.3 V VIN = 5 V VCC = 3.3 V VIN = 5 V VOUT = 1.2 V

VOUT = 1.2 V CSS = 12 nF fSW = 300 kHz CSS = 12 nF fSW = 300 kHz

Figure 15. Load Transient Response Figure 16. Load Transient Response

VIN = 3 V to 9 V VCC = 3.3 V VOUT = 1.2 V VIN = 9 V to 3 V VCC = 3.3 V VOUT = 1.2 V

Line () , IOUT = 2 A fSW = 300 kHz IOUT = 2 A fSW = 300 kHz

Figure 17. Line Transient Response Figure 18. Transient Response

Product Folder Links: LM2743

CSS = tSS

PWM

ILIM

40 PA

10 PA

SYNCHRONOUS

DRIVER LOGIC

OV UV

10 Ps

DELAY

0.71V 0.434V

SHUT DOWN

LOGIC

UVLO

SD VCC SGND

FB EAO

BOOT

ISEN

PWGD

SS/TRACK

PGND

90 PA

SSDONE

Soft-Start

Comparator

+ Logic +

VREF = 0.6V

REF

PWM LOGIC

CLOCK &

RAMP

FREQ PGND

1VPP

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

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7 Detailed Description

7.1 Overview

The LM2743 is a voltage-mode, high-speed synchronous buck regulator with a PWM control scheme. It has

output shutdown (SD), input under-voltage lock-out (UVLO) mode and power good (PWGD) flag (based on over-

voltage and under-voltage detection). The over-voltage and under-voltage signals are logically OR'ed to drive the

power good signal and provide a logic signal to the system if the output voltage goes out of regulation. Current

limit is achieved by sensing the voltage VDS across the low side MOSFET.

7.2 Functional Block Diagram

7.3 Feature Description

7.3.1 Start Up and Soft-Start

When VCC exceeds 2.76V and the shutdown pin (SD) sees a logic high, the soft-start period begins. Then an

internal, fixed 10 µA source begins charging the soft-start capacitor. During soft-start the voltage on the soft-start

capacitor CSS is connected internally to the non-inverting input of the error amplifier. The soft-start period lasts

until the voltage on the soft-start capacitor exceeds the LM2743 reference voltage of 0.6V. At this point the

reference voltage takes over at the non-inverting error amplifier input. The capacitance of CSS determines the

length of the soft-start period, and can be approximated by:

where

Product Folder Links: LM2743

0.65 = VOUT1 RT1

RT1 + RT2

Master Power

Supply

VOUT1 = 5V

SS/TRACK

LM2743

RT2

1 k:

RT1

150:

RFB2

10 k:

RFB1

5 k:

VOUT2 = 1.8V

VSS = 0.65V

VFB

RFADJ = -5.93 + 3.06 107

fSW + 0.24 1012

(fSW)2

RFB1

VOUT =RFB1 RFB2

+VFB

(VFB = 0.6V)

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Feature Description (continued)

• CSS is in µF and tSS is in ms (1)

During soft start the Power Good flag is forced low and it is released when the FB pin voltage reaches 70% of

0.6V. At this point the chip enters normal operation mode, and the output overvoltage and undervoltage

monitoring starts.

7.3.2 Normal Operation

While in normal operation mode, the LM2743 regulates the output voltage by controlling the duty cycle of the

high side and low side MOSFETs (see Typical Application Diagram). The equation governing output voltage is:

(2)

The PWM frequency is adjustable between 50 kHz and 1 MHz and is set by an external resistor, RFADJ, between

the FREQ pin and ground. The resistance needed for a desired frequency is approximately:

(3)

Where fSW is in Hz and RFADJ is in kΩ.

7.3.3 Tracking a Voltage Level

The LM2743 can track the output of a master power supply during soft-start by connecting a resistor divider to

the SS/TRACK pin. In this way, the output voltage slew rate of the LM2743 will be controlled by the master

supply for loads that require precise sequencing. When the tracking function is used no soft-start capacitor

should be connected to the SS/TRACK pin. Otherwise, a CSS value of at least 1 nF between the soft-start pin

and ground should be used.

Figure 19. Tracking Circuit

One way to use the tracking feature is to design the tracking resistor divider so that the master supply’s output

voltage (VOUT1) and the LM2743’s output voltage (represented symbolically in Figure 19 as VOUT2, that is, without

explicitly showing the power components) both rise together and reach their target values at the same time. For

this case, the equation governing the values of the tracking divider resistors RT1 and RT2 is:

(4)

Product Folder Links: LM2743

1.8V

VOUT1

VOUT2

VOUT2 =VOUT1 RT1

RT1 + RT2

1.8V

VOUT1

VOUT2

LM2743

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Feature Description (continued)

The current through RT1 should be about 3 mA to 4 mA for precise tracking. The final voltage of the SS/TRACK

pin should be set higher than the feedback voltage of 0.6 V (say about 0.65 V as in the above equation). If the

master supply voltage was 5 V and the LM2743 output voltage was 1.8 V, for example, then the value of RT1

needed to give the two supplies identical soft-start times would be 150 Ω. A timing diagram for the equal soft-

start time case is shown in Figure 20.

Figure 20. Tracking with Equal Soft-Start Time

7.3.4 Tracking Voltage Slew Rate

The tracking feature can alternatively be used not to make both rails reach regulation at the same time but rather

to have similar rise rates (in terms of output dV/dt). This method ensures that the output voltage of the LM2743

always reaches regulation before the output voltage of the master supply. Because the output of the master

supply is divided down, in order to track properly the output voltage of the LM2743 must be lower than the

voltage of the master supply. In this case, the tracking resistors can be determined based on the following

equation:

(5)

For the example case of VOUT1 =5VandVOUT2 = 1.8 V, with RT1 set to 150 Ωas before, RT2 is calculated from

the above equation to be 265 Ω. A timing diagram for the case of equal slew rates is shown in Figure 21.

Figure 21. Tracking with Equal Slew Rates

7.3.5 Sequencing

The start up/soft-start of the LM2743 can be delayed for the purpose of sequencing by connecting a resistor

divider from the output of a master power supply to the SD pin, as shown in Figure 22.

Product Folder Links: LM2743

t = 5 ms

VOUT1

VOUT2

1.8V

VSD-IH

1.08V

RS1

SRSD = SROUT1 RS1 + RS2

Master Power

Supply

VOUT1

LM2743

RS1 RFB2

RFB1

VOUT2

VFB

RS2

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Feature Description (continued)

Figure 22. Sequencing Circuit

A desired delay time tDELAY between the startup of the master supply output voltage and the LM2743 output

voltage can be set based on the SD pin low-to-high threshold VSD-IH and the slew rate of the voltage at the SD

pin, SRSD:

tDELAY = VSD-IH / SRSD (6)

Note again, that in Figure 22, the output voltage of the LM2743 has been represented symbolically as VOUT2,.

without explicitly showing the power components.

VSD-IH is typically 1.08V and SRSD is the slew rate of the SD pin voltage. The values of the sequencing divider

resistors RS1 and RS2 set the SRSD based on the master supply output voltage slew rate, SROUT1, using the

following equation:

(7)

For example, if the master supply output voltage slew rate was 1V/ms and the desired delay time between the

startup of the master supply and LM2743 output voltage was 5ms, then the desired SD pin slew rate would be

(1.08V/5 ms) = 0.216 V/ms. Due to the internal impedance of the SD pin, the maximum recommended value for

RS2 is 1 kΩ. To achieve the desired slew rate, RS1 would then be 274 Ω. A timing diagram for this example is

shown in Figure 23.

Figure 23. Delay for Sequencing

Product Folder Links: LM2743

8 PA

17 PA

Bias Enable

Soft-Start Enable

1.25V

10k

LM2743

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Feature Description (continued)

7.3.6 SD Pin Impedance

When connecting a resistor divider to the SD pin of the LM2743 some care has to be taken. Once the SD voltage

goes above VSD-IH, a 17-µA pull-up current is activated as shown in Figure 24. This current is used to create the

internal hysteresis (≊170 mV); however, high external impedances will affect the SD pin logic thresholds as well.

The external impedance used for the sequencing divider network should preferably be a small fraction of the

impedance of the SD pin for good performance (around 1 kΩ).

Figure 24. SD Pin Logic

7.3.7 MOSFET Gate Drivers

The LM2743 has two gate drivers designed for driving N-channel MOSFETs in a synchronous mode. Note that

unlike most other synchronous controllers, the bootstrap capacitor of the LM2743 provides power not only to the

driver of the upper MOSFET, but the lower MOSFET driver too (both drivers are ground referenced, i.e. no

floating driver). To fully turn the top MOSFET on, the BOOT voltage must be at least one gate threshold greater

than VIN when the high-side drive goes high. This bootstrap voltage is usually supplied from a local charge pump

structure. But looking at the Typical Application schematic, this also means that the difference voltage VCC - VD1,

which is the voltage the bootstrap capacitor charges up to, must be always greater than the maximum tolerance

limit of the threshold voltage of the upper MOSFET. Here VD1 is the forward voltage drop across the bootstrap

diode D1. This therefore may place restrictions on the minimum input voltage and/or type of MOSFET used.

The most basic charge bootstrap pump circuit can be built using one Schottky diode and a small capacitor, as

shown in Figure 25. The capacitor CBOOT serves to maintain enough voltage between the top MOSFET gate and

source to control the device even when the top MOSFET is on and its source has risen up to the input voltage

level. The charge pump circuitry is fed from VCC, which can operate over a range from 3.0V to 6.0V. Using this

basic method the voltage applied to the gates of both high-side and low-side MOSFETs is VCC - VD. This method

works well when VCC is 5 V±10%, because the gate drives will get at least 4.0V of drive voltage during the worst

case of VCC-MIN = 4.5 V and VD-MAX = 0.5 V. Logic level MOSFETs generally specify their on-resistance at VGS =

4.5 V. When VCC = 3.3 V ±10%, the gate drive at worst case could go as low as 2.5 V. Logic level MOSFETs are

not specified to turn on, or may have much higher on-resistance at 2.5 V. Sub-logic level MOSFETs, usually

specified at VGS = 2.5 V, will work, but are more expensive, and tend to have higher on-resistance. The circuit in

Figure 25 works well for input voltages ranging from 1 V up to 16 V and VCC = 5 V ±10%, because the drive

voltage depends only on VCC.

Product Folder Links: LM2743

BOOT

VIN

CBOOT

LM2743

LM78L05

VCC

BOOT

VIN

VCC

LM2743

CBOOT

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Feature Description (continued)

Figure 25. Basic Charge Pump (Bootstrap)

Note that the LM2743 can be paired with a low cost linear regulator like the LM78L05 to run from a single input

rail between 6.0 and 14 V. The 5-V output of the linear regulator powers both the VCC and the bootstrap circuit,

providing efficient drive for logic level MOSFETs. An example of this circuit is shown in Figure 26.

Figure 26. LM78L05 Feeding Basic Charge Pump

Figure 27 shows a second possibility for bootstrapping the MOSFET drives using a doubler. This circuit provides

an equal voltage drive of VCC - 3VD+ VIN to both the high-side and low-side MOSFET drives. This method should

only be used in circuits that use 3.3 V for both VCC and VIN. Even with VIN = VCC = 3.0 V (10% lower tolerance on

3.3 V) and VD= 0.5 V both high-side and low-side gates will have at least 4.5 V of drive. The power dissipation of

the gate drive circuitry is directly proportional to gate drive voltage, hence the thermal limits of the LM2743 IC will

quickly be reached if this circuit is used with VCC or VIN voltages over 5V.

Product Folder Links: LM2743

BOOT

VIN

VCC

LM2743

D2D3

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

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Feature Description (continued)

Figure 27. Charge Pump with Added Gate Drive

All the gate drive circuits shown in Figure 25 through Figure 27 typically use 100-nF ceramic capacitors in the

bootstrap locations.

7.3.8 Power Good Signal

The open drain output on the Power Good pin needs a pull-up resistor to a low voltage source. The pull-up

resistor should be chosen so that the current going into the Power Good pin is less than 1 mA. A 100-kΩresistor

is recommended for most applications.

The Power Good signal is an OR-gated flag which takes into account both output over-voltage and under-voltage

conditions. If the feedback pin (FB) voltage is 18% above its nominal value (118% x VFB = 0.708V) or falls 28%

below that value (72 %x VFB = 0.42V) the Power Good flag goes low. The Power Good flag can be used to signal

other circuits that the output voltage has fallen out of regulation, however the switching of the LM2743 continues

regardless of the state of the Power Good signal. The Power Good flag will return to logic high whenever the

feedback pin voltage is between 72% and 118% of 0.6V.

7.3.9 UVLO

The 2.76V turn-on threshold on VCC has a built in hysteresis of about 300 mV. If VCC drops below 2.42V, the chip

enters UVLO mode. UVLO consists of turning off the top and bottom MOSFETS and remaining in that condition

until VCC rises above 2.76V. As with shutdown, the soft-start capacitor is discharged through an internal

MOSFET, ensuring that the next start-up will be controlled by the soft-start circuitry.

7.3.10 Current Limit

Current limit is realized by sensing the voltage across the low-side MOSFET while it is on. The RDS(ON) of the

MOSFET is a known value; hence the current through the MOSFET can be determined as:

VDS = IOUT x RDS(ON) (8)

The current through the low-side MOSFET while it is on is also the falling portion of the inductor current. The

current limit threshold is determined by an external resistor, RCS, connected between the switching node and the

ISEN pin. A constant current of 40 µA is forced through RCS, causing a fixed voltage drop. This fixed voltage is

compared against VDS and if the latter is higher, the current limit of the chip has been reached. To obtain a more

accurate value for RCS you must consider the operating values of RDS(ON) and ISEN-TH at their operating

temperatures in your application and the effect of slight parameter differences from part to part. RCS can be found

by using the following equation using the RDS(ON) value of the low side MOSFET at it's expected hot temperature

and the absolute minimum value expected over the full temperature range for the for the ISEN-TH which is 25 µA:

RCS = RDSON-HOT x ILIM / 40 µA (9)

Product Folder Links: LM2743

IPK-CL = ILIM + (TSW - 200 ns) VIN - VO

ILIM

Normal Operation Current Limit

VIN ± 9.5V

RCS 10 mA

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Feature Description (continued)

For example, a conservative 15-A current limit in a 10-A design with a minimum RDS(ON) of 10 mΩwould require

a 6-kΩresistor. To prevent the ISEN pin from sinking too much current when the switch node goes above 9.5 V,

the value of the current limit setting resistor RCS should not be too low. The criterion is as follows,

(10)

where the 10 mA is the maximum current ISEN pin is allowed to sink. For example if VIN = 13.2 V, the minimum

value of RCS is 370 Ω. Because current sensing is done across the low-side MOSFET, no minimum high-side on-

time is necessary. The LM2743 enters current limit mode if the inductor current exceeds the current limit

threshold at the point where the high-side MOSFET turns off and the low-side MOSFET turns on. (The point of

peak inductor current, see Figure 28). Note that in normal operation mode the high-side MOSFET always turns

on at the beginning of a clock cycle. In current limit mode, by contrast, the high-side MOSFET on-pulse is

skipped. This causes inductor current to fall. Unlike a normal operation switching cycle, however, in a current

limit mode switching cycle the high-side MOSFET will turn on as soon as inductor current has fallen to the

current limit threshold. The LM2743 will continue to skip high-side MOSFET pulses until the inductor current peak

is below the current limit threshold, at which point the system resumes normal operation.

Figure 28. Current Limit Threshold

Unlike a high-side MOSFET current sensing scheme, which limits the peaks of inductor current, low-side current

sensing is only allowed to limit the current during the converter off-time, when inductor current is falling.

Therefore in a typical current limit plot the valleys are normally well defined, but the peaks are variable, according

to the duty cycle. The PWM error amplifier and comparator control the off-pulse of the high-side MOSFET, even

during current limit mode, meaning that peak inductor current can exceed the current limit threshold. Assuming

that the output inductor does not saturate, the maximum peak inductor current during current limit mode can be

calculated with the following equation:

(11)

Where TSW is the inverse of switching frequency fSW. The 200 ns term represents the minimum off-time of the

duty cycle, which ensures enough time for correct operation of the current sensing circuitry.

Product Folder Links: LM2743

(ILIM x RDS(ON)) ± (ISEN x RCS)

RCLF = RCS x VOUT

VIN ± 9.5V

RCS 10 mA

PLIM x RDS(ON) = RCS

ISEN

RCLF

VOUT

RCS SW

40 PA

ILIM ISEN

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Feature Description (continued)

In order to minimize the time period in which peak inductor current exceeds the current limit threshold, the IC

also discharges the soft-start capacitor through a fixed 90-µA sink. The output of the LM2743 internal error

amplifier is limited by the voltage on the soft-start capacitor. Hence, discharging the soft-start capacitor reduces

the maximum duty cycle D of the controller. During severe current limit this reduction in duty cycle will reduce the

output voltage if the current limit conditions last for an extended time. Output inductor current will be reduced in

turn to a flat level equal to the current limit threshold. The third benefit of the soft-start capacitor discharge is a

smooth, controlled ramp of output voltage when the current limit condition is cleared.

7.3.11 Foldback Current Limit

In the case where extra protection is used to help an output short condition, a current foldback resistor (RCLF)

should be considered, see Figure 29. First select the percentage of current limit foldback (PLIM):

PLIM = ILIM x P (12)

where P is a ratio between 0 and 1.

Figure 29. Foldback Current Limit Circuit

Obtain the RCS with the following equation:

(13)

where ISEN = 40 μA. If the switch node goes above 9.5 V the following criterion must be satisfied:

(14)

The equation for calculating the foldback resistance value is:

(15)

7.4 Device Functional Modes

7.4.1 Shutdown

If the shutdown pin is pulled low, (below 0.8 V) the LM2743 enters shutdown mode, and discharges the soft-start

capacitor through a MOSFET switch. The high and low-side MOSFETs are turned off. The LM2743 remains in

this state as long as VSD sees a logic low (see the Electrical Characteristics table). To assure proper IC start-up

the shutdown pin should not be left floating. For normal operation this pin should be connected directly to VCC or

to another voltage between 1.3 V to VCC (see the Electrical Characteristics table).

Product Folder Links: LM2743

D =

VOUT + VSWL

VIN - VSWH + VSWL

BOOT

ISEN

PGND

VCC

PWGD

FREQ

SS/TRACK

SGND

EAO

PGND

VCC = 3.3V VIN = 3.3V

VOUT = 1.2V@4A

RFB2

CC2 RC1

RCS

CSS

RFADJ

RCC

CCC

D1CBOOT

Q1 CIN1,2

CC1 RFB1

CO1,2

LM2743

RC2 CC3

RPULL-UP +

LM2743

www.ti.com

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

8 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component

specification, and TI does not warrant its accuracy or completeness. TI’s customers are

responsible for determining suitability of components for their purposes. Customers should

validate and test their design implementation to confirm system functionality.

8.1 Application Information

The LM2743 is a voltage-mode, high-speed synchronous buck regulator with a PWM control scheme. It is

designed for use in set-top boxes, thin clients, DSL/Cable modems, and other applications that require high-

efficiency buck converters. Use the following design procedure to select component values for the LM2743

device. Use the WEBENCH™ to generate a complete design.

8.2 Typical Applications

8.2.1 Synchronous Buck Converter Typical Application using LM2743

Figure 30. 3.3 V to 1.2 V at 4 A, fSW = 300 kHz

8.2.1.1 Design Requirements

The following section provides a step-by-step design guide of a voltage-mode synchronous buck converter using

the LM2743. This design converts 3.3 V (VIN) to 1.2 V (VOUT) at a maximum load of 4 A, with an efficiency of

89% and a switching frequency of 300 kHz. The same procedures can be followed to create many other designs

with varying input voltages, output voltages, and load currents.

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 Duty Cycle Calculation

The complete duty cycle for a buck converter is defined with the following equation:

(16)

where VSWL and VSWH are the respective forward voltage drops that develop across the low side and high side

MOSFETs. Assuming the inductor ripple current is 20% to 30% of the output current, therefore:

VSWL = IOUT x RDS(ON)LOW (Low-Side MOSFET) (17)

VSWH = IOUT x RDS(ON)HIGH (High-Side MOSFET) (18)

Product Folder Links: LM2743

L = VIN - VOUT

'IOUT x fSW x D

PCAP = (IRMS_RIP)2 x ESR

IRMS_RIP = IOUT x D(1 - D)

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Typical Applications (continued)

To calculate the maximum duty cycle use the estimated 'hot' RDS(on) value of the MOSFETs, the minimum input

voltage, and maximum load. As shown in Figure 31, the worst case maximum duty cycles of the LM2743 occurs

at 125°C junction temperature vs VCC (IC control section voltage). Ensure that the operating duty cycle is below

the curve in Figure 31, if this condition is not satisfied, the system will be unable to develop the required duty

cycle to derive the necessary system power and so the output voltage will fall out of regulation.

Figure 31. Maximum Duty Cycle vs VCC

TJ= 125°C

8.2.1.2.2 Input Capacitor

The input capacitors in a Buck converter are subjected to high stress due to the input current trapezoidal

waveform. Input capacitors are selected for their ripple current capability and their ability to withstand the heat

generated since that ripple current passes through their ESR. Input rms ripple current is approximately:

(19)

The power dissipated by each input capacitor is:

(20)

where n is the number of capacitors, and ESR is the equivalent series resistance of each capacitor. The equation

above indicates that power loss in each capacitor decreases rapidly as the number of input capacitors increases.

The worst-case ripple for a Buck converter occurs during full load and when the duty cycle (D) is 0.5. For this

3.3V to 1.2V design the duty cycle is 0.364. For a 4A maximum load the ripple current is 1.92A.

8.2.1.2.3 Output Inductor

The output inductor forms the first half of the power stage in a Buck converter. It is responsible for smoothing the

square wave created by the switching action and for controlling the output current ripple (ΔIOUT). The inductance

is chosen by selecting between tradeoffs in efficiency and response time. The smaller the output inductor, the

more quickly the converter can respond to transients in the load current. However, as shown in the efficiency

calculations, a smaller inductor requires a higher switching frequency to maintain the same level of output current

ripple. An increase in frequency can mean increasing loss in the MOSFETs due to the charging and discharging

of the gates. Generally the switching frequency is chosen so that conduction loss outweighs switching loss. The

equation for output inductor selection is:

(21)

Product Folder Links: LM2743

ESRMAX = 'VOUT

'IOUT

VIN(MAX) - VO

'IOUT = FSW x LACTUAL x D

L = 3.3V - 1.2V

0.4 x 4A x 300 kHz x1.2V

3.3V

LM2743

www.ti.com

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Typical Applications (continued)

(22)

L = 1.6 µH (23)

Here we have plugged in the values for output current ripple, input voltage, output voltage, switching frequency,

and assumed a 40% peak-to-peak output current ripple. This yields an inductance of 1.6 µH. The output inductor

must be rated to handle the peak current (also equal to the peak switch current), which is (IOUT + (0.5 x ΔIOUT)) =

4.8 A, for a 4 A design. The Coilcraft DO3316P-222P is 2.2 µH, is rated to 7.4-A peak, and has a direct current

resistance (DCR) of 12 mΩ.

After selecting an output inductor, inductor current ripple should be re-calculated with the new inductance value,

as this information is needed to select the output capacitor. Re-arranging the equation used to select inductance

yields the following:

(24)

VIN(MAX) is assumed to be 10% above the steady state input voltage, or 3.6V. The actual current ripple will then

be 1.2A. Peak inductor/switch current will be 4.6A.

8.2.1.2.4 Output Capacitor

The output capacitor forms the second half of the power stage of a Buck switching converter. It is used to control

the output voltage ripple (ΔVOUT) and to supply load current during fast load transients.

In this example the output current is 4 A and the expected type of capacitor is an aluminum electrolytic, as with

the input capacitors. Other possibilities include ceramic, tantalum, and solid electrolyte capacitors, however the

ceramic type often do not have the large capacitance needed to supply current for load transients, and tantalums

tend to be more expensive than aluminum electrolytic. Aluminum capacitors tend to have very high capacitance

and fairly low ESR, meaning that the ESR zero, which affects system stability, will be much lower than the

switching frequency. The large capacitance means that at the switching frequency, the ESR is dominant, hence

the type and number of output capacitors is selected on the basis of ESR. One simple formula to find the

maximum ESR based on the desired output voltage ripple, ΔVOUT and the designed output current ripple, ΔIOUT,

is:

(25)

In this example, in order to maintain a 2% peak-to-peak output voltage ripple and a 40% peak-to-peak inductor

current ripple, the required maximum ESR is 20 mΩ. The Sanyo 4SP560M electrolytic capacitor will give an

equivalent ESR of 14 mΩ. The capacitance of 560 µF is enough to supply energy even to meet severe load

transient demands.

8.2.1.2.5 MOSFETs

Selection of the power MOSFETs is governed by a tradeoff between cost, size, and efficiency. One method is to

determine the maximum cost that can be endured, and then select the most efficient device that fits that price.

Breaking down the losses in the high-side and low-side MOSFETs and then creating spreadsheets is one way to

determine relative efficiencies between different MOSFETs. Good correlation between the prediction and the

bench result is not specified, however. Single-channel buck regulators that use a controller IC and discrete

MOSFETs tend to be most efficient for output currents of 2A to 10A.

Losses in the high-side MOSFET can be broken down into conduction loss, gate charging loss, and switching

loss. Conduction loss, or I2R loss, is approximately:

PC= D ((IO)2x RDSON-HI x 1.3) (High-Side MOSFET) (26)

PC= (1 - D) x ((IO)2x RDSON-LO x 1.3) (Low-Side MOSFET) (27)

Product Folder Links: LM2743

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Typical Applications (continued)

In the above equations, the factor 1.3 accounts for the increase in MOSFET RDSON due to heating. Alternatively,

the 1.3 can be ignored and the RDSON of the MOSFET estimated using the RDSON Vs. Temperature curves in the

MOSFET datasheets.

Gate charging loss results from the current driving the gate capacitance of the power MOSFETs, and is

approximated as:

PGC = n x (VDD) x QGx fSW (28)

where ‘n’ is the number of MOSFETs (if multiple devices have been placed in parallel), VDD is the driving voltage

(see MOSFET Gate Drivers section) and QGS is the gate charge of the MOSFET. If different types of MOSFETs

are used, the nterm can be ignored and their gate charges simply summed to form a cumulative QG. Gate

charge loss differs from conduction and switching losses in that the actual dissipation occurs in the LM2743, and

not in the MOSFET itself.

Switching loss occurs during the brief transition period as the high-side MOSFET turns on and off, during which

both current and voltage are present in the channel of the MOSFET. It can be approximated as:

PSW = 0.5 x VIN x IOx (tr+ tf) x fSW (29)

where tRand tFare the rise and fall times of the MOSFET. Switching loss occurs in the high-side MOSFET only.

For this example, the maximum drain-to-source voltage applied to either MOSFET is 3.6V. The maximum drive

voltage at the gate of the high-side MOSFET is 3.1V, and the maximum drive voltage for the low-side MOSFET

is 3.3V. Due to the low drive voltages in this example, a MOSFET that turns on fully with 3.1V of gate drive is

needed. For designs of 5A and under, dual MOSFETs in SOIC-8 package provide a good trade-off between size,

cost, and efficiency.

8.2.1.2.6 Support Components

CIN2 - A small value (0.1-µF to 1-µF) ceramic capacitor should be placed as close as possible to the drain of the

high-side MOSFET and source of the low-side MOSFET (dual MOSFETs make this easy). This capacitor should

be X5R type dielectric or better.

RCC, CCC- These are standard filter components designed to ensure smooth DC voltage for the chip supply. RCC

should be 1 Ωto 10 Ω. CCC should 1 µF, X5R type or better.

CBOOT- Bootstrap capacitor, typically 100 nF.

RPULL-UP – This is a standard pull-up resistor for the open-drain power good signal (PWGD). The recommended

value is 10 kΩconnected to VCC. If this feature is not necessary, the resistor can be omitted.

D1- A small Schottky diode should be used for the bootstrap. It allows for a minimum drop for both high and low-

side drivers. The MBR0520 or BAT54 work well in most designs.

RCS - Resistor used to set the current limit. Since the design calls for a peak current magnitude (IOUT+ (0.5 x

ΔIOUT)) of 4.8 A, a safe setting would be 6A. (This is below the saturation current of the output inductor, which is

7 A.) Following the equation from the Current Limit section, a 1.3-kΩresistor should be used.

RFADJ - This resistor is used to set the switching frequency of the chip. The resistor value is calculated from

equation in Normal Operation section. For 300-kHz operation, a 97.6-kΩresistor should be used.

CSS - The soft-start capacitor depends on the user requirements and is calculated based on the equation given in

the section titled Start Up and Soft-Start. Therefore, for a 700-μs delay, a 12-nF capacitor is suitable.

8.2.1.2.7 Control Loop Compensation

The LM2743 uses voltage-mode (‘VM’) PWM control to correct changes in output voltage due to line and load

transients. One of the attractive advantages of voltage mode control is its relative immunity to noise and layout.

However VM requires careful small signal compensation of the control loop for achieving high bandwidth and

good phase margin.

Product Folder Links: LM2743

VIN x RO

GPS = VRAMP

sCORC + 1

a x s2 + b x s + c

fESR = 2SCOESR = 20.3 kHz

= 4.5 kHz

RO + RL

fDP = 1

2SLCO(RO + ESR)

VIN

ADC = VRAMP =3.3

1.0 = 10.4 dB

VRAMP

VREF

10 k:

CC1

CC2 RC1

RC2

CC3

LRL

VIN

LM2743

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SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Typical Applications (continued)

The control loop is comprised of two parts. The first is the power stage, which consists of the duty cycle

modulator, output inductor, output capacitor, and load. The second part is the error amplifier, which for the

LM2743 is a 9-MHz op-amp used in the classic inverting configuration. Figure 32 shows the regulator and control

loop components.

Figure 32. Power Stage and Error Amplifier

One popular method for selecting the compensation components is to create Bode plots of gain and phase for

the power stage and error amplifier. Combined, they make the overall bandwidth and phase margin of the

regulator easy to see. Software tools such as Excel, MathCAD, and Matlab are useful for showing how changes

in compensation or the power stage affect system gain and phase.

The power stage modulator provides a DC gain ADC that is equal to the input voltage divided by the peak-to-peak

value of the PWM ramp. This ramp is 1.0VP-P for the LM2743. The inductor and output capacitor create a

double pole at frequency fDP, and the capacitor ESR and capacitance create a single zero at frequency fESR. For

this example, with VIN = 3.3 V, these quantities are:

(30)

(31)

(32)

In the equation for fDP, the variable RLis the power stage resistance, and represents the inductor DCR plus the

on resistance of the top power MOSFET. ROis the output voltage divided by output current. The power stage

transfer function GPS is given by the following equation, and Figure 34 shows Bode plots of the phase and gain in

this example.

where

• a = LCO(RO+ RC)

• b = L + CO(RORL+ RORC+ RCRL)

• c = RO+ RL(33)

Product Folder Links: LM2743

GEA = AEA x

2SfZ1 + 1 s

2SfZ2 + 1

2SfP1 + 1 s

2SfP2 + 1

100 1k 10k 100k 1M

FREQUENCY (Hz)

-60

-44

-28

-12

GAIN (dB)

100 1k 10k 100k 1M

FREQUENCY (Hz)

-150

-120

-90

-60

-30

PHASE (o)

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Typical Applications (continued)

Figure 33. Gain vs Frequency Figure 34. Power Stage Gain and Phase

The double pole at 4.5 kHz causes the phase to drop to approximately -130° at around 10 kHz. The ESR zero, at

20.3 kHz, provides a +90° boost that prevents the phase from dropping to -180º. If this loop were left

uncompensated, the bandwidth would be approximately 10 kHz and the phase margin 53°. In theory, the loop

would be stable, but would suffer from poor DC regulation (due to the low DC gain) and would be slow to

respond to load transients (due to the low bandwidth.) In practice, the loop could easily become unstable due to

tolerances in the output inductor, capacitor, or changes in output current, or input voltage. Therefore, the loop is

compensated using the error amplifier and a few passive components.

For this example, a Type III, or three-pole-two-zero approach gives optimal bandwidth and phase.

In most voltage mode compensation schemes, including Type III, a single pole is placed at the origin to boost DC

gain as high as possible. Two zeroes fZ1 and fZ2 are placed at the double pole frequency to cancel the double

pole phase lag. Then, a pole, fP1 is placed at the frequency of the ESR zero. A final pole fP2 is placed at one-half

of the switching frequency. The gain of the error amplifier transfer function is selected to give the best bandwidth

possible without violating the Nyquist stability criteria. In practice, a good crossover point is one-fifth of the

switching frequency, or 60 kHz for this example. The generic equation for the error amplifier transfer function is:

(34)

In this equation, the variable AEA is a ratio of the values of the capacitance and resistance of the compensation

components, arranged as shown in Figure 32. AEA is selected to provide the desired bandwidth. A starting value

of 80,000 for AEA should give a conservative bandwidth. Increasing the value will increase the bandwidth, but will

also decrease phase margin. Designs with 45° to 60° are usually best because they represent a good trade-off

between bandwidth and phase margin. In general, phase margin is lowest and gain highest (worst-case) for

maximum input voltage and minimum output current. One method to select AEA is to use an iterative process

beginning with these worst-case conditions.

1. Increase AEA

2. Check overall bandwidth and phase margin

3. Change VIN to minimum and recheck overall bandwidth and phase margin

4. Change IOto maximum and recheck overall bandwidth and phase margin

The process ends when the both bandwidth and the phase margin are sufficiently high. For this example input

voltage can vary from 3.0 to 3.6 V and output current can vary from 0 to 4 A, and after a few iterations a

moderate gain factor of 101 dB is used.

The error amplifier of the LM2743 has a unity-gain bandwidth of 9 MHz. In order to model the effect of this

limitation, the open-loop gain can be calculated as:

Product Folder Links: LM2743

RC2 = 2SxCC3 x fP1 = 2.55 k:

RC1 = 2SxCC2 x fZ1 = 39.8 k:

CC3 = 2S x 10,000 x= 2.73 nF

fZ2 -1

fP1

CC2 =AEA x 10,000 - CC1 = 882 pF

fZ1

CC1 = AEA x 10,000 x fP2 = 27 pF

100 1k 10k 100k 1M

FREQUENCY (Hz)

-100

-70

-40

-10

PHASE (o)

100 1k 10k 100k 1M

FREQUENCY (Hz)

GAIN (dB)

GEA x OPG

HEA = 1 + GEA + OPG

2S x 9 MHz

OPG = s

LM2743

www.ti.com

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Typical Applications (continued)

(35)

The new error amplifier transfer function that takes into account unity-gain bandwidth is:

(36)

The gain and phase of the error amplifier are shown in Figure 36.

Figure 35. Gain vs Frequency Figure 36. Error Amplifier Gain and Phase

In VM regulators, the top feedback resistor RFB2 forms a part of the compensation. Setting RFB2 to 10 kΩ, ±1%

usually gives values for the other compensation resistors and capacitors that fall within a reasonable range.

(Capacitances > 1 pF, resistances < 1 MΩ) CC1, CC2, CC3, RC1, and RC2 are selected to provide the poles and

zeroes at the desired frequencies, using the following equations:

(37)

(38)

(39)

(40)

(41)

In practice, a good trade off between phase margin and bandwidth can be obtained by selecting the closest

±10% capacitor values above what are suggested for CC1 and CC2, the closest ±10% capacitor value below the

suggestion for CC3, and the closest ±1% resistor values below the suggestions for RC1, RC2. Note that if the

suggested value for RC2 is less than 100Ω, it should be replaced by a short circuit. Following this guideline, the

compensation components will be:

CC1 = 27 pF ±10%

CC2 = 820 pF ±10%

CC3 = 2.7 nF ±10%

Product Folder Links: LM2743

-180

-156

-132

-108

-84

-60

PHASE (o)

100 1k 10k 100k 1M

FREQUENCY (Hz)

-40

-20

GAIN (dB)

100 1k 10k 100k 1M

FREQUENCY (Hz)

GEA-ACTUAL x OPG

HEA = 1 + GEA-ACTUAL+ OPG

Z1 =

RC2 +1

sCC3

RC1 + RC2 1

sCC3

RC1

ZF = sCC1 x 10,000 + 1

sCC2

10,000 + 1

sCC1

sCC2

GEA-ACTUAL = ZF

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Typical Applications (continued)

RC1 = 39.2 kΩ±1%

RC2 = 2.55 kΩ±1%

The transfer function of the compensation block can be derived by considering the compensation components as

impedance blocks ZFand ZIaround an inverting op-amp:

(42)

(43)

(44)

As with the generic equation, GEA-ACTUAL must be modified to take into account the limited bandwidth of the error

amplifier. The result is:

(45)

The total control loop transfer function H is equal to the power stage transfer function multiplied by the error

amplifier transfer function.

H = GPS x HEA (46)

The bandwidth and phase margin can be read graphically from Bode plots of HEA are shown in Figure 38.

Figure 37. Gain vs Frequency Figure 38. Overall Loop Gain and Phase

The bandwidth of this example circuit is 59 kHz, with a phase margin of 60°.

Product Folder Links: LM2743

K = POUT

POUT + PTOTAL

x 100%

LM2743

www.ti.com

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Typical Applications (continued)

8.2.1.2.8 Efficiency Calculations

The following is a sample calculation.

A reasonable estimation of the efficiency of a switching buck controller can be obtained by adding together the

Output Power (POUT) loss and the Total Power (PTOTAL) loss:

(47)

The Output Power (POUT) for the Figure 30 design is (1.2 V x 4 A) = 4.8 W. The Total Power (PTOTAL), with an

efficiency calculation to complement the design, is shown below.

The majority of the power losses are due to low and high side of MOSFET’s losses. The losses in any MOSFET

are group of switching (PSW) and conduction losses(PCND).

PFET = PSW + PCND = 61.38 mW + 270.42 mW (48)

PFET = 331.8 mW (49)

The following equations show FET Switching Loss (PSW).

PSW = PSW(ON) + PSW(OFF) (50)

PSW = 0.5 x VIN x IOUT x (tr+ tf) x fSW (51)

PSW = 0.5 x 3.3 V x 4 A x 300 kHz x 31 ns (52)

PSW = 61.38 mW (53)

The FDS6898A has a typical turn-on rise time trand turn-off fall time tfof 15 ns and 16 ns, respectively. The

switching losses for this type of dual N-Channel MOSFETs are 0.061 W.

The following equations show FET Conduction Loss (PCND).

PCND = PCND1 + PCND2 (54)

PCND1 = (IOUT)2x RDS(ON) x k x D (55)

PCND2 = (IOUT)2x RDS(ON) x k x (1-D) (56)

RDS(ON) = 13 mΩand the factor is a constant value (k = 1.3) to account for the increasing RDS(ON) of a FET due to

heating.

PCND1 = (4A)2x 13 mΩx 1.3 x 0.364 (57)

PCND2 = (4A)2x 13 mΩx 1.3 x (1 - 0.364) (58)

PCND = 98.42 mW + 172 mW = 270.42 mW (59)

There are few additional losses that are taken into account:

The following equations show IC Operating Loss (PIC).

PIC = IQ_VCC x VCC, (60)

where IQ-VCC is the typical operating VCC current

PIC= 1.5 mA x 3.3V = 4.95 mW (61)

The following equations show FET Gate Charging Loss (PGATE).

PGATE = n x VCC x QGS x fSW (62)

PGATE = 2 x 3.3 V x 3 nC x 300 kHz (63)

PGATE = 5.94 mW (64)

Product Folder Links: LM2743

K = 4.8W

4.8W + 0.6W = 89%

K = POUT

POUT + PTOTAL

x 100%

PCAP = (1.924A)2 x 24 m:

IRMS_RIP = IOUT x D(1 - D)

PCAP = (IRMS_RIP)2 x ESR

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Typical Applications (continued)

The value n is the total number of FETs used and QGS is the typical gate-source charge value, which is 3 nC. For

the FDS6898A the gate charging loss is 5.94 mW.

The following equations show Input Capacitor Loss (PCAP).

where (65)

(66)

Here n is the number of paralleled capacitors, ESR is the equivalent series resistance of each, and PCAP is the

dissipation in each. So for example if we use only one input capacitor of 24 mΩ.

(67)

PCAP = 88.8 mW (68)

The following equation shows Output Inductor Loss (PIND).

PIND = I2OUT x DCR (69)

where DCR is the DC resistance. Therefore, for example

PIND = (4A)2x 11 mΩ(70)

PIND = 176 mW (71)

The following equations show Total System Efficiency.

PTOTAL = PFET + PIC + PGATE + PCAP + PIND (72)

(73)

(74)

Product Folder Links: LM2743

BOOT

ISEN

PGND

VCC

PWGD

FREQ

SS/TRACK

SGND

EAO

PGND

VCC = VIN= 3.3V

VOUT = 1.8V@2A

RFB2

CC2 RC1

RCS

CSS

RFADJ

RCC

CCC

D1CBOOT

Q1 CIN1,2

CC1 RFB1

CO1,2

LM2743

RC2 CC3

RPULL-UP +

LM2743

www.ti.com

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

Typical Applications (continued)

8.2.1.3 Application Curves

VCC = 3.3V VOUT = 1.2V VCC = 3.3 V VOUT = 1.2 V

IOUT = 4A CSS = 12nF fSW = 300 kHz IOUT = 4 A CSS = 12 nF fSW = 300 kHz

Figure 39. Start-Up (Full-Load) Figure 40. Shutdown (Full-Load)

8.2.2 Example Circuit 1

Figure 41. 3.3 V to 1.8 V at 2 A, fSW = 300 kHz

8.2.2.1 Design Requirements

This design converts 3.3 V (VIN) to 1.8 V (VOUT) at a maximum load of 2 A, with a switching frequency of 300

kHz.

8.2.2.2 Detailed Design Procedure

Follow the detailed design procedure in Detailed Design Procedure.

Product Folder Links: LM2743

BOOT

ISEN

PGND

VCC

PWGD

FREQ

SS/TRACK

SGND

EAO

PGND

VCC = 5V

VIN = 5V

VOUT = 2.5V@2A

RFB2

CC2 RC1

RCS

CSS

RFADJ

RCC

CCC

D1CBOOT

Q1 CIN1,2

CC1

RFB1

CO1,2

LM2743

RC2 CC3

RPULL-UP +

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

Typical Applications (continued)

8.2.2.3 Bill of Materials

Table 1. Bill of Materials

PART PART NUMBER TYPE PACKAGE DESCRIPTION VENDOR

U1 LM2743 Synchronous TSSOP-14 TI

Controller

Q1 FDS6898A Dual N-MOSFET SOIC-8 20 V, 10 mΩat 4.5 V, Fairchild

16 nC

D1 MBR0520LTI Schottky Diode SOD-123

L1 DO3316P-472 Inductor 4.7 µH, 4.8 Arms, 18 Coilcraft

mΩ

CIN1 16SP100M Aluminum Electrolytic 10mm x 6mm 100 µF, 16 V, 2.89 Sanyo

Arms

CO1 6SP220M Aluminum Electrolytic 10mm x 6mm 220 µF, 6.3 V, 3.1 Sanyo

Arms

CCC, CBOOT, VJ1206Y104KXXA Capacitor 1206 0.1 µF, 10% Vishay

CIN2, CO2

CC3 VJ0805Y332KXXA Capacitor 805 3300 pF, 10% Vishay

CSS VJ0805A123KXAA Capacitor 805 12 nF, 10% Vishay

CC2 VJ0805A821KXAA Capacitor 805 820 pF 10% Vishay

CC1 VJ0805A220KXAA Capacitor 805 22 pF, 10% Vishay

RFB2 CRCW08051002F Resistor 805 10.0 kΩ1% Vishay

RFB1 CRCW08054991F Resistor 805 4.99 kΩ1% Vishay

RFADJ CRCW08051103F Resistor 805 110 kΩ1% Vishay

RC2 CRCW08052101F Resistor 805 2.1 kΩ1% Vishay

RCS CRCW08052101F Resistor 805 2.1 kΩ1% Vishay

RCC CRCW080510R0F Resistor 805 10.0 Ω1% Vishay

RC1 CRCW08055492F Resistor 805 54.9 kΩ1% Vishay

RPULL-UP CRCW08051003J Resistor 805 100 kΩ5% Vishay

8.2.3 Example Circuit 2

Figure 42. 5 V to 2.5 V at 2A, fSW = 300kHz

8.2.3.1 Design Requirements

This design converts 5 V (VIN) to 2.5 V (VOUT) at a maximum load of 2 A, with a switching frequency of 300 kHz.

Product Folder Links: LM2743

BOOT

ISEN

PGND

VCC

PWGD

FREQ

SS/TRACK

SGND

EAO

PGND

VCC = 5V

VIN = 12V

VOUT = 3.3V@4A

RFB2

CC2 RC1

RCS

CSS

RFADJ

RCC

CCC

D1CBOOT

Q1 CIN1,2

CC1

RFB1

CO1,2

LM2743

RC2 CC3

RPULL-UP +

LM2743

www.ti.com

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

8.2.3.2 Detailed Design Procedure

Follow the detailed design procedure in Detailed Design Procedure.

8.2.3.3 Bill of Materials

Table 2. Bill of Materials

PART PART NUMBER TYPE PACKAGE DESCRIPTION VENDOR

U1 LM2743 Synchronous TSSOP-14 TI

Controller

Q1 FDS6898A Dual N-MOSFET SOIC-8 20 V, 10 mΩat 4.5 V, 16 Fairchild

D1 MBR0520LTI Schottky Diode SOD-123

L1 DO3316P-682 Inductor 6.8 µH, 4.4 Arms, 27 mΩCoilcraft

CIN1 16SP100M Aluminum 10mm x 6mm 100 µF, 16 V, 2.89 Arms Sanyo

Electrolytic

CO1 10SP56M Aluminum 6.3mm x 6mm 56 µF, 10 V, 1.7 Arms Sanyo

Electrolytic

CCC, CBOOT, VJ1206Y104KXXA Capacitor 1206 0.1 µF, 10% Vishay

CIN2, CO2

CC3 VJ0805Y182KXXA Capacitor 805 1800 pF, 10% Vishay

CSS VJ0805A123KXAA Capacitor 805 12 nF, 10% Vishay

CC2 VJ0805A821KXAA Capacitor 805 820 pF 10% Vishay

CC1 VJ0805A330KXAA Capacitor 805 33 pF, 10% Vishay

RFB2 CRCW08051002F Resistor 805 10.0 kΩ1% Vishay

RFB1 CRCW08053161F Resistor 805 3.16 kΩ1% Vishay

RFADJ CRCW08051103F Resistor 805 110 kΩ1% Vishay

RC2 CRCW08051301F Resistor 805 1.3 kΩ1% Vishay

RCS CRCW08052101F Resistor 805 2.1 kΩ1% Vishay

RCC CRCW080510R0F Resistor 805 10.0 Ω1% Vishay

RC1 CRCW08053322F Resistor 805 33.2 kΩ1% Vishay

RPULL-UP CRCW08051003J Resistor 805 100 kΩ5% Vishay

8.2.4 Example Circuit 3

Figure 43. 12 V to 3.3 V at 4 A, fSW = 300 kHz

8.2.4.1 Design Requirements

This design converts 12 V (VIN) to 3.3 V (VOUT) at a maximum load of 4 A, with a switching frequency of 300 kHz.

Product Folder Links: LM2743

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

8.2.4.2 Detailed Design Procedure

Follow the detailed design procedure in Detailed Design Procedure.

8.2.4.3 Bill of Materials

Table 3. Bill of Materials

PART PART NUMBER TYPE PACKAGE DESCRIPTION VENDOR

U1 LM2743 Synchronous TSSOP-14 TI

Controller

Q1 FDS6898A Dual N-MOSFET SOIC-8 20 V, 10 mΩat 4.5 V, 16 Fairchild

D1 MBR0520LTI Schottky Diode SOD-123

L1 DO3316P-332 Inductor 3.3 µH, 5.4 Arms, 15 mΩCoilcraft

CIN1 16SP100M Aluminum Electrolytic 10 mm x 6 mm 100 µF, 16 V, 2.89 Arms Sanyo

CO1 6SP220M Aluminum Electrolytic 10 mm x 6 mm 220 µF, 6.3 V 3.1 Arms Sanyo

CCC, CBOOT, VJ1206Y104KXXA Capacitor 1206 0.1 µF, 10% Vishay

CIN2, CO2

CC3 VJ0805Y222KXXA Capacitor 805 2200 pF, 10% Vishay

CSS VJ0805A123KXAA Capacitor 805 12 nF, 10% Vishay

CC2 VJ0805Y332KXXA Capacitor 805 3300 pF 10% Vishay

CC1 VJ0805A820KXAA Capacitor 805 82 pF, 10% Vishay

RFB2 CRCW08051002F Resistor 805 10.0 kΩ1% Vishay

RFB1 CRCW08052211F Resistor 805 2.21 kΩ1% Vishay

RFADJ CRCW08051103F Resistor 805 110 kΩ1% Vishay

RC2 CRCW08052611F Resistor 805 2.61 kΩ1% Vishay

RCS CRCW08054121F Resistor 805 4.12 kΩ1% Vishay

RCC CRCW080510R0F Resistor 805 10.0 Ω1% Vishay

RC1 CRCW08051272F Resistor 805 12.7 kΩ1% Vishay

RPULL-UP CRCW08051003J Resistor 805 100 kΩ5% Vishay

Product Folder Links: LM2743

Controller

VOUTGNDGNDVIN

Inductor

CIN

COUT

QHQL

Place controller as

close to the switches

LM2743

www.ti.com

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

9 Power Supply Recommendations

The LM2743 is a power management device. The power supply for the device is any DC voltage source within

the specified input range (see Design Requirements).

10 Layout

10.1 Layout Guidelines

In a buck regulator, the primary switching loop consists of the input capacitor and MOSFET switches. Minimixing

the area of this loop reduces the stray inductance, and minimizes noise and possible erratic operation. High

quality input capacitors should be placed as close as possible to the MOSFET switches, with the positive side of

the input capacitor connected directly to the high-side MOSFET drain, and the ground side of the capacitor

connected as close as possible to the low-side MOSFET switch ground connection. Connect all of the low power

ground connections directly to the SGND pin. Connect the VCC capacitor directly to the PGND pin.

10.2 Layout Example

Figure 44. Layout Recommendation

Product Folder Links: LM2743

LM2743

SNVS276H –APRIL 2004–REVISED OCTOBER 2015

www.ti.com

11 Device and Documentation Support

11.1 Community Resources

The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective

contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of

Use.

TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration

among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help

solve problems with fellow engineers.

Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and

contact information for technical support.

11.2 Trademarks

WEBENCH, E2E are trademarks of Texas Instruments.

All other trademarks are the property of their respective owners.

11.3 Electrostatic Discharge Caution

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

11.4 Glossary

SLYZ022 —TI Glossary.

This glossary lists and explains terms, acronyms, and definitions.

12 Mechanical, Packaging, and Orderable Information

The following pages include mechanical, packaging, and orderable information. This information is the most

current data available for the designated devices. This data is subject to change without notice and revision of

this document. For browser-based versions of this data sheet, refer to the left-hand navigation.

Product Folder Links: LM2743

PACKAGE OPTION ADDENDUM

www.ti.com 17-Mar-2017

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead/Ball Finish

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM2743MTC NRND TSSOP PW 14 94 TBD Call TI Call TI -40 to 125 2743

MTC

LM2743MTC/NOPB ACTIVE TSSOP PW 14 94 Green (RoHS

& no Sb/Br) CU NIPDAU | CU SN Level-1-260C-UNLIM -40 to 125 2743

MTC

LM2743MTCX/NOPB ACTIVE TSSOP PW 14 2500 Green (RoHS

& no Sb/Br) CU NIPDAU | CU SN Level-1-260C-UNLIM -40 to 125 2743

MTC

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability

information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined.

Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that

lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.

Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between

the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.

Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight

in homogeneous material)

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish

value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

PACKAGE OPTION ADDENDUM

www.ti.com 17-Mar-2017

Addendum-Page 2

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

LM2743MTCX/NOPB TSSOP PW 14 2500 330.0 12.4 6.9 5.6 1.6 8.0 12.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 23-Jun-2018

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

LM2743MTCX/NOPB TSSOP PW 14 2500 367.0 367.0 35.0

PACKAGE MATERIALS INFORMATION

www.ti.com 23-Jun-2018

Pack Materials-Page 2

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and ISO 26262), TI is not responsible for any failure to meet such industry standard requirements.

Where TI specifically promotes products as facilitating functional safety or as compliant with industry functional safety standards, such

products are intended to help enable customers to design and create their own applications that meet applicable functional safety standards

and requirements. Using products in an application does not by itself establish any safety features in the application. Designers must

ensure compliance with safety-related requirements and standards applicable to their applications. Designer may not use any TI products in

life-critical medical equipment unless authorized officers of the parties have executed a special contract specifically governing such use.

Life-critical medical equipment is medical equipment where failure of such equipment would cause serious bodily injury or death (e.g., life

support, pacemakers, defibrillators, heart pumps, neurostimulators, and implantables). Such equipment includes, without limitation, all

medical devices identified by the U.S. Food and Drug Administration as Class III devices and equivalent classifications outside the U.S.

TI may expressly designate certain products as completing a particular qualification (e.g., Q100, Military Grade, or Enhanced Product).

Designers agree that it has the necessary expertise to select the product with the appropriate qualification designation for their applications

and that proper product selection is at Designers’ own risk. Designers are solely responsible for compliance with all legal and regulatory

requirements in connection with such selection.

Designer will fully indemnify TI and its representatives against any damages, costs, losses, and/or liabilities arising out of Designer’s non-

compliance with the terms and provisions of this Notice.

Mailing Address: Texas Instruments, Post Office Box 655303, Dallas, Texas 75265

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