© Semiconductor Components Industries, LLC, 2006
April, 2006 − Rev. 8 1Publication Order Number:
MC33035/D
MC33035, NCV33035
Brushless DC
Motor Controller
The MC33035 is a high performance second generation monolithic
brushless DC motor controller containing all of the active functions
required to implement a full featured open loop, three or four phase
motor control system. This device consists of a rotor position decoder
for proper commutation sequencing, temperature compensated
reference capable of supplying sensor power, frequency
programmable sawtooth oscillator, three open collector top drivers,
and three high current totem pole bottom drivers ideally suited for
driving power MOSFETs.
Also included are protective features consisting of undervoltage
lockout, cycle−by−cycle current limiting with a selectable time
delayed latched shutdown mode, internal thermal shutdown, and a
unique fault output that can be interfaced into microprocessor
controlled systems.
Typical motor control functions include open loop speed, forward or
reverse direction, run enable, and dynamic braking. The MC33035 is
designed to operate with electrical sensor phasings of 60°/300° or
120°/240°, and can also efficiently control brush DC motors.
Features
•10 to 30 V Operation
•Undervoltage Lockout
•6.25 V Reference Capable of Supplying Sensor Power
•Fully Accessible Error Amplifier for Closed Loop Servo
Applications
•High Current Drivers Can Control External 3−Phase MOSFET
Bridge
•Cycle−By−Cycle Current Limiting
•Pinned−Out Current Sense Reference
•Internal Thermal Shutdown
•Selectable 60°/300° or 120°/240° Sensor Phasings
•Can Efficiently Control Brush DC Motors with External MOSFET
H−Bridge
•NCV Prefix for Automotive and Other Applications Requiring Site
and Control Changes
•Pb−Free Packages are Available
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AT
BT
Top Drive
Output
16
Bottom
Drive
Outputs
15
(Top View)
17
18
19
20
21
10
9
8
7
6
5
Sensor
Inputs
4
Oscillator
Current Sense
Noninverting Input
Reference Output
Output Enable
SC
SB
SA
60°/120°SelectFwd/Rev
Current Sense
Inverting Input
Gnd
VCC
CT
22
23
BB
CB
3
24
Brake
2
AB
1
VC
PIN CONNECTIONS
24
1
24
1
P SUFFIX
PLASTIC PACKAGE
CASE 724
DW SUFFIX
PLASTIC PACKAGE
CASE 751E
(SO−24L)
14
1312
11
Error Amp
Inverting Input
Error Amp
Noninverting Input
Error Amp Out/
PWM Input
Fault Output
See detailed ordering and shipping information in the packag
e
dimensions section on page 27 of this data sheet.
ORDERING INFORMATION
See general marking information in the device marking
section on page 27 of this data sheet.
DEVICE MARKING INFORMATION
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2
Motor
Enable
Q
S
CT
R
RT
Oscillator
Error Amp
PWM
Thermal
Shutdown
Reference
Regulator
Lockout
Undervoltage
Vin
Fwd/Rev
Q
R
S
Faster
SS
VM
Speed
Set
This device contains 285 active transistors.
Representative Schematic Diagram
Rotor
Position
Decoder
Output
Buffers
Current Sense
Reference
60°/120°
18
17
Brake
Fault N
N
7
22
3
6
5
4
8
11
12
13
10
14
2
1
24
21
20
19
9
15
2316
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MAXIMUM RATINGS
Rating Symbol Value Unit
Power Supply Voltage VCC 40 V
Digital Inputs (Pins 3, 4, 5, 6, 22, 23) − Vref V
Oscillator Input Current (Source or Sink) IOSC 30 mA
Error Amp Input Voltage Range (Pins 11, 12, Note 1) VIR −0.3 to Vref V
Error Amp Output Current (Source or Sink, Note 2) IOut 10 mA
Current Sense Input Voltage Range (Pins 9, 15) VSense −0.3 to 5.0 V
Fault Output Voltage VCE(Fault)20 V
Fault Output Sink Current ISink(Fault)20 mA
Top Drive Voltage (Pins 1, 2, 24) VCE(top) 40 V
Top Drive Sink Current (Pins 1, 2, 24) ISink(top) 50 mA
Bottom Drive Supply Voltage (Pin 18) VC30 V
Bottom Drive Output Current (Source or Sink, Pins 19, 20, 21) IDRV 100 mA
Power Dissipation and Thermal Characteristics
P Suffix, Dual In Line, Case 724
Maximum Power Dissipation @ TA = 85°C
Thermal Resistance, Junction−to−Air
DW Suffix, Surface Mount, Case 751E
Maximum Power Dissipation @ TA = 85°C
Thermal Resistance, Junction−to−Air
PD
RθJA
PD
RθJA
867
75
650
100
mW
°C/W
mW
°C/W
Operating Junction Temperature TJ150 °C
Operating Ambient Temperature Range (Note 3) MC33035
NCV33035 TA−40 to +85
−40 to +125 °C
Storage Temperature Range Tstg −65 to +150 °C
Stresses exceeding Maximum Ratings may damage the device. Maximum Ratings are stress ratings only. Functional operation above the
Recommended O p e r a t i n g Conditions is not implied. Extended exposure to stresses above the Recommended Operating Conditions may affect
device reliability.
ELECTRICAL CHARACTERISTICS (VCC = VC = 20 V, RT = 4.7 k, CT = 10 nF, TA = 25°C, unless otherwise noted.)
Characteristic Symbol Min Typ Max Unit
REFERENCE SECTION
Reference Output Voltage (Iref = 1.0 mA)
TA = 25°C
(Note 4)
Vref 5.9
5.82 6.24
−6.5
6.57
V
Line Regulation (VCC = 10 to 30 V, Iref = 1.0 mA) Regline − 1.5 30 mV
Load Regulation (Iref = 1.0 to 20 mA) Regload − 16 30 mV
Output Short Circuit Current (Note 5) ISC 40 75 − mA
Reference Under Voltage Lockout Threshold Vth 4.0 4.5 5.0 V
ERROR AMPLIFIER
Input Offset Voltage (Note 4) VIO − 0.4 10 mV
Input Offset Current (Note 4) IIO − 8.0 500 nA
Input Bias Current (Note 4) IIB −−46 −1000 nA
Input Common Mode Voltage Range VICR (0 V to Vref) V
Open Loop Voltage Gain (VO = 3.0 V, RL = 15 k) AVOL 70 80 − dB
Input Common Mode Rejection Ratio CMRR 55 86 − dB
Power Supply Rejection Ratio (VCC = VC = 10 to 30 V) PSRR 65 105 − dB
1. The input common mode voltage or input signal voltage should not be allowed to go negative by more than 0.3 V.
2. The compliance voltage must not exceed the range of −0.3 to Vref.
3. NCV33035: Tlow = − 40°C, Thigh = 125°C. Guaranteed by design. NCV prefix is for automotive and other applications requiring site and change
control.
4. MC33035: TA = −40°C to +85°C; NCV33035: TA = −40°C to +125°C.
5. Maximum package power dissipation limits must be observed.
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ELECTRICAL CHARACTERISTICS (continued) (VCC = VC = 20 V, RT = 4.7 k, CT = 10 nF, TA = 25°C, unless otherwise noted.)
Characteristic Symbol Min Typ Max Unit
ERROR AMPLIFIER
Output Voltage Swing
High State (RL = 15 k to Gnd)
Low State (RL = 15 k to Vref)VOH
VOL 4.6
−5.3
0.5 −
1.0
V
OSCILLATOR SECTION
Oscillator Frequency fOSC 22 25 28 kHz
Frequency Change with Voltage (VCC = 10 to 30 V) ΔfOSC/ΔV − 0.01 5.0 %
Sawtooth Peak Voltage VOSC(P) − 4.1 4.5 V
Sawtooth Valley Voltage VOSC(V) 1.2 1.5 − V
LOGIC INPUTS
Input Threshold Voltage (Pins 3, 4, 5, 6, 7, 22, 23)
High State
Low State VIH
VIL 3.0
−2.2
1.7 −
0.8
V
Sensor Inputs (Pins 4, 5, 6)
High State Input Current (VIH = 5.0 V)
Low State Input Current (VIL = 0 V) IIH
IIL −150
−600 −70
−337 −20
−150
μA
Forward/Reverse, 60°/120° Select (Pins 3, 22, 23)
High State Input Current (VIH = 5.0 V)
Low State Input Current (VIL = 0 V) IIH
IIL −75
−300 −36
−175 −10
−75
μA
Output Enable
High State Input Current (VIH = 5.0 V)
Low State Input Current (VIL = 0 V) IIH
IIL −60
−60 −29
−29 −10
−10
μA
CURRENT−LIMIT COMPARATOR
Threshold Voltage Vth 85 101 115 mV
Input Common Mode Voltage Range VICR − 3.0 − V
Input Bias Current IIB − −0.9 −5.0 μA
OUTPUTS AND POWER SECTIONS
Top Drive Output Sink Saturation (Isink = 25 mA) VCE(sat) − 0.5 1.5 V
Top Drive Output Off−State Leakage (VCE = 30 V) IDRV(leak) − 0.06 100 μA
Top Drive Output Switching Time (C L = 47 pF, RL = 1.0 k) ns
Rise Time tr− 107 300
Fall Time tf− 26 300
Bottom Drive Output Voltage
High State (VCC = 20 V, VC = 30 V, Isource = 50 mA)
Low State (VCC = 20 V, VC = 30 V, Isink = 50 mA) VOH
VOL (VCC −2.0)
−(VCC −1.1)
1.5 −
2.0
V
Bottom Drive Output Switching Time (CL = 1000 pF) ns
Rise Time tr− 38 200
Fall Time tf− 30 200
Fault Output Sink Saturation (Isink = 16 mA) VCE(sat) − 225 500 mV
Fault Output Of f−State Leakage (V CE = 20 V) IFLT(leak) − 1.0 100 μA
Under Voltage Lockout V
Drive Output Enabled (VCC or VC Increasing) Vth(on) 8.2 8.9 10
Hysteresis VH0.1 0.2 0.3
Power Supply Current
Pin 17 (VCC = VC = 20 V)
Pin 17 (VCC = 20 V, VC = 30 V)
Pin 18 (VCC = VC = 20 V)
Pin 18 (VCC = 20 V, VC = 30 V)
ICC
IC
−
−
−
−
12
14
3.5
5.0
16
20
6.0
10
mA
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Vsat, OUTPUT SATURATION VOLTAGE (V)
5.0 μs/DIV
AV = +1.0
No Load
TA = 25°C
, OUTPUT VOLTAGE (V)
O
4.5
3.0
1.5
1.0 μs/DIV
AV = +1.0
No Load
TA = 25°C
3.05
3.0
2.95
Gnd
Vref
IO, OUTPUT LOAD CURRENT (mA)
f, FREQUENCY (Hz)
56
1.0 k
220
200
180
160
140
120
100
80
60
−24
−16
− 8.0
0
8.0
16
24
32
40
48
10 M1.0 M100 k10 k
40
240
AVOL, OPEN LOOP VOLTAGE GAIN (dB)
EXCESS PHASE (DEGREES)
,
φ
Phase
Gain
TA, AMBIENT TEMPERATURE (°C)
−55
− 4.0
− 2.0
0
2.0
125
4.0
1007550250−25
f
OSC OSCILLATOR FREQUENCY CHANGE (%)
,Δ
100
CT = 1.0 nF
CT = 100 nF
1.0
RT, TIMING RESISTOR (kΩ)
100010010
0
10
f
OSC OSCILLATOR FREQUENCY (kHz)
,
CT = 10 nF
Figure 1. Oscillator Frequency versus
Timing Resistor Figure 2. Oscillator Frequency Change
versus Temperature
Figure 3. Error Amp Open Loop Gain and
Phase versus Frequency Figure 4. Error Amp Output Saturation
Voltage versus Load Current
Figure 5. Error Amp Small−Signal
Transient Response Figure 6. Error Amp Large−Signal
Transient Response
0
1.0 2.0
0
− 0.8
−1.6
1.6
0.8
5.04.03.00
VCC = 20 V
VC = 20 V
TA = 25°C
VCC = 20 V
VC = 20 V
RT = 4.7 k
CT = 10 nF
Source Saturation
(Load to Ground)
VCC = 20 V
VC = 20 V
TA = 25°C
Sink Saturation
(Load to Vref)
V
, OUTPUT VOLTAGE (V)
O
V
VCC = 20 V
VC = 20 V
VO = 3.0 V
RL = 15 k
CL = 100 pF
TA = 25°C
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, OUTPUT SATURATION VOLTAGE (V)Vsat
0
ISink, SINK CURRENT (mA)
016128.04.0
0.25
0.2
0.05
0
TA, AMBIENT TEMPERATURE (°C)
−25
−40
−20
−55 0
40
20
125100755025
, NORMALIZED REFERENCE VOLTAGE CHANGE (mV)ΔVref
0
Iref, REFERENCE OUTPUT SOURCE CURRENT (mA)
0
605040302010
−24
−20
− 4.0
− 8.0
− 12
− 16
Vref, REFERENCE OUTPUT VOLTAGE CHANGE (mV)Δ
Figure 7. Reference Output Voltage Change
versus Output Source Current Figure 8. Reference Output Voltage
versus Supply Voltage
Figure 9. Reference Output Voltage
versus Temperature Figure 10. Output Duty Cycle versus
PWM Input Voltage
Figure 11. Bottom Drive Response Time versus
Current Sense Input Voltage Figure 12. Fault Output Saturation
versus Sink Current
0
0
7.0
00
VCC, SUPPLY VOLTAGE (V)
6.0
40302010
5.0
4.0
3.0
2.0
1.0
Vref, REFERENCE OUTPUT VOLTAGE (V)
5.04.03.02.01.0
100
80
60
40
20
PWM INPUT VOLTAGE (V)
OUTPUT DUTY CYCLE (%)
0
CURRENT SENSE INPUT VOLTAGE (NORMALIZED TO Vth)
50
100
150
200
250
1.0 2.0 3.0 4.0 5.0 7.0 8.0 10
tHL, BOTTOM DRIVE RESPONSE TIME (ns)
No Load
TA = 25°C
VCC = 20 V
VC = 20 V
No Load
6.0 9.0
VCC = 20 V
VC = 20 V
TA = 25°C
VCC = 20 V
VC = 20 V
RT = 4.7 k
CT = 10 nF
TA = 25°C
VCC = 20 V
VC = 20 V
RL = 1
CL = 1.0 nF
TA = 25°C0.15
0.1
VCC = 20 V
VC = 20 V
TA = 25°C
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7
1.0
OUTPUT VOLTAGE (%)
Gnd
VC
− 2.0
40
0
IO, OUTPUT LOAD CURRENT (mA)
0
0
−1.0
2.0
806020
, OUTPUT SATURATION VOLTAGE (V)
sat
Sink Saturation
(Load to VC)
Source Saturation
(Load to Ground)
VCC = 20 V
VC = 20 V
TA = 25°C
V
VCC = 20 V
VC = 20 V
TA = 25°C
50 ns/DIV
VCC = 20 V
VC = 20 V
CL = 1.0 nF
TA = 25°C
100 ns/DIV
VCC = 20 V
VC = 20 V
RL = 1.0 k
CL = 15 pF
TA = 25°C
Figure 13. Top Drive Output Saturation
Voltage versus Sink Current Figure 14. Top Drive Output Waveform
Figure 15. Bottom Drive Output Waveform Figure 16. Bottom Drive Output Waveform
20
00
ISink, SINK CURRENT (mA)
10 30 40
0.4
0.8
1.2
Vsat, OUTPUT SATURATION VOLTAGE (V)
Figure 17. Bottom Drive Output Saturation
Voltage versus Load Current
50 ns/DIV
VCC = 20 V
VC = 20 V
CL = 15 pF
TA = 25°C
Figure 18. Power and Bottom Drive Supply
Current versus Supply Voltage
16
14
12
10
8.0
6.0
4.0
2.0
0
, POWER SUPPLY CURRENT (mA)
CC
, I
0 5.0 10 15 20 25 30
C
I
RT = 4.7 k
CT = 10 nF
Pins 3−6, 9, 15, 23 = Gnd
Pins 7, 22 = Open
TA = 25°C
VCC, SUPPLY VOLTAGE (V)
ICC
IC
100
0
100
0
100
0
OUTPUT VOLTAGE (%) OUTPUT VOLTAGE (%)
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PIN FUNCTION DESCRIPTION
Pin Symbol Description
1, 2, 24 BT, AT, CTThese three open collector Top Drive outputs are designed to drive the external
upper power switch transistors.
3 Fwd/Rev The Forward/Reverse Input is used to change the direction of motor rotation.
4, 5, 6 SA, SB, SCThese three Sensor Inputs control the commutation sequence.
7Output Enable A logic high at this input causes the motor to run, while a low causes it to coast.
8Reference Output This output provides charging current for the oscillator timing capacitor CT and a
reference for the error amplifier. It may also serve to furnish sensor power.
9Current Sense Noninverting Input A 100 mV signal, with respect to Pin 15, at this input terminates output switch
conduction during a given oscillator cycle. This pin normally connects to the top
side of the current sense resistor.
10 Oscillator The Oscillator frequency is programmed by the values selected for the timing
components, RT and CT.
11 Error Amp Noninverting Input This input is normally connected to the speed set potentiometer.
12 Error Amp Inverting Input This input is normally connected to the Error Amp Output in open loop
applications.
13 Error Amp Out/PWM Input This pin is available for compensation in closed loop applications.
14 Fault Output This open collector output is active low during one or more of the following
conditions: Invalid Sensor Input code, Enable Input at logic 0, Current Sense
Input greater than 100 mV (Pin 9 with respect to Pin 15), Undervoltage Lockout
activation, and Thermal Shutdown.
15 Current Sense Inverting Input Reference pin for internal 100 mV threshold. This pin is normally connected to
the bottom side of the current sense resistor.
16 Gnd This pin supplies a ground for the control circuit and should be referenced back
to the power source ground.
17 VCC This pin is the positive supply of the control IC. The controller is functional over a
minimum VCC range of 10 to 30 V.
18 VCThe high state (VOH) of the Bottom Drive Outputs is set by the voltage applied to
this pin. The controller is operational over a minimum VC range of 10 to 30 V.
19, 20, 21 CB, BB, ABThese three totem pole Bottom Drive Outputs are designed for direct drive of the
external bottom power switch transistors.
22 60°/120° Select The electrical state of this pin configures the control circuit operation for either
60° (high state) or 120°(low state) sensor electrical phasing inputs.
23 Brake A logic low state at this input allows the motor to run, while a high state does not
allow motor operation and if operating causes rapid deceleration.
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9
INTRODUCTION
The MC33035 is one of a series of high performance
monolithic DC brushless motor controllers produced by
Motorola. It contains all of the functions required to
implement a full−featured, open loop, three or four phase
motor control s ystem. In addition, the controller can be made
to operate DC brush motors. Constructed with Bipolar
Analog t echnology , i t of fers a h igh degree of p erformance and
ruggedness i n h ostile i ndustrial e nvironments. T he M C33035
contains a rotor position decoder for proper commutation
sequencing, a temperature c ompensated reference capable of
supplying a sensor power, a frequency programmable
sawtooth oscillator, a fully accessible e rror amplifier, a pulse
width modulator comparator, three open collector top drive
outputs, and three high current totem pole bottom driver
outputs ideally suited for driving power MOSFETs.
Included in the MC33035 are protective features
consisting of undervoltage lockout, cycle−by−cycle current
limiting with a selectable time delayed latched shutdown
mode, internal thermal shutdown, and a unique fault output
that can easily be interfaced to a microprocessor controller.
Typical motor control functions include open loop speed
control, forward or reverse rotation, run enable, and
dynamic braking. In addition, the MC33035 has a 60°/120°
select pin which configures the rotor position decoder for
either 60° or 120° sensor electrical phasing inputs.
FUNCTIONAL DESCRIPTION
A representative internal block diagram is shown in
Figure 19 with various applications shown in Figures 36, 38,
39, 43, 45, and 46. A discussion of the features and function
of each of the internal blocks given below is referenced to
Figures 19 and 36.
Rotor Position Decoder
An internal rotor position decoder monitors the three
sensor inputs (Pins 4, 5, 6) to provide the proper sequencing
of the top and bottom drive outputs. The sensor inputs are
designed to interface directly with open collector type Hall
Effect switches or opto slotted couplers. Internal pull−up
resistors are included to minimize the required number of
external components. The inputs are TTL compatible, with
their thresholds typically at 2.2 V. The MC33035 series is
designed t o control three phase motors and operate with four
of the most common conventions of sensor phasing. A
60°/120°Select (Pin 22) is conveniently provided and
affords the MC33035 to configure itself to control motors
having either 60°, 120°, 240° or 300° electrical sensor
phasing. With three sensor inputs there are eight possible
input code combinations, six of which are valid rotor
positions. The remaining two codes are invalid and are
usually caused by an open or shorted sensor line. With six
valid input codes, the decoder can resolve the motor rotor
position to within a window of 60 electrical degrees.
The Forward/Reverse input (Pin 3) is used to change the
direction of motor rotation by reversing the voltage across
the stator winding. When the input changes state, from high
to low with a given sensor input code (for example 100), the
enabled top and bottom drive outputs with the same alpha
designation are exchanged (AT to AB, BT to BB, CT to CB).
In effect, the commutation sequence is reversed and the
motor changes directional rotation.
Motor on/off control is accomplished by the Output
Enable (Pin 7). When left disconnected, an internal 25 μA
current source enables sequencing of the top and bottom
drive outputs. When grounded, the top drive outputs turn of f
and the bottom drives are forced low, causing the motor to
coast and the Fault output to activate.
Dynamic motor braking allows an additional margin of
safety to be designed into the final product. Braking is
accomplished by placing the Brake Input (Pin 23) in a high
state. This causes the top drive outputs to turn off and the
bottom drives to turn on, shorting the motor−generated back
EMF. The brake input has unconditional priority over all
other inputs. The internal 40 kΩ pull−up resistor simplifies
interfacing with the system safety−switch by insuring brake
activation if opened or disconnected. The commutation
logic truth table is shown in Figure 20. A four input NOR
gate is used to monitor the brake input and the inputs to the
three top drive output transistors. Its purpose is to disable
braking until the top drive outputs attain a high state. This
helps to prevent simultaneous conduction of the the top and
bottom power switches. In half wave motor drive
applications, the top drive outputs are not required and are
normally left disconnected. Under these conditions braking
will still be accomplished since the NOR gate senses the
base voltage to the top drive output transistors.
Error Amplifier
A high performance, fully compensated error amplifier
with access to both inputs and output (Pins 11, 12, 13) is
provided to facilitate the implementation of closed loop
motor speed control. The amplifier features a typical DC
voltage gain of 80 dB, 0.6 MHz gain bandwidth, and a wide
input common m ode v oltage range t hat e xtends f rom g round
to Vref. In most open loop speed control applications, the
amplifier is configured as a unity gain voltage follower with
the noninverting input connected to the speed set voltage
source. Additional configurations are shown in Figures 31
through 35.
Oscillator
The frequency of the internal ramp oscillator is
programmed by the values selected for timing components
RT and CT. Capacitor CT is charged from the Reference
Output (Pin 8) through resistor RT and discharged by an
internal discharge transistor. The ramp peak and valley
voltages are typically 4.1 V and 1.5 V respectively. To
provide a good compromise between audible noise and
output switching efficiency, an oscillator frequency in the
range of 2 0 t o 3 0 kHz is recommended. Refer to Figure 1 for
component selection.
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10
15
24
20
2
1
21
19
VM
Top
Drive
Outputs
Bottom
Drive
Outputs
CB
Current Sense
Reference Input
BB
AB
AT
BT
CT
Q
S
R
Oscillator
Error Amp
PWM
Thermal
Shutdown
Reference
Regulator
Lockout
Undervoltage
Q
R
S
Rotor
Position
Decoder
Brake Input
Figure 19. Representative Block Diagram
60°/120°Select
Output Enable
CT
RT
Vin
4
10
11
13
8
12
3
17
22
7
6
5
Forward/Reverse
Faster
Noninv. Input
SA
SC
SB
Sensor
Inputs
Error Amp Out
PWM Input
Sink Only
Positive True
Logic With
Hysteresis
=
Reference Output
16
Latch
Latch
23Gnd
14
9Current Sense Input
Fault Output
20 k
20 k
20 k
40 k
40 k
25 μA
VCC
VC
18
9.1 V
4.5 V
100 mV
40 k
Inputs (Note 2) Outputs (Note 3)
Sensor Electrical Phasing (Note 4) Top Drives Bottom Drives
SA60°
SBSCSA120°
SBSCF/R Enable Brake Current
Sense ATBTCTABBBCBFault
1
1
1
0
0
0
0
1
1
1
0
0
0
0
1
1
1
0
1
1
0
0
0
1
0
1
1
1
0
0
0
0
0
1
1
1
1
1
1
1
1
1
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
0
1
1
1
1
0
1
0
0
1
1
1
1
1
1
0
0
1
0
0
1
1
0
0
0
0
0
0
1
1
1
1
0
0
0
0
1
1
1
1
1
1
(Note 5)
F/R = 1
1
1
1
0
0
0
0
1
1
1
0
0
0
0
1
1
1
0
1
1
0
0
0
1
0
1
1
1
0
0
0
0
0
1
1
1
0
0
0
0
0
0
1
1
1
1
1
1
0
0
0
0
0
0
0
0
0
0
0
0
1
1
0
0
1
1
1
1
1
1
0
0
0
0
1
1
1
1
1
0
0
0
0
1
0
1
1
0
0
0
0
0
0
1
1
0
1
1
1
1
1
1
(Note 5)
F/R = 0
1
00
11
01
01
01
0X
XX
X0
0X
X1
11
11
10
00
00
00
0(Note 6)
Brake = 0
1
00
11
01
01
01
0X
XX
X1
1X
X1
11
11
11
11
11
10
0(Note 7)
Brake = 1
V V V V V V X 1 1 X 1 1 1 1 1 1 1 (Note 8)
V V V V V V X 0 1 X 1 1 1 1 1 1 0 (Note 9)
V V V V V V X 0 0 X 1 1 1 0 0 0 0 (Note 10)
MC33035, NCV33035
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11
V V V V V V X 1 0 1 1 1 1 0 0 0 0 (Note 11)
NOTES: 1. V = Any one of six valid sensor or drive combinations X = Don’t care.
2. The digital inputs (Pins 3, 4, 5, 6, 7, 22, 23) are all TTL compatible. The current sense input (Pin 9) has a 100 mV threshold with respect to Pin 15.
A logic 0 for this input is defined as < 85 mV, and a logic 1 is > 115 mV.
3. The fault and top drive outputs are open collector design and active in the low (0) state.
4. With 60°/120°select (Pin 22) in the high (1) state, configuration is for 60°sensor electrical phasing inputs. With Pin 22 in low (0) state, configuration
is for 120° sensor electrical phasing inputs.
5. Valid 60° or 120° sensor combinations for corresponding valid top and bottom drive outputs.
6. Invalid sensor inputs with brake = 0; All top and bottom drives off, Fault low.
7. Invalid sensor inputs with brake = 1; All top drives off, all bottom drives on, Fault low.
8. Valid 60° or 120°sensor inputs with brake = 1; All top drives off, all bottom drives on, Fault high.
9. Valid sensor inputs with brake = 1 and enable = 0; All top drives off, all bottom drives on, Fault low.
10. Valid sensor inputs with brake = 0 and enable = 0; All top and bottom drives off, Fault low.
11. All bottom drives off, Fault low.
Figure 20. Three Phase, Six Step Commutation Truth Table (Note 1)
Pulse Width Modulator
The use of pulse width modulation provides an energy
efficient method of controlling the motor speed by varying
the average voltage applied to each stator winding during the
commutation sequence. As CT discharges, the oscillator sets
both latches, allowing conduction of the top and bottom
drive outputs. The PWM comparator resets the upper latch,
terminating the bottom drive output conduction when the
positive−going ramp of CT becomes greater than the error
amplifier output. The pulse width modulator timing diagram
is shown in Figure 21. Pulse width modulation for speed
control appears only at the bottom drive outputs.
Current Limit
Continuous operation of a motor that is severely
over−loaded results in overheating and eventual failure.
This destructive condition can best be prevented with the use
of cycle−by−cycle current limiting. That is, each on−cycle
is treated as a separate event. Cycle−by−cycle current
limiting is accomplished by monitoring the stator current
build−up each time an output switch conducts, and upon
sensing an over current condition, immediately turning off
the switch and holding it off for the remaining duration of
oscillator ramp−up period. The stator current is converted to
a voltage by inserting a ground−referenced sense resistor RS
(Figure 36) in series with the three bottom switch transistors
(Q4, Q5, Q6). The voltage developed across the sense
resistor is monitored by the Current Sense Input (Pins 9 and
15), and compared to the internal 100 mV reference. The
current sense comparator inputs have an input common
mode range of approximately 3.0 V. If the 100 mV current
sense threshold is exceeded, the comparator resets the lower
sense latch and terminates output switch conduction. The
value for the current sense resistor is:
RS+0.1
Istator(max)
The Fault output activates during an over current condition.
The dual−latch PWM configuration ensures that only one
single output conduction pulse occurs during any given
oscillator cycle, whether terminated by the output of the
error amp or the current limit comparator.
MC33035, NCV33035
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12
Figure 21. Pulse Width Modulator Timing Diagram
Current
Sense Input
Capacitor CT
Error Amp
Out/PWM
Input
Latch Set"
Inputs
Top Drive
Outputs
Bottom Drive
Outputs
Fault Output
Reference
The on−chip 6.25 V regulator (Pin 8) provides charging
current for t h e o s cillator timing capacitor, a reference for the
error amplifier, and can supply 2 0 mA of current suitable for
directly powering sensors in low voltage applications. In
higher voltage applications, it may become necessary to
transfer the power dissipated by the regulator of f the IC. This
is easily accomplished with the addition of an external pass
transistor as shown in Figure 22. A 6.25 V reference level
was chosen to allow implementation of the simpler NPN
circuit, where Vref − VBE exceeds the minimum voltage
required by Hall Effect sensors over temperature. With
proper transistor selection and adequate heatsinking, up to
one amp of load current can be obtained.
Figure 22. Reference Output Buffers
The NPN circuit is recommended for powering Hall or opto sensors, where
the output voltage temperature coefficient is not critical. The PNP circuit is
slightly more complex, but is also more accurate over temperature. Neither
circuit has current limiting.
To
Control
Circuitry
6.25 V
Sensor
Power
≈5.6 V
MPS
U51A
Vin
MPS
U01A
Vin
To Control Circuitry
and Sensor Power
6.25 V
UVLO
17
39
REF
8
0.1
REF
8
18
UVLO
17
18
Undervoltage Lockout
A triple Undervoltage Lockout has been incorporated to
prevent damage to the IC and the external power switch
transistors. Under low power supply conditions, it
guarantees that the IC and sensors are fully functional, and
that there is sufficient bottom drive output voltage. The
positive power supplies to the IC (VCC) and the bottom
drives ( V C) are each monitored by separate comparators that
have their thresholds at 9.1 V. This level ensures sufficient
gate drive necessary to attain low RDS(on) when driving
standard power MOSFET devices. When directly powering
the Hall sensors from the reference, improper sensor
operation can result if the reference output voltage falls
below 4.5 V. A third comparator is used to detect this
condition. If one or more of the comparators detects an
undervoltage condition, the Fault Output is activated, the top
drives are turned off and the bottom drive outputs are held
in a low state. Each of the comparators contain hysteresis to
prevent oscillations when crossing their respective
thresholds.
Fault Output
The open collector Fault Output (Pin 14) was designed to
provide diagnostic information in the event of a system
malfunction. It has a sink current capability of 16 mA and
can directly drive a light emitting diode for visual indication.
Additionally, it is easily interfaced with TTL/CMOS logic
for use in a microprocessor controlled system. The Fault
Output is active low when one or more of the following
conditions occur:
1) Invalid Sensor Input code
2) Output Enable at logic [0]
3) Current Sense Input greater than 100 mV
4) Undervoltage Lockout, activation of one or more of
the comparators
5) Thermal Shutdown, maximum junction temperature
being exceeded
This u nique o utput c an also be u sed t o d istinguish b etween
motor start−up or sustained operation in an overloaded
condition. With the addition of an RC network between the
Fault Output and the enable input, it is possible to create a
time−delayed latched shutdown for overcurrent. The added
circuitry shown in Figure 23 makes easy starting of motor
systems which have high inertial loads by providing
additional starting torque, while still preserving overcurrent
protection. This task is accomplished by setting the current
limit t o a h igher t han n ominal value for a p redetermined t ime.
During an excessively long overcurrent condition, capacitor
CDLY will charge, causing the enable input to cross its
threshold t o a l ow s tate. A l atch i s t hen formed b y t he p ositive
feedback loop from the Fault Output to the Output Enable.
Once set, by the Current Sense Input, it can only be reset by
shorting CDLY or cycling the power supplies.
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13
Drive Outputs
The three top drive outputs (Pins 1, 2, 24) are open
collector NPN transistors capable of sinking 50 mA with a
minimum breakdown of 30 V. Interfacing into higher
voltage applications is easily accomplished with the circuits
shown in Figures 24 and 25.
The three totem pole bottom drive outputs (Pins 19, 20,
21) are particularly suited for direct drive of N−Channel
MOSFETs or NPN bipolar transistors (Figures 26, 27, 28
and 29). Each output is capable of sourcing and sinking up
to 100 mA. Power for the bottom drives is supplied from VC
(Pin 18). This separate supply input allows the designer
added flexibility in tailoring the drive voltage, independent
of VCC. A zener clamp should be connected to this input
when driving power MOSFETs in systems where VCC is
greater than 20 V so as to prevent rupture of the MOSFET
gates.
The control circuitry ground (Pin 16) and current sense
inverting input (Pin 15) must return on separate paths to the
central input source ground.
Thermal Shutdown
Internal thermal shutdown circuitry is provided to protect
the IC in the event the maximum junction temperature is
exceeded. When activated, typically at 170°C, the IC acts as
though the Output Enable was grounded.
tDLY [RDLY CDLY InǒVref –(I
IL enable RDLY)
Vth enable – (IIL enable RDLY)Ǔ
Figure 23. Timed Delayed Latched
Over Current Shutdown
24
20
2
1
21
REF
UVLO
Reset
POS
DEC
4
8
3
17
22
7
6
5
14
VM
CDLY
25 μA
Load
Figure 24. High Voltage Interface with
NPN Power Transistors
T ransistor Q 1 is a common base stage used to level shift from VCC to the
high motor voltage, VM. The collector diode is required if VCC is present
while VM is low.
Q2
[RDLY CDLY Inǒ6.25 – (20 x 10–6 RDLY)
1.4 – (20 x 10–6 RDLY)Ǔ
24
20
2
1
21
Rotor
Position
Decoder
14
VM
19
Q1
VCC
Q3
Q4
RDLY
18
MC33035, NCV33035
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14
Figure 25. High Voltage Interface with
N−Channel Power MOSFETs Figure 26. Current Waveform Spike Suppression
The addition of the RC filter will eliminate current−limit instability caused by th
e
leading edge spike on the current waveform. Resistor RS should be a low in
-
ductance type.
Load
24
20
2
1
21
Rotor
Position
Decoder
14 VM = 170 V
19
VCC = 12 V
Q4
1
24
5
6
MOC8204
Optocoupler
1N4744
1.0 k
4.7 k
1.0 M
VBoost
15
20
21
19
Brake Input
23
9
RS
R
C
40 k
100 mV
Figure 27. MOSFET Drive Precautions Figure 28. Bipolar Transistor Drive
t
+
0
−
IB
Base Charge
Removal
C
C
C
Series gate resistor Rg will dampen any high frequency oscillations caused
by the MOSFET input capacitance and any series wiring induction in the
gate−source circuit. Diode D is required if the negative current into the Bot-
tom Drive Outputs exceeds 50 mA.
The totem−pole output can furnish negative base current for enhanced tran-
sistor turn−off, with the addition of capacitor C.
15
20
21
19
Brake Input
23
9
D = 1N5819
40 k
100 mV
Rg
Rg
Rg
D
D
D
15
20
21
19
Brake Input
23
9
40 k
100 mV
MC33035, NCV33035
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15
Figure 29. Current Sensing Power MOSFETs Figure 30. High Voltage Boost Supply
D
GS
RS
MK
SENSEFET
Virtually lossless current sensing can be achieved with the implementation of
SENSEFET power switches.
VPin9[
RS@Ipk @RDS(on)
rDM(on) )RS
Power Ground:
To Input Source Return
If: SENSEFET = MPT10N10M
RS = 200 Ω, 1/4 W
Then : VPin 9 ≈ 0.75 Ipk
16 Gnd
Control Circuitry Ground (Pin 16) and Current Sense Inverting Input (Pin 15)
must return on separate paths to the Central Input Source Ground.
15
20
21
19
9
100 mV
This circuit generates VBoost for Figure 25.
1.0/200 V
VBoost
*
22
1
*
1N5352A
MC1555
5
2
6
0.001 18 k
3
VM + 12
VCC = 12 V
4
VM = 170 V
R
S
Q
* = MUR115
8
Boost Current (mA)
VM + 4.0 40
760
20
VM + 8.0
Boost Voltage (V)
0
Figure 31. Differential Input Speed Controller Figure 32. Controlled Acceleration/Deceleration
R4
R2
R1
R3
13
VB
VA
REF
PWM
EA
8
7
11
12
VPin13 +VAǒR3)R4
R1)R2ǓR2
R3
*ǒR4
R3
VBǓResistor R1 with capacitor C sets the acceleration time constant while R2
controls the deceleration. The values of R1 and R2 should be at least te
n
times greater than the speed set potentiometer to minimize time constan
t
variations with different speed settings.
R1
EA
R2
8
PWM
C
Enable
Increase
Speed
7
12
11
13
REF
25 μA
25 μA
MC33035, NCV33035
http://onsemi.com
16
PWM
EA
8
7
11
The SN74LS145 is an open collector BCD to One of Ten decoder. When con-
nected as shown, input codes 0000 through 1001 steps the PWM in incre-
ments of approximately 10% from 0 to 90% on−time. Input codes 1010
through 1111 will produce 100% on−time or full motor speed.
Figure 33. Digital Speed Controller Figure 34. Closed Loop Speed Control
16
VCC
Gnd Q0
240.4 k
8
P0
BCD
Inputs
Q9
Q8
Q7
Q6
Q5
Q4
Q3
Q2
Q1
P3
P2
P1
100 k
1
51.3 k
3
4
5
6
7
63.6 k
77.6 k
92.3 k
108 k
9126 k
11
145 k
166 k
10
5.0 V
SN74LS145
REF
15
14
13
12 25 μA
13
12
13
REF
PWM
EA
8
7
11
12
The rotor position sensors can be used as a tachometer. By differentiatin
g
the positive−going edges and then integrating them over time, a voltag
e
proportional t o speed can be generated. The error amp compares this vo
lt-
age to that of the speed set to control the PWM.
0.22
1.0 M
0.1
100 k
0.01
10 k
10 k
1.0 M
To Sensor
Input (Pin 4) 25 μA
13
REF
PWM
EA
8
7
11
12
This circuit can control the speed of a cooling fan proportional to the dif ference
between the sensor and set temperatures. The control loop is closed as the
forced air cools the NTC thermistor. For controlled heating applications, ex-
change the positions of R1 and R2.
Figure 35. Closed Loop Temperature Control
T
R1
R6
R5
R2
R3
R4
VB+
Vref
ǒR5
R6
)1
Ǔ
R3§§ R5øR
6
VPin3+VrefǒR3)R4
R1)R2ǓR2
R3
*ǒR4
R3
VBǓ
25 μA
MC33035, NCV33035
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17
SYSTEM APPLICATIONS
Three Phase Motor Commutation
The three phase application shown in Figure 36 is a
full−featured open loop motor controller with full wave, six
step drive. The upper power switch transistors are
Darlingtons while the lower devices are power MOSFETs.
Each of these devices contains an internal parasitic catch
diode that is used to return the stator inductive energy back
to the power supply. The outputs are capable of driving a
delta or wye connected stator, and a grounded neutral wye
if split supplies are used. At any given rotor position, only
one top and one bottom power switch (of different totem
poles) is enabled. This configuration switches both ends of
the stator winding from supply to ground which causes the
current flow to be bidirectional or full wave. A leading edge
spike is usually present on the current waveform and can
cause a current−limit instability. The spike can be eliminated
by adding an RC filter in series with the Current Sense Input.
Using a low inductance type resistor for RS will also aid in
spike reduction. Care must be taken in the selection of the
bottom power switch transistors so that the current during
braking does not exceed the device rating. During braking,
the peak current generated is limited only by the series
resistance of the conducting bottom switch and winding.
Ipeak +VM)EMF
Rswitch )Rwinding
If the motor is running at maximum speed with no load, the
generated back EMF can be as high as the supply voltage,
and at the onset of braking, the peak current may approach
twice the motor stall current. Figure 37 shows the
commutation waveforms over two electrical cycles. The
first cycle (0° to 360°) depicts motor operation at full speed
while the second cycle (360° to 720°) shows a reduced speed
with about 50% pulse width modulation. The current
waveforms reflect a constant torque load and are shown
synchronous to the commutation frequency for clarity.
Figure 36. Three Phase, Six Step, Full Wave Motor Controller
RS
R
C
Q5
Q6
Q4
VM
S
Motor
A
Q3
S
C
B
Q1
Q2
Enable
Q
S
CT
R
RT
Oscillator
Error Amp
PWM
Thermal
Shutdown
Reference
Regulator
Lockout
Undervoltage
VM
Fwd/Rev
Q
R
S
Faster
Speed
Set
Rotor
Position
Decoder
60°/120°
Brake
4
8
3
17
22
7
6
5
18
13
11
12
10
24
20
2
1
21
14
9
19
15
Fault
Ind.
Gnd 16 23
25 μA
ILimit
N
N
MC33035, NCV33035
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18
Figure 37. Three Phase, Six Step, Full Wave Commutation Waveforms
Rotor Electrical Position (Degrees)
100 000001
011111110
100
000001011111110
720660600540
480420360300240180120600
SA
SB
SC
Code
SC
SB
Code
SA
Sensor Inputs
60°/120°
Select Pin
Open
Sensor Inputs
60°/120°
Select Pin
Grounded
AB
BB
Q2 + Q6
CB
Q2 + Q4Q3 + Q4Q3 + Q5Q1 + Q5Q1 + Q6
Bottom Drive
Outputs
Q2 + Q6Q2 + Q4Q3 + Q4Q3 + Q5
Motor Drive
Current B
Fwd/Rev = 1
C
−
O
+
−
O
+
Conducting
Power Switch
Transistors Q1 + Q5
Top Drive
Outputs
Q1 + Q6
A
BT
AT
CT
−
O
+
100 110 001011 001011110100010 010 101101
Reduced Speed ( ≈ 50% PWM)Full Speed (No PWM)
MC33035, NCV33035
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19
Figure 38 shows a three phase, three step, half wave motor
controller. This configuration is ideally suited for
automotive and other low voltage applications since there is
only one power switch voltage drop in series with a given
stator winding. Current flow is unidirectional or half wave
because only one end of each winding is switched.
Continuous braking with the typical half wave arrangement
presents a motor overheating problem since stator current is
limited only by the winding resistance. This is due to the lack
of upper power switch transistors, as in the full wave circuit,
used to disconnect the windings from the supply voltage
VM. A unique solution is to provide braking until the motor
stops and then turn off the bottom drives. This can be
accomplished b y using the Fault Output in conjunction with
the Output Enable as an over current timer. Components
RDLY and CDLY are selected to give the motor sufficient time
to stop before latching the Output Enable and the top drive
AND gates low. When enabling the motor , the brake switch
is closed and the PNP transistor (along with resistors R1 and
RDLY) are used to reset the latch by discharging CDLY. The
stator flyback voltage is clamped by a single zener and three
diodes.
Figure 38. Three Phase, Three Step, Half Wave Motor Controller
Motor
9
24
20
Q
S
CT
R
RT
Oscillator
Gnd
ILimit
Error Amp
PWM
Thermal
Shutdown
Reference
Regulator
Lockout
Undervoltage
VM
4
2
1
21
16
10
11
13
8
12
3
17
22
7
6
5
Fwd/Rev
Q
R
S
19
Faster
60°/120°
SS
VM
Speed
Set
Rotor
Position
Decoder
18
Brake
15
14
23
CDLY
RDLY
R2
R1
25 μA
N
N
MC33035, NCV33035
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20
Three Phase Closed Loop Controller
The MC33035, by itself, is only capable of open loop
motor speed control. For closed loop motor speed control,
the MC33035 requires an input voltage proportional to the
motor speed. Traditionally, this has been accomplished by
means o f a tachometer to generate the motor speed feedback
voltage. Figure 39 shows an application whereby an
MC33039, powered from the 6.25 V reference (Pin 8) of the
MC33035, i s used to generate the required feedback voltage
without the need of a costly tachometer. The same Hall
sensor signals used by the MC33035 for rotor position
decoding are utilized by the MC33039. Every positive or
negative going transition of the Hall sensor signals on any
of the sensor lines causes the MC33039 to produce an output
pulse of defined amplitude and time duration, as determined
by the external resistor R1 and capacitor C1. The output train
of pulses at Pin 5 of the MC33039 are integrated by the error
amplifier of the MC33035 configured as an integrator to
produce a DC voltage level which is proportional to the
motor speed. This speed proportional voltage establishes the
PWM reference level at Pin 13 of the MC33035 motor
controller and closes the feedback loop. The MC33035
outputs drive a TMOS power MOSFET 3−phase bridge.
High currents can be expected during conditions of start−up,
breaking, and change of direction of the motor.
The system shown in Figure 39 is designed for a motor
having 120/240 degrees Hall sensor electrical phasing. The
system can easily be modified to accommodate 60/300
degree Hall sensor electrical phasing by removing the
jumper (J2) at Pin 22 of the MC33035.
Figure 39. Closed Loop Brushless DC Motor Control
Using The MC33035 and MC33039
Motor
TP2
0.05/1.0 W
0.1 33
TP1
1.0 k
VM (18 to 30 V)
1000
0.1
1.1 k
Close Loop
0.1
1.0 M
0.01
Speed
Faster
4.7 k
F/R
Brake
1.0 k
470
470
470
1N5819
1.1 k 1.1 k
1.0 k
1
2
3
4
8
7
6
5
1
2
3
4
9
5
6
7
8
10
24
23
22
21
20
19
18
17
16
15
MC33035
MC33039
1.0 M
R1
750 pF
C1
10 k
SS
J2
100 k
100
11
12
14
13
5.1 k
Enable J1330
47 μF
1N5355B
18 V
2.2 k
0.1
1N4148
Latch On
Fault
Fault
Reset
N
N
2.2 k
MC33035, NCV33035
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21
Sensor Phasing Comparison
There are four conventions used to establish the relative
phasing of the sensor signals in three phase motors. With six
step drive, an input signal change must occur every 60
electrical degrees; however, the relative signal phasing is
dependent upon the mechanical sensor placement. A
comparison o f the conventions in electrical degrees is shown
in Figure 40. From the sensor phasing table in Figure 41,
note that the order of input codes for 60° phasing is the
reverse o f 3 0 0 °. This means the MC33035, when configured
for 60° sensor electrical phasing, will operate a motor with
either 60° or 300°sensor electrical phasing, but resulting in
opposite directions of rotation. The same is true for the part
when it is configured for 120°sensor electrical phasing; the
motor will operate equally, but will result in opposite
directions of rotation for 120°for 240° conventions.
Figure 40. Sensor Phasing Comparison
Rotor Electrical Position (Degrees)
300°
240°
720660600540480420360300240180120600
SB
SA
120°
60°
SC
SA
SB
SC
SC
SB
SA
SC
SB
SA
Sensor Electrical Phasing
Sensor Electrical Phasing (Degrees)
60°120°240°300°
SASBSCSASBSCSASBSCSASBSC
100101110111
110100100110
1111101011 0 0
011010001000
001011011001
000001010011
Figure 41. Sensor Phasing Table
In this data sheet, the rotor position is always given in
electrical degrees since the mechanical position is a function
of the number of rotating magnetic poles. The relationship
between the electrical and mechanical position is:
Electrical Degrees +Mechanical Degreesǒ#Rotor Poles
2Ǔ
An increase in the number of magnetic poles causes more
electrical revolutions for a given mechanical revolution.
General purpose three phase motors typically contain a four
pole rotor which yields two electrical revolutions for one
mechanical.
Two and Four Phase Motor Commutation
The MC33035 is also capable of providing a four step
output that can be used to drive two or four phase motors.
The truth table in Figure 42 shows that by connecting sensor
inputs SB and SC together, it is possible to truncate the
number of drive output states from six to four. The output
power switches are connected to BT, CT, BB, and CB.
Figure 43 shows a four phase, four step, full wave motor
control application. Power switch transistors Q1 through Q8
are Darlington type, each with an internal parasitic catch
diode. With four step drive, only two rotor position sensors
spaced at 90 electrical degrees are required. The
commutation waveforms are shown in Figure 44.
Figure 45 shows a four phase, four step, half wave motor
controller. I t has the same features as the circuit in Figure 38,
except for the deletion of speed control and braking.
MC33035 (60°/120° Select Pin Open)
Inputs Outputs
Sensor Electrical
Spacing* = 90°
Top Drives Bottom Drives
SASBF/R BTCTBBCB
1
1
0
0
0
1
1
0
1
1
1
1
1
0
1
1
1
1
0
1
0
0
0
1
1
0
0
0
1
1
0
0
0
1
1
0
0
0
0
0
1
1
1
0
0
1
1
1
0
1
0
0
0
0
1
0
*With MC33035 sensor input SB connected to SC.
Figure 42. Two and Four Phase, Four Step,
Commutation Truth Table
MC33035, NCV33035
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22
CT
RT
VM
Enable
Fwd/Rev
10
13
12
11
8
17
7
22
3
6
5
4
9
19
20
Lockout
21
Rotor
Undervoltage
24
1
Motor
2
Reference
Thermal
Oscillator
Gnd 16
Q5
Q1
Q2
Q6
Q7
Q3
Q4
Q8
VM
R
CRS
Position
Decoder
Shutdown
Regulator
Error Amp
PWM
ILimit
S
RQ
R
S
Q
A
B
D
C
S
S
18
15
14
23
25 A
μ
Fault
Ind.
N
N
Figure 43. Four Phase, Four Step, Full Wave Motor Controller
MC33035, NCV33035
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23
Conducting
Power Switch
Transistors
A
SA
SB
Code
Q3 + Q5
Rotor Electrical Position (Degrees)
Fwd/Rev = 1
−
O
+
−
−
D
C
+
O
−
O
+
B
+
CB
O
BB
CT
BT
Q2 + Q8
Q1 + Q7
Q4 + Q6
Q3 + Q5
Q2 + Q8
Q1 + Q7
Q4 + Q6
0001111000011010
Motor Drive
Current
Bottom Drive
Outputs
Top Drive
Outputs
Sensor Inputs
60°/120°
Select Pin
Open
180 270 360 450 540 630 720090
Figure 44. Four Phase, Four Step, Full Wave Motor Controller
Full Speed (No PWM)
MC33035, NCV33035
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24
R
VM
RS
Motor
S
N
CT
RT
VM
Enable
Fwd/Rev
10
13
12
11
8
17
7
22
3
6
5
4
9
19
20
Lockout
21
Rotor
Undervoltage
24
1
2
Reference
Thermal
Oscillator
Gnd 16
Position
Decoder
Shutdown
Regulator
Error Amp
PWM
ILimit
S
RQ
R
S
Q
18
15
14
23
Brake
25 A
μ
Fault
Ind.
C
N
S
Figure 45. Four Phase, Four Step, Half Wave Motor Controller
MC33035, NCV33035
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25
Brush Motor Control
Though the MC33035 was designed to control brushless
DC motors, it may also be used to control DC brush type
motors. Figure 46 shows an application of the MC33035
driving a MOSFET H−bridge affording minimal parts count
to operate a brush−type motor. Key to the operation is the
input sensor code [100] which produces a top−left (Q1) and
a bottom−right (Q3) drive when the controller’s
forward/reverse pin i s a t l ogic [ 1]; t op−right ( Q4), b ottom−left
(Q2) drive is realized when the Forward/Reverse pin is at
logic [0]. This code supports the requirements necessary for
H−bridge drive accomplishing both direction and speed
control.
The controller functions in a normal manner with a pulse
width modulated frequency of approximately 25 kHz.
Motor speed is controlled by adjusting the voltage presented
to the noninverting input of the error amplifier establishing
the PWM’s slice or reference level. Cycle−by−cycle current
limiting of the motor current is accomplished by sensing the
voltage (100 mV) across the RS resistor to ground of the
H−bridge motor current. The over current sense circuit
makes i t possible to reverse the direction of the motor, using
the normal forward/reverse switch, on the fly and not have
to completely stop before reversing.
LAYOUT CONSIDERATIONS
Do not attempt to construct any of the brushless motor
control circuits on wire−wrap or plug−in prototype
boards. High frequency printed circuit layout techniques
are imperative to prevent pulse jitter. This is usually caused
by excessive noise pick−up imposed on the current sense or
error amp inputs. The printed circuit layout should contain
a ground plane with low current signal and high drive and
output bu ffer grounds returning on separate paths back to the
power supply input filter capacitor VM. Ceramic bypass
capacitors (0.1 μF) connected close to the integrated circuit
at VCC, VC, Vref and the error amp noninverting input may
be required depending upon circuit layout. This provides a
low impedance path for filtering any high frequency noise.
All high current loops should be kept as short as possible
using heavy copper runs to minimize radiated EMI.
MC33035, NCV33035
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26
9
24
20
Q
S
0.005
R
10 k
Oscillator
Gnd
ILimit
Error Amp
PWM
Thermal
Shutdown
Reference
Regulator
Lockout
Undervoltage
+12 V
4
2
1
21
16
10
11
13
8
12
3
17
22
7
6
5
Fwd/Rev
Q
R
S
19
Faster
Rotor
Position
Decoder
18
Brake
15
14
23
25 μA
Figure 46. H−Bridge Brush−Type Controller
RS
1.0 k
0.001
22
22
DC Brush
Motor M
+12 V
1.0 k
1.0 k
Q1*
Q2*
Q4*
Q3*
Enable
10 k
Fault
Ind.
20 k
MC33035, NCV33035
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27
ORDERING INFORMATION
Device Operating Temperature Range Package Shipping†
MC33035DW
TA = −40°C to +85°C
SO−24L 30 Units / Rail
MC33035DWG SO−24L
(Pb−Free) 30 Units / Rail
MC33035DWR2 SO−24L 1000 Tape & Reel
MC33035DWR2G SO−24L
(Pb−Free) 1000 Tape & Reel
MC33035P Plastic DIP 15 Units / Tube
MC33035PG Plastic DIP
(Pb−Free) 15 Units / Tube
NCV33035DWR2* TA = −40°C to +125°CSO−24L 1000 Tape & Reel
NCV33035DWR2G* SO−24L
(Pb−Free) 1000 Tape & Reel
†For information on tape and reel specifications, including part orientation and tape sizes, please refer to our Tape and Reel Packaging
Specification Brochure, BRD8011/D.
*NCV33035: T low = −40C, Thigh = +125C. Guaranteed by design. NCV prefix is for automotive and other applications requiring site and change
control.
PDIP−24
P SUFFIX
CASE 724
1
24
MC33035P
AWLYYWWG
MARKING DIAGRAMS
A = Assembly Location
WL = Wafer Lot
YY = Year
WW = Work Week
G = Pb−Free Package
24
1
MC33035DW
AWLYYWWG
SO−24
DW SUFFIX
CASE 751E
24
1
NCV33035DW
AWLYYWWG
MC33035, NCV33035
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28
PACKAGE DIMENSIONS
P SUFFIX
PLASTIC PACKAGE
CASE 724−03
ISSUE D
MIN MINMAX MAX
INCHES MILLIMETERS
DIM
1.265
0.270
0.175
0.020
0.060
0.012
0.140
15°
0.040
0.050 BSC
0.100 BSC
0.300 BSC
1.27 BSC
2.54 BSC
7.62 BSC
A
B
C
D
E
F
G
J
K
L
M
N
31.25
6.35
3.69
0.38
1.02
0.18
2.80
0°
0.51
32.13
6.85
4.44
0.51
1.52
0.30
3.55
15°
1.01
1.230
0.250
0.145
0.015
0.040
0.007
0.110
0°
0.020
NOTES:
1. CHAMFERED CONTOUR OPTIONAL.
2. DIMENSION L TO CENTER OF LEADS WHEN
FORMED PARALLEL.
3. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
4. CONTROLLING DIMENSION: INCH.
112
1324
-A-
-B-
C
K
N
-T-
SEATING
PLANE
GEF
D 24 PL
J 24 PL
M
NOTE 1
L
0.25 (0.010) T A
M M
0.25 (0.010) T B
M M
DW SUFFIX
PLASTIC PACKAGE
CASE 751E−04
(SO−24L)
ISSUE E
T
0.010 (0.25) A B
MS S MIN MINMAX MAX
MILLIMETERS INCHES
DIM
A
B
C
D
F
G
J
K
M
P
R
15.25
7.40
2.35
0.35
0.41
0.23
0.13
0°
10.05
0.25
15.54
7.60
2.65
0.49
0.90
0.32
0.29
8°
10.55
0.75
0.601
0.292
0.093
0.014
0.016
0.009
0.005
0°
0.395
0.010
0.612
0.299
0.104
0.019
0.035
0.013
0.011
8°
0.415
0.029
1.27 BSC 0.050 BSC
NOTES:
1. DIMENSIONING AND TOLERANCING PER ANSI
Y14.5M, 1982.
2. CONTROLLING DIMENSION: MILLIMETER.
3. DIMENSIONS A AND B DO NOT INCLUDE
MOLD PROTRUSION.
4. MAXIMUM MOLD PROTRUSION 0.15 (0.006)
PER SIDE.
5. DIMENSION D DOES NOT INCLUDE DAMBAR
PROTRUSION. ALLOWABLE DAMBAR
PROTRUSION SHALL BE 0.13 (0.005) TOTAL IN
EXCESS OF D DIMENSION AT MAXIMUM
MATERIAL CONDITION.
-A-
-B-
112
24 13
-T-
C
K
SEATING
PLANE
R X 45°
G 22 PL
P 12 PL
0.010 (0.25) B
M M
F
J
M
D 24 PL
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