LMV797MMX/NOPB - NATIONAL SEMICONDUCTOR

LMV796,LMV797

LMV796/LMV796Q/LMV797 17 MHz, Low Noise, CMOS Input, 1.8V Operational

Amplifiers

Literature Number: SNOSAU9C

LMV796/LMV796Q/LMV797

February 11, 2011

17 MHz, Low Noise, CMOS Input, 1.8V Operational

Amplifiers

General Description

The LMV796/LMV796Q (Single) and the LMV797 (Dual) low

noise, CMOS input operational amplifiers offer a low input

voltage noise density of 5.8 nV/ while consuming only

1.15 mA (LMV796/LMV796Q) of quiescent current. The

LMV796/LMV796Q and LMV797 are unity gain stable op

amps and have gain bandwidth of 17 MHz. The LMV796/

LMV796Q/LMV797 have a supply voltage range of 1.8V to

5.5V and can operate from a single supply. The LMV796/

LMV796Q/LMV797 each feature a rail-to-rail output stage ca-

pable of driving a 600Ω load and sourcing as much as 60 mA

of current.

The LMV796/LMV796Q family provides optimal performance

in low voltage and low noise systems. A CMOS input stage,

with typical input bias currents in the range of a few femtoAm-

peres, and an input common mode voltage range, which

includes ground, make the LMV796/LMV796Q and the

LMV797 ideal for low power sensor applications.

The LMV796/LMV796Q/LMV797 are manufactured using

National’s advanced VIP50 process. The LMV796/LMV796Q

are offered in 5–pin SOT23 package. The LMV797 is offered

in 8–pin MSOP package.

Features

(Typical 5V supply, unless otherwise noted)

■Input referred voltage noise 5.8 nV/

■Input bias current 100 fA

■Unity gain bandwidth 17 MHz

■Supply current per channel

—LMV796/LMV796Q 1.15 mA

—LMV797 1.30 mA

■Rail-to-rail output swing

—@ 10 kΩ load 25 mV from rail

—@ 2 kΩ load 45 mV from rail

■Guaranteed 2.5V and 5.0V performance

■Total harmonic distortion 0.01% @ 1kHz, 600Ω

■Temperature range −40°C to 125°C

■LMV796Q is an Automotive grade product that is AEC-

Q100 grade 1 qualified and is manufactured on an auto-

motive grade flow.

Applications

■Photodiode amplifiers

■Active filters and buffers

■Low noise signal processing

■Medical instrumentation

■Sensor interface applications

■Automotive

Typical Application

20183569

Photodiode Transimpedance Amplifier 20183539

Input Referred Voltage Noise vs. Frequency

LMV796/LMV796Q/LMV797 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers

Absolute Maximum Ratings (Note 1)

If Military/Aerospace specified devices are required,

please contact the National Semiconductor Sales Office/

Distributors for availability and specifications.

ESD Tolerance (Note 2)

Human Body Model 2000V

Machine Model 200V

Charge-Device Model 1000V

VIN Differential ±0.3V

Supply Voltage (V+ – V−)6.0V

Input/Output Pin Voltage V+ +0.3V, V− −0.3V

Storage Temperature Range −65°C to 150°C

Junction Temperature (Note 3) +150°C

Soldering Information

Infrared or Convection (20 sec) 235°C

Wave Soldering Lead Temp (10 sec) 260°C

Operating Ratings (Note 1)

Temperature Range (Note 3) −40°C to 125°C

Supply Voltage (V+ – V−)

−40°C ≤ TA ≤ 125°C 2.0V to 5.5V

0°C ≤ TA ≤ 125°C 1.8V to 5.5V

Package Thermal Resistance (θJA) (Note 3)

5-Pin SOT-23 180°C/W

8-Pin MSOP 236°C/W

2.5V Electrical Characteristics

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 2.5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at

the temperature extremes.

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 4)

Max

(Note 5)Units

VOS Input Offset Voltage 0.1 ±1.35

±1.65 mV

TC VOS Input Offset Voltage Temperature Drift LMV796/LMV796Q (Note 6) −1.0 μV/°C

LMV797 (Note 6) −1.8

IBInput Bias Current VCM = 1.0V

(Note 7, Note 8)

−40°C ≤ TA ≤ 85°C 0.05 1

25 pA

−40°C ≤ TA ≤ 125°C 0.05 1

100

IOS Input Offset Current VCM = 1.0V

(Note 8) 10 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 1.4V 80

94 dB

PSRR Power Supply Rejection Ratio

2.0V ≤ V+ ≤ 5.5V, VCM = 0V 80

100

1.8V ≤ V+ ≤ 5.5V, VCM = 0V 80 98

CMVR Common Mode Voltage Range CMRR ≥ 60 dB

CMRR ≥ 55 dB

−0.3

-0.3

1.5

1.5 V

AVOL Open Loop Voltage Gain

VOUT = 0.15V to 2.2V,

RLOAD = 2 kΩ to V+/2

LMV796/LMV796Q 85

LMV797 82

VOUT = 0.15V to 2.2V,

RLOAD = 10 kΩ to V+/2

110

VOUT

Output Voltage Swing High

RLOAD = 2 kΩ to V+/2 25 75

mV from

either rail

RLOAD = 10 kΩ to V+/2 20 65

Output Voltage Swing Low

RLOAD = 2 kΩ to V+/2 30 75

RLOAD = 10 kΩ to V+/2 15 65

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LMV796/LMV796Q/LMV797

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 4)

Max

(Note 5)Units

IOUT Output Current

Sourcing to V−

VIN = 200 mV (Note 9)

Sinking to V+

VIN = –200 mV (Note 9)

ISSupply Current per Amplifier

LMV796/LMV796Q 0.95 1.30

1.65 mA

LMV797

per channel

1.1 1.50

1.85

SR Slew Rate AV = +1, Rising (10% to 90%) 8.5 V/μs

AV = +1, Falling (90% to 10%) 10.5

GBW Gain Bandwidth 14 MHz

enInput Referred Voltage Noise Density f = 1 kHz 6.2 nV/

inInput Referred Current Noise Density f = 1 kHz 0.01 pA/

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RLOAD = 600Ω 0.01 %

5V Electrical Characteristics

Unless otherwise specified, all limits are guaranteed for TA = 25°C, V+ = 5V, V− = 0V, VCM = V+/2 = VO. Boldface limits apply at

the temperature extremes.

Symbol Parameter Conditions Min

(Note 5)

Typ

(Note 4)

Max

(Note 5)Units

VOS Input Offset Voltage 0.1 ±1.35

±1.65 mV

TC VOS Input Offset Voltage Temperature Drift LMV796/LMV796Q (Note 6) −1.0 μV/°C

LMV797 (Note 6) −1.8

IBInput Bias Current VCM = 2.0V

(Note 7, Note 8)

−40°C ≤ TA ≤ 85°C 0.1 1

25 pA

−40°C ≤ TA ≤ 125°C 0.1 1

100

IOS Input Offset Current VCM = 2.0V

(Note 8)

10 fA

CMRR Common Mode Rejection Ratio 0V ≤ VCM ≤ 3.7V 80

100 dB

PSRR Power Supply Rejection Ratio

2.0V ≤ V+ ≤ 5.5V, VCM = 0V 80

100

1.8V ≤ V+ ≤ 5.5V, VCM = 0V 80 98

CMVR Common Mode Voltage Range CMRR ≥ 60 dB

CMRR ≥ 55 dB

−0.3

-0.3

AVOL Open Loop Voltage Gain

VOUT = 0.3V to 4.7V,

RLOAD = 2 kΩ to V+/2

LMV796/LMV796Q 85

LMV797 82

VOUT = 0.3V to 4.7V,

RLOAD = 10 kΩ to V+/2

110

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LMV796/LMV796Q/LMV797

VOUT

Output Voltage Swing High

RLOAD = 2 kΩ to V+/2 35 75

mV from

either rail

RLOAD = 10 kΩ to V+/2 25 65

Output Voltage Swing Low

RLOAD = 2 kΩ to V+/2

LMV796/LM796Q 42 75

LMV797 45 80

RLOAD = 10 kΩ to V+/2 20 65

IOUT Output Current

Sourcing to V−

VIN = 200 mV (Note 9)

Sinking to V+

VIN = –200 mV (Note 9)

ISSupply Current per Amplifier

LMV796/LMV796Q 1.15 1.40

1.75 mA

LMV797per channel 1.30 1.70

2.05

SR Slew Rate AV = +1, Rising (10% to 90%) 6.0 9.5 V/μs

AV = +1, Falling (90% to 10%) 7.5 11.5

GBW Gain Bandwidth 17 MHz

enInput Referred Voltage Noise Density f = 1 kHz 5.8 nV/

inInput Referred Current Noise Density f = 1 kHz 0.01 pA/

THD+N Total Harmonic Distortion + Noise f = 1 kHz, AV = 1, RLOAD = 600Ω 0.01 %

Note 1: Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Operating Ratings indicate conditions for which the device is

intended to be functional, but specific performance is not guaranteed. For guaranteed specifications and the test conditions, see the Electrical Characteristics

tables.

Note 2: Human Body Model is 1.5kΩ in series with 100pF. Machine Model is 0Ω in series with 200pF.

Note 3: The maximum power dissipation is a function of TJMAX, θJA. The maximum allowable power dissipation at any ambient temperature is

PD = (TJMAX - TA) / θJA. All numbers apply for packages soldered directly onto a PC Board.

Note 4: Typical values represent the parametric norm at the time of characterization.

Note 5: Limits are 100% production tested at 25°C. Limits over the operating temperature range are guaranteed through correlations using the statistical quality

control (SQC) method.

Note 6: Offset voltage average drift is determined by dividing the change in VOS by temperature change.

Note 7: Positive current corresponds to current flowing into the device.

Note 8: This parameter is guaranteed by design and/or characterization and is not tested in production.

Note 9: The short circuit test is a momentary test, the short circuit duration is 1.5ms.

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LMV796/LMV796Q/LMV797

Connection Diagrams

5-Pin SOT23

20183501

Top View

8-Pin MSOP

20183502

Top View

Ordering Information

Package Part Number Package Marking Transport Media NSC Drawing

5-Pin SOT23 LMV796MF AT3A 1k Units Tape and Reel MF05A

LMV796MFX 3k Units Tape and Reel

5-Pin SOT23 LMV796QMF AD7A 1k Units Tape and Reel MF05A

LMV796QMFX 3k Units Tape and Reel

8-Pin MSOP LMV797MM AU3A 1k Units Tape and Reel MUA08A

LMV797MMX 3.5k Units Tape and Reel

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LMV796/LMV796Q/LMV797

Typical Performance Characteristics Unless otherwise specified, TA = 25°C, V– = 0, V+ = Supply Voltage

= 5V, VCM = V+/2.

Supply Current vs. Supply Voltage (LMV796/LMV796Q)

20183505

Supply Current vs. Supply Voltage (LMV797)

20183581

VOS vs. VCM

20183509

VOS vs. VCM

20183551

VOS vs. VCM

20183511

VOS vs. Supply Voltage

20183512

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LMV796/LMV796Q/LMV797

Slew Rate vs. Supply Voltage

20183529

Input Bias Current vs. VCM

20183562

Input Bias Current vs. VCM

20183587

Sourcing Current vs. Supply Voltage

20183520

Sinking Current vs. Supply Voltage

20183519

Sourcing Current vs. Output Voltage

20183550

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LMV796/LMV796Q/LMV797

Sinking Current vs. Output Voltage

20183554

Positive Output Swing vs. Supply Voltage

20183517

Negative Output Swing vs. Supply Voltage

20183515

Positive Output Swing vs. Supply Voltage

20183516

Negative Output Swing vs. Supply Voltage

20183514

Positive Output Swing vs. Supply Voltage

20183518

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LMV796/LMV796Q/LMV797

Negative Output Swing vs. Supply Voltage

20183513

Time Domain Voltage Noise

20183582

Input Referred Voltage Noise vs. Frequency

20183539

Overshoot and Undershoot vs. CLOAD

20183530

THD+N vs. Peak-to-Peak Output Voltage (VOUT)

20183526

THD+N vs. Peak-to-Peak Output Voltage (VOUT)

20183504

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LMV796/LMV796Q/LMV797

THD+N vs. Frequency

20183574

THD+N vs. Frequency

20183575

Open Loop Gain and Phase with Capacitive Load

20183541

Open Loop Gain and Phase with Resistive Load

20183573

Closed Loop Output Impedance vs. Frequency

20183532

Crosstalk Rejection

20183580

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LMV796/LMV796Q/LMV797

Small Signal Transient Response, AV = +1

20183538

Large Signal Transient Response, AV = +1

20183537

Small Signal Transient Response, AV = +1

20183533

Large Signal Transient Response, AV = +1

20183534

Phase Margin vs. Capacitive Load (Stability)

20183545

Phase Margin vs. Capacitive Load (Stability)

20183546

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LMV796/LMV796Q/LMV797

Positive PSRR vs. Frequency

20183527

Negative PSRR vs. Frequency

20183528

CMRR vs. Frequency

20183556

Input Common Mode Capacitance vs. VCM

20183576

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LMV796/LMV796Q/LMV797

Application Information

ADVANTAGES OF THE LMV796/LMV797

Wide Bandwidth at Low Supply Current

The LMV796 and LMV797 are high performance op amps that

provide a unity gain bandwidth of 17 MHz while drawing a low

supply current of 1.15 mA. This makes them ideal for provid-

ing wideband amplification in portable applications.

Low Input Referred Noise and Low Input Bias Current

The LMV796/LMV797 have a very low input referred voltage

noise density (5.8 nV/ at 1 kHz). A CMOS input stage en-

sures a small input bias current (100 fA) and low input referred

current noise (0.01 pA/ ). This is very helpful in maintain-

ing signal fidelity, and makes the LMV796 and LMV797 ideal

for audio and sensor based applications.

Low Supply Voltage

The LMV796 and the LMV797 have performance guaranteed

at 2.5V and 5V supply. The LMV796 family is guaranteed to

be operational at all supply voltages between 2.0V and 5.5V,

for ambient temperatures ranging from −40°C to 125°C, thus

utilizing the entire battery lifetime. The LMV796 and LMV797

are also guaranteed to be operational at 1.8V supply voltage,

for temperatures between 0°C and 125°C. This makes the

LMV796 family ideal for usage in low-voltage commercial ap-

plications.

RRO and Ground Sensing

Rail-to-rail output swing provides maximum possible dynamic

range at the output. This is particularly important when oper-

ating at low supply voltages. An innovative positive feedback

scheme is used to boost the current drive capability of the

output stage. This allows the LMV796 and the LMV797 to

source more than 40 mA of current at 1.8V supply. This also

limits the performance of the LMV796 family as comparators,

and hence the usage of the LMV796 and the LMV797 in an

open-loop configuration is not recommended. The input com-

mon-mode range includes the negative supply rail which

allows direct sensing at ground in single supply operation.

Small Size

The small footprint of the LMV796 and the LMV797 package

saves space on printed circuit boards, and enables the design

of smaller electronic products, such as cellular phones,

pagers, or other portable systems. Long traces between the

signal source and the op amp make the signal path suscep-

tible to noise. By using the physically smaller LMV796 or

LMV797 package, the op amp can be placed closer to the

signal source, reducing noise pickup and increasing signal

integrity.

CAPACITIVE LOAD TOLERANCE

The LMV796 and LMV797 can directly drive 120 pF in unity-

gain without oscillation. The unity-gain follower is the most

sensitive configuration to capacitive loading. Direct capacitive

loading reduces the phase margin of amplifiers. The combi-

nation of the amplifier’s output impedance and the capacitive

load induces phase lag. This results in either an under-

damped pulse response or oscillation. To drive a heavier

capacitive load, the circuit in Figure 1 can be used.

In Figure 1, the isolation resistor RISO and the load capacitor

CL form a pole to increase stability by adding more phase

margin to the overall system. The desired performance de-

pends on the value of RISO. The bigger the RISO resistor value,

the more stable VOUT will be. Increased RISO would, however,

result in a reduced output swing and short circuit current.

20183561

FIGURE 1. Isolation of CL to Improve Stability

INPUT CAPACITANCE AND FEEDBACK CIRCUIT

ELEMENTS

The LMV796 family has a very low input bias current (100 fA)

and a low 1/f noise corner frequency (400 Hz), which makes

it ideal for sensor applications. However, to obtain this per-

formance a large CMOS input stage is used, which adds to

the input capacitance of the op amp, CIN. Though this does

not affect the DC and low frequency performance, at higher

frequencies the input capacitance interacts with the input and

the feedback impedances to create a pole, which results in

lower phase margin and gain peaking. This can be controlled

by being selective in the use of feedback resistors, as well as,

by using a feedback capacitance, CF. For example, in the in-

verting amplifier shown in Figure 2, if CIN and CF are ignored

and the open loop gain of the op amp is considered infinite

then the gain of the circuit is −R2/R1. An op amp, however,

usually has a dominant pole, which causes its gain to drop

with frequency. Hence, this gain is only valid for DC and low

frequency. To understand the effect of the input capacitance

coupled with the non-ideal gain of the op amp, the circuit

needs to be analyzed in the frequency domain using a

Laplace transform.

20183564

FIGURE 2. Inverting Amplifier

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LMV796/LMV796Q/LMV797

For simplicity, the op amp is modeled as an ideal integrator

with a unity gain frequency of A0 . Hence, its transfer function

(or gain) in the frequency domain is A0/s. Solving the circuit

equations in the frequency domain, ignoring CF for the mo-

ment, results in an expression for the gain shown in Equation

(1)

It can be inferred from the denominator of the transfer function

that it has two poles, whose expressions can be obtained by

solving for the roots of the denominator and are shown in

Equation 2.

(2)

Equation 2 shows that as the values of R1 and R2 are in-

creased, the magnitude of the poles, and hence the band-

width of the amplifier, is reduced. This theory is verified by

using different values of R1 and R2 in the circuit shown in

Figure 1 and by comparing their frequency responses. In Fig-

ure 3 the frequency responses for three different values of

R1 and R2 are shown. When both R1 and R2 are 1 kΩ, the

response is flattest and widest; whereas, it narrows and peaks

significantly when both their values are changed to 10 kΩ or

30 kΩ. So it is advisable to use lower values of R1 and R2 to

obtain a wider and flatter response. Lower resistances also

help in high sensitivity circuits since they add less noise.

20183559

FIGURE 3. Gain Peaking Caused by Large R1, R2

A way of reducing the gain peaking is by adding a feedback

capacitance CF in parallel with R2. This introduces another

pole in the system and prevents the formation of pairs of com-

plex conjugate poles which cause the gain to peak. Figure 4

shows the effect of CF on the frequency response of the cir-

cuit. Adding a capacitance of 2 pF removes the peak, while a

capacitance of 5 pF creates a much lower pole and reduces

the bandwidth excessively.

20183560

FIGURE 4. Gain Peaking Eliminated by CF

AUDIO PREAMPLIFIER WITH BAND PASS FILTERING

With low input referred voltage noise, low supply voltage and

current, and a low harmonic distortion, the LMV796 family is

ideal for audio applications. Its wide unity gain bandwidth al-

lows it to provide large gain for a wide range of frequencies

and it can be used to design a preamplifier to drive a load of

as low as 600Ω with less than 0.01% distortion. Two amplifier

circuits are shown in Figure 5 and Figure 6. Figure 5 is an

inverting amplifier, with a 10 kΩ feedback resistor, R2, and a

1kΩ input resistor, R1, and hence provides a gain of −10.

Figure 6 is a non-inverting amplifier, using the same values

of R1and R2, and provides a gain of 11. In either of these cir-

cuits, the coupling capacitor CC1 decides the lower frequency

at which the circuit starts providing gain, while the feedback

capacitor CF decides the frequency at which the gain starts

dropping off. Figure 7 shows the frequency response of the

inverting amplifier with different values of CF.

20183565

FIGURE 5. Inverting Audio Preamplifier

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LMV796/LMV796Q/LMV797

20183566

FIGURE 6. Non-inverting Audio Preamplifier

20183558

FIGURE 7. Frequency Response of the Inverting

Audio Preamplifier

TRANSIMPEDANCE AMPLIFIER

CMOS input op amps are often used in transimpedance ap-

plications as they have an extremely high input impedance.

A transimpedance amplifier converts a small input current into

a voltage. This current is usually generated by a photodiode.

The transimpedance gain, measured as the ratio of the output

voltage to the input current, is expected to be large and wide-

band. Since the circuit deals with currents in the range of a

few nA, low noise performance is essential. The LMV796/

LMV797 are CMOS input op amps providing wide bandwidth

and low noise performance, and are hence ideal for tran-

simpedance applications.

Usually, a transimpedance amplifier is designed on the basis

of the current source driving the input. A photodiode is a very

common capacitive current source, which requires tran-

simpedance gain for transforming its miniscule current into

easily detectable voltages. The photodiode and the

amplifier’s gain are selected with respect to the speed and

accuracy required of the circuit. A faster circuit would require

a photodiode with lesser capacitance and a faster amplifier.

A more sensitive circuit would require a sensitive photodiode

and a high gain. A typical transimpedance amplifier is shown

in Figure 8. The output voltage of the amplifier is given by the

equation VOUT = −IINRF. Since the output swing of the amplifier

is limited, RF should be selected such that all possible values

of IIN can be detected.

The LMV796/LMV797 have a large gain-bandwidth product

(17 MHz), which enables high gains at wide bandwidths. A

rail-to-rail output swing at 5.5V supply allows detection and

amplification of a wide range of input currents. A CMOS input

stage with negligible input current noise and low input voltage

noise allows the LMV796/LMV797 to provide high fidelity am-

plification for wide bandwidths. These properties make the

LMV796/LMV797 ideal for systems requiring wide-band tran-

simpedance amplification.

20183569

FIGURE 8. Photodiode Transimpedance Amplifier

As mentioned earlier, the following parameters are used to

design a transimpedance amplifier: the amplifier gain-band-

width product, A0; the amplifier input capacitance, CCM; the

photodiode capacitance, CD; the transimpedance gain re-

quired, RF; and the amplifier output swing. Once a feasible

RF is selected using the amplifier output swing, these num-

bers can be used to design an amplifier with the desired

transimpedance gain and a maximally flat frequency re-

sponse.

An essential component for obtaining a maximally flat re-

sponse is the feedback capacitor, CF. The capacitance seen

at the input of the amplifier, CIN, combined with the feedback

capacitor, RF, generate a phase lag which causes gain-peak-

ing and can destabilize the circuit. CIN is usually just the sum

of CD and CCM. The feedback capacitor CF creates a pole,

fP in the noise gain of the circuit, which neutralizes the zero in

the noise gain, fZ, created by the combination of RF and CIN.

If properly positioned, the noise gain pole created by CF can

ensure that the slope of the gain remains at 20 dB/decade till

the unity gain frequency of the amplifier is reached, thus en-

suring stability. As shown in Figure 9, fP is positioned such

that it coincides with the point where the noise gain intersects

the op amp’s open loop gain. In this case, fP is also the overall

−3 dB frequency of the transimpedance amplifier. The value

of CF needed to make it so is given by Equation 3. A larger

value of CF causes excessive reduction of bandwidth, while

a smaller value fails to prevent gain peaking and instability.

(3)

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LMV796/LMV796Q/LMV797

20183584

FIGURE 9. CF Selection for Stability

Calculating CF from Equation 3 can sometimes return unrea-

sonably small values (<1 pF), especially for high speed ap-

plications. In these cases, it is often more practical to use the

circuit shown in Figure 10 in order to allow more reasonable

values. In this circuit, the capacitance CF′ is (1+ RB/RA) times

the effective feedback capacitance, CF. A larger capacitor can

now be used in this circuit to obtain a smaller effective ca-

pacitance.

For example, if a CF of 0.5 pF is needed, while only a 5 pF

capacitor is available, RB and RA can be selected such that

RB/RA = 9. This would convert a CF′ of 5 pF into a CF of

0.5 pF. This relationship holds as long as RA << RF.

20183571

FIGURE 10. Obtaining Small CF from Large CF′

LMV796 AS A TRANSIMPEDANCE AMPLIFIER

The LMV796 was used in the designs for a number of ampli-

fiers with varying transimpedance gains and source capaci-

tances. The gains, bandwidths and feedback capacitances of

the circuits created are summarized in Table 1. The frequency

responses are presented in Figure 11 and Figure 12. The

feedback capacitances are slightly different from the formula

in Equation 3, since the parasitic capacitance of the board and

the feedback resistor RF had to be accounted for.

TABLE 1.

Transimpedance, ATI CIN CF−3 dB Frequency

470000 50 pF 1.5 pF 350 kHz

470000 100 pF 2.0 pF 250 kHz

470000 200 pF 3.0 pF 150 kHz

47000 50 pF 4.5 pF 1.5 MHz

47000 100 pF 6.0 pF 1 MHz

47000 200 pF 9.0 pF 700 kHz

20183577

FIGURE 11. Frequency Response for ATI = 470000

20183578

FIGURE 12. Frequency Response for ATI = 47000

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LMV796/LMV796Q/LMV797

HIGH GAIN WIDEBAND TRANSIMPEDANCE AMPLIFIER

USING THE LMV797

The LMV797 dual, low noise, wide bandwidth, CMOS input

op amp IC can be used for compact, robust and integrated

solutions for sensing and amplifying wide-band signals ob-

tained from sensitive photodiodes. One of the two op amps

available can be used to obtain transimpedance gain while

the other can be used for amplifying the output voltage to fur-

ther enhance the transimpedance gain. The wide bandwidth

of the op amps (17 MHz) ensures that they are capable of

providing high gain for a wide range of frequencies. The low

input referred noise (5.8 nV/ ) allows the amplifier to de-

liver an output with a high SNR (signal to noise ratio). The

small 8-pin MSOP footprint saves space on printed circuit

boards and allows ease of design in portable products.

The circuit shown in Figure 13, has the first op amp acting as

a transimpedance amplifier with a gain of 47000, while the

second stage provides a voltage gain of 10. This provides a

total transimpedance gain of 470000 with a −3 dB bandwidth

of about 1.5 MHz, for a total input capacitance of 50 pF. The

frequency response for the circuit is shown in Figure 14

20183586

FIGURE 13. 1.5 MHz Transimpedance Amplifier

with ATI = 470000

20183579

FIGURE 14. 1.5 MHz Transimpedance Amplifier

Frequency Response

SENSOR INTERFACES

The low input bias current and low input referred noise of the

LMV796 and LMV797 make them ideal for sensor interfaces.

These circuits are required to sense voltages of the order of

a few μV and currents amounting to less than a nA hence, the

op amp needs to have low voltage noise and low input bias

current. Typical applications include infra-red (IR) thermom-

etry, thermocouple amplifiers and pH electrode buffers. Fig-

ure 15 is an example of a typical circuit used for measuring

IR radiation intensity, often used for estimating the tempera-

ture of an object from a distance. The IR sensor generates a

voltage proportional to I, which is the intensity of the IR radi-

ation falling on it. As shown in Figure 15, K is the constant of

proportionality relating the voltage across the IR sensor (VIN)

to the radiation intensity, I. The resistances RA and RB are

selected to provide a high gain to amplify this voltage, while

CF is added to filter out the high frequency noise.

20183572

FIGURE 15. IR Radiation Sensor

17 www.national.com

LMV796/LMV796Q/LMV797

Physical Dimensions inches (millimeters) unless otherwise noted

5-Pin SOT-23

NS Package Number MF05A

8-Pin MSOP

NS package Number MUA08A

www.national.com 18

LMV796/LMV796Q/LMV797

Notes

19 www.national.com

LMV796/LMV796Q/LMV797

Notes

LMV796/LMV796Q/LMV797 17 MHz, Low Noise, CMOS Input, 1.8V Operational Amplifiers

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