September 1999 - Linear Technology

LINEAR TECHNOLOGY

SEPTEMBER 1999 VOLUME IX NUMBER 3

Four New Amplifiers Serve

Many Applications

, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,

FilterCAD, Hot Swap, LinearView, Micropower SwitcherCAD, No Latency ∆Σ, No R

SENSE

, Operational Filter, OPTI-LOOP,

PolyPhase, PowerSOT, SwitcherCAD and UltraFast are trademarks of Linear Technology Corporation. Other product

names may be trademarks of the companies that manufacture the products.

IN THIS ISSUE…

Four New Amplifiers

Serve Many Applications ................. 1

Issue Highlights .............................. 2

LTC

in the News… .......................... 2

DESIGN FEATURES

JFET Op Amps Equal Low Noise

Bipolars and Have Picoamp

Current Noise .................................. 3

Alexander Strong

1813: 100MHz, 750V/

s Amplifier

Draws Only 3mA ............................. 5

George Feliz

Micropower, Precision Current Sense

Ampliﬁer Operates from 2.5V to 60V

....................................................... 7

Richard Markell, Glen Brisebois and

Jim Mahoney

Triple 300MHz Current Feedback

Ampliﬁers Drive RGB/Component

Video and LCD Displays ................ 12

Brian Hamilton

LTC1702/LTC1703 Switching

Regulator Controllers Set a New

Standard for Transient Response

..................................................... 16

Dave Dwelley

An 8-Channel, High-Accuracy,

No Latency

∆Σ

™ 24-Bit ADC ........... 21

Michael K. Mayes

Micropower 5-Lead SOT-23 Switching

Regulators Extend Battery Life in

Space-Sensitive Applications ........ 24

Bryan Legates

Versatile Ring Tone Generator Finds

Uses in Motor Drivers and Amplifiers

..................................................... 28

Dale Eagar

DESIGN IDEAS

35 Watt Isolated DC/DC Converter

Replaces Modules at Half the Cost

..................................................... 30

Robert Sheehan

Comparator Circuit Provides

Automatic Shutdown of the LT1795

High Speed ADSL Power Ampliﬁer

..................................................... 33

Tim Regan

(More Design Ideas on pages 35–37;

complete list on page 30)

New Device Cameos ....................... 38

Design Tools ................................. 39

Sales Offices ................................. 40

This issue of Linear Technology

features a quartet of exciting new

amplifiers from LTC: The LT1792/

LT1793 low noise JFET op amps, the

LT1813 high slew rate op amp, the

LT1787 high-side current sense

amplifier and the LT1399/LT1399HV

triple current feedback amplifiers.

The LT1792/LT1793 single JFET

op amps offer both very low voltage

noise and low current noise, provid-

ing the lowest total noise over a wide

range of transducer impedances.

These op amps are unconditionally

stable for gains of one or more, even

with capacitive loads of 1000pF. Their

low offset voltage and high DC gain

allow the LT1792/LT1793 to ﬁt into

precision applications, especially

those involving high impedance,

capacitive transducers.

The LT1813 is a 100MHz, 750V/µs

dual operational ampliﬁer. Requiring

only 3mA of supply current, it uses

LTC’s advanced, low voltage comple-

mentary bipolar process and a few

design tricks to exceed the perfor-

mance of its older siblings. A key

ﬁgure of merit for amplifiers is the

ratio of gain bandwidth to supply

current (expressed as MHz/mA). The

new process employed by the LT1813

forsakes high supply voltage operation

for a 3×–4× increase in MHz/mA.

Blazing speed from such a modest

amount of supply current (3mA) is

extremely attractive for low power

applications. The LT1813 extends the

frequency response of applications

such as ﬁlters, instrumentation

ampliﬁers and buffers.

The LT1787 employs precision

technology to build a superior high-

side current sense amplifier. The

LT1787 will find uses in cellular

phones, portable instruments and

wireless telecom devices for precisely

monitoring the current into or out of

a battery. The LT1787 monitors bidi-

rectional currents via the voltage

across an external sense resistor. It

features a minute input offset voltage

of 40µV with a full-scale input of up to

500mV. This translates to a dynamic

range of over 12 bits. The LT1787HV

features a 60V maximum input, which

allows it to be used in telecom and

industrial applications that require

the sensing of higher voltages. The

device is self-powered from the sup-

ply that it monitors and requires only

60µA supply current.

For video and computer display

applications, LTC introduces the

LT1399/LT1399HV triple current

feedback ampliﬁers. These devices

contain three independent 300MHz

CFAs, each with a shutdown pin.

Each CFA has 0.1dB gain ﬂatness of

150MHz and a slew rate of 800V/µs.

The LT1399 operates on supplies from

4V to ±6V. The LT1399HV operates

on supplies ranging from 4V to ±7.5V.

Each ampliﬁer can be enabled in 30ns

and disabled in 40ns, making them

ideal in spread-spectrum and por-

table equipment applications. With

the addition of a small series resistor,

the parts can drive large capacitive

loads. This feature, combined with

the LT1399HV’s high voltage opera-

tion, makes it ideal for driving LCD

displays.

Linear Technology Magazine • September 1999

EDITOR’S PAGE

Issue Highlights

In addition to the four new ampli-

fiers described on page one, this issue

of Linear Technology features a vari-

ety of power and data converter

products.

In the power area, we introduce the

LTC1702/LTC1703. The LTC1702 is

the ﬁrst in a new family of low voltage,

high speed switching regulator con-

trollers. It is designed to operate from

the standard 5V logic rail and gener-

ate two lower voltage, high current

regulated outputs. Running at a ﬁxed

550kHz switching frequency, each

side of the LTC1702 features a volt-

age feedback architecture using a

25MHz gain-bandwidth op amp as

the feedback ampliﬁer, allowing loop-

crossover frequencies in excess of

50kHz. Large onboard MOSFET driv-

ers allow the LTC1702 to drive high

current external MOSFETs efﬁciently

at 550kHz and beyond, allowing the

use of small external inductors and

capacitors while maintaining excel-

lent output ripple and transient

response. The LTC1703 is a modiﬁed

LTC1702 with a 5-bit DAC control-

ling the output voltage at side 1. The

DAC conforms to the Intel Mobile VID

speciﬁcation.

Two other new switchers debuted

in this issue are the micropower

LT1615 and LT1617. These devices

can be used in a number of topolo-

gies, including boost, SEPIC and

positive-to-negative. With an input

voltage range of 1.2V to 15V, they are

ideal for a wide range of applications

and work with a variety of input

sources. An internal 36V switch al-

lows them to easily generate output

voltages of up to ±34V without the

use of costly transformers. The

LT1615 is designed to regulate posi-

tive output voltages, whereas the

LT1617 is designed to directly regu-

late negative output voltages without

the need for level-shifting circuitry.

Both parts use a current-limited, ﬁxed

off-time control scheme, which helps

achieve high efﬁciency operation over

a wide range of load currents. Both

devices use tiny, low proﬁle inductors

and capacitors to minimize the over-

all system footprint and cost.

A new ∆Σ ADC is also featured in

this issue: the LTC2408 is the off-

spring of the LTC2400 24-bit ∆Σ

converter, introduced in our Novem-

ber 1998 issue. The

LTC2408

combines the high performance

LTC2400 with an 8-channel analog

input multiplexer. The resulting de-

vice offers many unique features. The

single-cycle settling characteristics

lead to simpliﬁed multiplexer hook-

up and channel selection, without the

added overhead seen with other

converters. The exceptional noise per-

formance of the device eliminates the

need for a programmable gain amp-

liﬁer (PGA).

The unprecedented 10ppm

absolute accuracy of the device allows

measurement of microvolt signals

superimposed upon large DC volt-

ages. A wide range of sensor inputs

and voltage levels can be applied si-

multaneously to the LTC2408.

Our final Design Feature spotlights

some unusual applications for the

LT1684 ring tone generator, intro-

duced in the June issue. The LT1684

was speciﬁcally designed for OEM

telephone equipment. Because of the

versatility of the device, the LT1684

ring tone chip ﬁnds itself at home in

motor drives, digital input ampliﬁed

speakers, alarm systems and sine

wave UPS systems.

Our Design Ideas section features

an isolated 35 watt DC/DC converter

designed to replace “half-brick” power

modules at half the cost and a

comparator circuit that provides

automatic shutdown for the LT1795

high speed ADSL power amplifier.

We conclude with a trio of New

Device Cameos.

LTC in the News…

On July 20, Linear Technology Cor-

poration announced its financial

results for fiscal year 1999. Robert

H. Swanson, Chairman & CEO,

stated, “We had record levels of

sales and profits as we finished the

year strongly. As we commence our

first fiscal year in the new millen-

nium, we are encouraged by the

increasing opportunity for high

performance analog circuits.” The

Company reported net sales for the

fourth quarter of $140,524,000 (a

6% increase over net sales for the

fourth quarter of the previous year).

Net income for the fourth quarter

was $54,179,000 compared with

$49,503,000 a year ago. For the

year, the Company’s net sales were

$506,669,000 (up 5% over the pre-

vious year), with net income of

$194,293,000 (up 7% versus the

prior year).

The Company was also featured

in “America’s Favorite Stocks,” an

article in the July issue of Money

magazine that detailed which stocks

are most popular among invest-

ment clubs in the United States.

Linear Technology ranked 94th in

terms of number of clubs holding

the stock, and 68th in terms of the

value of Linear Technology stock

held by clubs.

Linear Technology was included

in Business Week’s annual Global

1,000 rankings. The Company

ranked 289th among U.S. compa-

nies and 570th among companies

internationally.

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

http://www.linear-tech.com/ezone/zone.html

Articles, Design Ideas, Tips from the Lab…

Linear Technology Magazine • September 1999

DESIGN FEATURES

JFET Op Amps Equal Low Noise

Bipolars and Have Picoamp

Current Noise by Alexander Strong

The LT1792 and LT1793 are single

JFET op amps that offer both very low

voltage noise (4nV/√Hz for the LT1792

and 6nV/√Hz for the LT1793) and low

current noise (10fA/√Hz for the

LT1792 and 0.8fA/√Hz for the

LT1793), providing the lowest total

noise over a wide range of transducer

impedance. Traditionally, op amp

users have been faced with a choice:

which op amp will have the lowest

noise for the transducer at hand. For

high transducer impedance, the

LT1792/LT1793 JFET op amps will

win over the lowest voltage noise

bipolar op amps due to lower current

noise. The current noise (2qI

) of an

ampliﬁer is a function of the input

bias current (I

). For lower trans-

ducer impedance, bipolar op amps

usually win over typical JFET op amps

due to lower voltage noise for the

same tail current of the differential

input pair. The LT1792/LT1793 op

amps are designed to have voltage

noise that approaches that of bipolar

op amps. All of these op amps are

unconditionally stable for gains of

one or more, even with capacitive

loads of 1000pF. The low offset volt-

age of 250µV and high DC gain of four

million allow the LT1792/LT1793 to

ﬁt into precision applications. Voltage

noise, slew rate and gain-bandwidth

product are 100% tested. All of the

speciﬁcations are maintained in the

SO-8 package.

The combination of low voltage and

current noise offered by the LT1792/

LT1793 makes them useful in a wide

range of applications, especially with

high impedance, capacitive transduc-

ers such as hydrophones, precision

accelerometers and photo diodes. The

total noise in such systems is the gain

times the square root of the sum of

the op amp input–referred voltage

noise squared, the thermal noise of

the transducer (4kTR) and the op

amp’s bias current noise times the

transducer resistance squared (2qI

× R

). Figure 1 shows total input

voltage noise versus source resistance.

In a low source resistance application

(<5k), the op amp’s voltage noise will

dominate the total noise. In this region

of low source resistance, the LT1792/

LT1793 JFET op amps are way ahead

of other JFET op amps; only very low

noise bipolar op amps such as the

LT1007 and LT1028 have the edge.

As source resistance increases from

5k to 50k, the LT1792/LT1793 will

match the best bipolar or JFET op

amp for noise performance, since the

thermal noise of the transducer (4kTR)

will dominate the total noise. A fur-

ther increase in source resistance, to

above 50k, brings us to the region

where the op amp’s current noise

(2qI

× R

SOURCE

) will dominate the

total noise. At these high source

resistances, the LT1792/LT1793 will

outperform the lowest noise bipolar

op amp due to the inherently low

current noise of FET input op amps.

In some conditions it may be neces-

sary to add a capacitor in parallel

with a source resistor to cancel the

pole that is caused by the source

impedance and the input capacitance

(14pF for the LT1792 and 1.5pF for

the LT1793). Observe what happens

to noise with source resistances over

100k; the overall noise for the LT1792

and LT1793 actually decreases.

The high input impedance JFET

front end makes the LT1792 and

LT1793 suitable for applications

where very high charge sensitivity is

required. Figure 2 illustrates the

LT1792 and LT1793 in inverting and

noninverting modes of operation. A

charge ampliﬁer is shown in the

inverting mode example; here the gain

depends on the principle of charge

conservation at the input of the

ampliﬁer. The charge across the trans-

ducer capacitance, C

, is transferred

to the feedback capacitor, C

, result-

ing in a change in voltage, dV, equal to

dQ/C

, resulting in a gain of C

For unity gain, the C

should equal

the transducer capacitance plus the

input capacitance of the ampliﬁer and

should equal R

. In the non-

inverting mode example, the

transducer current is converted to a

change in voltage by the transducer

capacitance; this voltage is then buff-

TOTAL SOURCE RESISTANCE (Ω)

100

10k

1k 100M 1G

LT1169 • F02

100k

100

10M10k 1M

INPUT NOISE VOLTAGE (nV/√Hz)

= A

n2(OP AMP)

+ 4kTR

+ 2

SOURCE RESISTANCE = 2R

= R

* PLUS RESISTOR

†

PLUS RESISTOR  1000pF CAPACITOR

RESISTOR NOISE ONLY

LT1793

LT1007*

LT1007

†

LT1792/

†

LT1793

–

LT1792

LT1007

LT1792*

(AND RESISTOR

NOISE)

LT1793*

retemaraP2971TL3971TLstinU

)xaM(65.037.0Vm

)xaM(05401Ap

)zHk1(2.46 zH√/Vn

)zHk1(018.0zH√/Af

PWBG

)zHk001= 65 zHM

2.42.4Am

Table 1. LT1792/LT1793 specifications

Figure 1. Comparison of LT1792/LT1793 and

LT1007 input voltage noise vs source

resistance

Linear Technology Magazine • September 1999

DESIGN FEATURES

ered by the ampliﬁer, with a gain of

1␣ + R2/R1. A DC path is provided by

, which is either the transducer

impedance or an external resistor.

Since R

is usually several orders of

magnitude greater than the parallel

combination of R1 and R2, R

is added

to balance the DC offset caused by

the noninverting input bias current

and R

. The input bias currents,

although small at room temperature,

can create signiﬁcant errors over

increasing temperature, especially

with transducer resistances of up to

1000M or more. The optimum value

for R

is determined by equating the

thermal noise (4kTR

) to the current

noise times R

, (2qI

) R

, resulting in

= 2V

= kT/q = 26mV at

25°C). A parallel capacitor, C

, is used

to cancel the phase shift caused by

the op amp input capacitance and R

The LT1792 has the lowest voltage

noise (4nV/√Hz) of the two, which

makes it the best choice for trans-

ducer impedances of 5k or less. For

transducer impedances over 100M,

the LT1793, with a typical input bias

current of only 3pA, will have lower

output noise than the LT1792. The

LT1793 has the additional advantage

of very high input resistance (10

ohms). Unlike most JFET op amps,

the LT1792 and LT1793 have input

bias currents that remains almost

constant over the entire common

mode range. The speciﬁcations for

the LT1792 and LT1793 are summa-

rized in Table 1.

The low noise of the LT1792 and

LT1793 is achieved by maximizing

the g

of the input pair. The polygate

JFETs have a higher g

-to-area ratio

than standard, single-gate JFETs.

This is done by maximizing the tail

current and the size of the input JFET

geometeries. Forty percent of the total

supply current is used as the tail

current for the LT1792 and LT1793.

These op amps are best used with

very high impedance transducers. The

low noise hydrophone ampliﬁer in

Figure 3 is an application where the

LT1792 excels. The AC current out-

put of the hydrophone is converted to

a voltage output by the 100M input

resistor (R8). This signal is ampliﬁed

OUTPUT

CSRSCB ≅ CS

RB = RS

RS > R1 OR R2

TRANSDUCER

OUTPUT

CBRB

CB = CFCS

RB = RFRS

CSRS

TRANSDUCER

–

LT1792 LT1792

Q = CV; = I = C

dt dV

–

5V TO 15V

–5V TO –15V

0.47µF

100M R6

100k

LT1792

1/2

LT1464

3.9k

C1*

R1*

100M

200Ω

HYDRO-

PHONE

OUTPUT

DC OUTPUT ≤ 2.5mV FOR T

< 70°C

OUTPUT VOLTAGE NOISE = 128nV/√Hz AT 1kHz (GAIN = 20)

C1 ≈ C

≈ 100pF TO 5000pF; R4C2 > R8C

; *OPTIONAL

by the R3/R2 ratio. DC leakage cur-

rents at the output of the hydrophone

are subtracted by the servo action of

the feedback ampliﬁer. This ampliﬁer

need not have the low voltage noise of

the LT1792; therefore, it can be cho-

sen to minimize the overall system

supply current. The LT1464 has less

than an order of magnitude of supply

current of the LT1792 and LT1793

and picoampere input bias current.

This allows the time constant of this

loop to be set using high value resis-

tors and less expensive low value

capacitors.

The LT1792 and LT1793 op amps

are in a class by themselves when

amplifying low level signals from high

impedance sources. The design and

process have been optimized to pro-

duce both low power consumption

and low current and voltage noise.

Most competing JFET op amps will

have higher voltage noise or much

higher supply current. Practically all

bipolar op amps will have higher cur-

rent noise. No other op amp will deliver

the noise performance for a given

supply current. For applications

where low noise and power are issues,

the LT1792 and LT1793 are the best

choices.

Figure 2. LT1792/LT1793 inverting and noninverting gain conﬁgurations

Figure 3. Low noise hydrophone ampliﬁer with DC servo

Authors can be contacted

at (408) 432-1900

Linear Technology Magazine • September 1999

DESIGN FEATURES

LT1813: 100MHz, 750V/

s Amplifier

Draws Only 3mA by George Feliz

Introduction

The LT1813 is a 100MHz dual opera-

tional ampliﬁer that has been

optimized for supply voltages under

12V. It features an easy-to-use voltage

feedback topology with high imped-

ance inputs, yet it slews 750V/µs

with only 3mA supply current. DC

performance has not been neglected

—the device has a 1.5mV maximum

and a 400nA maximum I

Performance

A summary of important speciﬁca-

tions of the LT1813, compared to its

higher voltage brethren, is shown in

Table 1. A key ﬁgure of merit is the

ratio of gain bandwidth to supply

current (GBW/I

SUPPLY

, expressed in

units of MHz/mA). The new process

employed by the LT1813 forsakes high

supply voltage operation for a 3×–4×

increase in MHz/mA compared to the

LT1361 and LT1364. Blazing speed

from such a modest amount of supply

current is extremely attractive for low

power applications. The LT1813 also

propagates the family traits of

matched, high input impedance in-

puts and low V

, I

and input

noise. The improved common mode

input range of the LT1813 adds to its

utility in low supply voltage applica-

tions. Stability with capacitive load-

ing is another distinctive and desirable

feature. Although the LT1813 is not

stable with unlimited capacitive loads,

it is stable with nearly two orders of

magnitude more capacitance than

competitors’ high speed ampliﬁers.

The small-signal transient response

in unity gain with C

LOAD

=100pF,

500pF and 1000pF is shown in

Figure␣ 1.

The LT1813 extends the frequency

response of applications such as active

ﬁlters, instrumentation ampliﬁers and

buffers. Figure 2 shows the LT1813

converting a single-ended signal to a

differential drive for the LTC1417

14-bit analog-to-digital converter

(ADC). Note that the top ampliﬁer

provides unity voltage gain, but the

ampliﬁer is conﬁgured in a noise-gain

of 2 to match the phase response of

the bottom ampliﬁer, which has a

gain of –1. The ﬁlter in front of the

ADC reduces broadband noise. The

spurious free dynamic range (SFDR)

of this circuit is –79dB for a 425kHz,

P-P

input.

Circuit Design

A simpliﬁed schematic of the circuit

is shown in Figure 3. The circuit looks

similar to a current feedback ampli-

ﬁer, but both inputs are high

100pF

500pF

1000pF

100mV/DIV

200ns/DIV

3181TL4631TL1631TL

htdiwdnaBniaG zHM001zHM05zHM73

reifilpmAreptnerruCylppuS Am0.3Am0.6Am8.3

/WBGI

YLPPUS

Am/zHM3.33Am/zHM3.8Am/zHM7.9

etaRwelS sµ/V057sµ/V054sµ/V053

egnaRedoMnommoCtupnI V0.4±V2.3–,V4.3+V2.3–,V4.3+

gniwStuptuO V0.4±V1.4±V0.4±

tnerruCtuptuO Am06Am54Am83

)xaM( Vm5.1Vm5.1Vm0.1

)xaM( Aµ0.4Aµ0.2Aµ0.1

)xaM( An004An053An052

LOV

)niM( Vm/V5.1Vm/V5.3Vm/V3

egatloVesioNtupnI zH/Vn8zH/Vn9zH/Vn9

tneruCesioNtupnI zH/Ap1zH/Ap1zH/Ap9.0

DAOL

Fp0001 ∞∞

(egatloVylppuSxaMV

otV

–

) V6.21V63V63

Table 1. Comparison of dual, high speed op amps (V

= ±5V, 25°C)

OUT

= ±3V)

Figure 1. LT1813 in a gain-of-one conﬁguration, no R

; C

= 100pF, 500pF or 1000pF

Linear Technology Magazine • September 1999

DESIGN FEATURES

impedance as in a traditional voltage

feedback ampliﬁer. A complementary

cascade of emitter followers, Q1–Q4,

buffers the noninverting input and

drives one side of resistor R1. The

other side of the resistor is driven by

Q5–Q8, which form a buffer for the

inverting input. The input voltage

appears across the resistor, generat-

ing currents in Q3 and Q4 that are

mirrored by Q9–Q11 and Q13–Q15

into the high impedance node. Tran-

sistors Q17–Q24 form the output

stage. Bandwidth is set by R1, the

’s of Q3, Q4, Q7 and Q8 and the

compensation capacitor, C

The voltage drops of Q1–Q4 and

the diodes Q10 and Q14 set the input

common mode range of the ampliﬁer.

The emitters of Q3 and Q4 follow the

noninverting input. As the input

approaches either supply rail, the

limiting voltage is determined by the

saturation of Q3 or Q4, which occurs

at approximately a V

plus a V

SAT

from the supply rail. Typically, the

input common mode range is 1V from

either supply rail, and is guaranteed

by the CMRR speciﬁcation to be 1.5V

from either rail. This excellent input

range is achieved without compro-

mising the output impedance of the

mirrors Q9–Q11 and Q13–Q15,

because Q25 and Q26 provide ﬂoat-

ing bias points for cascode devices Q9

and Q13. Lower bandwidth processes

cannot successfully use this tech-

nique and maintain high bandwidth,

due to phase shift in the mirror.

The current available to slew com-

pensation capacitor C

is proportional

to the voltage that appears across R1.

This method of “slew boost” achieves

low distortion due to its inherent lin-

earity with input step size. Large slew

currents can be generated without

increasing quiescent current. A low

value for R1 reduces the input noise

voltage to 8nV/√Hz and helps reduce

input offset voltage and drift. The

LT1813 is built with small-geometry,

multi-GHz transistors that produce

abundant bandwidth with meager

operating currents and allow for fur-

ther reduction of idling supply current.

The output stage buffers the high

impedance node from the load by

–

1k 100Ω

100Ω

500pF

LT1813

LTC1417 SERIAL

OUT

14 BITS

V+

–IN

V–

Q5 Q6

Q2 Q1 +IN

Q10 Q25

Q14 Q26

Q13

Q15

Q11

Q12

Q17

Q18

Q16

Q22

Q24

Q19

Q21Q20

Q23

RCCC

OUT

providing current gain. The simplest

output stage would be two pairs of

complementary emitter followers,

which would provide a current gain of

Beta

NPN

× Beta

PNP

. Unfortunately, this

gain is insufﬁcient for driving even

modest loads. Adding another emitter-

follower or a Darlington conﬁguration

reduces output swing and creates

instability with large capacitive loads.

The solution used on the LT1813

was to create a pair of composite

transistors formed by transistors

Q19–Q21 and Q22–Q24. The current

mirrors attached to the collectors of

emitter followers Q19 and Q22 pro-

vide additional current gain. The ratio

of transistor geometries Q20 to Q21

and Q23 to Q24 increase the current

gain by approximately ﬁfteen. There

Figure 2. Single-ended to differential ADC buffer: 2V

P-P

input at 425kHz yields –79dB SFDR

Figure 3. LT1813 simpliﬁed schematic

continued on page 15

Linear Technology Magazine • September 1999

DESIGN FEATURES

Micropower, Precision Current Sense

Ampliﬁer Operates from 2.5V to 60V

Introduction

The LT1787 is a precision, high-side

current sense ampliﬁer designed for

monitoring of the current either into

or out of a battery or other element

capable of sourcing or sinking cur-

rent. The LT1787 features a miniscule

40µV (typical) input offset voltage with

a 128mV full-scale input voltage. (The

part is generally used at ±128mV full-

scale, although it is specified for

500mV minimum full-scale.) This

translates to a 12-bit dynamic range

in resolving currents. A hefty 60V

maximum input voltage speciﬁcation

allows the part to be used not only in

low voltage battery applications but

also in telecom and industrial appli-

cations where higher voltages may be

present.

The device is self-powered from the

supply that it is monitoring and

requires only 60µA of supply current.

The power supply rejection ratio of

the LT1787 is in excess of 130dB.

The LT1787 allows the use of a

user-selectable sense resistor, the

value of which depends on the cur-

rent to be monitored. This allows the

input voltage to be optimized to the

128mV value, maximizing dynamic

range. The part has a ﬁxed voltage

gain of eight from input to output.

Additional LT1787 features include

provisions for input noise ﬁltering

(both differential and common mode)

and the ability to operate over a very

wide supply range of 2.5V to 60V. The

part is available in both 8-lead SO

and MSOP packages.

Operation of the LT1787

Figure 1 shows a function diagram of

the LT1787 integrated circuit. When

current is ﬂowing from V

S+

to V

S–

, a

sense voltage of V

SENSE

= I

SENSE

•

SENSE

is generated. Because ampli-

ﬁer A1’s positive and negative inputs

are forced equal by feedback, V

SENSE

also appears across the R

side with

the higher V

potential. Hence, for the

situation where V

S+

is greater than

S–

, V

SENSE

appears across R

G2A

and

G2B

. This current ﬂows through Q2

and becomes the output current, I

OUT

Q1 is kept off by the ampliﬁer and

does not contribute to I

OUT

For split-supply operation, where

S+

and V

range from ±2.5V to

±30V; with V

BIAS

at ground, V

OUT

becomes (I

OUT

• R

OUT

). In this case,

and for the above described direction

of sense current, V

OUT

is positive or

“above” V

BIAS

; thus Q2 is sourcing

current.

When current ﬂows from V

S–

S+

, input V

S–

is at a higher potential

than V

S+

, so V

SENSE

will appear across

G1A

and R

G1B

. The current, I

OUT

SENSE

/(R

G1A

+ R

G1B

), will be conducted

through Q1, while Q2 remains off.

OUT

then duplicates itself through a

one-to-one current mirror at V

OUT

is negative or “below” V

BIAS

; Q1

sources current and the mirror sinks

the current at the V

OUT

node.

The output voltage across R

OUT

related to the input sense voltage by

the following relationships:

VOUT – VBIAS = IOUT • ROUT

VSENSE = ISENSE • RSENSE

IOUT = VSENSE/RG,

RG = RG1A + RG1B = RG2A + RG2B

RG(TYP) = 2.5k

VOUT – VBIAS = (ROUT) (VSENSE)/

RG, ROUT/RG = 8

ROUT(TYP) = 20k

VOUT – VBIAS = 8 (VSENSE),

VSENSE = VS+ – VS–

VOUT = 8 (VSENSE) + VBIAS

Selection of R

SENSE

Maximum sense current can vary for

each application. To sense the widest

dynamic range, it is necessary to select

an external sense resistor that ﬁts

each application. 12-bit dynamic

range performance can be achieved

regardless of the maximum current to

be sensed, whether it is 10mA or 10A.

The correct R

SENSE

value is derived so

that the product of the maximum

sense current and the sense resistor

value is equal to the desired maxi-

mum sense voltage (usually 128mV).

For instance, the value of the sense

resistor to sense a maximum current

of 10mA is 128mV/10mA = 12.8Ω.

Since the LT1787 is capable of 12-bit

resolution, the smallest measurable

current is 10mA/4096 counts =

2.44µA/count. In terms of sense volt-

age, this translates to 128mV/4096

= 31.25µV/count. Other current

ranges can be accommodated by a

simple change in value in the sense

resistor. Care should also be taken to

ensure that the power dissipated in

the sense resistor, I

MAX.2

• R

SENSE

does not exceed the maximum power

rating of the resistor.

by Richard Markell, Glen Brisebois and Jim Mahoney

SENSE

1787 F 01

G2A

G1A

OUT

BIAS

OUT

S–

–+

Q1 Q2

1:1 MIRROR

FIL

–

S+

FIL

Figure 1. LT1787 function diagram

Linear Technology Magazine • September 1999

DESIGN FEATURES

Application Circuits

Dual-Supply,

Bidirectional Current Sense

Figure 2 shows the schematic dia-

gram of the LT1787 operated with

dual supplies. This circuit can sense

current in either direction, positive

current ﬂow being from V

S+

to V

S–

where the output ranges from 0V to

1.024V for V

SENSE

= 0mV to 128mV,

and negative current ﬂow being from

S–

to V

S+

, where the output varies

from 0V to –1.024V for V

SENSE

= 0mV

to –128mV. Figure 3 shows the out-

put voltage versus V

SENSE

for this

conﬁguration.

Operation with Bias

If a negative supply is not available, a

voltage reference may be connected

to the V

BIAS

pin of the LT1787. The

value of the reference is not critical, it

simply biases the output of the part to

a new “zero” point (for V

SENSE

= 0V);

zero is now at V

BIAS

= V

REF

, which, for

the case of Figure 4, is equal to 1.25V.

This conﬁguration can be used for

both unipolar and bipolar current

sensing, with V

OUT

ranging either

“above” or “below” V

BIAS

, depending

on the direction of the current ﬂow.

This can be seen from the graph shown

in Figure 5. (Note that the reference

must be able to both source and sink

1µF

–5V

SENSE

15V

OUTPUT

LT1787

FIL

S+

BIAS

OUT

FIL

–

S–

DNC

1000pF

CHARGER/

LOAD

Figure 2. Split-supply, bidirectional operation

1µF

RBIAS

20k

RSENSE

3.3V

60V 3.3V

LT1787HV

FIL+

VS+

VBIAS

VOUT

FIL–

VS–

DNC

VEE LT1634-1.25

1000pF

CHARGER/

LOAD

OUTPUT

Figure 4. Bidirectional operation with a reference

–

FIL

–

S+

DNC

FIL

S–

BIAS

OUT

LTC1404

LT1495

LT1634-1.25

36k

100pF

1Ω

LT1787

Figure 6. Operation with a buffer

SENSE VOLTAGE (VS+ – VS–) (mV)

OUTPUT VOLTAGE (V)

1.5

1.0

0.5

–0.5

–1.0

–1.5

–128 –64 032

–96 –32 64 96 128

VS = ±1.65V TO ±30V

TA = –40°C TO 85°C

Figure 3. V

OUT

vs V

SENSE

in bipolar mode

SENSE VOLTAGE (V

S+

– V

S–

) (mV)

OUTPUT VOLTAGE (V)

2.75

2.25

1.75

1.25

0.75

0.25

–0.25

–128 –64 032

–96 –32 64 96 128

= 3.3V TO 60V

= –40°C TO 85°C

Figure 5. V

OUT

vs V

SENSE

in bipolar mode

4500

4000

3500

3000

2500

2000

1500

1000

500

0–200 –100 0 100

COUNTS

INPUT CURRENT (mA)

SINGLE 5V SUPPLY

REF

= 1.25V

I–V CONVERTER

USING INTERNAL R

OUT

–250 –150 –50 50

Figure 7. Output counts vs input current

for Figure 6’s circuit

Linear Technology Magazine • September 1999

DESIGN FEATURES

current from the V

BIAS

pin—refer to

the block diagram in Figure 1.)

Operation with a Buffer

Figure 6 uses a rail-to-rail op amp,

the LT1495, as an I/V converter to

buffer the LT1787’s output. The

LT1634-1.25 reference is used to bias

the LT1787’s output so that zero cur-

rent is now represented by a 1.25V

output. This allows the device to moni-

tor current in either direction while

the circuit operates on a single sup-

ply. This also allows lower voltage

operation, since V

OUT

of the LT1787 is

held constant by the op amp. Figure

7 shows input current versus output

counts (from the LTC1404 A/D con-

verter), showing excellent linearity.

Single-Supply Current Sense

The circuit in Figure 8 provides good

accuracy near full-scale, but has a

limited dynamic range. In this circuit,

the LT1787 is operated from a single

supply of 2.5V minimum to 60V maxi-

mum. Current is allowed to ﬂow

through R

SENSE

in both directions but

is measured in a single direction only,

with current ﬂow from V

S+

to V

S–

. In

this connection, V

BIAS

and V

are

grounded. The output voltage (V

OUT

)

of the LT1787 for this circuit is equal

to 8 • V

SENSE

, for V

SENSE

= V

S+

– V

S–

0mV to 128mV. The dynamic range

limits of this circuit can be seen in the

graph shown in Figure 9.

Operation with

an A/D Converter

Figure 10 shows the details of an

LT1787 connected to a LTC1404 12-

bit serial A/D converter. Details of the

circuit are similar to those shown

previously in Figures 2 and 8 and in

the text detailing these circuits. The

main difference in the applications is

that the circuits in Figures 2 and 8

provide an analog output voltage pro-

portional to the current, whereas the

circuit shown in Figure 10 digitizes

that analog voltage to provide a digi-

tal output. Figures 11 and 12 show

the output of the LTC1404 A/D con-

verter. The data in Figure 11 was

collected with V

operated from –5V;

in other words, both the LT1787 and

the LTC1404 used a negative supply

of –5V. Similarly, the data in Figure

12 was taken with V

connected to

ground.

Connecting the optional section of

the schematic (still operating the cir-

cuit from a single supply) allows the

A/D’s reference to “bias up” the

LT1787 exactly as shown in Figure 5.

Of course, the graph of the output

would then be recast as similar to

Figure 11 (counts versus V

SENSE

Auto Shutdown

Linear Regulator

Figure 13 shows the details of a linear

regulator with high-side current

sensing and latched shutdown capa-

bility. The circuit shuts down power

to the load when the current reaches

its overcurrent trip point. Power can

then be restored only by cycling the

main power off and on again. This

circuit features the LT1787 and the

LTC1440 precision comparator with

on-chip reference. The LT1528 is a 3A

low dropout linear regulator with

shutdown.

The circuit uses the LT1787, U1,

as a precision current sensor; the

gain of the LT1787 allows the use of a

0.05Ω sense resistor, which dissi-

pates a mere 0.312W of power. The

LTC1440 ultralow power compara-

tor, U2, with its internal reference, is

used as a precision trigger, followed

by a 74HC00 connected as an RS ﬂip-

ﬂop (U3C and U3D) to latch the error

condition until power is removed and

reapplied. The other two NAND gates

0.1µF

SENSE

OUT

2.5V TO

60V

LOAD

LT1787HV

FIL

S+

BIAS

OUT

FIL

–

S–

DNC

OUTPUT VOLTAGE (V)

0.30

0.25

0.20

0.15

0.10

0.05

0IDEAL

SENSE

(mV)

0 5 10 15 20 25 30

FIL

–

S+

DNC

FIL

S–

OUT

BIAS

4LTC1404

1Ω 1%

= 5V

LT1787

10k

10µF

16V

10µF

16V

10µF

16V

CONV

CLK

OUT

CLOCKING

CIRCUITRY

OPTIONAL:

CONNECT TO V

BIAS

;

DISCONNECT V

BIAS

FROM GROUND

REF

GND

OUT

(±1V)

~20k

1000

800

600

400

200

–200

–400

–600

–800

–1000

–128 –96 –64 –32 0 32 64 96 128

VSENSE (mV)

COUNTS

VCC = 5V

VEE = –5V

Figure 8. Output voltage referred to ground—

unidirectional sensing mode

Figure 9. Expanded scale of V

OUT

vs V

SENSE

unidirectional current sensing mode

Figure 10. Connection to an A/D converter (current-to-counts converter) Figure 11. LT1787 input to LTC1404 ADC

Linear Technology Magazine • September 1999

DESIGN FEATURES

are used to provide a power-on reset

of the RS ﬂip-ﬂop. The 1M resistor,

R7, and the 0.33µF capacitor, C3, at

pin 2 of U3A provide a long enough

time constant to properly initialize

the ﬂip-ﬂop.

As shown, the circuit’s trip point

for shutdown is just under 2.5A. This

may be changed by altering the value

of current sense resistor R1. Consult

the LT1787 data sheet for details on

how to alter this resistor to sense

different current ranges.

Battery Fuel Gauge

The industry standard method for

gauging battery charge is to keep

track of the endpoints, the full charge

and discharged states, and, in

between, to measure how much dis-

charge has occurred since the last

full charge (and vice versa). In an

automobile, this would be analogous

to having only a “full” reading and an

“empty” reading on the gas tank, and,

in between, to keep track of mileage.

Applying this strategy to batteries is

called “Coulomb-counting”; it is

achieved by measuring (and

numerically accumulating in a

microcontroller) the current ﬂow from

the battery over time. Keeping track

of the history of battery currents and

voltages allows the present state of

the battery charge to be determined.

Hence the need for an accurate cur-

rent sense ampliﬁer like the LT1787.

Figure 14 shows a schematic for

measuring battery current using the

LT1787 and measuring battery volt-

age using the micropower LT1635

with 200mV internal reference. The

LT1787 is conﬁgured for single-sup-

ply, bidirectional operation, with the

2.0V reference coming from U2B, the

LT1635 (created by amplifying its in-

ternal 200mV reference by a gain of

ten.) Note that the reference voltage is

fed to the ADC, so its absolute value

is not critical, except in that it will

form the center point for the battery

voltage measurement and will thus

determine the valid battery input volt-

age range. Resistors R1 and R2 form

a divide-by-ﬁve, bringing the battery

voltage down from ~10.8V to ~2.1V to

put it within the input range of a

downstream ADC. Op amp U2A with

resistors R3 and R4 level shift this by

the reference voltage and apply a gain

of ﬁve. If a 12-bit ADC with a 5V

reference is used, the following equa-

tions apply:

VBATT = 4 (VREF ) + VB

= (5/4095) (4(Ch1) + Ch2)

IBATT = VS / RS

= (1/8) (VI – VREF) / 0.05

= 2.5 (VI – VREF)

= 2.5 (5/4095) (Ch0 – Ch1)

where Ch0, Ch1 and Ch2 are in counts

from 0 to 4095.

1000

800

600

400

200

00 20 40 60 80 100 120

SENSE

(mV)

COUNTS

= 5V

= –5V

–

SHDN

OUTPUT

SENSE

GND

U4 LT1528

FIL

–

S+

DNC

FIL

S–

OUT

BIAS

R1 0.05Ω

U1 LT1787

5V C1 0.1µFR2

20k

R4 3.24k

R5 20k

2.4M

430Ω

1µF

–

HYST

REF

–

OUT

GND

U2 LTC1440

0.1µF

0.01µF

1N5712

R10

180k

U3C

U3B

U3A

U3D

10k

12 11

OUT

3.3V/2.5A

100µF

R11

330Ω

1µF

LATCHED SHUTDOWN

POWER-ON

RESET

U3 = 74HC00

0.33µF

Figure 12. LT1787 input to LTC1404 ADC—

single supply

Figure 13. LT1787 auto shutdown with latch

Linear Technology Magazine • September 1999

DESIGN FEATURES

Figure 15 shows a typical discharge

and charge cycle for a 10.8V, 4A-hour

Li-Ion battery.

Conclusion

The LT1787 high-side current sense

ampliﬁer provides an easy-to-use

method of sensing current with 12-

bit resolution for a multiplicity of

application areas. The part can oper-

FIL

–

S–

DNC

FIL

S+

OUT

BIAS

0.05Ω

U1 LT1787

C1 0.22µF

–

BATT

SONY LIP9020

10.8V, 4AH

Li-Ion BATTERY

STAR GROUND

AT NEGATIVE

BATTERY

TERMINAL

BATT

LOAD CHARGE

(12.5V MAX)

R9 1k ADC CH0

0.22µF

R8 1k

ADC CH1

REF

C2 100pF

R5 9.09k

U2B

1/2 LT1635

U2A

1/2 LT1635 6

100pF

249k 1%

ADC CH2

249k

BATT

200mV

R7 1k

SENSE V

BATT

THE POSITIVE BATTERY

TERMINAL – V

–2

–4 0 30 60 90 120 150 180

TIME (MINUTES)

VOLTAGE (V)CURRENT (A)

VBATT

IBATT

ate to 60V, making it ideal for higher

voltage topologies such as might be

used in telecom or industrial applica-

tions. Additionally, the part can ﬁnd

homes in battery-powered, handheld

equipment and computers, where the

need for gauging the amount of cur-

rent consumed and/or the amount of

charge remaining in the battery is

critical.

Figure 14. Battery “fuel gauging” system

Figure 15. Discharge and charge cycle

of a 10.8V Li-Ion battery

http://www.linear-tech.com/ezone/zone.html

Articles, Design Ideas, Tips from the Lab…

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

Linear Technology Magazine • September 1999

DESIGN FEATURES

Triple 300MHz Current Feedback

Ampliﬁers Drive RGB/Component

Video and LCD Displays by Brian Hamilton

Introduction

With the advent of HDTV and DVD

video, there is a renewed focus on

RGB/component video to maximize

picture quality. LCD displays have

also entered the mainstream for high

end and portable computer displays.

Both of these applications require

high speed triple ampliﬁers for rout-

ing and conditioning video signals.

LCD displays also require voltage

swings of over 10V, with fast settling,

into large capacitive loads. With these

applications in mind, Linear Tech-

nology has introduced the LT1399

and LT1399HV triple current feed-

back ampliﬁers. The LT1399 and

LT1399HV contain three independent

300MHz current feedback ampliﬁers,

each with a shutdown pin. Each

ampliﬁer has exceptional 0.1dB gain

ﬂatness of 150MHz and a slew rate of

800V/µs. Output drive current is a

minimum of 80mA over temperature.

The LT1399 operates on all sup-

plies from a single 4V to ±6V. The

LT1399HV supports higher supply

voltages and will operate on supplies

ranging from a single 4V to ±7.5V.

Each of the three ampliﬁers draws

4.6mA when active. When disabled,

each ampliﬁer draws zero supply cur-

rent and its output becomes high

impedance. Each ampliﬁer can be

enabled in 30ns and disabled in 40ns,

making the LT1399 and LT1399HV

ideal in spread-spectrum and por-

table equipment applications.

With the addition of a small series

resistor at the output, the LT1399

and LT1399HV are capable of driv-

ing large capacitive loads. The

LT1399HV’s high voltage operation,

when combined with its ability to

drive capacitive loads, makes it ideal

for driving LCD displays.

What’s Inside

Figure 1 is a simpliﬁed schematic for

one of the three ampliﬁers found in

the LT1399/LT1399HV. Tying EN low

allows current to ﬂow through J1,

Q1–Q4 and R1. Q3 and Q6 mirror this

current on top, while Q5 and Q7

mirror the current on the bottom. Q6

and Q7 thus act as current sources

for input-stage transistors Q8–Q11.

+IN is a high-impedance input, driv-

ing the bases of Q8 and Q9. The

emitters of these transistors then drive

the bases of Q10 and Q11, which

have their emitters tied together and

form a buffered representation of +IN.

This node is the inverting input –IN.

Any current ﬂowing into or out of –IN

modulates the collector currents of

Q12 and Q14. This, in turn, modu-

lates the collector currents of Q13

and Q15, which drive the high imped-

ance node. Transistors Q16–Q21 and

resistors R2 and R3 form the output

stage that buffers the signal at the

high impedance node from the output.

Using the LT1399HV

to Drive LCD Displays

Driving present-day XGA and UXGA

LCD displays can be a difﬁcult prob-

lem because they require drive

voltages of up to 12V, usually present

a capacitive load of over 300pF and

require fast settling. The LT1399HV

is particularly well suited for driving

these LCD displays because it is

capable of swinging more than ±6V

on ±7.5V supplies and, with a small

series resistor R

at the output, can

drive large capacitive loads with good

settling time. This circuit topology

can be seen in Figure 2. As seen in

Figure 3, at a gain of three, with a

16.9Ω output series resistor and a

330pF load, the LT1399HV is capable

of settling to 0.1% in 30ns for a 6V

step. Similarly, as seen in Figure 4, a

12V output step will settle in 70ns.

+IN –IN OUT

–

Q3 Q12

Q14Q5

Q10

Q11

Q13 Q16

Q18

HI-Z

Q15

Q19

Q17

Q20

Q21

–

VIN

CLOAD

VOUT

1/3

LT1399HV

Figure 1. LT1399 simpliﬁed schematic (one ampliﬁer) Figure 2. Adding an output series resistor for

driving capacitive loads

Linear Technology Magazine • September 1999

DESIGN FEATURES

A 3-Input Video Multiplexer

and Cable Driver

Figure 5 shows a low cost, 3-input

video MUX cable driver. The scope

photo in Figure 6 displays the cable

output of a 30MHz square wave driv-

ing 150Ω. In this circuit, the active

ampliﬁer is loaded by the sum of R

and R

of each disabled ampliﬁer.

Resistor values have been chosen to

keep the total back termination at

75Ω while maintaining a gain of one

at the 75Ω load. Figure 7 shows the

envelope of the output signal as the

multiplexer is switched from channel

A to channel B. Channel A is being

driven by a 2V

P-P

, 3.58MHz sine wave.

The switching envelope of the output

is well behaved and the switching

time is approximately 32ns.

Buffered RGB to

Color-Difference Matrix

Two LT1399s can be used to create

buffered color-difference signals from

RGB inputs. In the application shown

in Figure 8, a total of four ampliﬁers

is used to create color-difference sig-

nals. The luminance signal Y is created

using ampliﬁers A2 and A3. The

remaining color-difference signals

each use a single ampliﬁer and the

newly created Y output to perform the

appropriate difference function.

The R input arrives via 75Ω coax

and is routed to 1082Ω resistor R8

and the noninverting input of LT1399

ampliﬁer A1. There is also an 80.6Ω

termination resistor, R11, which

yields a 75Ω input impedance at the

R input when considered in parallel

with R8. R8 connects to the inverting

input of a second ampliﬁer, A2, which

also sums the weighted G and B in-

puts to create a –0.5 • Y output.

Ampliﬁer A3 then takes the –0.5 • Y

output and ampliﬁes it by a gain of

minus two, resulting in the Y output.

Ampliﬁer A1 is conﬁgured in a nonin-

verting gain-of-two conﬁguration, with

the bottom of the gain resistor R2 tied

to the Y output. The output of ampliﬁer

A1 thus results in the color-differ-

ence output R – Y.

The B input is similar to the R

input. It arrives via 75Ω coax and is

routed to 2940Ω resistor R10 and the

noninverting input of ampliﬁer A4.

There is also a 76.8Ω termination

resistor R13, which yields a 75Ω input

impedance when considered in paral-

lel with R10. R10 also connects to the

inverting input of ampliﬁer A2, add-

ing the B contribution to the Y signal

as discussed above. Ampliﬁer A4 is

conﬁgured in a noninverting gain-of-

two conﬁguration with the bottom of

the gain resistor R4 tied to the Y

output. The output of ampliﬁer A4

thus results in the color-difference

output B – Y.

The G input also arrives via 75Ω

coax and adds its contribution to the

Y signal via a 549Ω resistor R9 that is

tied to the inverting input of ampliﬁer

A2. There is also an 86.6Ω termina-

tion resistor, R12, which yields a 75Ω

termination when considered in par-

–

1/3 LT1399

200Ω

324Ω

EN A

IN A

–

1/3 LT1399

200Ω

324Ω

EN B

IN B

CHANNEL

SELECT

97.6Ω

–

1/3 LT1399

200Ω

324Ω

EN C

IN C

97.6Ω

75Ω

OUT

75Ω

CABLE

1399 TA01

= 324

Ω

= 162

Ω

= 16.9

Ω

= 330pF

1V/DIV

OUT

2V/DIV

20ns/DIV

Figure 3. LT1399/LT1399HV large-signal pulse response

driving 330pF (typical LCD loading)

= 324

Ω

= 162

Ω

= 16.9

Ω

= 330pF

2V/DIV

OUT

5V/DIV

50ns/DIV

Figure 4. LT1399HV output-voltage swing is

>12V with ±7V supplies

Figure 5. 3-input video mux/cable driver

Linear Technology Magazine • September 1999

DESIGN FEATURES

allel with R9. Using superposition, it

is straightforward to determine the

output of ampliﬁer A2. Although

inverted, it sums the R, G and B

signals in the standard proportions of

0.3R, 0.59G and 0.11B, which are

used to create the Y signal. Ampliﬁer

A3 then inverts and ampliﬁes the

signal by two, resulting in the Y out-

put. Two additional LT1399 ampliﬁers

remain unused, available for addi-

tional signal conditioning as needed.

Buffered Color-Difference

to RGB Matrix

The LT1399 can also be used to cre-

ate buffered RGB outputs from

color-difference signals. As seen in

Figure 9, the R output is a back-

terminated 75Ω signal created using

resistor R5 and LT1399 ampliﬁer A1

conﬁgured for a gain of two via 324Ω

resistors R3 and R4. The noninvert-

ing input of ampliﬁer A1 is connected

via 1k resistors R1 and R2 to the Y

and R – Y inputs, respectively, result-

ing in cancellation of the Y signal at

the ampliﬁer input. The remaining R

signal is then ampliﬁed by A1.

The B output is also a back-termi-

nated 75Ω signal created using

resistor R16 and ampliﬁer A3

conﬁgured for a gain of two via 324Ω

resistors R14 and R15. The non-

inverting input of ampliﬁer A3 is

connected via 1kΩ resistors R12 and

R13 to the Y and B – Y inputs respec-

tively, resulting in cancellation of the

Y signal at the ampliﬁer input. The

remaining B signal is then ampliﬁed

by A3.

The G output is the most compli-

cated of the three. It is a weighted

sum of the Y, R – Y and B – Y inputs.

The Y input is attenuated via resis-

tors R6 and R7 such that ampliﬁer

A2’s noninverting input sees 0.83Y.

Using superposition, we can calcu-

late the positive gain of A2 by assuming

that R8 and R9 are grounded. This

results in a gain of 2.41 and a contri-

bution at the output of A2 of 2Y. The

R – Y input is ampliﬁed by A2 and

resistors R8 and R10, giving a gain of

–1.02. This results in a contribution

at the output of A2 of 1.02Y – 1.02R.

The B – Y input is ampliﬁed by A2 and

resistors R9 and R10, giving a gain of

–0.37. This results in a contribution

at the output of A2 of 0.37Y – 0.37B.

If we now sum the three contribu-

tions at the output of A2, we get:

A2OUT = 3.40Y – 1.02R – 0.37B

It is important to remember though,

that Y is a weighted sum of R, G, and

B such that:

Y = 0.3R + 0.59G + 0.11B.

If we substitute for Y at the output

of A2, we then get:

A2OUT = (1.02R – 1.02R) + 2G +

(0.37B – 0.37B) = 2G

The back-termination resistor R16

then halves the output of A2, result-

ing in the G output.

OUTPUT

200mV/DIV

5nS/DIV

= 150

Ω

= R

= 340

Ω

f = 30MHz

EN A

EN B V

5V V

INA

= V

INB

P-P

AT 3.58MHz

OUTPUT

5V/DIV

10ns/DIV

–

1/3 LT1399

–

1/3 LT1399

–

1/3 LT1399

324Ω

162ΩR5

324Ω

R10

2940Ω

549Ω

R11

80.6Ω

R12

86.6Ω

R13

76.8Ω

ALL RESISTORS 1%

= ±5V

1082Ω

75Ω

SOURCES

324Ω

B – Y

R – Y

–

1/3 LT1399

Figure 6. 30MHz square wave response Figure 7. 3-input video mux switching response

Figure 8. Buffered RGB to color-difference matrix

Linear Technology Magazine • September 1999

DESIGN FEATURES

–

1/3 LT1399

B-Y

R-Y

R10

324Ω

R11

75Ω

205Ω

316Ω

845Ω

–

1/3 LT1399

R14

324Ω

R16

75Ω

R12

R13

R15

324Ω

ALL RESISTORS 1%

= ±5V

–

1/3 LT1399

324Ω

75Ω

324Ω

–

+A1

1/3 LT1399

1N4148

4.7µF

1N4148 R9

75Ω

324Ω

6V TO 12V

324Ω

2.32Ω

75Ω

VIN

1300Ω

1000Ω

107Ω

160Ω

VOUT

Single-Supply RGB

Video Ampliﬁer

The LT1399 can be used with a single

supply voltage of 6V or more to drive

ground-referenced RGB video. As seen

in Figure 10, two 1N4148 diodes, D1

and D2, have been placed in series

with the output of the ampliﬁer A1,

but within the feedback loop formed

by resistor R8. These diodes effec-

tively level-shift A1’s output downward

by 2 diodes, allowing the circuit out-

put to swing to ground.

Ampliﬁer A1 is used in a positive

gain conﬁguration. The feedback

resistor R8 is 324Ω. The gain resistor

is created from the parallel com-

bination of R6 and R7, giving a

Thevenin-equivalent 80.4Ω connected

to 3.75V. This gives an AC gain of ﬁve

from the noninverting input of ampli-

ﬁer A1 to the cathode of D2. However,

the video input is also attenuated

before arriving at A1’s positive input.

Assuming a 75Ω source impedance

for the signal driving V

, the Thev-

enin-equivalent signal arriving at A1’s

positive input is 3V + (0.4 • V

), with

a source impedance of 714Ω. The

combination of these two inputs gives

an output at the cathode of D2 of 2 •

with no additional DC offset. The

75Ω back termination resistor R9

halves the signal again such that

OUT

equals a buffered version of V

It is important to note that the

4.7µF capacitor C1 is required to

maintain the voltage drop across

diodes D1 and D2 when the circuit

output drops low enough that the

diodes might otherwise be reverse

biased. This means that this circuit

works ﬁne for continuous video input,

but will require that C1 be charged

after a period of inactivity at the input.

Conclusion

Linear Technology has introduced the

LT1399 and LT1399HV triple 300MHz

current feedback ampliﬁers. Both of

these products are well suited for use

in component video applications. The

higher supply voltage rating of the

LT1399HV makes it an excellent

choice for LCD driver applications.

Both products feature 4.6mA of sup-

ply current per ampliﬁer, 300MHz

–3dB bandwidth, an exceptional

0.1dB gain ﬂatness of 150MHz, 800V/

µs slew rate and a shutdown pin for

each channel.

Figure 9. Buffered color-difference to RGB matrix Figure 10. Single-supply RGB video ampliﬁer (one of three channels)

is no output swing penalty as the

swing is limited at the collectors of Q9

and Q13. The dynamics of the com-

posites are not as benign as those of

emitter followers, so compensation is

required and is provided by C1 and

C2.

The stability with capacitive loads

is provided by the R

, C

network

between the output stage and the

LT1813, continued from page 6

gain node. When the ampliﬁer is driv-

ing a light or moderate load, the output

can follow the high impedance node

and the network is bootstrapped and

has no effect. When driving a heavy

load such as a capacitor or small-

value resistor, the network is

incompletely bootstrapped and adds

to the compensation provided by C

The added capacitance provided by

slows down the ampliﬁer and the

zero created by R

adds phase margin

to increase stability.

Conclusion

The combination of a high slew rate,

DC accuracy and a frugal 3mA-per-

ampliﬁer supply current make the

LT1813 a compelling choice for low

voltage and low power, high speed

applications.

Linear Technology Magazine • September 1999

DESIGN FEATURES

LTC1702/LTC1703 Switching Regulator

Controllers Set a New Standard for

Transient Response

Introduction

The LTC1702 dual switching regula-

tor controller uses a high switching

frequency and precision feedback cir-

cuitry to provide exceptional output

regulation and transient response

performance. Running at a ﬁxed

550kHz switching frequency, each

side of the LTC1702 features a volt-

age feedback architecture using a

25MHz gain-bandwidth op amp as

the feedback ampliﬁer, allowing loop-

crossover frequencies in excess of

50kHz. Large onboard MOSFET driv-

ers allow the LTC1702 to drive high

current external MOSFETs efﬁciently

at 550kHz and beyond. The high

switching frequency allows the use of

small external inductors and capaci-

tors while maintaining excellent

output ripple and transient response,

even as load currents exceed the 10A

level. The dual-output LTC1702 is

packaged in a space-saving 24-pin

narrow SSOP, minimizing board space

consumed.

Mobile PCs using the most recent

Intel Pentium

III processors require

LTC1702-level performance coupled

with a DAC-controlled voltage at the

core supply output. The LTC1703 is

designed speciﬁcally for this applica-

tion and consists of a modiﬁed

LTC1702 with a 5-bit DAC control-

ling the output voltage at side 1. The

DAC conforms to the Intel Mobile VID

speciﬁcation. Figure 6 shows an

example of a complete mobile Pen-

tium III power supply solution using

the LTC1703. The LTC1703 is pack-

aged in the 28-pin SSOP package,

conserving valuable PC board real

estate in cramped mobile PC designs.

LTC1702/LTC1703

Architecture

The LTC1702/LTC1703 each consist

of two independent switching regula-

tor controllers in one package. Each

controller is designed to be wired as a

voltage feedback, synchronous step-

down regulator, using two external

N-channel MOSFETs per side as

power switches (Figure 1). A small

external charge pump (D

and C

Figure 1) provides a boosted supply

voltage to keep M1 turned fully on.

The switching frequency is set

internally at 550kHz. A user-program-

mable current limit circuit uses the

synchronous MOSFET switch, M2,

as a current sensing element, elimi-

nating the need for an external low

value current sensing resistor. The

LTC1702/LTC1703 are designed to

operate from a 5V or 3.3V input sup-

ply, provided either by the main

off-line supply in an AC powered sys-

tem or a primary switching regulator

in battery powered systems. Maxi-

mum input voltage is 7V.

Synchronous operation maximizes

efﬁciency at full load, where resistive

drops in the switching MOSFET and

the synchronous rectiﬁer dominate

the power losses. As the load drops

and switching losses become a larger

factor, the LTC1702/LTC1703 auto-

matically shifts into discontinuous

mode, where the synchronous recti-

ﬁer MOSFET turns off before the end

of a switching cycle to prevent reverse

current ﬂow in the inductor. As the

load current continues to decrease,

the LTC1702/LTC1703 switches

modes again and enters Burst Mode™,

where it will only switch as required

to keep the output in regulation, skip-

ping cycles whenever possible to

reduce switching losses to a bare

minimum. With no output load in

Burst Mode, the supply current for

the entire system drops to the 3mA

quiescent current drawn by each side

of the LTC1702/LTC1703. Each side

can be shutdown independently; with

both sides shut down, the LTC1702/

LTC1703 enters a sleep mode where

it draws less than 50µA.

Inside the LTC1702/LTC1703

The LTC1702/LTC1703 features

peerless regulation and transient

response, due to both to its high

switching frequency and a carefully

designed internal architecture (Figure

2). Much of the transient response

improvement comes from a new feed-

back ampliﬁer design. Unlike

conventional switching regulator

designs, the LTC1702/LTC1703 use

a true 25MHz gain-bandwidth op amp

as the feedback ampliﬁer (FB in Fig-

ure 2). This allows the use of an

optimized compensation scheme that

can tailor the loop response more

precisely that the traditional RC from

COMP to ground. A “type 3” feedback

circuit (Figure 3) typically allows the

by Dave Dwelley

BOOST

PGND

OUT

EXT

1µF

LTC1702/

LTC1703

Figure 1. LTC1702/LTC1703 switching architecture

Pentium is a registered trademark of Intel Corp.

Linear Technology Magazine • September 1999

DESIGN FEATURES

loop to be crossed over beyond 50kHz

while maintaining good stability, sig-

niﬁcantly enhancing load transient

response. Two additional high speed

comparators (MIN and MAX in Figure

2) run in parallel with the main feed-

back ampliﬁer, providing virtually

instantaneous correction to sudden

changes in output voltage. In a typi-

cal application, the LTC1702/

LTC1703 will correct the duty cycle

and have the output voltage headed

back in the right direction the very

next switching cycle after a transient

load is applied.

The positive input of the feedback

op amp is connected to an onboard

reference trimmed to 800mV ±3mV.

DC output error due to the reference

and the feedback ampliﬁer are inside

0.5% and DC load and line regulation

are typically better than 0.1%, pro-

viding excellent DC accuracy. The

800mV reference level allows the

LTC1702/LTC1703 to provide regu-

lated output voltages as low as 0.8V

without additional external compo-

nents. This reference performance,

combined with the high speed inter-

nal feedback ampliﬁer and properly

chosen external components, allows

the LTC1702 to provide output regu-

lation tight enough for virtually any

microprocessor, today or in the future.

For those Intel processors that don’t

know what voltage they want until

they actually get powered up, the

LTC1703 with its onboard 5-bit VID

output voltage control is the best

solution.

Another architecture trick inside

the LTC1702/LTC1703 reduces the

required input capacitance with vir-

tually no performance penalty. The

LTC1702/LTC1703 includes a single

master clock, which drives the two

sides such that side 1 is 180° out of

phase from side 2. This technique,

known as 2-phase switching, has the

effect of doubling the frequency of the

switching pulses seen by the input

capacitor and signiﬁcantly reducing

their RMS value. With 2-phase switch-

ing, the input capacitor is sized as

required to support a single side at

maximum load. As the load increases

at the other side, it tends to cancel,

rather than add to, the RMS current

seen by the input capacitor; hence,

no additional capacitance needs to be

added.

External Components

The other half of the performance

equation is made up by the external

components used with the LTC1702/

LTC1703. The 550kHz clock frequency

and the low 5V input voltage allow the

use of external inductors in the 1µH

range or lower (L

EXT

in Figure 1) while

still keeping inductor ripple current

under control. This low inductance

BURST

LOGIC

SOFT

START

90% DC

GAIN

RUN/SS

COMP

500mV

800mV 760mV 840mV

IMAX

DRIVE

LOGIC

100µs

DELAY

OSC

550kHz

+–

FB MIN MAX

920mV

FLT

DIS

FCB

1702 BD

BOOST

FROM

OTHER

HALF

SD TO THIS

HALF

SD TO

ENTIRE CHIP

FAULT

PVCC

10µs

DELAY

FROM

OTHER

HALF

PGND

PGOODD

40k

VID0

VCC

40k

VID1

VCC

40k

VID2

VCC

40k

VID3

VCC

40k

VID4

VCC

R11

RB1

SENSE

FB1

SWITCH

CONTROL

LOGIC

LTC1703 ONLY

ILIM

Figure 2. LTC1702/LTC1703 block diagram

Linear Technology Magazine • September 1999

DESIGN FEATURES

value helps in two ways: it reduces

the energy stored in the inductor dur-

ing each switching cycle, reducing

the physical core size required; and it

raises the attainable di/dt at the out-

put of the circuit, decreasing the time

that it takes for the circuit to correct

for sudden changes in load current.

This, in turn, reduces the amount of

output capacitance (C

OUT

in Figure 1)

required to support the output volt-

age during a load transient. Together

with the reduced capacitance at the

input due to the LTC1702/LTC1703’s

2-phase internal switching, this

signiﬁcantly reduces the amount of

total capacitance needed, compared

to a conventional design running at

300kHz or less.

Each side of an LTC1702/LTC1703

circuit requires a pair of N-channel

power MOSFETs to complete the

power switching path. These are cho-

sen for low R

DS(ON)

and minimum gate

charge, to minimize conductive losses

with heavy loads and switching losses

at lighter loads. MOSFET types that

work well with the LTC1702/LTC1703

include the IRF7805 from Interna-

tional Rectiﬁer, the Si9802 and Si9804

from Siliconix and the FDS6670A from

Fairchild.

The compensation components

round out the list of external parts

required to complete an LTC1702/

LTC1703 circuit. Because the

LTC1702/LTC1703 uses an op amp

as the feedback ampliﬁer, the com-

pensation network is connected

between the COMP pin (at the output

of the op amp) and the FB pin (the

inverting input) as a traditional op

amp integrator (Figure 3). A bias

resistor is added to set the DC output

voltage and two pole/zero pairs are

added to the circuit to compensate for

phase shift caused by the inductor/

output capacitor combination. Cur-

rent limit and soft start time for each

side are programmed with a single

resistor (R

IMAX

) at each I

MAX

pin and a

single capacitor (C

) at each RUN/

SS pin. Optional FAULT (LTC1702/

LTC1703) and PWRGD (LTC1702

only) ﬂags are available to provide

status information to the host system.

Applications

Dual Outputs from a 5V Supply

A typical LTC1702 application is

shown in Figure 4. The input is taken

from the 5V logic supply. Side 1 is set

up to provide 1.8V at 10A and side 2

is set to supply 3.3V at a lower 3A load

level. System efﬁciency peaks at

greater than 90% at each side. This

circuit shows examples of both high

power and lower power output designs

possible with the LTC1702 controller.

Side 1 uses a pair of ultralow R

DS(ON)

Fairchild FDS6670A SO-8 MOSFETs

0.8V

OUT

COMP

–

FB FB

BOOST1

TG1

SW1

BG1

MAX1

COMP1

FB1

RUN/SS1

BOOST2

TG2

SW2

BG2

MAX2

COMP2

FB2

RUN/SS2

LTC1702

GND PGND

OUT1

1.8V/10A

OUT1

470µF

×2

R11

10k

0.1%

C11

330pF

7.96k

0.1%

C21

680pF

C31

560pF

EXT1

1µH 12A

EXT2

2.2µH 6A

R31 4.7k

R21 13k

QT1

QB1

CP2

MBR0530T

CP1

MBR0530T

CP1

1µFC

CP2

1µF

D2 MBR330T

IMAX1

22k

SS1

0.1µF

SS2

0.1µF

10Ω

470µF

×2

QT2

QB2

OUT2

3.3V/3A

OUT2

470µF

C12

120pF

C22

270pF

R22 20k

1.62k

0.1%

R12

4.99k

0.1%

C32

820pF

R32 2.2k

QT1, QB1: FAIRCHILD FDS6670A (207) 775-4502

QT2, QB2: 1/2 SILICONIX Si9802 (800) 544-5565

IMAX2

47k

EXT1

EXT2

:MURATA LQT12535C1ROM12 (814) 237-1431

COILTRONICS UP2B-2R2 (561) 241-7876

Figure 3. Type-3 feedback loop

Figure 4. Dual outputs from a 5V supply

Linear Technology Magazine • September 1999

DESIGN FEATURES

and a large 1µH/12A Murata surface

mount inductor. C

consists of two

470µF, low ESR tantalum capacitors

to support side 1 at full load, and

OUT1

uses two more of the same to

provide better than 5% regulation

with 0A–10A transients.

Side 2 uses a single SO-8 package

with two smaller MOSFETs inside

(the Siliconix Si9402) and a smaller

2.2µH/6A inductor. C

OUT2

is a single

470µF tantalum to support 0A–3A

transients while maintaining better

than 5% regulation. As the load cur-

rent at side 2 increases, the LTC1702

2-phase switching actually reduces

the RMS current in C

, removing the

need for additional capacitance at the

input beyond what side 1 requires.

Both sides exhibit exceptional tran-

sient response (Figure 5). The entire

circuit can be laid out in less than 2

square inches when a double-sided

PC board is used.

2-Step Converter for

Notebook Computers

Figure 6 is a complete power supply

for a typical notebook computer using

the next generation of Intel mobile

Pentium III processor. The circuit uses

the LTC1628 to generate 5V and 3.3V

from the input battery and uses the

LTC1703 to generate the processor

core voltage (with 5-bit VID control)

and the CPU I/O ring supply voltage.

Both the LTC1628 and the LTC1703

use 2-phase switching to minimize

capacitance required by the circuit;

the entire 4-output circuit requires

barely 2000µF while generating 60W

of output power.

The 2-step conversion used in this

circuit provides improved transient

response compared to the traditional

single-step approach where each

SIDE 1

OUT

= 1.8V

= 5V

LOAD

= 0A–10A–0A

2% MAX DEVIATION

SIDE 2

OUT

= 3.3V

= 5V

LOAD

= 0A–3A–0A

2.2% MAX DEVIATION

2-Step Conversion

As microprocessor operating volt-

ages continue to decrease, power

conversion for CPU core power is

becoming a daunting challenge. A

core power supply must have fast

transient response, good efﬁciency

and low heat generation in the vicin-

ity of the processor. These factors

will soon force a move away from

1-step power conversion directly

from battery or wall adapter to pro-

cessor, to 2-step conversion, where

the CPU core power is obtained from

the 5V or 3.3V supply.

Several beneﬁts result from 2-step

conversion: more symmetrical tran-

sient response, lower heat generation

in the vicinity of the processor and

easy modiﬁcation for lower proces-

sor voltages in the future. Peak

currents taken from the battery are

also reduced, which leads to

improved battery chemical efﬁciency

that can often compensate for the

slight difference in electrical efﬁ-

ciency measured using laboratory

power supplies. Battery life in a real

notebook computer is virtually

identical for 1-step and 2-step

architectures.

The duty cycle for a step-down

switching regulator is given by the

ratio of V

OUT

to V

. In 1-step power

conversion, the duty cycle must be

very low because the step-down ratio

is large. This gives a very fast inductor

current rise time and a much slower

current decay time. The inductor size

must be large enough to keep the

current under control during the

ramp-up. Fast current rise and slow

current decay mean that the tran-

sient response of the regulator is good

for load increases but poor for load

decreases. The lower, constant input

voltage for a 2-step conversion pro-

cess yields a more symmetrical

transient response and allows smaller,

lower cost external components to be

used. Because there is less switching

loss due to the lower voltage swings,

the switching frequency may also be

increased.

Thermal concerns are also eased

with the 2-step approach. To mini-

mize high current PCB trace lengths,

the core supply must be located near

the processor. Core-voltage-level

1-step converters usually run at mid-

80% efﬁciencies, while the second

step of a 2-step solution (like the

LTC1703) runs near 90% efﬁciency,

minimizing heat generation near the

processor.

The biggest argument against

2-step conversion is the perceived

drop in efﬁciency. “Off the cuff” cal-

culations give a false impression

that efﬁciency decreases. In fact,

accurate calculations of efﬁciency

for 2-step power conversion based

on actual circuit measurements

show efﬁciency numbers within 1%

of 1-step, high efﬁciency converters.

As time goes forward, microproces-

sor fabrication lithography will

continue to shrink and force still

lower CPU core operating voltages

and higher operating currents; 1.1V

supplies and 15A operating cur-

rents are already on the horizon for

portable systems. These demands

will render the traditional 1-step

conversion approaches unworkable

as a result of inﬁnitesimal duty cycles

and severely skewed transient

behavior.

For more information on 2-step conversion,

see www.linear-tech.com/ezone/2-step.html

Figure 5a. Transient response, side 1 Figure 5b. Transient response, side 2

Linear Technology Magazine • September 1999

DESIGN FEATURES

BOOST1

BG1

TG1

SW1

MAX1

FCB

RUN/SS1

COMP1

SGND

FB1

SENS

VID0

VID1

MAX2

BOOST2

BG2

TG2

SW2

PGND

FAULT

RUN/SS2

COMP2

FB2

VID4

VID3

VID2

LTC1703

OUT4

1.5V

12A

1µF180µF, 4V

× 6

L3, 0.8µH

ETQP6F0R8L

MBRD-

835L

QB1A

IRF7811

QB1B

IRF7811

QT1A

IRF7811 QT1B

IRF7811 D6

MBR0520LT1

1µF

18.7k, 1%

R18, 1M

0.22µF

200pF 15pF

100k

220pF 10k

1µF24.9k, 1% D7

MBR0520LT1

QT2

NDS8926

QB2

NDS8926

L4 2.2µF

DO3316P-222

1µF150µF

× 2

POINT A

L5, 0.33µH

DO3316P-331HC

OUT3

1.8V

180µF

8.06k

10.2k

2200pF

10Ω

1µF

100k

15pF 220pF

0.22µF

1µF

VID0

VID1

VID2

VID3

VID4

COREVENABLE

1.8VENABLE

FAULT

0.1µF

NOTE: PLACE

LTC1703 CLOSE

TO PROCESSOR

RUN/SS1

SENSE1

–

OSENSE1

FREQSET

STBYMD

FCB

TH1

SGND

3.3V

OUT

TH2

OSENSE2

SENSE2

–

SENSE2

FLTCPL

TG1

SW1

BOOST1

BG1

EXTV

INTV

PGND

BG2

BOOST2

SW2

TG2

RUNSS2

LTC1628

5VENABLE

STBYMD

STDBY3.3V

STDBY5V

3.3VENABLE

0.1µF

330pF

33k

56pF

33k

1000pF

0.1µF

D2 CMDSH-3

1µF

0.01µF

4.7µF

D1 CMDSH-3

0.22µF

IRF7807

MBRS130T3

4.6µH

ETQP6F4R6H

MBRD835L

IRF7805

IRF7805 L1

2.9µH

ETQP6F2R9L

0.1µF

50V

0.004Ω

150µF

× 2

180µF

0.1µF

50V 22µF

50V

7V TO

20V

OUT1

105k

20.0k

63.4k

100pF

47pF

10µF

6.3V

10µF

6.3V

OUT2

3.3V

0.1µF

50V

1000pF

POINT

0.01Ω

voltage is derived directly from the

battery voltage. 2-step also allows the

use of smaller external components

without paying an efﬁciency or per-

formance penalty and it eases layout

and thermal management concerns.

See the “2-Step Conversion” sidebar

for more information.

Conclusion

The LTC1702 and LTC1703 achieve

DC and AC regulation performance

that tops the best switching regulator

controllers available today. As logic

densities continue to climb, more

applications are appearing where the

input voltage is limited to below 7V

and the output voltage is low, the

output current is high and multiple

outputs are required. The LTC1702

and LTC1703 provide the best com-

bination of regulation performance,

high efﬁciency, small size and low

system cost for such applications,

whether they appear in advanced

notebook computers or complex logic

systems.

Figure 6. 4-output notebook computer power supply

1–D7: MOTOROLA

(800) 441-2447

Q1–Q5, QT1A/1B, QB1A/1B:

INTERNATIONAL RECTIFIER

(310) 322-3331

QT2, QB2: FAIRCHILD

(207) 775-4503

L1–L3: PANASONIC

(201) 348-7522

L4, L5: COILTRONICS

(561) 241-7876

Linear Technology Magazine • September 1999

DESIGN FEATURES

An 8-Channel, High-Accuracy,

No Latency ∆∆

∆∆

∆ΣΣ

ΣΣ

Σ 24-Bit ADC by Michael K. Mayes

Introduction

Recently, Linear Technology intro-

duced the world’s most accurate,

simplest to use, 24-bit analog-to-digi-

tal converter, the LTC2400. With its

on-chip oscillator, 120dB line-fre-

quency rejection, user-transparent

offset/full-scale calibration, 10 parts-

per-million (ppm) total unadjusted

error and 1.5µV

RMS

noise, the

LTC2400 has become a key building

block in many system designs. The

LTC2400’s ease of use and high per-

formance enable faster design cycles

and better performance than other

∆Σ converters.

This article introduces the

LTC2408, a device combining the high

performance LTC2400 ADC core with

an 8-channel analog input multiplexer

(see Figure 1). This device offers many

unique features. The single-cycle set-

tling characteristics lead to simpliﬁed

multiplexer hook up and channel

selection, without the added overhead

required other converters. The excep-

tional noise performance of the device

eliminates the need for a program-

mable gain ampliﬁer (PGA). This allows

direct

digitization of a variety of volt-

age levels. Its 10ppm absolute

accuracy ensures a minimum perfor-

mance in excess of 16 bits. A unique

analog modulator implementation al-

lows measurement of microvolt

signals superimposed upon large DC

voltages. A wide range of sensor inputs

and voltage levels can be applied

simultaneously to the LTC2408. These

signals can extend below ground,

above V

or anywhere in between,

with the same 10ppm absolute accu-

racy.

Single-Cycle Settling

Ensures No Latency

Many applications requiring 16-bit to

24-bit resolution use delta-sigma (∆Σ)

ADCs. These applications typically

measure slow-moving signals, such

as those found in temperature mea-

surement, weight scales, strain-gage

transducers, gas analyzers, battery

monitoring circuits and DVMs. One

advantage delta-sigma converter

architectures offer over conventional

ADCs is on-chip digital ﬁltering. For

the low frequency applications

described above, this ﬁlter is designed

to provide rejection of line frequencies

at 50Hz or 60Hz and their harmonics.

A disadvantage of conventional

digital ﬁlters, prior to the release of

the LTC2400, was digital ﬁlter set-

tling time. If the input signal changes

abruptly, the conversion result is

invalid for the following 3–4 conver-

sion cycles (see Figure 2a). This makes

multiplexing the input difﬁcult. The

LTC2400 does not exhibit a ﬁlter set-

tling time; hence, it is easy to multiplex

(see Figure 2b); There is a one-to-one

correspondence between the conver-

sion result and the applied input

signal. Each conversion result is

independent from the previous con-

version result. The 10ppm total error

is maintained for each conversion

cycle, even in the extreme case of

sequentially measuring 0V and 5V on

adjacent channels.

The Advantages of

Not Using a Programmable

Gain Ampliﬁer (PGA)

The exceptional noise performance of

the LTC2408 (1.5µV

RMS

) corresponds

to an effective resolution of 21.6 bits

for a 5V input range. Low level input

signals within a 100mV range achieve

better than 16-bit effective resolution

without the use of a PGA. On the

other hand, conventional ∆Σ ADCs

are signiﬁcantly noisier than the

LTC2408 for a 5V input range. These

converters require internal PGAs in

order to improve the noise perfor-

mance for low level input voltages.

The LTC2408 offers several sig-

niﬁcant advantages over those

converters requiring a PGA. One

advantage the LTC2408 offers is the

ability to measure small signals

(microvolts) superimposed upon large

DC voltages (volts). For example (see

Figure 3), a 100mV signal sitting on

2V (2V to 2.1V) can be measured with

the same accuracy and noise perfor-

mance as a 100mV signal sitting on

ground (0V to 0.1V). Conversely, an

ADC operating with a programmable

gain of 50 is limited to an input range

of 0V to 0.1V with a 5V reference (see

CH0

CH1

CH2

CH3

CH4

CH5

CH6

CH7

8-CHANNEL

MUX 24-BIT ∆-Σ ADC

CSADC

CSMUX

CLK/SCK

ADC

MUX

OUT

REF

–

LTC2408

19, 25

74 3

2.7V–5.5V

2, 8

1µF

GND

1, 5, 16, 18

22, 27, 28

Figure 1. LTC2408 block diagram

Linear Technology Magazine • September 1999

DESIGN FEATURES

Figure 3). It cannot digitize any signal

larger than 100mV full-scale.

A second advantage the LTC2408

offers is full-scale accuracy. Since the

total unadjusted error is less than

10ppm, the absolute accuracy of any

input voltage within the 0V to 5V

range is within 10ppm or 16 bits.

Alternatively, devices using PGAs

exhibit full-scale errors limited by the

matching of internal components. The

user is burdened with removal of these

errors. The user must first apply the

system’s full-scale voltage to the device

and then perform a system calibra-

tion.

The use of a PGA in conventional

∆Σ adds complexity. Each channel

requires a system full-scale and off-

set calibration. Each channel may

have a different PGA gain and input-

signal range settings, corresponding

to different offset and full-scale cali-

bration coefﬁcients. This requires

programming and maintaining con-

ﬁguration/status registers, gain/

offset registers and channel/PGA-gain

registers. The LTC2408 does not re-

quire any registers. The offset and

full-scale error corrections are per-

formed during each conversion cycle

and are transparent to the user.

“Microvolts on Volts”

In order to measure a small level

signal (microvolts) superimposed

upon a large signal (volts), the con-

verter must exhibit extremely good

DC performance. The device must

have very low offset and full-scale

errors and excellent linearity perfor-

mance in order to accurately digitize

small signals with large ﬁxed DC lev-

els. Additionally, the temperature

coefﬁcients of offset, full-scale and

linearity errors must be low. The

LTC2408’s offset error is less than

1ppm and its offset drift is less than

N N+1

CONVERT CONVERT CONVERT CONVERT CONVERT

N XXXX XXXX XXXX N+1

DATA OUT

INVALID DATA

OR INDETERMINATE RESULTS

N N+1 N+2

CONVERT N+1 CONVERT N+2

N N+1 N+2

(VALID

DATA OUT)

ADC

OPERATION

0.01ppm/°C (see Figure 4a). The full-

scale error is less than 4ppm while its

drift is less than 0.02ppm/°C (see

Figure 4b). Combined with an inte-

gral nonlinearity error of 4ppm, the

LTC2408 can consistently resolve low

level signals in the microvolt range,

regardless of the ﬁxed DC level (within

the 0V to V

REF

range).

The accuracy, noise performance,

and temperature stability of the

LTC2408 enable the converter to

measure input signals from a multi-

tude of sensors (see Figure 5). In

addition to the LTC2408’s ability to

measure signals from 0 to V

REF

, the

device also has overrange/underrange

capabilities. The device can measure

an input signal 100mV below ground

and 100mV above V

REF

, even if V

REF

equal to V

A Simple 4-Wire

SPI Interface

Interfacing to the LTC2408 is simple.

The individual CS and CLK signals

(see Figure 6) can be common to both

the ADC and multiplexer or driven

independently to allow separate con-

REF

(5V)

100mV

USE UPPER 8 BITS

TO DETERMINE RANGE

REF

(5V)

100mV

INFORMATION

NOT AVAILABLE

5.0

2.5

–2.5

–5.0–55 –30 –5 20 45 75 95 120

TEMPERATURE (°C)

FULL-SCALE ERROR (PPM)

= 5V

REF

= 5V

= 0V

5.0

2.5

–2.5

–5.0–55 –30 –5 20 45 75 95 120

TEMPERATURE (°C)

FULL-SCALE ERROR (PPM)

= 5V

REF

= 5V

Figure 2a. Effect of conventional digital ﬁlter settling time

Figure 2b. The LTC2408 has no digital ﬁlter settling time.

Figure 3. Full range without PGA (left); limited range with PGA (right)

Figure 4a. Offset error drift

Figure 4b. Full-scale error drift

Linear Technology Magazine • September 1999

DESIGN FEATURES

–

5.9µV/°C

40.6µV/°C

5V ±µV

REF

12k

100Ω

350Ω

10k

BIPOLAR SENSOR

±100mV

TO V

+ 100mV

CH0

CH1

REF

–

CH2

CH3

CH4

CH5

CH6

CH7

MICROVOLTS AROUND 0V

(WITH FIXED GAIN TO REDUCE NOISE)

MICROVOLTS AROUND V

COLD-JUNCTION TEMPERATURE

MEASUREMENTS (MICROVOLTS)

MICROVOLTS AT MIDSCALE

INPUT VOLTAGE—10k SERIES RESISTOR

FOR ISOLATION/FAULT PROTECTION

"LIVE-AT-ZERO" INPUT

OVERRANGE

DIRECT TEMPERATURE

MEASUREMENTS

REF

2.7V–5.5V

LTC2408

CHOPPER-

STABILIZED

OP AMP

AMPLIFIED

LOW LEVEL

THERMOCOUPLES

(TYPES R, S)

Pt RTD

HALF-BRIDGE

DIRECT THERMOCOUPLE

VOLTAGE CONVERSION

(TYPES J, K)

SDO

CLK

DIN

VIN

EN D2 D1 D0 DON'T CARE

EOC "0" MSB MSB

66ms BUILT-IN

SETTLING TIME

FOR THE MUX

HI-Z

INTERNAL

OFFSET CAL

(66ms)

CONVERT

(66ms)

trol of the ADC and the mux. D

serially programmed to select the

desired input channel; SDO is the

serial output data of the converter.

and SDO may be shared by using

an external driver with a high imped-

ance output state. Since the LTC2408

exhibits single-cycle settling, there is

no overhead associated with digital

ﬁlter settling time. At the conclusion

of each conversion, a new channel

may be selected by a 4-bit serial input

word, or the same channel can be

retained by not shifting in a new

word. A new input channel may be

selected up to 66ms after the data-

output read has been completed. This

66ms period may be used to allow the

input signal to settle or offer the user

ﬂexibility in the timing of the mux

channel selection.

Conclusion

The LTC2408 is a highly accurate No

Latency ∆Σ converter capable of digi-

tizing a variety of input signals. Its

exceptional noise performance allows

direct digitization of sensors. The

device can measure microvolts on

one channel and volts on another, all

with 10ppm accuracy. The LTC2408

requires no user calibration or PGA,

and there is no overhead associated

with the input multiplexer. The

LTC2408’s exceptional accuracy,

ease-of-use and eight input channels

make it an ideal multichannel ADC

for complete system monitoring.

Figure 5. The LTC2408 simultaneously measures many input devices.

Figure 6. Mux/ADC timing and look ahead

Linear Technology Magazine • September 1999

DESIGN FEATURES

Micropower 5-Lead SOT-23

Switching Regulators Extend Battery

Life in Space-Sensitive Applications

by Bryan Legates

Introduction

The LT1615 and LT1617 are designed

for portable electronics that need a

power solution with a minimum foot-

print and long battery life. These

devices can be used as step-up or

boost converters, single-ended pri-

mary inductance converters (SEPIC)

or positive-to-negative converters.

With an input voltage range of 1.2V to

15V, these devices are ideal for a wide

range of applications and work with a

variety of input sources. An internal

36V switch allows the two devices to

easily generate output voltages of up

to ±34V without the use of costly

transformers. The LT1615 is designed

to regulate positive output voltages,

whereas the LT1617 is designed to

directly regulate negative output volt-

ages without the need for level-shifting

circuitry. Both parts use a current-

limited, ﬁxed off-time control scheme,

which helps achieve high efﬁciency

operation over a wide range of load

currents. With a no-load quiescent

current of only 20µA (with the output

in regulation) and a shutdown quies-

cent current of 0.5µA, these devices

squeeze the most life out of any bat-

tery application. Both devices use

tiny, low proﬁle inductors and

capacitors to minimize the overall

system footprint and cost.

The LT1615 and LT1617 are pin

compatible with two other members

of the PowerSOT™ family, the LT1613

and LT1611, respectively. This allows

the same board layout to be used to

evaluate the performance of multiple

devices. The LT1613 and LT1611 are

both current mode, constant fre-

quency devices, capable of producing

larger output currents than the

LT1615 and LT1617.

LT1615 2-Cell to 3.3V

Boost Converter

A popular supply for many portable

electronic devices, a 2-cell alkaline to

3.3V converter with the LT1615 can

deliver 60mA of load current. The

circuit is shown in Figure 1 and the

system efﬁciency appears in Figure 2.

The efﬁciency peaks at 84% with a

fresh 2-cell battery and averages 78%

over the entire 1.5V to 3V input volt-

age range. Switching waveforms with

an input voltage of 2.8V and a 60mA

load appear in Figure 3. This photo

illustrates the Burst Mode operation

of the LT1615, as the system delivers

energy to the output in short bursts

about every 15µs. The device is in

standby (drawing only 20µA of quies-

cent current) for about 12µs of each

15µs burst cycle, which greatly

increases the overall converter

efﬁciency.

LT1615 1-Cell Li-Ion

to 15V Boost Converter

The internal 36V switch of the LT1615

makes it an attractive choice for

applications that need a high output

voltage at a relatively low current.

Figure 4’s circuit is a typical system

that provides 15V at 15mA from a

single-cell Li-Ion battery. Efﬁciency,

shown in Figure 5, reaches 82% from

a fully charged battery.

LT1615

1.5V TO 3V

4.7µHD1

SHDN

604k

1M C2

22µF

OUT

3.3V

60mA

1615 TA03

GND

4.7µF

C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150

C2: TAIYO YUDEN JMK325BJ226 (408) 573-4150

L1: MURATA LQH3C4R7M24 (814) 237-1431

D1: MOTOROLA MBR0520 (800) 441-2447

15 4.7pF

0.1 1 10 100

LOAD CURRENT (mA)

EFFICIENCY (%)

= 1.5V

= 3.0V

OUT

20mV/DIV

AC COUPLED

500mA/DIV

2V/DIV

5µs/DIV

Figure 1. 2-cell to 3.3V boost converter

Figure 2. 3.3V boost converter efﬁciency

reaches 84%

Figure 3. Switching waveforms of 3.3V boost converter with 60mA load

Linear Technology Magazine • September 1999

DESIGN FEATURES

LT1615 33V Boost Converter

The converter in Figure 6 displays the

exceptional input and output voltage

range of the LT1615. A 33V output is

easily generated from a wide ranging

input voltage using a simple boost

topology. A small, 1µF ceramic

capacitor is all that is needed at the

output, making the total footprint

much smaller than other systems that

need much larger tantalum capaci-

tors. The efﬁciencies for inputs of

3.3V, 5V and 12V are shown in Figure

7. With a 12V input, this converter

can deliver up to 1.32W (40mA at

33V) of power at an efﬁciency of 85%,

all from a tiny SOT-23 package.

LT1615 1-Cell Li-Ion

to 3.3V SEPIC

Lithium-ion (Li-Ion) is the battery of

choice for systems needing the most

energy with the lightest weight, but

with a cell voltage ranging from 4.2V

down to 2.5V, a simple boost topology

cannot be used to provide a 3.3V

OUT

20mV/DIV

AC COUPLED

200mA/DIV

5V/DIV

2µs/DIV

output. Figure 8’s circuit is a SEPIC

converter that can easily do the job,

providing 100mA of load current. The

circuit shown uses two separate

inductors, but a single, dual-winding

inductor (a 1:1 transformer) can be

substituted. Figure 9 shows the

switching waveforms for this SEPIC

converter with an input voltage of

2.7V and a 50mA load. Notice that

this circuit uses the same basic Burst

Mode operation as the boost con-

verter, but the inductor current is

LT1615

2.5V TO 4.2V

10µHD1

SHDN

88.7k

1M C2

4.7µF

15V

15mA

1615 TA03

GND

4.7µF

C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150

C2: TAIYO YUDEN EMK316BJ475 (408) 573-4150

L1: MURATA LQH3C100KZ4 (814) 237-1431

D1: MOTOROLA MBR0520 (800) 441-2447

0.1 1 10 100

LOAD CURRENT (mA)

EFFICIENCY (%)

= 2.5V

= 4.2V

LT1615

3.3V TO 12V

10µHD1

SHDN

76.8k

2M C2

1µF

33V

8mA

1615 TA03

GND

4.7µF

C1: TAIYO YUDEN EMK316BJ475 (408) 573-4150

C2: TAIYO YUDEN GMK316BJ105 (408) 573-4150

L1: MURATA LQH3C100KZ4 (814) 237-1431

D1: MOTOROLA MBR0540 (800) 441-2447

0.1 1 10 100

LOAD CURRENT (mA)

EFFICIENCY (%)

= 5V

= 3.3V

= 12V

LT1615

2.5V TO

4.2V

10µHD1

1µF

SHDN

604k

1M C2

10µF

OUT

3.3V

100mA

1615 TA07

GND

4.7µF

C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150

C2: TAIYO YUDEN JMK316BJ106 (408) 573-4150

C3: TAIYO YUDEN JMK107BJ105 (408) 573-4150

L1, L2: MURATA LQH3C100K24 (814) 237-1431

D1: MOTOROLA MBR0520 (800) 441-2447

10µH

15 4.7pF

Figure 4. Li-Ion to 15V boost converter

Figure 6. 33V boost converter

Figure 5. Efﬁciency of Figure 4’s

boost converter

Figure 7. 33V boost converter efﬁciency

Figure 8. Li-Ion to 3.3V SEPIC converter Figure 9. Switching waveforms of the 3.3V SEPIC converter with 50mA load

Linear Technology Magazine • September 1999

DESIGN FEATURES

about one half that of the 3.3V boost

circuit whose waveforms are shown

in Figure 3. For the SEPIC, the switch

current is split equally between the

two inductors, with both inductors

providing current to the load when

the switch is turned off. Typical efﬁ-

ciency for this converter is 70%.

LT1615 ±20V

Dual-Output Converter

Figure 10 shows a single-inductor,

dual-output converter ideal for use in

applications needing both a positive

and a negative voltage. The positive

output is generated using a tradi-

tional boost converter, whereas the

negative output is generated using an

inverting charge pump. Regulation is

achieved by sensing the positive out-

put, but by using identical output

capacitors and rectifying diodes, the

negative output is also very well regu-

lated. For a 2× difference in output

currents, the positive and negative

output voltages differ less than 3%;

for a 10× difference, they differ less

than 5%. This converter provides 8mA

total output current from a 1.5V input

(two fully discharged alkaline batter-

ies) and 12mA total output current

from a 2.5V input (a fully discharged

single-cell Li-Ion battery). Increasing

the value of L1 to 22µH increases the

available output current by about

15%. If even larger load currents are

needed, the same converter can be

implemented using the LT1613 in

place of the LT1615. If the accuracy of

the negative output is more critical

than the accuracy of the positive out-

put, try the same topology using the

LT1617 to regulate the negative out-

put. Efﬁciency for this dual-output

converter is quite good, reaching 77%

with a single Li-Ion battery. See Fig-

ure 11 for efﬁciency curves at several

different input voltages.

LT1615 ±20V

Dual-Output Converter

with Load Disconnect

One drawback to the circuit shown in

Figure 10 is that during shutdown,

the positive output is one diode drop

below the input voltage. This is an

undesirable condition for many sys-

tems, and can be easily corrected with

the circuit in Figure 12. This is a dual-

output converter where both outputs

are developed using charge pumps,

so that both are disconnected from

the input when the LT1615 is turned

off. An additional beneﬁt is that cross

regulation is improved because both

outputs are generated in the same

manner. For a 5× difference in output

currents, the positive and negative

output voltages differ less than 1%;

for a 10× difference, they differ less

than 2%. The improvements this cir-

cuit provides do come at a cost: slightly

lower efﬁciency. Figure 13 shows that

the efﬁciency curves are about 3%

lower for load currents greater than

1mA, but the efﬁciency still reaches a

respectable 74%.

LT1615

1.5V TO 5V

10µHD1

SHDN

130k

2M C2

1µF

20V

4mA

–20V

4mA

1615 TA04

GND

1µF

4.7µF

C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150

C2, C3, C4: TAIYO YUDEN TMK316BJ105 (408) 573-4150

L1: MURATA LQH3C100K24 (814) 237-1431

D1, D2, D3: MOTOROLA MBR0530 (800) 441-2447

15 4.7pF

1µF

0.1 1 10 100

LOAD CURRENT (mA)

EFFICIENCY (%)

= 3V

= 1.8V

= 4.2V

LT1615

1.5V TO 5V

10µHD1

SHDN

130k

2M C2

1µF

20V

4mA

–20V

4mA

1615 TA05

GND

1µF

4.7µF

C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150

C2, C3, C4: TAIYO YUDEN TMK316BJ105 (408) 573-4150

C5: TAIYO YUDEN LMK212BJ105 (408) 573-4150

L1: MURATA LQH3C100K24 (814) 237-1431

D1, D2, D3, D4: MOTOROLA MBR0530 (800) 441-2447

15 4.7pF

1µFC5

1µF

0.1 1 10 100

LOAD CURRENT (mA)

EFFICIENCY (%)

= 3V

= 1.8V

= 4.2V

Figure 10. ±20V dual-output converter

Figure 12. ±20V dual-output converter with load disconnect

Figure 11. Efﬁciency of Figure 10’s circuit

Figure 13. Efﬁciency of Figure 12’s circuit

Linear Technology Magazine • September 1999

DESIGN FEATURES

NFB

LT1617

2.5V TO 4.2V

22µH

0.22µF

SHDN

24.9k

267k C2

4.7µF

–15V

15mA

1615 TA07

GND

4.7µF

C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150

C2: TAIYO YUDEN EMK316BJ475 (408) 573-4150

C3: TAIYO YUDEN TMK1316BJ224 (408) 573-4150

L1, L2: MURATA LQH3C220K24 (814) 237-1431

D1: MOTOROLA MBR0530 (800) 441-2447

22µH

0.1 1 10 100

LOAD CURRENT (mA)

EFFICIENCY (%)

= 2.5V

= 4.2V

VIN SW

NFB

LT1617

VIN

10µH

0.22µF

SHDN

24.9k

619k C2

1µF

–33V

20mA

1615 TA07

GND

4.7µF

C1: TAIYO YUDEN LMK316BJ475 (408) 573-4150

C2: TAIYO YUDEN GMK316BJ105 (408) 573-4150

C3: TAIYO YUDEN GMK1212BJ224 (408) 573-4150

L1: MURATA LQH3C100K24 (814) 237-1431

D1: MOTOROLA MBR0540 (800) 441-2447

0.1 1 10 100

LOAD CURRENT (mA)

EFFICIENCY (%)

= 5V

LT1617 1-Cell Li-Ion

to –15V Inverting Converter

Many electronic systems need a nega-

tive supply but have only a positive

input voltage to work with. A well

regulated, positive-to-negative con-

verter can be easily designed using

the LT1617. A Li-Ion to –15V invert-

ing converter capable of providing

15mA of load current is detailed in

Figure 14. Efﬁciency for this inverter,

shown in Figure 15, peaks at 76%.

LT1617 5V to –33V

Inverting Charge Pump

For the previous inverting converter,

the maximum voltage seen by the

power switch is equal to the sum of

the input and output voltages; this,

along with the 36V switch rating,

limits the output voltage that can be

provided using the inverting topol-

ogy. If higher negative voltages are

needed, use an inverting charge pump,

in which the maximum voltage seen

by the switch is equal to the output

voltage. Figure 16 shows a –33V, 20mA

inverting charge pump that can pro-

vide 20mA of load current. Efﬁciency

reaches 74%, as seen in Figure 17.

Conclusion

The applications presented show the

versatility of the LT1615 and LT1617.

These devices are capable of produc-

ing a wide range of positive and

negative outputs from a variety of

input sources. Their tiny SOT-23

packages, along with small external

components, combine to minimize

footprint and cost in space-conscious

applications.

References

1. Pietkiewicz, Steve. “SOT-23 Switch-

ing Regulators Deliver Low Noise

Outputs in a Small Footprint.” Linear

Technology IX:1 (February 1999),

pp.11–13, 23.

Figure 14. Li-Ion to –15V inverting converter

Figure 16. –33V inverting charge pump

Figure 15. Li-Ion to –15V inverting converter

efﬁciency

Figure 17. 5V to –33V inverting charge pump

efﬁciency

for

the latest information

on LTC products,

visit

www.linear-tech.com

Linear Technology Magazine • September 1999

DESIGN FEATURES

Versatile Ring Tone Generator Finds

Uses in Motor Drivers and Amplifiers

Overview

The LT1684 was speciﬁcally designed

for OEM telephone equipment. Its

function is to interface between the

digital control logic and the high volt-

age analog phone line. When used in

the design of a phone system, the

LT1684 allows software to control the

frequency, voltage and cadence of the

ring signal. Because of similarities of

application to telephone systems, the

LT1684 ring tone chip ﬁnds itself at

home in motor drives, digital input

ampliﬁed speakers, alarm systems

and sine wave UPS systems.

Introducing the LT1684

The LT1684 provides the tool set to

easily implement a digital, pulse width

modulated (PWM) signal to DC-

coupled voltage converter (at high

currents), while providing isolation,

switching frequency ﬁltering and

output protection. Figure 1 is a sys-

tem-level diagram of the LT1684 in

action. By controlling a pair of exter-

nal MOSFETs, the LT1684 utilizes

their inherent robustness while pro-

viding control of the output voltage

and current. In its telecom applica-

tion circuit, the LT1684 provides up

to ±240V of smooth, clean output at

up to 200mA of output current. Higher

output voltages are obtained by cas-

coding MOSFETs, while higher

currents are readily achieved by using

the LT1166 MOSFET automatic bias

generator chip as a companion.

The Circuit

Figure 2 is the schematic of the

LT1684 implementing a digital-PWM-

to-ringing-telephone converter. This

is something like a high power, high

voltage, isolated, output ﬁltered D/A

converter. Like its DAC counterpart,

the LT1684 has a precision reference,

switches and an output ampliﬁer.

Unlike its DAC counterpart, it includes

post-conversion ripple ﬁltering, iso-

lation and a robust high voltage

output.

In addition to the isolation, ﬁlter-

ing and ampliﬁcation, the LT1684

provides the gate-bias control and

gate voltage protection for the two

external MOSFETs. Providing such a

plethora of functions from a single,

monolithic IC requires the use of a

somewhat tricky circuit. This circuit

by Dale Eagar

LT1684

STUFF

DIGITAL

STUFF

VDIGITAL

DIGITAL GROUND

PWM

5V + HIGH VOLTAGE

–HIGH VOLTAGE HIGH

VOLTAGE

GROUND

ISOLATION

BARRIER

HIGH

VOLTAGE

LOAD

IN A GATE

IN B

6.8nF

IRF9610

IRF610

R10

100k

µC

LIM

OUT

REF

COMP1

COMP2

AMPIN LIM

–

GATE

–

100pF

1N4001

20pF

100pF

ISOLATION

100Ω

10k

1N5817

0.1µF

1µF

R6 5.1k

R4 2k

300k

6.8nF R7

100k

100Ω

RING-TONE

OUTPUT

+100V

–

–100V

LT1684

4700pF

1684 TA01

FB1

FB1: FERRONICS FMB1601 (716) 388-1020

–

Figure 1. The LT1684 uses differential pulse width modulation to provide isolation for digitally

controlled analog power solutions.

Figure 2. Typical LT1684 digital-PWM-to-ringing-telephone converter application

Figure 3. The basic 2nd order lowpass MFB

ﬁlter, as implemented by the LT1684

Linear Technology Magazine • September 1999

DESIGN FEATURES

is arrived at by applying a circuit

transformation to a simple ﬁlter cir-

cuit. This transformation is performed

on the basic 2nd order lowpass mul-

tiple feedback ﬁlter (MFB) circuit and

ends up looking somewhat like the

ﬁlter/ampliﬁer shown in Figure 3.

The ﬁlter/ampliﬁer components in

Figure 2 are R3–R5 and C3 and C4.

Looking backwards through the

circuit transformation, these compo-

nents, in fact, form the 2nd order

MFB ﬁlter shown in Figure 3. The

values chosen for these components

in Figure 2 implement a 2nd order

Butterworth MFB lowpass ﬁlter with

a cutoff frequency of 100Hz and a DC

gain of 100. These were chosen to

provide ±80V of output swing with

PWM duty factors of 10% to 90%,

while ﬁltering the 10kHz PWM ripple

to meet telephone speciﬁcations.

PWM

1000pF

10k

300k

0.1µF

470pF

5.1k

100k

6800pF

MTP-

2N50E

MTP-

2N50E

120V

–120V

100Ω

IN A

IN B

OUT

AMPIN

COMP1

LIM

OUT

REF

COMP2

LIM

–

GATE

–

LT1684

FB1

100pF

20pF

180µH

3.9k

1nF

2N3904

2N3906

2N3904

100Ω

47Ω

100Ω

IRF9240

–100V

1µF

0.22Ω

IRF230

100V

2k 1µH

SENSE

LIM+

OUT

LIM–

SENSE

–

BOTTOM

TOP

LT1166

TYPICAL POWER SLICE

(1 OF 13 IN PARALLEL)

100Ω

FB1: FERRONICS FMB1601 (716) 388-1020

5kW

LOAD

Stealing the LT1684

for Use In Other Applications

The LT1684, used as shown in Figure

1, outputs a ring signal that meets

Belcore speciﬁcation. This means we

can ring a phone 22,000 feet away.

The LT1684 ﬁts another role, where

22,000 feet of separation would be a

nice minimum. This is the applica-

tion of the LT1684 in the scaleable

power ampliﬁer, as detailed in Figure

4. This ampliﬁer can be used to drive

motors, simulate the power company

in sine wave UPS systems and oper-

ate large audio drivers. Because of its

scaleable nature, this design can be

used at any power level. The circuit in

Figure 4 is shown implementing a

5000W bits-to-decibels converter.

When this converter is implemented

with the appropriate audio drivers

and enclosures, the output sound

pressure level can be signiﬁcant—so

signiﬁcant, in fact, that the author

suggests giving it a wide berth of at

least 22,000 feet.

–

3kHz LPF

TRIANGLE IN

10kHz

5V 5V

–5V

LT1671 LT1784

100pF R1

10k

100pF

10k

TERM

120Ω

IN A

IN B

LT1684

–1V

ANALOG

0.8V

–0.8V

RS485

Figure 4. 5kW PWM-to-analog converter

Figure 5. A remote, isolated, analog input ampliﬁer using a robust RS485 driver and a

terminated, twisted-pair line

continued on page 35

Linear Technology Magazine • September 1999

35 Watt Isolated DC/DC Converter

Replaces Modules at Half the Cost

by Robert Sheehan

Introduction

The choice between building or buy-

ing an isolated DC/DC converter can

be a complex decision. If you use an

off-the-shelf, module you are con-

strained by what the module makers

offer in their catalogs. In many cases,

this may not precisely meet the re-

quirements for a particular project.

Also, while simple to use, the cost of

these modules can be significantly

higher than the cost of “rolling your

own.” The complexity of the DC/DC

design can be daunting and leads

many to the decision to buy. Demon-

stration circuit DC227 provides a

DC/DC solution that can serve the

needs of many “standard” module

applications and offers the designer

the option of customizing the design

to suit any slightly unusual system

requirements. The power supply now

becomes merely another collection of

parts in the system.

Features

Demonstration circuit DC227 is a

board level replacement for “half-

brick” DC/DC converters. It can

provide 5V or 3.3V at up to 7A from an

isolated 48V (36V to 72V) input. The

isolation voltage is 500VDC with an

option for 1500VDC. The circuit has

low input capacitance, fast turn-on

time, low shutdown power consump-

tion and overtemperature protection.

Continuous short-circuit protection

eliminates any restriction on maxi-

mum capacitive load. The output

overvoltage circuit provides protec-

tion for open or short circuits on the

output power or sense lines. The stan-

dard footprint allows the circuit to ﬁt

directly into the module’s socket. Fig-

ure 1 shows a typical layout for a

2.28" by 2.40" circuit board.

DC227A-A is designed for 500VDC

isolation and lowest cost; it uses a

standard Coiltronics VERSA-PAC™

transformer and a Pulse Engineering

inductor for the output ﬁlter. DC227A-

B has 1500V isolation and uses a

semicustom transformer, also from

Coiltronics. DC227A-C has 500VDC

isolation and achieves the highest

efﬁciency using a Panasonic type PCC-

S1 inductor for the output ﬁlter. The

efﬁciency curves in Figures 2–5 are

quite competitive, reaching 85% for

the DC227A-C with a 5V output. The

efﬁciency at 3.3V out is somewhat

lower, due to the ﬁxed losses of the

output rectiﬁer.

Circuit Description

This single-ended forward converter

operates at a nominal switching fre-

quency of 200kHz. Referring to the

schematic in Figure 6, pulse width

modulation is controlled by U1, an

LT1247 current mode PWM control-

ler. Transformer T2 and optocoupler

Q7 provide galvanic isolation. C2 is a

Figure 1. Control (left) and power component (right) views of demonstration circuit DC227, a complete 35W DC/DC converter

in a 2.28" by 2.40" footprint

VERSA-PAC is a trademark of Coiltronics, Inc.

DESIGN IDEAS

35 Watt Isolated DC/DC Converter

Replaces Modules at Half the Cost

................................................... 30

Robert Sheehan

Comparator Circuit Provides

Automatic Shutdown of the LT1795

High Speed ADSL Power Ampliﬁer

................................................... 33

Tim Regan

SMBus Controlled

CCFL Power Supply ..................... 35

Jim Williams

Triple Output TFT-LCD Bias Supply

Uses All Ceramic Capacitors ....... 36

Gary Shockey

Low Noise Boost VCO Power Supply for

Portable Applications .................. 37

Ted Henderson

Linear Technology Magazine • September 1999

local bypass cap to reduce common

mode–induced current.

To achieve fast start-up time, a

hysteretic buck regulator is used for

the bias supply power. U2, an LT1431

shunt voltage regulator, provides con-

trol for this function, with Q1 acting

as the switch element; L2 and C21

provide output ﬁltering. Q2 and Q4

protect the circuit during a hot plug,

making this a very robust design; it is

also impervious to output short cir-

cuits. The input surge voltage is

limited to 80V by the rating of Q1–Q4.

The main switching power path

through T2 comprises L1 and C18 as

the input ﬁlter, Q6 as the primary

switch, D7 as the secondary rectiﬁer

and L3 and C14, C16, C17 and C20

as the secondary ﬁlter. Transient volt-

50 01234567

EFFICIENCY (%)

OUT

(A)

= 36V

= 48V

= 72V

50 01234567

EFFICIENCY (%)

IOUT (A)

VIN = 36V

VIN = 48V

VIN = 72V

50 01234567

EFFICIENCY (%)

IOUT (A)

VIN = 36V

VIN = 48V

VIN = 72V

50 01234567

EFFICIENCY (%)

OUT

(A)

= 36V

= 48V

= 72V

age suppressor D8 is used to protect

Schottky diode D7 during large-sig-

nal transient conditions. Power is

transferred during the on cycle of Q6

and integrated by the output ﬁlter,

just as in a buck regulator. The input

ﬁlter component values for L1 and

C18 are optimal and should not be

changed without careful evaluation.

C19 damps the input ﬁlter and will

provide adequate stability for large

values of input inductance. See LTC

Application Note 19 for a discussion

of input ﬁlter stability analysis.

Output voltage feedback is con-

trolled using U3, another LT1431

shunt voltage regulator, as an error

ampliﬁer. In the event of a fault on the

output power or sense lines, Z1/Q5

will override U3 and provide overvolt-

age protection. R10 and R21 are sized

to handle any overvoltage condition.

During an output short-circuit con-

dition, the LT1247 is able to decrease

the on time of Q6 to less than 200ns.

This results in good control of the

output short-circuit current, keeping

power dissipation to a manageable

level.

The demonstration circuit uses

surface mount devices for Q6 and D7.

For elevated temperature operation

at the full rated load, TO-220 devices

can by mounted on a standard half-

brick heat sink.

For –48V inputs that require hot

swap capability, the LT1640H nega-

tive voltage HotSwap™ controller

provides a seamless interface.

Demonstration circuit DC223A-B

using the LT1640HCS8 is the recom-

mended solution for use with the

DC227A.

Conclusion

At 35 watts, the topology presented

here is one of the most common used

by the module manufacturers. This is

only one solution for isolated power,

and opens up many possibilities for

other input and output voltage

combinations. For lower power, dem-

onstration circuit DC211 using the

LT1425 isolated ﬂyback switching

regulator is designed for 10 watts.

Demonstration circuit DC259 using

the LT1339 adds synchronous recti-

ﬁcation, providing a high efﬁciency

solution for 50 watts. See the DC/DC

Converter Module section of LTC’s

Volume 1 1999 New Products Catalog

for additional information.

Figure 3. DC227A-A/B 5V output efﬁciency

(typical)

Figure 4. DC227A-C 3.3V output efﬁciency

(typical)

Figure 5. DC227A-A/B 3.3V output efﬁciency

(typical)

Figure 2. DC227A-C 5V output efﬁciency

(typical)

http://www.linear-tech.com/ezone/zone.html

Articles, Design Ideas, Tips from the Lab…

Linear Technology Magazine • September 1999

DESIGN IDEAS

COLLECTOR

COMP

MID

GND-F

TOP

REF

GND-S

U2 LT1431CS8

COMP

SENSE

REF

OUTPUT

GND

U1 LT1247CS8

COLLECTOR

COMP

MID

GND-F

REF

TOP

GND-S

U3 LT1431CS8

36V–72V

C19

12µF

100V R18

100k

R16

10k

FMMTA56

R11

FMMTA56

R12

20k

FMMTA06

BAT54

FMMD914

C12

1000pF

1000V

100k

D2 BAT54

R19

10ΩD5

BAS21

R13

47k

1.30k R3

RT1

NTC

10k AT 25°C

3300pF C1

4700pF

0.01µF

845Ω

8.2k

0.1µF

1µF

–V

ON/OFF

4.7µH

FMMTA06

R25 4.7Ω

R23

16.2k

C10

0.01µF

C18

2.2µF

100V

1mH C15

470pF

200V

R24

4.75k Q6

MTB20N20E

3.9Ω

FMMD914

22.1Ω

1500pF

1SMB36A

MBRB-

2545CT

(DUAL)

2200pF

9.3µH 7.2A

C14, C16,

C17, C20

220µF, 10V

×4

R14

220Ω

1/4W

OUT

–V

OUT

5.6V

R31

100Ω

R30

FMMT3904

+SENSE

TRIM

–SENSE

R21

100Ω

1/2W

R27

15.4k

0.1%

R28

7.41k

0.1%

JP3

R20

10k

R22

4.99k

0.1%

R10

100Ω

1/2W

R26

100Ω

0.047µF

C11

0.022µF

C13

0.022µF

FMMD914

MOC207

R29

330Ω

1000pF, 1000V

2200pF SAFETY

RECOGNIZED

JP1

JP2

R33

33Ω

1/4W

REPMUJDNA2T3L2C

A-A722CDNOITALOSICDV0051PJ;0021-5PVSCINORTLIOC36635-EPGNIREENIGNEESLUPTAGXK201Y8081JVNOMIRTIV

B-A722CDNOITALOSICDV00512PJ;2X-18241-20XTCSCINORTLIOC36635-EPGNIREENIGNEESLUPCG-K22R7X5403MHGATARUM

C-A722CD NOITALOSICDV005 YCNEICIFFEHGIH 1PJ;0021-5PVSCINORTLIOCAH2R7FAPQTECINOSANAPTAGXK201Y8081JVNOMIRTIV

NOTES:

1. CHANGE Z1 TO A 3.3V ZENER FOR 3.3V OUT

2. JP3: OPEN = 3.3V; SHORTED = 5V

CHASSIS

R15 1k

C21

10µF

25V

R17

10Ω, 1/4W

L1: COILCRAFT D01608C-472 (847) 639-6400

L2: COILCRAFT D03316P-105

T1: DALE LPE-3325-A142 (605) 665-1627

C14, C16, C17, C20: AVXTPSE227M010R0100 (803) 946-0362

C18: ITW PAKTRON 225100ST3827T (804) 239-6941

C19: SANYO 100MV12GX (619) 661-6835

Figure 6. 35W isolated DC/DC converter schematic diagram

Linear Technology Magazine • September 1999

DESIGN IDEAS

Comparator Circuit Provides Automatic

Shutdown of the LT1795 High Speed

ADSL Power Ampliﬁer

by Tim Regan

Introduction

Data transmission standards such

as HDSL2, ADSL (both Full Rate and

G.Lite) and VDSL (collectively known

as xDSL) require the combined speed,

output power and dynamic range

capabilities of the LT1795 to drive the

telephone line. In a typical data com-

munications installation, in both the

telephone central office and office

building sites, hundreds of telephone-

wire pairs are brought together into a

line multiplexer. These multiplexers

compact the individual line driver and

receiver circuits to save space. Eight

lines per PC card are often imple-

mented. This tight partitioning raises

the challenges of both power and heat

management in each installation.

A simple comparator circuit can be

used to monitor the activity of an

individual phone line and completely

shut down the line driver when not in

use. This provides a means of imp-

lementing activity-based power

consumption with reduced overall

heat generation.

Controlling Power Dissipation

The LT1795 is a dual, high speed,

current feedback amplifier with high

output current capability. With a

50MHz Gain Bandwidth product,

900V/ms slew rate and an output

stage that can source and sink 500mA,

the LT 1795 is ideal for use as a low

distortion differential line driver in

very high data rate modem applica-

tions.

For the LT1795 to obtain full perfor-

mance, a fair amount of quiescent

operating current is required—typi-

cally 60mA. When supplied by ±15V

power supplies to obtain maximum

dynamic output range, the quiescent

power dissipation with no load is typi-

cally 1.8 watts (30V × 60mA). In these

applications, however, full perfor-

mance is not required at all times or

for all applications. To address this

issue, the LT1795 provides the ability

to completely shut down the driver or

to tailor the quiescent operating cur-

rent to match the actual requirements

of a particular application. Figure 1

illustrates how this control is

implemented.

Pins 10 and 11 of the LT1795 com-

bine to control the operating current

of the ampliﬁers. If the Shutdown

Reference (pin 11) is grounded and

the Shutdown input (pin 10) is driven

to a voltage greater than two diode

drops above ground, both ampliﬁers

are biased to “full speed ahead” with

maximum AC performance and also

maximum quiescent power dissipa-

tion. The current through the two

diodes shown in Figure 1 is internally

limited to 300µA and results in 60mA

of quiescent current for the ampliﬁ-

ers. Resistance can be added between

the Shutdown Reference pin and

ground to limit the maximum operat-

ing current of the ampliﬁers. This

programmability can optimize the

trade-off between AC performance

(primarily slew rate and bandwidth)

and quiescent package power dissi-

pation. Many applications require the

peak current drive capability of the

LT1795 but do not need the full band-

width and slew performance. Details

of this control can be found on the

LT1795 data sheet.

If the Shutdown input is grounded,

both ampliﬁers are disabled and the

total quiescent current drain for the

package is reduced to only 200µA.

This feature can be used to save a

substantial amount of power when

the line drivers are not in use at all

times. A simple comparator circuit,

as shown in Figure 2, provides a

timed, automatic shutdown when no

input signals are applied to the

ampliﬁers.

Timed Automatic Shutdown

In this circuit, an LT1795 is conﬁg-

ured as a unity gain differential driver

for a 100Ω transformer-coupled wire

pair. The two comparators in an

LT1720 package monitor the signals

on each of the input lines to the driver

ampliﬁer. If the signal on either input

exceeds the threshold set by the sen-

sitivity adjustment (shown to provide

a range from 65mV to 500mV peak),

the output of one or the other of the

comparators goes high immediately

and puts the LT1795 into action. The

LT1720 comparators feature a propa-

gation delay of only 4.5ns, allowing

them to respond to input signals well

beyond 10MHz in frequency.

The LT1720 outputs are wire-ORed

through a simple timing network.

When either output is high, timing

capacitor C

charges quickly to 5V

through the diodes shown. While a

signal is present, one or the other

comparator output keeps C

charged.

The voltage on the timing capacitor is

buffered by a third comparator, an

LTC1440, to provide a sharp 0V to 5V

shutdown/enable signal to the

LT1795.

SHUTDOWN

REFERENCE

ADJUST

0µA–

300µA

TO INTERNAL

AMPLIFIER

CIRCUITRY

5V ON

OFF

Figure 1. Shutdown/I

adjust for the LT1795

Linear Technology Magazine • September 1999

DESIGN IDEAS

–

–e

49.9Ω

1k 1k

100Ω

1/2 LT1795

1:1

±e

OUT

15V

2, 19

–15

S/D REF

10 S/D

1k 1k

0.1µ

6.2k

100k

IN4148

100k

IN4148

50k*

1µF

1/2 LT1720

510Ω

1µF

2.61k 649k

HYST

LTC1440

SHUTDOWN

1.2V

INPUT SENSITIVITY ADJUSTMENT

65mV TO 500mV PEAK

1.3V

of the LTC1440—that is, when no

further signals are applied to the input

of the circuit—the output of the

LTC1440 snaps low and immediately

shuts down the LT1795 power ampli-

ﬁer. The discharge voltage of C

fairly slow moving but the input hys-

teresis set between pins 4 and 5 of the

LTC1440 allows a clean ON-to-OFF

transition.

With the component values pro-

vided in Figure 2, the circuit “wakes

up” and properly ampliﬁes the input

signal in approximately 50µs. Most of

this delay is in the charging of C

and

the propagation delay of 8µs through

the LTC1440. With the Shutdown pin

driven high, the LT1795 is fully up to

speed in only 1µs. For a DSL applica-

tion, this wake-up time occurs during

the initial line “training-up” interval

of the data transmission sequence.

When the input signal is removed,

discharges in approximately 65ms

to put the driver into a low power

dissipation state until the next data

transmission. This time-out interval

and wake-up time can be easily tai-

lored through the selection of C

and

and R

. If the two timing resis-

tors are of equal value, the OFF time

interval is set by the relationship: t =

0.713 • R

• C

Conclusion

This simple comparator sensing and

timing circuit provides automatic con-

trol over the power dissipation of a

high speed power ampliﬁer line driver.

With no signal present there is no

wasted quiescent power. This same

system enhancement can also be

achieved through direct logic control

of the LT1795 shutdown feature. This

would require a 0V to 5V control

signal for each line, which is set or

cleared in synchronization with each

data transmission interval.

The LTC1440 is a CMOS compara-

tor with a built-in voltage reference

and programmable hysteresis. The

1.18V reference is used for DC bias-

ing to set a stable input threshold for

the circuit. The input current of this

comparator, 10pA, does not present a

signiﬁcant load on the timing

capacitor.

When the signals of both inputs to

the driver ampliﬁer drop below the

threshold, both outputs of the LT1720

go low. While low, the timing capaci-

tor discharges exponentially through

the two 100k resistors, R

and R

. If

the voltage on C

is given time to

discharge below the 1.18V threshold

Figure 2. Automatic shutdown of the LT1795 power ampliﬁer

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

Linear Technology Magazine • September 1999

DESIGN IDEAS

SMBus Controlled CCFL Power Supply

by Jim Williams

Figure 1 shows a cold cathode fluo-

rescent lamp (CCFL) power supply

that is controlled via the popular

SMBus interface. The LT1786 CCFL

switching regulator receives the

SMBus instruction. The IC converts

this instruction to a current, which

appears at the I

OUT

pin. This current,

routed to the I

CCFL

pin, provides a set

point for switching regulator opera-

tion. The resultant duty cycle at the

pin pulls current through L2. L2,

acting as a switched current sink,

drives a resonant Royer converter

composed of Q1–Q2, C1 and L1. The

high voltage sine wave produced at

L2’s secondary drives the floating

lamp.

Current flow into the Royer con-

verter is monitored by the IC at pin 13

(“Royer” in Figure 1).

Royer current

correlates tightly with lamp current,

which, in turn, is proportional to

intensity. The IC compares the Royer

current to the SMBus-derived cur-

rent, closing a lamp-intensity control

loop. The SMBus permits wide-range

regulated lamp-intensity control and

allows complete IC shutdown. Opti-

mal display and lamp characteristics

permit 90% efficiency. The circuit is

calibrated by correlating SMBus

instruction codes with attendant RMS

lamp current. Detailed information

on circuit operation and measure-

ment techniques appears in the

references below.

BAT

8V TO 28V

CCFL

DIO

CCFL V

AGND

SHDN

SMBSUS

ADR

CCFL V

BULB

BAT

ROYER

OUT

SCL

SDA

CCFL

PGND

LT1786F

LAMP

10 6

750Ω

100µH

321 5

C7, 1µF

C1*

0.068µF

1000pF

220k

100k

C3A

2.2µF

35V

2.2µF

3V ≤ V

≤ 6.5V

SMBus

HOST

C3B

2.2µF

35V

27pF

3kV

Q2* Q1*

SHUTDOWN

C1 MUST BE A LOW LOSS CAPACITOR, C1 = WIMA MKI OR MKP-20

(914) 347-2474

= PANASONIC ECH-U

(201) 348-7522

Q1, Q2 = ZETEX ZTX849

OR ROHM 2SC5001

0µA TO 50µA I

CCFL

CURRENT GIVES

0mA TO 6mA LAMP CURRENT

FOR A TYPICAL DISPLAY.

FOR ADDITIONAL CCFL/LCD CONTRAST APPLICATION CIRCUITS,

REFER TO THE LT1182/83/84/84F DATA SHEET

BAT85

8D1

1N5818

L1 = COILTRONICS CTX210605

L2 = COILTRONICS CTX100-4

*DO NOT SUBSTITUTE COMPONENTS

COILTRONICS (561) 241-7876

(516) 543-7100

(800) 955-7646

Figure 1. 90% efficient floating CCFL with 2-wire SMBus lamp-current control

References:

1. Williams, Jim. Linear Technology

Application Note 65: A Fourth

Generation of LCD Backlight

Technology. November 1995.

2. LT1786F Data Sheet. Linear

Technology Corp. 1998.

Local historians can’t be certain, but this may be

the only IC pin ever named after a person.

Analog Inputs Welcome

The scaleable ampliﬁcation system

detailed in Figure 4 can be driven

with analog inputs while still main-

taining full isolation. Such a system

is detailed in Figure 5, where the

analog input is ﬁltered (to prevent

ailiasing) and converted to PWM. Fig-

ure 5 goes on to show the use of an

RS485 differential driver to drive a

twisted pair line. The receiver end of

the twisted pair line is terminated

with a resistor and put across the

isolation barrier. This provides very

good ESD protection on both ends of

the line.

Conclusion

The LT1684 is useful in a wide variety

of applications. The LT1684 is a highly

integrated solution for use in any

system that requires digital control of

high output voltage or high output

power.

Ring Tone, continued from page 29

Linear Technology Magazine • September 1999

DESIGN IDEAS

Triple Output TFT-LCD Bias Supply

Uses All Ceramic Capacitors by Gary Shockey

Current power supply require-

ments for TFT-LCD panels call for an

8V or 10V main supply plus two or

more auxiliary outputs. The overall

layout must be small and meet tight

height requirements of under 2mm.

Bulky inductors and capacitors must

be eliminated if the design is to meet

space requirements. The circuit de-

scribed in this design idea features

the new LT1949 and delivers 8V at

200mA from 3.3V while generating

auxiliary 24V and –8V outputs capable

of 10mA of output current.

The LT1949 is a boost switching

regulator that comes in the MSOP-8

package and has an integrated 1.1A

NPN power transistor. For this appli-

cation to be small and have a low

profile, the boost converter must

switch at a high frequency, which

allows compact inductors to be used,

and must be able to work with ceramic

output capacitors. The LT1949 does

both of these things. The switching

frequency is fixed at 600kHz and the

external compensation pin allows for

loop characteristics to be tuned so

that tiny ceramic output capacitors

can be used.

To understand circuit operation,

refer to Figure 1. The LT1949 gener-

ates the 8V output in the normal

boost mode configuration, while using

charge pumps for the 24V and –8V

outputs. During boost operation, the

SW pin is switching between V

OUT

and ground. When at V

OUT

, capacitor

C6 is charged to V

OUT

through D5.

When the SW pin flies to ground, C6

holds its charge, causing D6 to be

forward biased, charging C5 to –8V.

The positive 24V output is developed

in a similar fashion except that V

OUT

is tripled. Figure 2 details the tran-

sient response of V

OUT

to an 80mA to

200mA load step.

OUT

100mV/DIV

INDUCTOR

CURRENT

500mA/DIV

LOAD

STEP 200mA

80mA

50µs/DIV

Figure 2. Transient response for an 80mA to 200mA load step

Figure 1. 3.3V to 8V/200mA DC/DC converter with auxiliary 24V and –8V outputs

SHDN

LBI

LBO

VIN SW

VCGND

LT1949

24V/10mA

VOUT

8V/200mA

–8V/10mA

4.7µF

10µF

40.2k

7.5k

47k

680pF

10µF

10µH

D1 D2 D3 D4

0.1µFC8

0.1µFC9

0.1µF

VIN

3.3V

4.7µFD5

SHUTDOWN

L1: SUMIDA CDRH628B-100

(847) 956-0666

C1, C2: TAIYO YUDEN TMK432BJ106MN X7R 1210

(408) 573-4150

C4: TAIYO YUDEN TMK325BJ475MN X5R 1210

C5, C6: TAIYO YUDEN LMK316BJ475ML X5R 1206

C7–C9: 0.1µF, 50V CERAMIC

D1–D6: ZETEX FMMD7000 DUAL DIODE

(516) 543-7100

D7: MBRM120LT3

R3: 5.76k FOR 10V/–10V/30V OUTPUTS

http://www.linear-tech.com/ezone/zone.html

Articles, Design Ideas, Tips from the Lab…

Linear Technology Magazine • September 1999

DESIGN IDEAS

Low Noise Boost VCO Power Supply

for Portable Applications

Introduction

Many portable RF products use volt-

age controlled oscillators (VCOs) to

generate the RF carrier frequency.

These applications often require low

noise VCO power supply voltages that

are greater than the primary battery

supply. A DC/DC converter powering

a low noise linear regulator is often

used. Unfortunately, there are sev-

eral disadvantages to this solution.

The DC/DC converter tends to pro-

duce noise that may not be rejected

by the regulator, resulting in regula-

tor output noise far greater than the

thermal noise levels. The linear regu-

lator may require a large output

compensation capacitor with specific

ESR requirements. The board area

for both devices and support compo-

nents can be large. The LTC1682

charge pump DC/DC voltage con-

verter has been designed to minimize

these issues. The charge pump and

linear regulator have been mutually

optimized for minimum regulator out-

put noise. The linear regulator was

designed to operate with several dif-

ferent types of output capacitors

including small, low value, low ESR

ceramic capacitors. The charge pump

voltage converter and low dropout

linear regulator are combined on one

die and assembled in an extremely

small MS8 package. Both fixed and

adjustable output voltage versions

are available to cover the widest pos-

sible output voltage range.

VCO Power Supply

Figure 1 shows the LTC1682 generat-

ing a 4.2V low noise power supply for

a 900MHz VCO with an input voltage

range of 2.5V to 4.4V. Figure 2 shows

the close-in phase noise of the VCO

operating open loop and Figure 3

shows the typical peak-to-peak noise

voltage at V

OUT

Conclusion

The new LTC1682 family of charge

pump DC/DC voltage converters rep-

resents a complete low noise solution

for VCO power supplies. A wide input

voltage range of 1.8V to 4.4V, low

dropout voltage, low quiescent cur-

rent, low external parts count and

small board area make these devices

ideal for portable applications.

36k

SHUTDOWN

15k

100k

4.7µF

0.22µF

2.5V TO

4.4V

4.7µF

4.2V

OUT

SHDN

GND

CPO

–

LTC1682

VCO

MURATA

MQE001-902

1µF

4.7µF1000pF

1000pF1000pF 4.7µF

VCO

OUTPUT

902MHz

AMPLITUDE

10dB/DIV

CENTER = 902MHz

SPAN = 100kHz

SWP = 10s

RESBW = 1kHz

VBW = 30Hz

REF = 0dBm V

OUT

200µV/DIV

100µs/DIV

OUT

= 4.7µF

OUT

= 10mA

OUT

= 4.0V

BW = 10Hz–2MHz

Figure 1. 4.2V VCO power supply

Figure 2. Close-in phase noise Figure 3. Output noise voltage

by Ted Henderson

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

For more information on parts featured in this issue, see

http://www.linear-tech.com/go/ltmag

Linear Technology Magazine • September 1999

NEW DEVICE CAMEOS

LT1762/LT1763 Low Noise,

Micropower, Low Dropout

Regulators Save Current in

Battery-Powered Applications

The LT1762 and LT1763 are low noise,

low dropout linear regulators. The

LT1762 is rated for 150mA of output

current, whereas the LT1763 is rated

for 500mA. The typical dropout volt-

age for either regulator at the rated

output current is 300mV. The reg-

ulators are designed for use in

battery-powered systems, with 25µA

operating current for the LT1762 and

30µA for the LT1763; both regulators

feature a 0.1µA shutdown state. Qui-

escent current is well controlled for

these devices; it does not rise in drop-

out as is the case with many other

regulators.

The LT1762 and LT1763 regula-

tors feature low noise operation. With

the addition of an external 0.01µF

bypass capacitor, output voltage noise

over the 10Hz to 100kHz bandwidth

is reduced to 30µV

RMS

for both

regulators. Both regulators can oper-

ate with small capacitors, as low as

2.2µF for the LT1762 and 4.7µF for

the LT1763. Small ceramic capaci-

tors can be used with either device

without the need for additional series

resistance as is common with other

regulators. Internal protection cir-

cuitry on both regulators includes

reverse-battery protection, current

limiting, thermal limiting and reverse-

current protection.

Both regulators are available in

ﬁxed output voltages of 2.5V, 3V,

3.3V and 5V, or as an adjustable

device with an output voltage range of

1.22V to 20V. The LT1762 regulators

are packaged in the 8-lead MSOP

package and the LT1763 regulators

are available in the 8-lead SO package.

LTC2050:

Zero-Drift Operational

Ampliﬁer in SOT-23

The LTC2050 is the latest zero-drift

operational ampliﬁer in the LTC fam-

ily. Available in the SOT-23 and SO-8,

the LTC2050 permits single-supply

operation down to 2.7V. The op amp

consumes 800µA of current and the

6-lead SOT-23 and SO-8 packages

include a shutdown pin that drops

supply current below 10µA. Input

common mode range has been

extended from the negative rail to

within 1V (typical) of the positive rail.

The LTC2050’s speciﬁcations rival

those of the other members of the

zero-drift op amp family, with typical

offset voltages of 1µV, offset drifts of

10nV/°C, DC to 10Hz noise of

1.5µV

P-P

, and a gain-bandwidth of

3MHz. The LTC2050 uses the indus-

try-standard op amp pinout and

requires no external components.

LTC1563-2/LTC1563-3:

Easy-to-Use 3V, Rail-to-Rail,

DC Accurate, Active RC

Lowpass Filter Family

The LTC1563-2 and LTC1563-3 are

active RC 4th order lowpass ﬁlters

suitable for systems with resolution

of 16 bits or more. They operate on a

supply voltage as ranging from 2.7V

to ±5V. They support cutoff frequen-

cies from 5kHz to 256kHz with

rail-to-rail input and output. Each

part comes in the narrow SSOP-16

package (SO-8 footprint).

The LTC1563-2 and LTC1563-3

are extremely easy to use: unlike con-

ventional 4th order, discrete RC active

lowpass filters, which require the cal-

culation of a minimum of six different

external resistors and four different

external capacitors, they require only

six equal-valued resistors to produce

a unity gain, 4th order Butterworth

(LTC1663-2) or Bessel (LTC1563-3)

filter. The calculation of the resistor

values is trivial—complex algorithms

or filter design software is not needed.

By simply allowing the six external

resistors to be of different values,

gain and other transfer functions (for

example, Chebyshev, Gaussian tran-

sitional and linear-phase equiripple)

are achieved. Cascading two devices

forms an 8th order lowpass ﬁlter.

Simple resistor-value tables make the

design of any all-pole lowpass ﬁlter

elementary. Complicated design al-

gorithms are a thing of the past.

The LTC1563-2 and LTC1563-3

also feature excellent DC offset (typi-

cally less than 1mV); they are DC

accurate and their broadband noise

ranges from 30µV

RMS

to 60µV

RMS

de-

pending on the cutoff frequency. For

cutoff frequencies below 25.6kHz, the

parts have a low power mode where

the supply current is typically 1mA.

For higher frequencies, the supply

current is 10mA typically. A shut-

down mode is also provided to limit

the supply current to less than

10µA.

New Device Cameos

For further information on any

of the devices mentioned in this

issue of Linear Technology, use

the reader service card or call

the LTC literature service

number:

1-800-4-LINEAR

Ask for the pertinent data sheets

and Application Notes.

Linear Technology Magazine • September1999

DESIGN TOOLS

Applications on Disk

FilterCAD™

2.0 CD-ROM — This CD is a powerful ﬁlter

design tool that supports all of Linear Technology’s high

performance switched capacitor ﬁlters. Included is Fil-

terView™, a document navigator that allows you to

quickly ﬁnd Linear Technology monolithic ﬁlter data

sheets, the FilterCAD manual, application notes, design

notes and

Linear Technology

magazine articles. It

does

not

have to be installed to run FilterCAD. It is not

necessary to use FilterView to view the documents, as

they are standard .PDF ﬁles, readable with any version

of Adobe Acrobat™. FilterCAD runs on Windows

3.1 or

Windows 95. FilterView requires Windows 95. The

FilterCAD program itself is also available on the web and

will be included on the new LinearView™ CD.

Available at no charge.

Noise Disk — This IBM-PC (or compatible) program

allows the user to calculate circuit noise using LTC op

amps, determine the best LTC op amp for a low noise

application, display the noise data for LTC op amps,

calculate resistor noise and calculate noise using specs

for any op amp. Available at no charge

SPICE Macromodel Disk — This IBM-PC (or compat-

ible) high density diskette contains the library of LTC op

amp SPICE macromodels. The models can be used with

any version of SPICE for general analog circuit simula-

tions. The diskette also contains working circuit examples

using the models and a demonstration copy of PSPICE™

by MicroSim. Available at no charge

SwitcherCAD™ — The SwitcherCAD program is a pow-

erful PC software tool that aids in the design and

optimization of switching regulators. The program can

cut days off the design cycle by selecting topologies,

calculating operating points and specifying component

values and manufacturer’s part numbers. 144 page

manual included. $20.00

SwitcherCAD supports the following parts: LT1070 se-

ries: LT1070, LT1071, LT1072, LT1074 and LT1076.

LT1082. LT1170 series: LT1170, LT1171, LT1172 and

LT1176. It also supports: LT1268, LT1269 and LT1507.

LT1270 series: LT1270 and LT1271. LT1371 series:

LT1371, LT1372, LT1373, LT1375, LT1376 and LT1377.

Micropower SwitcherCAD™ — The MicropowerSCAD

program is a powerful tool for designing DC/DC convert-

ers based on Linear Technology’s micropower switching

regulator ICs. Given basic design parameters,

MicropowerSCAD selects a circuit topology and offers

you a selection of appropriate Linear Technology switch-

ing regulator ICs. MicropowerSCAD also performs circuit

simulations to select the other components which sur-

round the DC/DC converter. In the case of a battery

supply, MicropowerSCAD can perform a battery life

simulation. 44 page manual included. $20.00

MicropowerSCAD supports the following LTC micro-

power DC/DC converters: LT1073, LT1107, LT1108,

LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,

LT1300, LT1301 and LT1303.

Technical Books

1990 Linear Databook, Vol I —This 1440 page collec-

tion of data sheets covers op amps, voltage regulators,

references, comparators, ﬁlters, PWMs, data conver-

sion and interface products (bipolar and CMOS), in both

commercial and military grades. The catalog features

well over 300 devices. $10.00

1992 Linear Databook, Vol II — This 1248 page supple-

ment to the 1990 Linear Databook is a collection of all

products introduced in 1991 and 1992. The catalog

contains full data sheets for over 140 devices. The 1992

Linear Databook, Vol II is a companion to the 1990

Linear Databook, which should not be discarded.

$10.00

1994 Linear Databook, Vol III —This 1826 page supple-

ment to the 1990 and 1992 Linear Databooks is a

collection of all products introduced since 1992. A total

of 152 product data sheets are included with updated

selection guides. The 1994 Linear Databook Vol III is a

companion to the 1990 and 1992 Linear Databooks,

which should not be discarded. $10.00

1995 Linear Databook, Vol IV —This 1152 page supple-

ment to the 1990, 1992 and 1994 Linear Databooks is a

collection of all products introduced since 1994. A total

of 80 product data sheets are included with updated

selection guides. The 1995 Linear Databook Vol IV is a

companion to the 1990, 1992 and 1994 Linear Databooks,

which should not be discarded. $10.00

1996 Linear Databook, Vol V —This 1152 page supple-

ment to the 1990, 1992, 1994 and 1995 Linear Databooks

is a collection of all products introduced since 1995. A

total of 65 product data sheets are included with updated

selection guides. The 1996 Linear Databook Vol V is a

companion to the 1990, 1992, 1994 and 1995 Linear

Databooks, which should not be discarded. $10.00

1997 Linear Databook, Vol VI —This 1360 page supple-

ment to the 1990, 1992, 1994, 1995 and 1996 Linear

Databooks is a collection of all products introduced

since 1996. A total of 79 product data sheets are in-

cluded with updated selection guides. The 1997 Linear

Databook Vol VI is a companion to the 1990, 1992, 1994,

1995 and 1996 Linear Databooks, which should not be

discarded. $10.00

1990 Linear Applications Handbook, Volume I —

928 pages full of application ideas covered in depth by

40 Application Notes and 33 Design Notes. This catalog

covers a broad range of “real world” linear circuitry. In

addition to detailed, systems-oriented circuits, this hand-

book contains broad tutorial content together with liberal

use of schematics and scope photography. A special

feature in this edition includes a 22-page section on

SPICE macromodels. $20.00

1993 Linear Applications Handbook, Volume II —

Continues the stream of “real world” linear circuitry

initiated by the 1990 Handbook. Similar in scope to the

1990 edition, the new book covers Application Notes 40

through 54 and Design Notes 33 through 69. References

and articles from non-LTC publications that we have

found useful are also included. $20.00

1997 Linear Applications Handbook, Volume III —

This 976 page handbook maintains the practical outlook

and tutorial nature of previous efforts, while broadening

topic selection. This new book includes Application

Notes 55 through 69 and Design Notes 70 through 144.

Subjects include switching regulators, measurement

and control circuits, ﬁlters, video designs, interface,

data converters, power products, battery chargers and

CCFL inverters. An extensive subject index references

circuits in LTC data sheets, design notes, application

notes and

Linear Technology

magazines. $20.00

1998 Data Converter Handbook — This impressive

1360 page handbook includes all of the data sheets,

application notes and design notes for Linear

Technology’s family of high performance data converter

products. Products include A/D converters (ADCs), D/A

converters (DACs) and multiplexers—including the fast-

est monolithic 16-bit ADC, the 3Msps, 12-bit ADC with

the best dynamic performance and the ﬁrst dual 12-bit

DAC in an SO-8 package. Also included are selection

guides for references, op amps and ﬁlters and a glossary

of data converter terms. $10.00

Interface Product Handbook — This 424 page hand-

book features LTC’s complete line of line driver and

receiver products for RS232, RS485, RS423, RS422,

V.35 and AppleTalk

applications. Linear’s particular

expertise in this area involves low power consumption,

high numbers of drivers and receivers in one package,

mixed RS232 and RS485 devices, 10kV ESD protection

of RS232 devices and surface mount packages.

Available at no charge

Power Management Solutions Brochure — This 96

page collection of circuits contains real-life solutions for

common power supply design problems. There are over

70 circuits, including descriptions, graphs and perfor-

mance specifications. Topics covered include battery

chargers, desktop PC power supplies, notebook PC

power supplies, portable electronics power supplies,

distributed power supplies, telecommunications and

isolated power supplies, off-line power supplies and

power management circuits. Selection guides are pro-

vided for each section and a variety of helpful design

tools are also listed for quick reference.

Available at no charge.

Data Conversion Solutions Brochure␣ —␣ This 64 page

collection of data conversion circuits, products and

selection guides serves as excellent reference for the

data acquisition system designer. Over 60 products are

showcased, solving problems in low power, small size

and high performance data conversion applications—

with performance graphs and speciﬁcations. Topics

covered include ADCs, DACs, voltage references and

analog multiplexers. A complete glossary deﬁnes data

conversion speciﬁcations; a list of selected application

and design notes is also included.

Available at no charge

Telecommunications Solutions Brochure

—This 76

page collection of application circuits and selection

guides covers a wide variety of products targeted for

telecommunications. Circuits solve real life problems

for central office switching, cellular phones, high speed

modems, base station, plus special sections covering

–48V and Hot Swap

applications. Many applications

highlight new products such as Hot Swap controllers,

power products, high speed amplifiers, A/D converters,

interface transceivers and filters. Includes a telecom-

munications glossary, serial interface standards, protocol

information and a complete list of key application notes

and design notes. Available at no charge.

continued on page 40

DESIGN TOOLS

Information furnished by Linear Technology Corporation

is believed to be accurate and reliable. However, Linear

Technology makes no representation that the circuits

described herein will not infringe on existing patent rights.

Linear Technology Magazine • September 1999

LINEAR TECHNOLOGY CORPORATION

1630 McCarthy Boulevard

Milpitas, CA 95035-7417

(408) 432-1900 FAX (408) 434-0507

www.linear-tech.com

For Literature Only: 1-800-4-LINEAR

Acrobat is a trademark of Adobe Systems, Inc.; Windows

is a registered trademark of Microsoft Corp.; Macintosh

and AppleTalk are registered trademarks of Apple Com-

puter, Inc. PSPICE is a trademark of MicroSim Corp.

CD-ROM Catalog

LinearView — LinearView™ CD-ROM version 3.01 is

Linear Technology’s latest interactive CD-ROM. It al-

lows you to instantly access thousands of pages of

product and applications information, covering Linear

Technology’s complete line of high performance analog

products, with easy-to-use search tools.

The LinearView CD-ROM includes the complete product

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brary (Volumes I–VI) and the complete Applications

Handbook collection (Volumes I–III). Our extensive

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Linear Technology

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A powerful search engine built into the LinearView CD-

ROM enables you to select parts by various criteria,

such as device parameters, keywords or part numbers.

All product categories are represented: data conversion,

references, ampliﬁers, power products, ﬁlters and inter-

face circuits. Up-to-date versions of Linear Technology’s

software design tools, SwitcherCAD, Micropower

DESIGN TOOLS, continued from page 39

SwitcherCAD, FilterCAD, Noise Disk and Spice Macro-

model library, are also included. Everything you need to

know about Linear Technology’s products and applica-

tions is readily accessible via LinearView. LinearView

runs under Windows 95 and Macintosh

System 8.0 or

later. Available at no charge.

World Wide Web Site

Linear Technology Corporation’s customers can now

quickly and conveniently ﬁnd and retrieve the latest

technical information covering the Company’s products

on LTC’s internet web site. Located at www.linear-

tech.com, this site allows anyone with internet access

and a web browser to search through all of LTC’s

technical publications, including data sheets, applica-

tion notes, design notes,

Linear Technology

magazine

issues and other LTC publications, to ﬁnd information

on LTC parts and applications circuits. Other areas

within the site include help, news and information about

Linear Technology and its sales ofﬁces.

Other web sites usually require the visitor to download

large document ﬁles to see if they contain the desired

information. This is cumbersome and inconvenient. To

save you time and ensure that you receive the correct

information the ﬁrst time, the ﬁrst page of each data

sheet, application note and

Linear Technology

maga-

zine is recreated in a fast, download-friendly format.

This allows you to determine whether the document is

what you need, before downloading the entire ﬁle.

The site is searchable by criteria such as part numbers,

functions, topics and applications. The search is per-

formed on a user-deﬁned combination of data sheets,

application notes, design notes and

Linear Technology

magazine articles. Any data sheet, application note,

design note or magazine article can be downloaded or

faxed back. (ﬁles are downloaded in Adobe Acrobat PDF

format; you will need a copy of Acrobat Reader to view

or print them. The site includes a link from which you

can download this program.)

International

Sales Ofﬁces

FRANCE

Linear Technology S.A.R.L.

Immeuble “Le Quartz”

58 Chemin de la Justice

92290 Chatenay Malabry

France

Phone: 33-1-41079555

FAX: 33-1-46314613

GERMANY

Linear Technology GmbH

Oskar-Messter-Str. 24

D-85737 Ismaning

Germany

Phone: 49-89-962455-0

FAX: 49-89-963147

HONG KONG

Linear Technology Corp. Ltd.

Unit 2109, Metroplaza Tower 2

223 Hing Fong Road

Kwai Fong, N.T., Hong Kong

Phone: 852-2428-0303

FAX: 852-2348-0885

JAPAN

Linear Technology KK

5F NAO Bldg.

1-14 Shin-Ogawa-cho Shinjuku-ku

Tokyo, 162

Japan

Phone: 81-3-3267-7891

FAX: 81-3-3267-8510

KOREA

Linear Technology Korea Co., Ltd

Namsong Building, #403

Itaewon-Dong 260-199

Yongsan-Ku, Seoul 140-200

Korea

Phone: 82-2-792-1617

FAX: 82-2-792-1619

SINGAPORE

Linear Technology Pte. Ltd.

507 Yishun Industrial Park A

Singapore 768734

Phone: 65-753-2692

FAX: 65-752-0108

SWEDEN

Linear Technology AB

Sollentunavägen 63

S-191 40 Sollentuna

Sweden

Phone: 46-8-623-1600

FAX: 46-8-623-1650

TAIWAN

Linear Technology Corporation

Rm. 602, No. 46, Sec. 2

Chung Shan N. Rd.

Taipei, Taiwan, R.O.C.

Phone: 886-2-2521-7575

FAX: 886-2-2562-2285

UNITED KINGDOM

Linear Technology (UK) Ltd.

The Coliseum, Riverside Way

Camberley, Surrey GU15 3YL

United Kingdom

Phone: 44-1276-677676

FAX: 44-1276-64851

World Headquarters

Linear Technology Corporation

1630 McCarthy Boulevard

Milpitas, CA 95035-7417

Phone: (408) 432-1900

FAX: (408) 434-0507

U.S. Area

Sales Ofﬁces

NORTHEAST REGION

Linear Technology Corporation

3220 Tillman Drive, Suite 120

Bensalem, PA 19020

Phone: (215) 638-9667

FAX: (215) 638-9764

Linear Technology Corporation

15 Research Place

North Chelmsford, MA 01863

Phone: (978) 656-4750

FAX: (978) 656-4760

NORTHWEST REGION

Linear Technology Corporation

720 Sycamore Drive

Milpitas, CA 95035

Phone: (408) 428-2050

FAX: (408) 432-6331

SOUTHEAST REGION

Linear Technology Corporation

17000 Dallas Parkway, Suite 219

Dallas, TX 75248

Phone: (972) 733-3071

FAX: (972) 380-5138

Linear Technology Corporation

9430 Research Blvd.

Echelon IV Suite 400

Austin, TX 78759

Phone: (512) 343-3679

FAX: (512) 343-3680

Linear Technology Corporation

1080 W. Sam Houston Pkwy., Suite 225

Houston, TX 77043

Phone: (713) 463-5001

FAX: (713) 463-5009

Linear Technology Corporation

5510 Six Forks Road, Suite 102

Raleigh, NC 27609

Phone: (919) 870-5106

FAX: (919) 870-8831

CENTRAL REGION

Linear Technology Corporation

2010 E. Algonquin Road, Suite 209

Schaumburg, IL 60173

Phone: (847) 925-0860

FAX: (847) 925-0878

Linear Technology Corporation

Kenosha, WI 53144

Phone: (414) 859-1900

FAX: (414) 859-1974

SOUTHWEST REGION

Linear Technology Corporation

21243 Ventura Blvd., Suite 208

Woodland Hills, CA 91364

Phone: (818) 703-0835

FAX: (818) 703-0517

Linear Technology Corporation

15375 Barranca Parkway, Suite A-213

Irvine, CA 92618

Phone: (949) 453-4650

FAX: (949) 453-4765

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