February 1997 - Linear Technology

FEBRUARY 1997 VOLUME VII NUMBER 1

, LTC and LT are registered trademarks of Linear Technology Corporation. Adaptive Power, Burst Mode, C-Load,

LinearView, Micropower SwitcherCAD, PowerPath, SwitcherCAD and UltraFast are trademarks of Linear Technology

Corporation. Other product names may be trademarks of the companies that manufacture the products.

The New LT1425

Isolated Flyback Controller

by Kurk Matthews

Introduction

Low voltage circuitry, such as local

area networks (LAN), isolation ampli-

ﬁers and telephone interfaces,

frequently requires isolated power

supplies. The ﬂyback converter is of-

ten the choice for these low power

supplies because of its simplicity, size

and low parts count. Unfortunately,

designers are forced to add opto-

couplers and references in order to

achieve the desired output regulation

and transient response.

The new LT1425 provides a one-

chip solution for these and other

applications. The LT1425 is a 275kHz

current mode controller with an inte-

gral 1.25A switch designed primarily

to provide well regulated, isolated

voltages from 3V–20V sources. The

LT1425 is available in a 16-pin SO.

Features include a new error ampliﬁer

and load compensation circuitry that

eliminate the need for optocouplers

while maintaining output regulation

typically within a few percent.

Figure 1 shows a typical ﬂyback

LAN supply using the LT1425. Figure

1 also includes details on an alter-

nate transformer for a complete

PCMCIA type II height solution. The

output voltage is within 1% of –9V for

load currents of 0mA–250mA. Input

current is limited to 0.35 amps in the

event the output is short circuited.

The output voltage droops only 300mV

during a 50mA to 250mA load tran-

sient (see Figure 2). The off-the-shelf

transformers provide 500V

of isola-

tion. The high switching frequency

allows the use of small case size, low

cost, high value ceramic capacitors

on the input and output of the supply.

Isolated Feedback

The heart of the LT1425 is shown in

Figure 3. During S1’s off-time, the

voltage on the V

pin increases to V

+ (V

OUT

+ V

)/n, where n is the trans-

former turns ratio and V

is the output

diode voltage. Q1’s collector current

becomes I

CQ1

= (V

OUT

+ V

)/(n × R1).

R2 converts I

CQ1

into the input volt-

age for the transconductance feedback

ampliﬁer. C1 on the V

pin then inte-

grates the feedback ampliﬁer’s output

current. The voltage on the V

sets the current mode trip point.

Although we now have a means for

generating a feedback voltage, a few

problems remain. The feedback volt-

age is not present during S1’s on-time

or when the secondary current de-

cays to zero, which is often the case

with a discontinuous ﬂyback. To make

matters worse, T1’s leakage induc-

tance can cause large voltage spikes

at turn-off.

These issues are taken care of by

the error ampliﬁer enable block, which

incorporates enable-delay, collapse-

detect and minimum-enable-time

circuitry. Enable delay waits approxi-

mately 200ns after the switch turns

off before enabling the feedback

ampliﬁer, thus avoiding the leakage-

inductance spike. The collapse detect

continued on page 3

IN THIS ISSUE . . .

COVER ARTICLE

The New LT®1425

Isolated Flyback Controller ........ 1

Kurk Matthews

Issue Highlights ........................2

LTC in the News .........................2

DESIGN FEATURES

The LT1328: a Low Cost IrDA®

Receiver Solution for Data Rates

up to 4Mbps ...............................6

Alexander Strong

The LTC®1473 Dual PowerPath™

Switch Driver Simpliﬁes Portable

Power Management Design ........8

Jaime Tseng

The LTC1560-1: a 1MHz/500kHz

Continuous-Time, Low Noise,

Elliptic Lowpass Filter ............ 11

Nello Sevastopoulos

The LTC1594 and LTC1598:

Micropower 4- and 8-Channel

12-Bit ADCs ............................. 14

Kevin R. Hoskins and Marco Pan

LTC1474 and LTC1475 High

Efﬁciency Switching Regulators

Draw Only 10µA Supply Current

................................................17

Greg Dittmer

DESIGN IDEAS .................. 21–32

(Complete list on page 21)

DESIGN INFORMATION

Introducing the LT2078/LT2079

and LT2178/LT2179 Single

Supply, Micropower, Precision

Ampliﬁers in Surface Mount

Packages

................................................33

Raj Ramchandani

LTC1387 Single 5V

RS232/RS485 Multiprotocol

Transceiver .............................34

Y.K. Sim

New Device Cameos.................. 37

Design Tools ............................39

Sales Offices ............................40

LINEAR TECHNOLOGY

Linear Technology Magazine • February 1997

EDITOR’S PAGE

Issue Highlights

To mark the new year, we have a

collection of exciting new parts from

the design gurus at LTC. This issue’s

lead article features the new LT1425

isolated flyback converter, which pro-

vides a one-chip solution for low

voltage circuitry, such as local area

networks, isolation ampliﬁers and

telephone interfaces. The LT1425 is a

275kHz current mode controller with

an integral 1.25A switch, designed to

provide well regulated, isolated volt-

ages from 3V–20V sources.

Other power products featured in

this issue include the LTC1474 and

LTC1475 ultralow quiescent current,

high efﬁciency step-down switching

regulators. These regulators draw only

10µA at no load and require only four

external components to make a com-

plete, high efﬁciency (up to 92%)

step-down regulator. Low component

count and the parts’ tiny MSOP pack-

ages provide a minimum-area solution

to meet the limited space require-

ments of portable applications.

Another boon to the designers of

portable, battery-powered equipment

is the LTC1473 dual PowerPath switch

driver, which simplifies the design of

circuitry for switching between two

batteries or a battery and an AC

adapter. Presently, switching between

power sources is implemented with

discrete components—regulators,

comparators, references, glue logic,

MOSFET switches and drivers. These

solutions are expensive and occupy a

lot of printed circuit board space. The

LTC1473 drives low loss N-channel

MOSFET switches that direct power

in the main power path of a single or

dual rechargeable battery system, the

type found in most notebook comput-

ers and other portable equipment.

In the filter department, this issue

introduces the LTC1560-1, a high

frequency, continuous-time, low noise

ﬁlter. This device is a single-ended

input and output, 5th order elliptic

lowpass ﬁlter with a pin-selectable

cutoff frequency of 1MHz or 500kHz.

It requires no external components or

clocks and provides better than 60dB

LTC in the News…

Results of LTC’s second fiscal quarter

underscore the company’s standing as

an industry leader. Net sales for the

second fiscal quarter of 1997, ended

December 29, 1996, were $90,080,000.

Although this was a decrease of 6% over

net sales of $96,017,000 for the second

quarter of the previous year, this actu-

ally represented phenomenally good

performance, as we will see in just a

moment.

LTC reported net income for the quar-

ter of $31,631,000 or $0.40 per share,

a decrease of 8% from the second quar-

ter of the previous year. Sequentially,

the results for the second quarter were

essentially flat as compared to net sales

and income for the quarter ended Sep-

tember 29, 1996 of $90,063,000 and

$31,358,000 or $0.40 per share, re-

spectively. A cash dividend of $0.05 will

be paid on February 12, 1997 to share-

holders of record on January 24, 1997.

According to Robert H. Swanson,

president and CEO, “Although we en-

tered the quarter with reduced backlog,

we were able to achieve flat sequential

sales and profits with our return on

sales continuing to lead the industry. In

addition, cash and short-term invest-

ments grew by approximately $20 mil-

lion in the quarter. Customers’ demand

picked up moderately throughout the

quarter and our shorter lead times en-

abled us to ship some of their demand

within the quarter. We believe our mar-

ket is improving and we are optimistic

about the future.”

It’s becoming known around the

nation’s regional stock exchanges that

LTC is a leading pace setter and vital

indicator of the economic condition of

the electronics industry. For many

months, the Bloomberg Silicon Valley

Index and the Dean Witter Silicon Val-

ley Stock Index have included LTC’s

business performance in their assess-

ment of the industry.

Now, the New York-based Reuters

America News Service reports that the

Philadelphia Stock Exchange has added

two chip companies—one of which is

LTC—and an equipment maker to its

Semiconductor Index to replace three

stocks the exchange has removed. Late

last month, the exchange added Linear

Technology Corp. (LLTC), Xilinx Inc.

(XLNX.O) and Lam Research Corp.

(LRCX.O). To us at LTC, this move is

further proof that even in the realm of

high finance, “it’s a linear world.”

of stopband attenuation and 75dB

SNR, with only 0.3dB passband ripple.

In the data conversion area, we

debut the LTC1594 and LTC1598,

micropower 12-bit ADCs, which fea-

ture a 4- or 8-channel multiplexer,

respectively. These devices include

an auto shutdown feature that re-

duces power dissipation when the

converter is inactive. Nominal power

dissipation with the converter clocked

at 320kHz is typically 1.6mW. Each

ADC includes a simple, efﬁcient serial

interface that reduces interconnects

and, thereby, possible noise sources.

Reduced interconnections also reduce

board size and allow the use of pro-

cessors having fewer I/O pins, both of

which help reduce system costs.

For data communications, this is-

sue introduces the low cost LT1328

IrDA data receiver. This device con-

tains all the necessary circuitry to

convert current pulses from an exter-

nal photodiode to a digital TTL output

while rejecting unwanted lower fre-

quency interference. The LT1328 plus

six external components are all that

is required to make an IrDA-compat-

ible receiver. Power requirements for

the LT1328 are minimal: a single 5V

supply and 2mA of quiescent current.

This issue includes a varied selec-

tion of Design Ideas, including three

power supplies, a battery charger that

doubles as the main step-down con-

verter, a voltage controlled limiter for

video, a detector circuit for 470MHz

signals, and an evaluation of battery

life under a variety of load conditions.

Penultimately, we present Design

Information on the LT2078/LT2079

and LT2178/LT2179, improved

single-supply, precision surface

mount op amps, and the LT1387

single 5V multiprotocol transceiver.

We conclude with a selection of New

Device Cameos.

Linear Technology Magazine • February 1997

DESIGN FEATURES

–

1425_03.eps

Q1 Q2 2.6V

OUT

R3 

OCMP

CMPC

50k

2 A/V

T1

1:N

COMP LOGIC/

DRIVER

REF

1.224V ERROR

AMP

ENABLE

disables the feedback ampliﬁer when

the R

REF

voltage falls below 80% of the

1.224V reference. This natural col-

lapse of the feedback voltage occurs

sometime during the off-time in the

discontinuous ﬂyback mode (see Fig-

ure 4, Trace C) or when the switch

turns on in the continuous mode (see

Trace A). Finally, a 200ns minimum

enable time, which follows the enable

delay time, ensures that the error

ampliﬁer can pump up the V

node

during start up and other conditions

when V

OUT

is low.

This unique feedback system pro-

duces controlled output voltages while

maintaining fast dynamic response

not found in similar isolated ﬂyback

schemes. 200ns of leading edge, cur-

rent sense blanking is also included

to reject turn-on spikes.

Load Compensation

If the world were a perfect place, with

ideal transformers, diodes and ca-

pacitors, no additional compensation

would be required to maintain perfect

regulation. Unfortunately, as the load

current increases, the additional volt-

age drop due to secondary winding

resistance, the output diode and ca-

pacitor ESR results in decreased

output voltage. To compensate for

this change in output voltage, a cur-

rent is generated in Q2 (see Figure 3),

which is proportional to the average

primary current. Since primary cur-

rent changes with output load, the

effects of nonideal components are

minimized and regulation is possible

over a wide load range. R3 determines

the amount of load compensation.

Connecting R

CMPC

to ground defeats

the load compensation.

Figure 3. LT1425 isolated feedback block diagram

LT1425, continued from page 1

1425_01.eps

1

2

3

4

5

6

7

16

15

14

13

12

11

10

GND

N/C



REF



SYNC

SGND

GND

3.01k

R1 R2

MBRS130LT3

22.1k

0.1µF

LT1425

47pF

C3

10µF

25V

C4

10µF

25V

1.8k

OUT

COM

–9V

1000pF

C1

10µF

25V

INPUT

COM

C2

10µF

25V

0.1µF

100k

GND

SD

OCMP



CMPC



IN



PGND

GND

C1, C2, C3, C4 = MARCON THCS50E1E106Z CERAMIC

 CAPACITOR, SIZE 1812. (847) 696-2000



1TREMROFSNART

IRP

SNRUT OITARNOITALOSI EZIS

L(×W×)HI

YCNEICIFFE1D2D2R,1R6C,5C3R

ELAD

703A-1484-EPL63µH1:1:1CAV0057.01×5.11×mm3.6Am052%67DESUTONDESUTON74ΩFp033k3.31

SCINORTLIOC

38431-20XTC72µH1:1CAV00541×41×mm2.2Am002%078425N11LT0450RBM57ΩFp022k9.5



Figure 1. 5V to –9V/250mA isolated LAN supply

Figure 2. Transient response of LT1425 5V to

–9V converter

200mV/DIV

100mA/DIV

5ms/DIV

Linear Technology Magazine • February 1997

DESIGN FEATURES

of 10V–15V). The isolation voltage is

ultimately limited only by bobbin se-

lection and transformer construction.

The schematic shows details on build-

ing the transformer.

Figure 7 implements a 12V to

5V/1A step-down regulator with off-

the-shelf magnetics. The circuit uses

an external, cascoded 100V MOSFET

to extend the LT1425’s 35V maxi-

mum switch voltage limit. D1 and Q1

ensure the LT1425 does not start

until almost 9V, guaranteeing ad-

equate gate voltage for the MOSFET.

The MUR120 prevents the source from

rising above the gate at turn-off.

The circuit in Figure 8 achieves

even higher input voltages, this time

in the form of a –48V to 5V/2A iso-

lated telecom supply. The input

voltage is too high to directly run Q1

or the LT1425, so a bootstrap wind-

ing is used to provide feedback and

power for the IC after start-up. The

voltage to the V

pin is controlled by

D1, D2, Q2, Q3 and associated com-

ponents, which form the necessary

start-up circuitry with hysteresis.

Nothing happens until C1 charges

through R1 to 15V. At that point, Q2

turns on Q3, pulling the shutdown

pin high. Q3, in turn, latches Q2 on,

setting the turn-off voltage to ap-

proximately 11V. Switching begins

1

2

3

8

7

6

5



TOP VIEW

LT1424

SD

C

SYNC

SGND

CMPC

IN

SW

PGND

1425_05.eps

1425_06.eps

GND

N/C

LT1425

MBRS1100T3

T1*

REF

SYNC

SGND

GND

OCMP

CMPC

PIN 3 TO 4, 7 TURNS BIFILAR 34AWG

*PHILIPS EFD-15-3F3 CORE

GAP FOR PRIMARY

L = 40µH

0.12 INCH MARGIN TAPE



PIN 7 TO 8, 28 TURNS 40AWG

PIN 5 TO 6, 28 TURNS 40AWG

PIN 1 TO 2, 7 TURNS BIFILAR 34AWG

3 LAYERS 2 MIL

POLYESTER FILM

PGND

GND

3.01k

1N759

18.4k

0.1%

+15V

–15V

OUT

COM

7.32k

5V TO

15V

INPUT

COM

0.1µF

220pF

1µF

22µF

35V

15µF

35V

15µF

35V

1000pF

0.1µF

130 330pF

MBR0540LT1

Figure 5. LT1424 pinout

Figure 6. Fully isolated ±15V, ±60mA supply

D: ISW = 0.2A/DIV

C: VSW = 20V/DIV

B: ISW = 1A/DIV

A: VSW = 20V/DIV

1µs/DIV

Figure 4. Switch voltage and current for

Figure 1’s circuit with outputs of –9V/250mA

and –9V/30mA

The LT1424

The LT1424, devoted to ﬁxed output

voltage applications, is available in

an 8-pin SO package. The LT1424

retains the features of the LT1425

and incorporates the feedback, refer-

ence and load compensation resistors.

Figure 5 shows the LT1424 pinout.

Both the LT1424 and LT1425 include

shutdown and synchronization func-

tions. Consult the factory for further

information on the LT1424.

Typical Applications

Figure 6 shows a ±15V supply with

1.5kV of isolation. Output regulation

remains within ±3% over the entire

5V to 15V input voltage and ±60mA

output current range, even with one

output fully loaded and the other

unloaded (±1.5% with input voltages

and, before C1 has a chance to dis-

charge to 11V, the bootstrap winding

begins to supply power. If the output

is shorted, R2 prevents C1 from being

charged by the transformer’s leakage

energy, causing the supply to con-

tinually attempt to restart. This limits

input and output current during a

short circuit. Feedback voltage is fed

directly through a resistor divider to

the R

REF

pin. The sampling error am-

pliﬁer still works, but the load

compensation circuitry is bypassed.

This results in a ±5% load regulation

over line and load. A dedicated feed-

back winding referencing the feedback

voltage to the V

pin could be used to

include the load compensation func-

tion and improve regulation.

Conclusion

The LT1425 offers high performance

and accuracy without the additional

circuitry traditionally associated with

isolated DC to DC converters.

Linear Technology Magazine • February 1997

DESIGN FEATURES

220µF

10V

1425_07.eps

GND

N/C

LT1425

MBRS340T3

REF

SYNC

SGND

GND

OCMP

CMPC

PGND

GND

3.01k

25.5k

9.3k

MMFT1N10E

2.4k

12V

INPUT

COM

0.1µF

22µF

35V 220µF

10V 200Ω

OUT

COM

COILTRONIX

VP1-0190

TURNS RATIO 1 : 1 : 1 : 1 : 1 : 1

12µH PER WINDING

407-241-7876

1000pF

MUR120

Q1

2N3906

0.1µF

100Ω

1.8k

330pF

D1

1N755

7.5V

1425_08.eps

GND

N/C

BAV21

MUR120

LT1425

MBR745

T1*

REF

SYNC

SGND

GND

OCMP

CMPC

PIN 3 TO 4, 15 TURNS BIFILAR 31AWG

*PHILIPS EFD-15-3F3 CORE

GAP FOR PRIMARY

L = 100µH

PIN 7 TO 8, 6 TURNS QUADFILAR 29AWG

PIN 5 TO 6, 15 TURNS BIFILAR 33AWG

PIN 1 TO 2, 15 TURNS BIFILAR 31AWG

1 LAYER 2 MIL 

POLYESTER FILM

2 LAYERS 2 MIL 

POLYESTER FILM

PGND

GND

3.16k

Q2

2N3906 Q3

2N3904

Q1

IRF610

D1

7.5V

1N755

D2

7.5V

1N755

30.1k

R2

R1

24k 50Ω

510

10k

2.4k

100k

INPUT

COM

–36V TO

–72V

3.3µF

150pF

0.1µF

0.1µFC1

27µF

35V

150µF

6.3V

150µF

6.3V



OUT

COM

1000pF

470pF

Figure 7. 5V/1A step-down, isolated supply

Figure 8. 5V/2A telecommunications supply

Linear Technology Magazine • February 1997

DESIGN FEATURES

(5V)

1328_01.eps

C4

0.1µFC5

10µF

HIGH – SIR

LOW – FIR AND 4PPM

FILT

FILT LO

GND

LT1328

MODE

C3

1000pF

DATA

BIAS

TTL DATA OUT

C1

330pF

LIGHT IN

D1

BPU22NF

TEMIC

C2

10nF

1328_02.eps



10k



100Ω



DRIVER

Q3

2N7002

TRANSMIT

INPUT



6.8Ω

1/2W

Q4

2N7002

D2

HSDL-4220

The LT1328: a Low Cost IrDA Receiver

Solution for Data Rates up to 4Mbps

by Alexander Strong

Introduction

The need for ever increasing data

rates required by a vast array of de-

vices, such as notebook computers,

printers, mobile phones, pagers, elec-

tronic cameras and modems, has been

satisﬁed by the technology of infrared

data transmission. The Infrared Data

Association (IrDA

) standard, which

covers data rates from 2400bps to

4Mbps, is the overwhelming choice

for infrared data transmission. The

LT1328 is a photodiode receiver that

supports IrDA data rates up to 4Mbs,

as well as other modulation methods,

such as Sharp ASK and TV remote

control.

The LT1328, in the MSOP package,

contains all the necessary circuitry to

convert current pulses from an exter-

nal photodiode to a digital TTL output

while rejecting unwanted lower fre-

quency interference. The LT1328 plus

six external components is all that is

required to make the IrDA-compat-

ible receiver shown in Figure 1. An

IrDA- compatible transmitter can also

be implemented with only six compo-

nents, as shown in Figure 2. Power

requirements for the LT1328 are mini-

mal: a single 5V supply and 2mA of

quiescent current.

LT1328

Functional Description

Figure 3 is a block diagram of the

LT1328. Photodiode current from D1

is transformed into a voltage by feed-

back resistor R

. The DC level of the

preamp is held at V

BIAS

by the servo

action of the transconductance

ampliﬁer’s g

. The servo action only

suppresses frequencies below the

FILT

pole. This highpass ﬁlter-

ing attenuates interfering signals,

such as sunlight or incandescent or

ﬂuorescent lamps, and is selectable

at pin 7 for low or high data rates. For

high data rates, pin 7 should be held

low. The highpass ﬁlter breakpoint is

set by the capacitor C1 at f = 25/(2

× R

× C), where R

= 60k. The

330pF capacitor (C1) sets a 200kHz

corner frequency and is used for data

rates above 115kbps. For low data

rates (115kbps and below), the ca-

pacitance at pin 2 is increased by

taking pin 7 to a TTL high. This

switches C2 in parallel with C1,

lowering the highpass ﬁlter break-

point. A 10nF cap (C2) produces a

6.6kHz corner. Signals processed by

the preamp/g

ampliﬁer combina-

tion cause the comparator output to

swing low.

IrDA SIR

The LT1328 circuit in Figure 1 oper-

ates over the full 1cm to 1 meter range

of the IrDA standard at the stipulated

light levels. For IrDA data rates of

115kbs and below, a 1.6µs pulse width

is used for a zero and no pulse for a

one. Light levels are 40mW/sr (Watts

per steradian) to 500mW/sr. Figure 4

shows a scope photo for a transmitter

input (top trace) and the LT1328 out-

put (bottom trace). Note that the input

to the transmitter is inverted; that is,

transmitted light produces a high at

the input, which results in a zero at

the output of the transmitter. The

Mode pin (pin 7) should be high for

these data rates.

IrDA FIR

The second fastest tier of the IrDA

standard addresses 576kbps and

1.152Mbs data rates, with pulse

widths of 1/4 of the bit interval for

zero and no pulse for one. The

1.152Mbs rate, for example, uses a

pulse width of 217ns; the total bit

time is 870ns. Light levels are

100mW/sr to 500mW/sr over the 1cm

to 1 meter range. A photo of a trans-

mitted input and LT1328 output is

Figure 1. LT1328 IrDA receiver—typical application Figure 2. IrDA transmitter

Linear Technology Magazine • February 1997

DESIGN FEATURES

–

1328_03.eps

COMPARATOR

CELL

BIAS

FILTER

PHOTODIODE

FILTER LO

DATA OUT

MODE

GND

PREAMP

C1

330pF C2

10nF

shown in Figure 5. The LT1328 output

pulse width will be less than 800ns

wide over all of the above conditions

at 1.152Mbps. Pin 7 should be held

low for these data rates and above.

4ppm

The last IrDA encoding method is for

4Mbs and uses pulse position modu-

lation, thus its name: 4ppm. Two bits

are encoded by the location of a 125ns

wide pulse at one of the four positions

within a 500ns interval (2 bits ×

1/500ns = 4Mbps). Range and input

levels are the same as for 1.152Mbs.

Figure 6 shows the LT1328 reproduc-

tion of this modulation.

Conclusion

In summary, the LT1328 can be used

to build a low cost receiver compat-

ible with IrDA standards. Its ease of

use and ﬂexibility also allow it to

provide solutions to numerous other

photodiode receiver applications. The

tiny MSOP package saves on PC board

area.

Figure 3. LT1328 block diagram IrDA is a registered trademark of the Infrared Data

Association

Authors can be contacted

at (408) 432-1900

Figure 4. IrDA 115kbs modulation Figure 5. IrDA 1.152Mbs modulation Figure 6. IrDA 4ppm modulation

TRANSMITTER

INPUT

LT1328 OUTPUT

2ns/DIV

TRANSMITTER

INPUT

LT1328 OUTPUT

200ns/DIV

TRANSMITTER

INPUT

LT1328 OUTPUT

200ns/DIV

Linear Technology Magazine • February 1997

DESIGN FEATURES

The LTC1473 Dual PowerPath Switch

Driver Simpliﬁes Portable Power

Management Design by Jaime Tseng

Introduction

The LTC1473 is the latest addition to

Linear Technology’s new family of

power management controllers, which

simplify the design of circuitry for

switching between two batteries or a

battery and an AC adapter. Presently,

switching between power sources is

implemented with discrete compo-

nents—a mixture of regulators,

comparators, references, glue logic,

MOSFET switches and drivers. In-

variably, these solutions are expensive

and occupy a considerable amount of

printed circuit board space. Although

these circuits are frequently designed

in a hurry, the problems associated

with power path switching are often

subtle and daunting. For example,

switching from one battery to another

can produce huge inrush currents

between the batteries when their volt-

ages differ. In extreme cases, system

bypass capacitors can be destroyed if

tantalums are used. Slowing the

switch turn-on rate helps reduce the

inrush current, but may cause a pre-

cipitous drop in the system supply

voltage.

Solutions to these “real world” prob-

lems have been designed into our new

power path

switch driver. The

LTC1473 dual PowerPath™ switch

driver drives low loss N-channel MOS-

FET switches that direct power in the

main power path of a single or dual

rechargeable battery system, the type

found in most notebook computers

and other portable equipment.

Overview

The power management system in

Figure 1 shows the LTC1473 driving

two sets of back-to-back N-channel

MOSFET switches connecting the two

batteries to the system DC/DC regu-

lator. Each of the switches is controlled

by a TTL/CMOS compatible input

that interfaces directly with a power

management system microprocessor.

An internal boost regulator provides

the voltage to fully enhance the logic-

level N-channel MOSFET switches.

The LTC1473 uses a current sense

loop to limit current rushing in and

out of the batteries and the system

supply capacitor during switch-over

transitions or during a fault condi-

tion. A user programmable timer

monitors the time during which the

MOSFET switches are in current limit

and latches them off if the pro-

grammed time is exceeded. A unique

“2-diode logic mode” ensures system

start-up, regardless of which input

receives power ﬁrst.

BAT1

BAT2

SWA1

SWA2

HIGH

EFFICIENCY

DC/DC

SWITCHING

REGULATOR

3.3V

12V

DCIN

LTC1473

POWER

MANAGEMENT

µP

SWB2

SWB1 R

SENSE

STEP-UP

SWITCHING

REGULATOR

1µF

50V

GATE

DRIVER

GATE

DRIVER

1µF

50V

INRUSH

CURRENT

SENSING

AND LIMITING

4700pF

TIMER

1473_01.eps

IN1

IN2

DIODE

L1

1mH

Si9926

MB914LT1

MBRD340

Figure 1. Dual-battery PowerPath switch driver: V

regulator, inrush limiting and switch-gate drivers

Linear Technology Magazine • February 1997

DESIGN FEATURES

BAT1

BAT2

SWA1

SWA2

HIGH

EFFICIENCY

DC/DC

SWITCHING

REGULATOR

3.3V

1473_02.eps

12V

DCIN

LTC1473

POWER

MANAGEMENT

µP

SWB2

SWB1

ON OFF

SENSE

MBRD340

Si9926

Back-to-Back Switches

The back-to-back topology eliminates

the problems associated with the in-

herent body diodes in power MOSFET

switches and allows each switch pair

to block current ﬂow in either direc-

tion when the two switches are turned

off. The low loss, N-channel switch

pairs are housed in 8-pin SO and

SSOP packaging and are available

from a number of manufacturers. The

Si9926DY, for example, houses two

20V MOSFETs rated at 0.03 with

= 4.5V.

Inrush Current Limiting

The back-to-back topology also al-

lows for independent control of each

half of the switch pair, facilitating

bidirectional inrush current limiting.

The voltage across a single low value

resistor, R

SENSE

, is measured to de-

termine the instantaneous current

ﬂowing through the two main switch

pairs, SWA1/B1 and SWA2/B2. The

inrush current is then controlled by

the gate drivers until the transition

from one power source to the other

has been completed. The current ﬂow-

ing in and out of the two main power

sources and the DC/DC converter

input capacitor is dramatically

reduced.

Tantalum Capacitors

Tantalum capacitors, with their high

volumetric efﬁciency and low ESR,

are the dielectric of choice for low

impedance applications, such as ﬁl-

tering the input of a switching

regulator. However, because these ca-

pacitors are exposed to uncontrolled

energy supplies, they are subject to

failures caused by high inrush cur-

rents unless current surges are

restricted. The inrush-current limit-

ing feature of the LTC1473 makes it

feasible to use low proﬁle tantalum

surface mount capacitors in place of

bulkier electrolytic capacitors.

Built-In Step-Up Regulator

The gate drive for the two low loss N-

channel switches is supplied by a

micropower step-up regulator that

continuously generates 8.5V above

V+, up to 37V maximum. The V

supply provides sufﬁcient headroom

to ensure that the logic-level MOS-

FET switches are fully enhanced by

the gate drivers which, supply a regu-

lated 5.7V gate-to-source voltage, V

when turned on.

The power for the micropower boost

regulator is taken from external di-

odes connected to each power source.

The highest voltage potential is di-

rected to V

, where L1, an inexpensive

1mH surface mount inductor, is con-

nected. An internal diode directs the

current from L1 to the V

output

capacitor, C2.

Programmable Fault Timer

A fault-timer capacitor, C

TIMER

, is used

to program the time during which the

MOSFET switches are allowed to be

in current limit continuously. In the

event of a fault condition, the MOS-

FET current is limited by the inrush

current-limit loop. A MOSFET switch

operating in current limit is in a high

dissipation mode and can fail cata-

strophically if this condition is not

promptly terminated.

The fault-time delay is programmed

with an external capacitor connected

between the TIMER pin and ground.

At the instant the MOSFET switch

enters current limit, a 5µA current

source starts charging C

TIMER

through

the TIMER pin. When the voltage

across C

TIMER

reaches 1.2V, an inter-

nal latch is set and the MOSFET

switch is turned off. To reset the

latch, the gate-drive input of the

MOSFET switch is deselected.

The “2-Diode Mode”

Under normal operating conditions,

both halves of each switch pair are

turned on and off simultaneously.

For example, when the input power

source is switched from BAT1 to BAT2,

both gates of switch pair SWA1/B1

are turned off and both gates of switch

pair SWA2/B2 are turned on. The

back-to-back body diodes in switch

pair SWA1/B1 block current ﬂow into

or out of the BAT1 input connector.

In the “2-diode mode,” only the

ﬁrst half of each power path switch

pair, for example, SWA1 and SWA2, is

turned on, and the second half, that

is, SWB1 and SWB2, is turned off.

These two switch pairs now simply

act as two diodes connected to the

two main input power sources, as

illustrated in Figure 2. The power

path diode with the highest input

voltage passes current to the input of

the DC/DC converter to ensure that

the power management microproces-

sor is powered, even under start-up

Figure 2. LTC1473 dual PowerPath switch driver in “2-diode mode”

Linear Technology Magazine • February 1997

DESIGN FEATURES

or abnormal operating conditions.

After “good” power is reconnected to

one of the main inputs, the LTC1473

can be instructed to drive the appro-

priate switch pair on fully as the other

switch is turned off, restoring normal

operation.

Typical Application

A typical dual-battery system is shown

in Figure 3. The LTC1473 accepts

commands from a power manage-

ment microprocessor to select the

SAB1

GB1

SENSE+

SENSE-

GA2

SAB2

GB2

IN1

IN2

DIODE

TIMER

GND

LTC1473GA1

SENSE

0.04Ω

POWER

MANAGEMENT

µP

SUPPLY

MONITOR

BAT1

DCIN

BAT2

OUT

INPUT OF SYSTEM

HIGH EFFICIENCY DC/DC

SWITCHING REGULATOR

(LTC1435,ETC)

TIMER



4700PF 1µF

1µF

1mH

Si9926

1473_03.eps

MBRD340

MMBD2823LT1

MMBD914LT1

appropriate battery. The micropro-

cessor monitors the presence of

batteries and the AC adapter through

a supply monitor block, or, in the case

of some battery packs, through a

thermistor sensor. This block com-

prises a resistor divider and a

comparator for each supply. If the AC

adapter is present, the two switches

are turned off by the microprocessor

and the power is delivered to the

input of the system DC/DC switching

regulator via a Schottky diode.

Conclusion

The LTC1473 dual PowerPath switch

driver eases the design of the front

end of the power management sys-

tem. Designed to drive low cost

N-channel MOSFET switches and

packed with numerous protection fea-

tures in a narrow, 16-lead SSOP

package, the LTC1473 solves the

problems of cost, space and reliability

for power management system

designers.

Figure 3. Dual-battery power-management system

Authors can be contacted

at (408) 432-1900

Linear Technology Magazine • February 1997

DESIGN FEATURES

The LTC1560-1: a 1MHz/500kHz

Continuous-Time, Low Noise,

Elliptic Lowpass Filter by Nello Sevastopoulos

Introduction

The LTC1560-1 is a high frequency,

continuous-time, low noise ﬁlter. It is

a single-ended input, single-ended

output, 5th order elliptic lowpass ﬁl-

ter with a pin-selectable cutoff

frequency (f

) of 1MHz or 500kHz.

Several features distinguish the

LTC1560-1 from other commercially

available high frequency, continuous-

time monolithic ﬁlters:

❏Fifth order 1MHz or 500kHz

elliptic response in an SO-8

package

❏No external components or

clocks required

❏Better than 60dB stopband

attenuation

❏75dB signal-to-noise ratio (SNR)

❏0.3dB passband ripple

The LTC1560-1 delivers accurate

ﬁxed cutoff frequencies of 500kHz

and 1MHz without the need for inter-

nal or external clocks. Through a

simple mask change, other fre-

quencies from 450kHz to 1.3MHz can

be produced upon demand. The

LTC1560-1’s extremely small size

makes it suitable for compact designs

(see Figure1) and for a variety of ap-

plications, including communication

ﬁlters, antialiasing ﬁlters and smooth-

ing or reconstruction ﬁlters.

DC Performance and

Power Shutdown

The LTC1560-1 operates with ±5V

supplies and has a power shutdown

mode. The typical DC output swing of

the ﬁlter is from –3V to 3.5V. The

output DC offset of the ﬁlter is typi-

cally ±200mV. The operating power

supply range is ±4V to ±6V.

AC Performance

Frequency Response

The LTC1560-1 offers a pin-select-

able cutoff frequency of either 500kHz

(Figure 2; pin 5 tied to V

) or 1MHz

(pin 5 tied to V

–

). The detailed pass-

band frequency response of the 1MHz

ﬁlter is shown in Figure 3. In the

1MHz mode, the passband is ﬂat up

to 0.55 × f

with a typical ripple of

±0.2dB, increasing to ±0.3dB up to

0.9 × f

. The typical gain at f

–0.6dB. Referring to Figure 4, note

that the transition band has a gain of

–22dB at 1.44 × f

rolling off to –47dB

1

2

3

8

7

6

VOUT

SD

V+

0.5FC/FC

LTC1560-1

GND

VIN

GND

V–

1560_01.eps

1

2

3

8

7

6

VOUT

–5V, 1MHz

5V, 500kHz

0.1µF

(OR 5V)

0.01µF

0.01µF0.1µF

LTC1560-1

VIN

–5V

1560_02.eps

FREQUENCY (MHz)

–0.6

–0.1

–0.2

–0.3

–0.4

–0.5

0.6

0.5

0.4

0.3

0.2

0.1

GAIN (dB)

1.00.90.80.70.6

1560_03.eps

0.1 0.50.40.30.2

FREQUENCY (MHz)

–90

–40

–50

–60

–70

–80

–10

–20

–30

GAIN (dB)

10



1560_04.eps

0.1 1

= 500kHz

= 1MHz

Figure 1. LTC1560-1 in an SO-8 package Figure 2. 1MHz or 500kHz elliptic lowpass

ﬁlter with no external components Figure 4. Gain vs frequency of the 1MHz and

500kHz ﬁlters

Figure 3. Expanded passband ripple for the

1MHz ﬁlter

The LTC1560-1’s extremely

small size makes it well

suited for compact designs

and a variety of applications,

including communication

ﬁlters, antialiasing ﬁlters

and smoothing or

reconstruction ﬁlters.

The LTC1560-1 provides a power

shutdown option that signiﬁcantly

reduces current consumption when

the device is not being used. The ﬁlter

operation could be controlled by a

TTL input together with an inverter

applied to the SD pin (pin 7). A logic

high input turns the device on for

normal operation, whereas a logic low

puts the ﬁlter into its sleep mode, in

which it dissipates only 5mW of power.

(Leaving pin 7 open yields the default

mode of normal operation.)

Linear Technology Magazine • February 1997

DESIGN FEATURES

at 2 × f

. The stopband attenuation is

63dB starting from 2.43 × f

and

remains at least 60dB for input

frequencies up to 10MHz. When pro-

grammed for f

= 500kHz, the

frequency response remains the same,

with the exception of the gain at f

which is typically –1.3dB. Figure 4

compares the gain responses of the

1MHz and 500kHz ﬁlters.

Noise and

Distortion Performance

The LTC1560-1 architecture offers

not only low wide band noise but also

low total harmonic distortion (THD).

The combination of low noise and low

distortion means a wide dynamic

range. With a 1V

RMS

input signal, the

signal-to-noise ratio (SNR) is 69dB

and the THD + Noise is –63dB (0.07%).

The maximum SNR of 75dB is

achieved with a 2.1V

RMS

input signal.

This results in –46dB (0.5%) THD.

For the 500kHz device, the noise per-

formance is even better, with 77dB

SNR at a 1V

RMS

input.

–

1

2

3

8

7

6

OUT

LT1360

15V

–15V

–5V, 1MHz

5V, 500kHz

0.1µF

(OR 5V)

0.01µF

0.1µF

LTC1560-1

–5V

1560_05.eps

–

–15V

15V

0.1µF

1

2

3

8

7

6

VOUT

LT1360 8

0.1µF

(OR 5V)

0.01µF

8.1k

0.1µF

LTC1560-1

VIN

300pF

–5V

1560_06.eps

300pF

To achieve the full high frequency

performance from the ﬁlter, a small

resistor (about 200 ) should be added

at the output of the device to isolate

any capacitive load greater than 20pF.

Figure 5 shows a typical application

circuit to be used for any AC

performance measurements of the

LTC1560-1. Any high speed, high

slew-rate operational ampliﬁer such

as the LT1360 can serve as the buffer.

To correctly evaluate the high fre-

quency distortion performance of the

LTC1560-1 requires a very low dis-

tortion input signal, either from a

very high quality signal generator or,

if such a source is not available, from

a source that has been ﬁltered to

control its harmonic content.

Applications and

Experimental Results

The LTC1560-1 can be used not only

as a single device (as shown in Fig-

ures 2 and 5) but also as part of a

more complete frequency-shaping

system. Two representative examples

follow.

Highpass-Lowpass Filter

As a typical application in communi-

cation systems, where there is a need

to reject DC and some low frequency

signals, a 2nd order RC highpass

network can be inserted in front of the

LTC1560-1 to obtain a highpass-low-

pass response. Figures 6 and 7 depict

the network and its measured fre-

quency response, respectively. Notice

that the second resistor in the high-

pass ﬁlter is the input resistance of

the LTC1560-1, which is about 8.1k.

Delay-Equalized Elliptic Filter

Although elliptic ﬁlters offer high Q

and a sharp transition band, they

lack a constant group delay in the

passband, which implies more ring-

ing in the time-domain step response.

In order to minimize the delay ripple

in the passband of the LTC1560-1, an

allpass ﬁlter (delay equalizer) is cas-

caded with the LTC1560-1, as shown

in Figure 8. Figures 9 and 10 illus-

trate the eye diagrams before and

after the equalization, respectively.

An eye diagram is a qualitative

representation of the time-domain

response of a digital communication

system. It shows how susceptible the

system is to intersymbol interference

(ISI). Intersymbol interference is

caused by erroneous decisions in the

receiver due to pulse overlapping and

decaying oscillations of a previous

symbol. A pseudorandom 2-level se-

quence has been used as the input of

the LTC1560-1 to generate these eye

diagrams. The larger eye opening in

Figure 5. A typical circuit for evaluating the full performance of the LTC1560-1

Figure 6. A highpass-lowpass ﬁlter

Figure 7. Measured frequency response of

Figure 6’s circuit

Linear Technology Magazine • February 1997

DESIGN FEATURES

–

1

2

3

8

7

6

6.49k

6.65k

49.9Ω

22pF 0.1µF

0.1µF

22pF

9.75k

20k

40.2k

OUT

1/2 LT1364 16

1/2 LT1364

–15V

0.1µF

(OR 5V)

0.01µF

0.01µF0.1µF

LTC1560-1

–5V

1560_08.eps

Figure 10, an indication of the equal-

ization effect, leads to reduced ISI.

Note that in Figure 8, the equalizer

section has a gain of 2 for driving and

back-terminating 50 cable and load.

For a simple unterminated gain-of-1

equalizer, the 40.2k resistor changes

to 20k and the 49.9 r esistor is re-

Figure 8. Augmenting the LTC1560-1 for improved delay ﬂatness

Figure 9. 2-level eye diagram of the LTC1560-1 before equalization Figure 10. 2-level eye diagram of the equalized ﬁlter



for

the latest information

on LTC products, 

visit

www.linear-tech.com

Mojitaba Atarodi contributed significant portions

of this article.

moved from the circuit. The 22pF

capacitors are 1% or 2% dipped silver

mica or COG ceramic.

Conclusions

The LTC1560-1 is a 5th order elliptic

lowpass ﬁlter that features a 10-bit

gain linearity at signal ranges up to

1MHz. Being small and user friendly,

the LTC1560-1 is suitable for any

compact design. It is a monolithic

replacement for larger, more expen-

sive and less accurate solutions in

communications, data acquisitions,

medical instrumentation and other

applications.

Linear Technology Magazine • February 1997

DESIGN FEATURES

The LTC1594 and LTC1598:

Micropower 4- and 8-Channel

12-Bit ADCs by Kevin R. Hoskins and Marco Pan

Introduction

Data acquisition applications that

require low power dissipation fall into

two general areas: products that re-

quire highly efﬁcient power use, such

as battery-powered portable test

equipment and remotely located data

logging equipment, and products that

either operate in high temperature

environments or must not contribute

to increasing ambient temperature.

To help meet these requirements, Lin-

ear Technology has introduced the

LTC1594 and LTC1598.

Micropower ADCs

in Small Packages

The LTC1594 and LTC1598 are

micropower 12-bit ADCs that feature

a 4- and 8-channel multiplexer, re-

spectively. The LTC1594 is available

in a 16-pin SO package and the

LTC1598 is available in a 24-pin SSOP

package. Each ADC includes a simple,

efﬁcient serial interface that reduces

interconnects and, thereby, possible

sources of corrupting digital noise.

Reduced interconnections also reduce

board size and allow the use of pro-

cessors having fewer I/O pins, both of

which help reduce system costs. Small

packages also shorten the distance

between the ADC and its supply and

voltage reference bypass components.

This reduces lead inductance and

allows bypass components to operate

as efﬁciently as possible.

Conserve Power with

Auto Shutdown Operation

The LTC1594 and LTC1598 include

an auto shutdown feature that re-

duces power dissipation when the

converter is inactive (whenever the

CS signal is a logic high). Nominal

power dissipation while either con-

verter is clocked at 320kHz is typically

1.6mW. The curve in Figure 1 indi-

cates the amount of current drawn by

this MUXed 12-bit ADC family for

sample rates up to 16.8ksps. As an

example, when converting at 4ksps,

the dissipation is just 450µW and

270µW for the 5V and 3V parts,

respectively.

Supply Flexibility:

2.7V or 5V

To increase applications ﬂexibility,

the LTC1594 and LTC1598 are also

available as 3V parts (LTC1594L and

LTC1598L), which are tested for 2.7V

operation. The LTC1594L and

LTC1598L typically draw 160µA at

maximum conversion rate, one-half

of the supply current drawn by the 5V

parts. Nominal power dissipation

while either converter is clocked at

200kHz (10.5ksps) is typically 800µW.

SAMPLE FREQUENCY (kHz)

0.1

SUPPLY CURRENT (µA)

100

1000

1 10 100

1598_01.eps

= 25°C

= 5V

REF

= 5V

CLK

= 320kHz

1

2

3

4

5

6

7

8

9

10

11

24

23

22

21

20

19

18

17

16

15

14

13



CH5

CH6

CH7

GND

CLK

CS MUX



COM

GND

CS ADC

OUT



CH4

CH3

CH2

CH1

CH0



MUXOUT

ADCIN

REF



CLK

LTC1598

ANALOG INPUTS

0V TO 5V

RANGE

R4, 7.5k R2, 7.5k

1µF

1598_02.eps



DATA IN

CHIP SELECT

CLOCK

DATA OUT

–

1µF

C6

0.015µF

C4

0.03µF

C2

0.1µF

R1, 7.5k R3, 7.5k

C3

0.03µF

C5

0.015µF

–

C1

0.1µF

1/2

LT1368

1/2

LT1368

Figure 1. Supply current vs sample rate

Figure 2. A simple data acquisition system takes advantage of the LTC1598’s MUXOUT/ADCIN pins to ﬁlter analog signals prior to ADC conversion.

Linear Technology Magazine • February 1997

DESIGN FEATURES

Good DC Performance

The DC specs include excellent dif-

ferential nonlinearity (DNL) of

±3/4LSB, as required by pen-screen

and other monitoring applications.

No missing codes are guaranteed over

temperature.

Versatile, Flexible Serial I/O

The serial interface found on the

LTC1594 and LTC1598 is designed

for ease of use, ﬂexibility, minimal

interconnections and I/O compatibil-

ity with QSPI, SPI, MICROWIRE™

and other serial interfaces. The MUX

and the ADC have separate chip se-

lect (CS) and serial clock inputs, which

adds versatility. The remaining serial

interface signals are data input (D

)

and data output (D

OUT

). The maxi-

mum serial clock frequencies are

320kHz and 200kHz for the 5V and

3V parts, respectively.

Latch-up Proof MUX Inputs

The LTC1594’s and LTC1598’s input

MUXes are designed to handle input

voltages that exceed the nominal in-

put range, GND to the supply voltage,

without latch-up. Although an over-

driven, unselected channel may

corrupt a selected, correctly driven

channel, no latch-up occurs and cor-

rect conversion results resume when

the offending input voltage is removed.

The MUX inputs remain latch-up proof

for input currents up to ±200mA over

temperature.

Individual ADC

and MUX Chip Selects

Enhance Flexibility

The LTC1594 and LTC1598 feature

separate chip selects for ADC and

MUX. This allows the user to select a

particular channel once for multiple

conversions. This has the following

beneﬁts: ﬁrst, it eliminates the over-

head of sending D

word for the same

channel each time for each conver-

sion; second, it avoids possible

glitches that may occur if a slow-

settling antialiasing ﬁlter is used; and

third, it sets the gain once for mul-

tiple conversions if the MUXOUT/

ADCIN loop is used to create a pro-

grammable gain ampliﬁer (PGA).

MUXOUT/ADCIN

Loop Economizes

Signal Conditioning

The MUXOUT and ADCIN pins form a

very ﬂexible external loop that allows

PGA and/or processing analog input

signals prior to conversion. This loop

is also a cost effective way to perform

the conditioning, because only one

circuit is needed instead of one for

each channel. Figure 2 shows the

loop being used to antialias ﬁlter sev-

eral analog inputs. The output signal

of the selected MUX channel, present

on the MUXOUT pin, is applied to R1

of the Sallen-Key ﬁlter. The ﬁlter band

limits the analog signal and its out-

put is applied to ADCIN. The LT1368

rail-to-rail op amps used in the ﬁlter

will, when lightly loaded as in this

application, swing to within 8mV of

the positive supply voltage. Since only

one circuit is used for all channels,

each channel sees the same ﬁlter

characteristics.

1598_03.eps

CH0

CH1

CH2

CH3

CH4

CH5

CH6

CH7

20

21

22

23

24

1

2

64R

32R

16R

8R

4R

2R

R

–

8 COM

18 MUXOUT

GND

4, 9

10

6

5, 14

11

CS ADC

CS MUX

CLK

DOUT

DIN

12

NC

12-BIT

SAMPLING

ADC

8-CHANNEL

MUX

1

2

3

4

5

6

7

16

15

14

13

12

11

10

LTC1391

1µF

ADCIN

17 16 15, 19 1µF

0.1µF

1µF

VREF VCC

–

CH0

CH1

CH2

CH3

CH4

CH5

CH6

CH7

V+

D

V–

DOUT

DIN

CS

CLK

GND

1/2 LT1368

LTC1598

µP/µC

Figure 3. Using the MUXOUT/ADCIN loop of the LTC1598 to form a PGA with eight gains in a noninverting conﬁguration

MICROWIRE is a trademark of National Semiconductor

Corp.

Linear Technology Magazine • February 1997

DESIGN FEATURES

Using MUXOUT/

ADCIN Loop as PGA

Figure 3 shows the LTC1598’s

MUXOUT/ADCIN loop and an LT1368

being used to create a single-channel

PGA with eight noninverting gains.

Combined with the LTC1391, as

shown in Figure 3, the system can

expand to eight channels and eight

gains for each channel. Using the

LTC1594, the PGA is reduced to four

gains. The output of the LT1368 drives

the ADCIN and the resistor ladder.

The resistors above the selected MUX

channel form the feedback for the

LT1368. The loop gain for this ampli-

ﬁer is (R

1/R

2) + 1. R

1 is the

summation of the resistors above the

selected MUX channel and R

2 is the

summation of the resistors below the

selected MUX channel. If CH0 is se-

lected, the loop gain is 1 since R

1 is

0. Table 1 shows the gain for each

MUX channel. The LT1368 dual rail-

to-rail op amp is designed to operate

with 0.1µF load capacitors. These

capacitors provide frequency compen-

sation for the ampliﬁers, help reduce

the ampliﬁers’ output impedance and

improve supply rejection at high fre-

quencies. Because the LT1368’s I

low, the R

of the selected channel

will not affect the loop gain given by

the formula above. In the case of the

inverting conﬁguration of Figure 4,

the selected channel’s R

will be

added to the resistor that sets the

loop gain.

8-Channel, Differential,

12-Bit A/D System Using

the LTC1391 and LTC1598

The LTC1598 can be combined with

the LTC1391 8-channel, serial-inter-

face analog multiplexer to create a

differential A/D system. Figure 5

shows the complete 8-channel, dif-

ferential A/D circuit. The system uses

the LTC1598’s MUX as the nonin-

verting input multiplexer and the

LTC1391 as inverting input multi-

plexer. The LTC1598’s MUXOUT

drives the ADCIN directly. The

inverting multiplexer’s output is ap-

plied to the LTC1598’s COM input.

The LTC1598 and LTC1391 share the

CS, D

, and CLK control signals.

1598_04.eps

CH0

CH1

CH2

CH3

CH4

CH5

CH6

CH7

20

21

22

23

24

1

2

128R

64R

32R

16R

8R

4R

2R

–

8 COM GND

4, 9

10

6

5, 14

7

CS ADC

CS MUX

CLK



OUT

12

NC

12-BIT

SAMPLING

ADC

ADCINMUXOUT

18 17 16 15, 19 1µF

0.1µF

REF

–

1/2 LT1368

128R

LTC1598

1598_05.eps

CH0

CH1

CH2

CH3

CH4

CH5

CH6

CH7

20

21

22

23

24

1

2

–

8-CHANNEL

MUX

8 COM GND

4, 9

10

6

5, 14

7

CS ADC

CS MUX

CLK



OUT

12

NC

12-BIT

SAMPLING

ADC

ADCINMUXOUT

18 17 16 15, 19 1µF

REF

1

2

3

4

5

6

7

16

15

14

13

12

11

10

LTC1391

CH0

CH7

CLK

OUT

CH0

CH1

CH2

CH3

CH4

CH5

CH6

CH7

V+

D

V–

OUT



CS

CLK

GND

LTC1598

Figure 4. Using the MUXOUT/ADCIN loop of the LTC1598 to form a PGA with

eight inverting gains

Figure 5. Using the LTC1598 and LTC1391 as an 8-channel, differential 12-bit ADC system:

opening the indicated connection and shorting the dashed connection daisy-chains the

external and internal MUXes, increasing channel-selection flexibility.

continued on page 20

XUM

lennahC

gnitrevninoN

niaG

gnitrevnI

niaG

011–

122–

244–

388–

46161–

52323–

64646–

7821821–

Table 1. PGA gain for each MUX channel of

Figures 3 and 4

Linear Technology Magazine • February 1997

DESIGN FEATURES

–

VFB

5Ω

RSENSE

(OPTIONAL)

LBI

1µA

RUN

LBOUT

1SHOT

VFB

1.23V

REFERENCE

LBI

GND

READY

1.23V

1474_01.eps

5µS

100mV VIN

VIN

VOUT

SENSE

1×25×

CONNECTION NOT PRESENT IN LTC1474

CONNECTION PRESENT IN LTC1474 ONLY

OUT

LTC1474 and LTC1475 High Efﬁciency

Switching Regulators Draw Only

10µA Supply Current by Greg Dittmer

Introduction

Maximizing battery life, one of the key

design requirements for all battery-

powered products, is now easier with

Linear Technology’s new family of

ultralow quiescent current, high

efﬁciency step-down regulator ICs,

the LTC1474 and LTC1475. The

LTC1474/LTC1475 are step-down

regulators with on-chip P-channel

MOSFET power switches. These regu-

lators draw only 10µA supply current

at no load while maintaining the out-

put voltage. With the on-chip switch

(1.3 at V

= 10V), only four external

components are necessary to make a

complete, high efﬁciency (up to 92%)

step-down regulator. Low component

count and the LTC1474/LTC1475’s

tiny MSOP packages provide a mini-

mum-area solution to meet the limited

space requirements of portable appli-

cations. Wide supply voltage range

(3V–18V) and 100% duty cycle capa-

bility for low dropout allow maximum

energy to be extracted from the bat-

tery, making the LTC1474/LTC1475

ideal for moderate current (up to

300mA) battery-powered applications.

The peak inductor current is pro-

grammable via an optional current

sense resistor to allow the design to

be optimized for a particular applica-

tion and to provide short-circuit pro-

tection and excellent start-up

behavior. Other features include Burst

Mode™ operation to maintain high

efﬁciency over almost four decades of

load current, an on-chip low-battery

comparator and a shutdown mode to

further reduce supply current to 6µA.

The LTC1475 provides on/off control

with push-button switches for use in

handheld products.

The LTC1474/LTC1475 are avail-

able in adjustable output voltage

versions, in 8-pin MSOP and SO

packages.

Figure 1. LTC1474/LTC1475 functional block diagram

Linear Technology Magazine • February 1997

DESIGN FEATURES

1474_02.eps

MODE

RUN

ONE-SHOT

1.23V REFERENCE

INDUCTOR

CURRENT

CURRENT

COMPARATOR

VOLTAGE

COMPARATOR

LOW-BATTERY

COMPARATOR

SHUTDOWNSLEEP5µS OFF-TIMEON

OFFOFFOFFON (100µA)

OFFOFFON (10µA)ON (5µA)

OFFON (5µA)

ON (4µA)

ON (1µA)

High Performance on a

Microampere Budget

The functional block diagram, shown

in Figure 1, provides a study in power

management. LTC1474/LTC1475

control the output voltage by charg-

ing the output capacitor in short burst

cycles using Burst Mode operation.

The peak current in each burst cycle

(up to 400mA) is set by the external

sense resistor. As the load increases,

the frequency of the burst cycles in-

creases (up to a maximum of 170kHz)

to maintain the charge in the output

capacitor. The burst cycle begins when

the output voltage falls below the

lower threshold of the voltage com-

parator (V). The P-channel power

switch turns on to ramp the inductor

current up until either the current

comparator (C) trips on peak current

or the voltage comparator trips on the

upper voltage threshold. At this time

the one-shot is triggered and begins

the 5µs off-time period, during which

the switch is turned off and inductor

current ramps down. If, at the end of

the off-time, the output voltage is

below the upper voltage comparator

threshold, the switch is turned on

again to begin another cycle. If the

upper voltage threshold is exceeded,

however, the switch remains off and

the output capacitor supplies the load

current. The switch remains off until

the load current discharges the out-

put capacitor below the lower voltage

threshold.

The ultralow supply current and

very high efﬁciency at light loads are

achieved by powering only those func-

tions that are necessary at any given

time. Figure 2 is a summary of the

current used by each of the functions

in each of the different operating

modes. During sleep mode, when the

output capacitor is supplying the load,

only the 1.23V reference, the voltage

comparator and low-battery compara-

tor are on; together they draw only

10µA of supply current to perform

their functions. These three functions

are on at all times except during

shutdown. During shutdown, the volt-

age comparator is turned off to save

an additional 5µA. The current com-

parator, which, as a result of its speed

requirement necessarily draws more

current, is only turned on during the

switch on-time, when it is needed to

monitor the switch current. When the

current or voltage comparator trips,

the current comparator is turned off

and the one-shot timer is triggered,

drawing 10µA during its 5µs time-

out period. When the one-shot times

out it turns off, reducing the supply

current to the 10µA needed for the

voltage comparator, reference and low-

battery comparator, until a new burst

cycle begins.

In Control of

Inductor Current

Excessive peak inductor current can

be a liability. Lower peak current

offers the advantages of smaller volt-

age ripple (∆V = I

PEAK

× ESR), lower

noise and less stress on alkaline bat-

teries and other circuit components.

Also, lower peaks allow the use of

inductors with smaller physical size.

The LTC1474/LTC1475 provide ﬂex-

ibility by allowing the peak switch/

inductor current to be programmed

with an optional sense resistor to

provide just enough to meet the load

requirement. Without a sense resis-

tor (that is, with pins 6 and 7 shorted)

the current limit defaults to its maxi-

mum of 400mA. Using the default

current limit eliminates the need for a

sense resistor and associated de-

coupling capacitor.

A sense MOSFET (a portion of the

main power MOSFET) is used to di-

vert a sample (about 5%) of the switch

current through the internal 5 sense

resistor. The internal current com-

parator monitors the voltage drop

across the series combination of the

internal and external sense resistors

and trips when this voltage drop ex-

ceeds 100mV. This results in a peak

current of I

PEAK

= 0.1/(0.25 + R

SENSE

)

+ 0.2 × (V

– V

OUT

)/L. The second

term in the above equation is the

result of overshoot of the peak cur-

rent due to delays in the current

comparator and must be taken into

account at lower inductances and

higher supply voltages to guarantee

that maximum current ratings of the

inductor and switch are not exceeded.

Note that worst case will occur during

a short circuit, when V

OUT

= 0.

3.3V/200mA

Step-Down Regulator

A typical application circuit using the

LTC1474 is shown in Figure 3. This

circuit supplies a 200mA load at 3.3V

with an input supply range of 4V–18V

(3.3V at no load). The 0.1 sense

resistor reduces the peak current to

about 285mA, which is the minimum

level necessary to meet the 200mA

load current requirement with a

Figure 2. Supply current breakdown during each operational mode

Linear Technology Magazine • February 1997

DESIGN FEATURES

100µH inductor. The peak can be

reduced further if a higher value in-

ductor is used. Since the output

capacitor dominates the output volt-

age ripple, an AVX TPS series low ESR

(150m ) output capacitor is used to

provide a good compromise between

size and low ESR. With this capacitor

the output ripple is less than 50mV.

Efﬁciency Considerations

The efﬁciency curves for the 3.3V/

200mA regulator at various supply

voltages are shown in Figure 4. Note

the ﬂatness of the curves over the

upper three decades of load current

and that the efﬁciency remains high

down to extremely light loads. Efﬁ-

ciency at light loads depends on low

quiescent current. The curves are ﬂat

because all signiﬁcant sources of loss

except for the 10µA standby cur-

rent—I

R losses in the switch, catch

diode losses, gate charge losses to

turn on the switch and burst cycle DC

supply current losses—are identical

during each burst cycle. The only

variable is the rate at which the burst

cycles occur. Since burst frequency is

proportional to load, the loss as a

percentage of load remains relatively

constant. The efﬁciency drops off as

the load decreases below about 1mA

because the non-load-dependent

10µA standby current loss then con-

stitutes a more signiﬁcant percentage

of the output power. This loss is pro-

portional to V

and thus its effect is

more pronounced at higher V

Care must be used in selecting the

catch diode to maximize both low and

high current efﬁciency. Low reverse

leakage current is critical for maxi-

mizing low current efﬁciency because

the leakage can potentially approach

the magnitude of the LTC1474/

LTC1475 supply current. Low for-

ward drop is critical for high current

efﬁciency because loss is proportional

to forward drop. These are conﬂicting

parameters, but the MBR0530 0.5A

Schottky diode used in the Figure 3 is

a good compromise. Lower induc-

tances also help by minimizing DCR

without increasing the inductor size.

However, lower inductances also re-

duce the maximum available output

power for a given I

PEAK

due to the ﬁxed

5µs off-time and may also increase

the peak current overshoot due to

high di/dt (see formula for I

PEAK

LTC1475 Push-Button

On/Off Operation

The LTC1475 provides the option of

push-button control of run and shut-

down modes for handheld products.

In contrast to the LTC1474’s run/

shutdown mode, which is controlled

by a voltage level at the RUN pin

(ground = shutdown, open/high =

run), the LTC1475 run/shutdown

mode is controlled by an internal S/R

ﬂip-ﬂop that is set (run mode) by a

momentary ground at the RUN pin

and reset (shutdown mode) by a mo-

mentary ground at the LBI pin (see

Figure 5). This provides simple on/off

control with two push-button

switches. The simplest implementa-

tion of this function is shown in Figure

6, with normally open push-button

switches connected to the RUN and

LBI pins. Note that because the switch

on LBI is normally open, it doesn’t



10pF

1.69M

100k

1474_03.eps

RUN

SENSE

LBO

0.1µF10µF

25V

1000pF

LBI

LTC1474

LBI LBO

MBR0530

GND

OUT



3.3V/200mA V



4V-18V

RUN

D:

L:

OUT

:

MBR0530

SUMIDA CDRH74

TPSC107006R0150

THC50EIE106Z

0.1Ω

100µF

6.3V

L

100µH

LOAD CURRENT (mA)

100

EFFICIENCY (%)

200

1474_04.eps

0.02 0.2 2 20

 L= 100 µH

V

OUT

= 3.3V

R

SENSE

= 0.1 Ω

= 5V V

= 10V

= 15V

LBI

RUN

MODE RUN SHUTDOWN RUN

RUN

LTC1474

LTC1475

RUN OVERRIDES SHUTDOWN 

WHILE RUN IS LOW

100k

10µF

1474_05.eps

SENSE

OFF

OUT

LBO

LTC1475

BATT

GND

ON RUN

LBI/SD

1M 2.2M

100µH



100µF

Figure 3. LTC1474 3.3V/200mA step-down regulator Figure 4. Efﬁciency vs load for Figure 3’s

circuit

Figure 5. Comparison of RUN/SHUTDOWN

operation for the LTC1474 and LTC1475 Figure 6. LTC1475 step-down regulator with push-button on/off control

Linear Technology Magazine • February 1997

DESIGN FEATURES

affect the normal operation of this

input to the low-battery comparator.

With a resistor divider network con-

nected to the LBI to monitor the input

supply voltage level, the voltage at

this pin will normally be above the

low-battery trip threshold of 1.23V.

When this pin is pulled below 0.7V by

depressing the switch, the internal

ﬂip-ﬂop is reset to invoke shutdown.

Figure 7 shows an example of push-

button on/off control of a LTC1475

microcontroller application with a

single push button. The push button

is connected to the microcontroller as

a discrete input so that the

microcontroller can monitor the state

of the push button. The LTC1475 LBI

pin is connected to one of the

microcontroller’s open-drain discrete

outputs so that it can force the

LTC1475 off when it detects a

depressed push button. Because the

LTC1475 supplies power to

the microcontroller, once the micro-

controller is off, it can no longer turn

the LTC1475 back on. However, since

the push button is also connected

directly to the RUN pin, the LTC1475

can be turned back on directly from

the push button without the micro-

controller. The LTC1475 then powers

up the microcontroller. The discrete

inputs of most microcontrollers have

a reverse biased diode between the

input and supply; thus a blocking

diode with less than 1µA leakage is

necessary to prevent the powered

down microcontroller from pulling

down on the RUN pin.

Conclusion

The LTC1474 and LTC1475 ultralow

quiescent current step-down regula-

tor ICs provide a perfect solution for

low to moderate current (up to 300mA)

battery-powered applications where

high efﬁciency and maximizing bat-

tery life are critical. The 10µA no-load

supply current requirement ensures

that little battery energy is wasted on

the regulator. The internal P-channel

power switch, MSOP package and the

need for as few as four additional

components result in a very compact

solution, and the current program-

mability and wide supply-voltage

range provide the ﬂexibility neces-

sary to optimize the design for a variety

of applications.

100k

100µF

100µH

10µF

1M 2.2M

1474_06.eps

SENSE

OUT

LBO

LTC1475

BATT

GND

ON/OFF

MMBD914LT1

0.1µF

µC

RUN

LBI/SD

Figure 7. A single push-button controls on/off for the LTC1475 regulator and microcontroller.

This arrangement simultaneously

selects the same channel on each

multiplexer and maximizes the

system’s throughput. The dotted-line

connection daisy-chains the MUXes

of the LTC1391 and LTC1598 to-

gether. This conﬁguration provides

the ﬂexibility to select any channel

in the noninverting input MUX with

respect to any channel in the invert-

ing input MUX. This allows any

combination of signals applied to the

inverting and noninverting MUX in-

puts to be routed to the ADC for

conversion.

Conclusion

With their serial interfaces, small

packages, and auto shutdown, the

LTC1594(L) and LTC1598(L) achieve

very low power consumption while

occupying very little circuit board

area. Their outstanding DC speciﬁca-

tions make them the choice for

applications that beneﬁt from low

power, battery conserving operation,

multichannel inputs and space and

component saving signal condition-

ing loop.

LTC1594/LTC1598, continued from page 16

Linear Technology Magazine • February 1997

DESIGN IDEAS

–

DI1440_01.EPS

CMOS

ONE-SHOT

(CD4047)

LTC1440

2X9.1MΩ

27kΩ

λ/4

λ/2.5

D1

1N5711

DETECTOR

= 50Ω

D2

1N5711

100pF

REFERENCE

100pF

Biased Detector Yields High Sensitivity

with Ultralow Power Consumption

RF ID tags, circuits that detect a

“wake-up” call and return a burst of

data, must operate on very low quies-

cent current for weeks or months, yet

have enough battery power in reserve

to answer an incoming call. For small-

est size, most operate in the ultrahigh

frequency range, where the design of

a micropower receiver circuit is prob-

lematic. Familiar techniques, such as

direct conversion, super regeneration

or superheterodyne, consume far too

much supply current for long battery

life. A better method involves a

technique borrowed from simple ﬁeld-

strength meters: a tuned circuit and

a diode detector.

Figure 1 shows the complete cir-

cuit, which was tested at 470MHz.

This circuit contains a couple of im-

provements over the standard

L/C-with-whip ﬁeld-strength meter.

Tuned circuits aren’t easily con-

structed or controlled at UHF, so a

transmission line is used to match

the detector diode (1N5711) to a 6"

whip antenna. The 0.4-wavelength

section presents an efﬁcient, low im-

pedance match to the base of the

quarter-wave whip, but transforms

the received energy to a relatively

high voltage at the diode for good

sensitivity.

Biasing the detector diode improves

the sensitivity by an additional 10dB.

The forward threshold is reduced to

essentially zero, so a very small volt-

age can generate a meaningful output

change. The detector diode’s bias point

is monitored by an LTC1440 ultralow

power comparator, and by a second

diode, which serves as a reference.

When a signal at the resonant fre-

quency of the antenna is received,

Schottky diode D1 rectiﬁes the

incoming carrier and creates a nega-

tive-going DC bias shift at the

noninverting input of the compara-

tor. Note that the bias shift is sensed

at the base of the antenna where the

impedance is low, rather than at the

Schottky where the impedance is high.

This introduces less disturbance into

the tuned antenna and transmission-

line system. The falling edge of the

comparator triggers a one-shot, which

temporarily enables answer-back and

other pulsed functions.

Total current consumption is

approximately 5µA. Monolithic one-

shots draw signiﬁcant load current,

but the venerable ‘4047 is about the

best in this respect. Alternatively, a

discrete one-shot constructed from a

quad NAND gate draws negligible

power.

Sensitivity is excellent. The ﬁn-

ished circuit can detect 200mW

radiated from a reference dipole at

100'. Range, of course, depends on

operating frequency, antenna orien-

tation and surrounding obstacles; in

the clear, a more reasonable distance,

such as 10', can be covered at 470MHz

with only a few milliwatts.

All selectivity is provided by the

antenna itself. Add a quarter-wave

stub (shorted with a capacitor) to the

base of the antenna for better selec-

tivity and improved rejection of low

frequency signals.

Figure 1. Micropower ﬁeld detector for use at 470MHz

by Mitchell Lee

DESIGN IDEAS

Biased Detector Yields High

Sensitivity with Ultralow Power

Consumption .......................... 21

Mitchell Lee

LT1256 Voltage-Controlled

Amplitude Limiter .................. 22

Frank Cox

New IC Features Reduce EMI from

Switching Regulator Circuits

................................................23

John Seago

Dual Output Voltage Regulator

................................................26

Peter Guan

Free Digital Panel Meters from the

Oppressive Yoke of Batteries

................................................27

Mitchell Lee

Battery Charger IC Can Also Serve

as Main Step-Down Converter

................................................28

Arie Ravid

Combine a Switching Regulator

and an UltraFast™ Linear

Regulator for a High Performance

3.3V Supply

................................................29

Craig Varga

What Efﬁciency Curves

Don’t Tell................................. 30

San-Hwa Chee

Linear Technology Magazine • February 1997

DESIGN IDEAS

LT1256 Voltage-Controlled

Amplitude Limiter by Frank Cox

Amplitude-limiting circuits are use-

ful where a signal should not exceed

a predetermined maximum ampli-

tude, such as when feeding an A/D or

a modulator. A clipper, which com-

pletely removes the signal above a

certain level, is useful for many appli-

cations, but there are times when it is

not desirable to lose information. For

instance, when video signals have

amplitude peaks that exceed the

dynamic range of following processing

stages, simply clipping the peaks at

the maximum level will result in the

loss of all detail in the areas where

clipping takes place. Often these well

illuminated areas are the primary

subject of the scene. Because these

peaks usually correspond to the

highest level of luminosity, they are

referred to as “highlights.” One way to

preserve some of the detail in the

highlights is to automatically reduce

the gain (compress) at high signal

levels.

The circuit in Figure 1 is a voltage-

controlled breakpoint ampliﬁer that

can be used for highlight compres-

sion. When the input signal reaches a

predetermined level (the breakpoint),

the ampliﬁer gain is reduced. As both

the breakpoint and the gain for sig-

nals greater than the breakpoint are

voltage programmable, this circuit is

useful for systems that adapt to chang-

ing signal levels. Adaptive highlight

compression ﬁnds use in CCD video

cameras, which have a very large

dynamic range. Although this circuit

was developed for video signals, it can

be used to adaptively compress any

signal within the 40MHz bandwidth

of the LT1256.

The LT1256 video fader is con-

nected to mix proportional amounts

of input signal and clipped signal to

provide a voltage-controlled variable

gain. The clipped signal is provided

by a discrete circuit consisting of

three transistors. Q1 acts as an emit-

ter follower until the input voltage

exceeds the voltage on the base of Q2

(the breakpoint voltage or V

). When

the input voltage is greater than V

Q1 is off and Q2 clamps the emitters

of the two transistors to V

plus a

. Q3, an NPN emitter follower,

buffers the output and drops the volt-

age a V

and thus the DC level of the

input signal is preserved to the extent

allowed by the V

matching and tem-

perature tracking of the transistors

used. The breakpoint voltage at the

base of Q2 must remain constant

when this transistor is turning on or

the signal will be distorted. The

LT1363 maintains a low output

impedance well beyond video frequen-

cies and makes an excellent buffer.

Figure 2 is a multiple-exposure

photograph of a single line of mono-

chrome video, showing four different

levels of compression ranging from

fully limited signal to unprocessed

input signal. The breakpoint is set to

40% of the peak amplitude to clearly

show the effect of the circuit; nor-

mally only the top 10% of video would

be compressed.

–

75Ω

VIDEO

OUT

1.5k

LT1256

LT1363

1.5k

Q1 Q2

–5V

100Ω

75Ω

–5V

100Ω

510Ω1.5k

BREAK POINT

VOLTAGE

100k

LT1004-2.5

TO V

FS

LT1256, PIN 12

TO V

CONTROL



LT1256, PIN 3

10k

Q1, Q2 =2N4957

 Q3 =2N2857

Figure 1. Voltage-controlled amplitude limiter

Figure 2. Multiple-exposure photograph of a

single line of monochrome video, showing

four different levels of compression

Linear Technology Magazine • February 1997

DESIGN IDEAS

New IC Features Reduce EMI from

Switching Regulator Circuits by John Seago

One disadvantage of using a switch-

ing regulator is that it generates

electronic noise, known as EMI (elec-

tromagnetic interference). This noise

can be conducted or radiated, and it

can affect other circuits in your prod-

uct or interfere with the operation of

nearby products. The LTC1436-PLL,

LTC1437, LTC1439 and LTC1539

have features that can be used to

suppress this interference.

Frequently, EMI problems don’t

show up until the integration phase

of product development. By using this

EMI suppression capability, a resis-

tor- or capacitor-value change may

be all that is required to solve an

interference problem. The LTC1436-

PLL shown in the circuit of Figure 1

produces a switched 5V, 3A output

and a 3.3V, 0.1A linear output. The

circuit is conﬁgured to provide either

switch-frequency synchronization or

switch-frequency modulation. Also,

transistor Q3 ensures constant fre-

quency at very low output current

levels, thus eliminating audio frequen-

cies and maintaining high efﬁciency

using the internal Adaptive Power™

circuitry.

Switch-Frequency

Synchronization

Switching regulator noise results from

switching high currents on and off.

This creates high energy levels at the

switching frequency and all of its

harmonics. A common EMI-control

technique is to synchronize the

switching frequency to an external

clock so that all harmonic frequencies

can be controlled. The LTC1436-PLL

uses a phase-locked loop for syn-

chronization to avoid the loss of slope

compensation common to other syn-

chronizing techniques. In addition,

the input to the VCO in the phase-

locked loop is available at the PLL LPF

(phase-locked loop lowpass ﬁlter) pin

so, that a lowpass ﬁlter can be used to

control how fast the loop acquires

lock.

Switch-Frequency Modulation

Access to the VCO input also makes it

possible to modulate the regulator’s

switching frequency. Through fre-

quency modulation, the peak energy

of the fundamental is spread over the

frequency range of modulation, thus

decreasing the peak energy level at

any one frequency. This frequency

spreading action increases with each

harmonic, so that the second har-

monic has twice the bandwidth and

the third harmonic has three times

the bandwidth until all the harmon-

ics blend together, decreasing the

signal strength at all frequencies. This

can be seen in the spectrum analyzer

plots shown in Figures 2, 3 and 4.

1

2

3

4

5

6

7

8

9

10

11

24

23

22

21

20

19

18

17

16

15

14

13



PLL LPF

OSC



RUN/SS



SFB

SGND

PROG



OSENSE



SENSE

–



SENSE



AUXON

AUXFB

PLLIN

POR

BOOST

TGL

SW

TGS



INTV



BG

PGND

EXTV



AUXDR

DI1436_01.eps

5V

POR

3.3V

0.1A

GND

LTC1436-PLL

10k

R6

47k

5.5V TO

24V R7

10Ω

100Ω

R2, 10k

MOD

PLL

0.01µF

C3, 0.1µF

C8

0.1µF

R11

47k

R10

35.7k

C15

3.3µF

C11, C12: KEMET T495X226M035AS

C13, C14: AVX TPSD107M010R0065

L1: SUMIDA CDRH125-10

Q1 + Q2: SILICONIX Si4936DY (DUAL FET)

Q3: INTERNATIONAL RECTIFIER IRLML2803

R8: IRC LR2010-01-R033-J

* SEE FIGURE 6

C9

4.7µF

C10

0.1µF

D1

MBRS0530 D2

MBRS130L

C4

330pF

C2, 47pF

C5, 47pF

C7, 0.001µF

R5

100Ω

INTV

C6

100pF

C13, C14

100µF

10V

×2

C12

22µF

35V

L1

10µH

R9

20k

R8

0.033Ω

C11

22µF

35V

Q4

MMBT2907ALT1

SWITCH-

FREQUENCY

MODULATOR*

Figure 1. 2-output LTC1436-PLL test circuit

Linear Technology Magazine • February 1997

DESIGN IDEAS

Figure 2 shows the full load output

noise level from the circuit of Figure

1, before and after switch-frequency

modulation. The black trace shows

the normal output noise from 1kHz to

1MHz with the VCO at minimum fre-

quency, whereas the colored trace

shows output noise after modulation

around the center frequency. The

228kHz unmodulated switch-fre-

quency output noise decreased more

than 30dB through modulation

between 270kHz and 370kHz. Fig-

ures 3 and 4 show a 10dB to 15dB

attenuation in full-load output volt-

age noise from 1MHz to 30MHz after

modulation.

The VCO in the LTC1436-PLL has

an input range from 0V to 2.4V. As

shown in Figure 5, the switch fre-

quency can be modulated at least

±30% around the center frequency f

The ideal modulating signal varies an

equal amount above and below the

center frequency voltage of 1.2V, with

a constant slope. The reference cir-

cuit of Figure 6 develops a 100Hz

sawtooth voltage from 0.9V to 1.5V

that modulates the LTC1436-PLL in

Figure 1 to generate the plots shown

in Figures 2, 3 and 4. Modulator

circuit complexity is largely deter-

mined by functional requirements.

For most applications, a precision

modulating signal is not required,

because high order harmonics blend

together. Consequently, modulating

frequency, slope and peak-to-peak

voltage are not critical.

Audio Frequency Suppression

The Adaptive Power feature of the

LTC1436-PLL signiﬁcantly reduces

audio frequency generation, while

maintaining good efﬁciency under

very light load conditions. Figure 7

shows the audio frequencies gener-

ated by the highly efﬁcient cycle

skipping mode of the LTC1436-PLL.

Figure 8 shows the decrease in au-

dio frequencies resulting from

Adaptive Power operation. Figure 9

shows efﬁciency curves of both the

cycle skipping and Adaptive Power

modes along with the traditional,

forced continuous mode of operation.

Figure 2. Output noise before and after switch-frequency modulation

Figure 3. Output high frequency noise before switch-frequency modulation

Figure 4. Output high frequency noise after switch-frequency modulation

Linear Technology Magazine • February 1997

DESIGN IDEAS

Cycle skipping is the most efﬁcient

mode during light-load operation,

where the output capacitor supplies

load current most of the time and is

replenished by bursts of energy at a

rate determined by the load. When

load current is low enough, the burst

rate falls into the audio-frequency

range, which can cause problems.

With the addition of Q3, an inexpen-

sive SOT-23 size MOSFET, the

Adaptive Power circuitry inside the

LTC1436-PLL takes control during

light load conditions, turning off high

current MOSFETs Q1 and Q2. Q3

and D2 are then used in a conven-

tional constant frequency buck mode,

eliminating the power loss caused by

charging and discharging the large

input capacitance of both power

MOSFETs.

The conventional way of avoiding

audio-frequency interference is the

forced current mode, where both high

current MOSFETs continue to oper-

ate at full frequency and normal duty

PLL LPF

(V)

0.7f

1.3f

FREQUENCY

2.5

DI1436_05.eps

0 2.01.51.00.5

OSC

= 47pF

OSC

= 100pF

–

DI1436_06.eps



DISCH

THRESH

CONT

GND

TRIG

OUT

RESET

1

2

3

0.1µF

TLC555

LTC1436-PLL

PIN 17

(5V)

LTC1436-PLL

PIN 6

(GND)

1.5V

0.9V ~10ms

1.2M

220Ω

0.1µF

2N3904

100k

150k

510k

LT1077

LTC1436-PLL

PIN 1

(MOD)

8

7

6

OUTPUT CURRENT

100

EFFICIENCY (%)

10A1mA 10mA 1A100mA

(3)

(2)

(1)

10V IN

5V OUT

1. CYCLE SKIPPING OPERATION: 

 VARIABLE FREQUENCY 

 COMPONENTS AT LOWER 

 OUTPUT CURRENTS

2. Adaptive Power MODE: 

 CONSTANT FREQUENCY WITH 

 AUTOMATIC SWITCHOVER TO 

 SMALL MOSFET Q3

3. FORCED CONTINUOUS OPERATION: 

 CONSTANT FREQUENCY USING 

 LARGE MOSFETS Q1 AND Q2

cycle under all load conditions. This

causes the peak-to-peak inductor

current to ﬂow, even under no load

conditions. The synchronous buck

topology allows the top switch, Q1, to

put current into the output capacitor,

followed by the bottom switch, Q2,

taking current out of the output ca-

pacitor while regulating the output

voltage under no-load conditions.

Although constant frequency is main-

tained, high current I

R losses and

high gate charge losses continue

under light load conditions. LTC1436-

PLL features forced-current operation

to provide the fast transient response

required for high di/dt loads like the

Intel Pentium

processor.

Cycle skipping, Adaptive Power and

forced current operation are all avail-

able on the LTC1436-PLL, so that the

best operating mode can be selected

for each application.

Conclusion

The family of LTC1436-PLL parts of-

fers very effective EMI suppression

features. Reviewing the spectrum

analyzer plots in Figures 2, 3 and 4

shows that the output voltage noise

amplitude can be signiﬁcantly re-

duced at frequencies to 30MHz with

switch-frequency modulation. The

Adaptive Power mode provides high

efﬁciency, constant-frequency opera-

tion at very low output currents and

avoids audio frequency operation.

Figure 5. Operating frequency vs V

PLLPF

Figure 6. Switch-frequency modulator

Figure 7. Audio frequencies in output noise

during cycle-skipping operation

Figure 8. Output noise with Adaptive Power

mode operation

Figure 9. Efﬁciency curves for light load currents

Pentium is a registered trademark of Intel Corp.

10Hz 10kHz

–20dBm

DI1436_08.eps

–120dBm

–100dBm

–80dBm

–60dBm

–40dBm

20kHz

10V



OUT

AT 3mA

BW = 100Hz

10Hz 10kHz

–20dBm

–40dBm

–60dBm

–80dBm

–100dBm

DI1436_07.eps

–120dBm 20kHz

10V



OUT

AT 3mA

BW = 100Hz

Linear Technology Magazine • February 1997

DESIGN IDEAS

Dual-Output Voltage Regulator

by Peter Guan

DI1263_01.eps

TDRIVE

PWR V

PINV

BINH

*COILTRONICS CTX0212801

LTC1266-3.3

1000pF

SENSE–

BDRIVE

PGND

OUT

SGND

SHDN

SENSE+



180pF



3300pF

1µF

C2 = 0.47µF

C3 = 10µF

C4 = 10µF

C1 = 0.47µF

SENSE



0.02Ω



470Ω

OUT

= 3.3V/5A

OUT

= 12V/60mA

16 Si9410DY D1

MBRS140T3



100µF

20V

× 2

OUT



220µF

10V

× 2

L*

5µH

Si9410DY

C1–

C1+

C2–

C2+

LTC1263



FROM µP

GND

OUT

SHDN

The LTC1266-3.3 and LTC1263 are

perfect complements for one another.

The combination of these two parts

provides two regulated outputs of

3.3V/5A and 12V/60mA from an in-

put range of 4.75V to 5.5V. These two

outputs are perfect for notebook and

palmtop computers with micropro-

cessors that burn several amps of

current from a regulated 3.3V sup-

ply, ﬂash memories that consume

milliamps of current from a regulated

12V supply and interface and logic

components that still run off the 5V

supply. In fact, this quick and easy

combination may well be the aspirin

for many of the headaches caused by

the rigorous power supply demands

in today’s electronics.

The LTC1263, using only four ex-

ternal components (two 0.47µF charge

capacitors, one 10µF bypass capaci-

tor and a 10µF output capacitor),

generates the regulated 12V/60mA

output from a 5V input using a charge

pump tripler. During every period of

the 300kHz oscillator, the two charge

capacitors are ﬁrst charged to V

and then stacked in series, with the

bottom plate of the bottom capacitor

shorted to V

and the top plate of the

top capacitor connected to the output

capacitor. As a result, the output

capacitor is slowly charged up from

5V to 12V. The 12V output is regu-

lated by a gated oscillator scheme

that turns the charge pump on when

OUT

is below 12V and turns it off

when it exceeds 12V.

The LTC1266-3.3 then uses the 5V

input along with the 12V output from

the LTC1263 and various external

components, including bypass capaci-

tors, sense resistors and Schottky

diodes, to switch two external N-chan-

nel MOSFETs and a 5µH inductor to

charge and regulate the 3.3V/5A out-

put. The charging scheme for this

part, however, is very different from

that of the LTC1263. The LTC1266-

3.3 ﬁrst charges the output capacitor

by turning on the top N-channel

MOSFET, allowing current to ﬂow

from the 5V input supply and through

the inductor. By monitoring the

Figure 1. 5V to 3.3V/5A and 12V/60mA supply

continued on page 32

Linear Technology Magazine • February 1997

DESIGN IDEAS

DIDPM_01.eps

LT1303

SHDNONOFF

10µF

25V

1.8VDC–

6VDC

10µF

25V

10µF

25V

PGNDGND

R2

1.2k

R1

8.2k

MBR0520L

Q1

2N3906

10µF

25V

10µF

25V

10.7µH

COILTRONICS

VP1-0190

DIGITAL

PANEL

METERS

10µF

25V

5 × MBR0520L

10µF

25V

Free Digital Panel Meters from the

Oppressive Yoke of Batteries by Mitchell Lee

Digital panel meters (DPMs) have

dropped in price to well under $10 for

3-1/2 digit models, even in single-

piece quantities. These make excellent

displays for many instruments, but

suffer from one major ﬂaw: they re-

quire a ﬂoating power supply, usually

in the form of a 9V battery. This

renders inexpensive meters useless

for most applications because no one

wants multiple 9V batteries in their

product.

The circuit shown in Figure 1 pow-

ers up to ﬁve meters from a single

1.8V to 6V source. The source need

not be ﬂoating, yet all ﬁve outputs are

fully ﬂoating, isolated and indepen-

dent in every respect. The circuit

consists of an LT1303 micropower,

high efﬁciency DC/DC converter driv-

ing a 5-output ﬂyback converter. An

off-the-shelf surface mount coil,

Coiltronics’ Versa-Pac™ VP1-0190, is

used as the transformer. This device

is hipot tested to 500V

RMS

—more than

adequate for most applications.

Feedback is extracted from the

primary by Q1, which samples the

ﬂyback pedestal during the switch off

time. Typical DPMs draw approxi-

mately 1mA supply current. The

primary is also loaded with 1mA for

optimum regulation and ripple. Pri-

mary snubbing components, a

necessity in most ﬂyback circuits, are

obviated by the primary feedback rec-

tiﬁer and smoothing capacitor.

Although this circuit has been set up

for 9V output (9.3V, to be exact), some

DPMs need 5V or 7V. Use a 4.3k or

6.2k r esistor in place of R1 for these

voltages. The output voltage is set by

R1 = (VOUT – 0.7)/1mA.

Do not attempt to regulate the out-

put beyond 10V or you will exceed the

maximum switch rating of the LT1303.

The LT1111 is better suited for higher

voltage applications.

Output ripple measures 200mV

P–P

and can be proportionately reduced

by increasing the output capacitance.

If more ripple is acceptable, the out-

put capacitors can be reduced in

value. A shutdown feature is avail-

able on the LT1303, useful where a

“sleep” function is included to save

power.

With each output loaded at 1mA,

the input current is 16.5mA on a 5V

supply. This ﬁgure rises to about

45mA on a 1.8V (2-cell) input. If the

system is battery operated and if the

battery voltage does not exceed 7V,

operate the circuit directly from the

battery for best efﬁciency. In line-

operated equipment, use a regulated

5VDC or 3.3VDC supply.

Figure 1. LT1303 ﬂyback regulator provides fully ﬂoating and isolated 9V supplies to

ﬁve independent digital panel meters. Substitute 4.3k for R1 if 5V meters are used.

Versa-Pac is a trademark of Coiltronics, Inc.

Linear Technology Magazine • February 1997

DESIGN IDEAS

Battery Charger IC Can Also Serve

as Main Step-Down Converter by Arie Ravid

1

2

3

4

5

6

7

16

15

14

13

12

11

10

GND

CC1



CC2



PROG

C

BAT

GND

GND

GND

SW

BOOST

GND

OVP

SENSE

GND

GND

U1

LTC1510CS16

C9*

0.1µF

CR7* 

1N914

CR6* 

1N914

CR1

1N5819

R9

100k

Q4

VN2222

CHARGE/TRICKLE

R8

16.2k

R7*R6

300Ω

R5

R4

4.99k

BAT1

C5

1µF

C4

0.1µF

Q3

VN2222

C3

22µF

25V

CR3 1N5819

C2† 

10µF

Q1*

MPS3906

R3

1.21K

CR5 1N5819

Q2*

Si9433

R2*

R1

100k

SYSTEM

ON/OFF

SWITCH

C1 

0.22µF

L1**

33µH

L2

2.2mH

L3

10mH

CR4

1N5817

SYSTEM

LOAD

C7

100µF

C6

100µFU2

LT1300CS8

C8*

0.1µF

SELECT SENSE

PGND GND

SHDN I

LIM

CR2 

1N914

NC NC

 *SEE TEXT

 **COILTRONICS CTX33-2

 †TOKIN OR MARCOM CERAMIC SURFACE MOUNT

Using a power adapter with the

highest feasible output voltage is at-

tractive to portable system designers

for a couple of reasons. Lower current

is required to maintain the same sys-

tem power, which translates into a

smaller cable and input connector. If

the adapter output voltage is consid-

erably higher than the battery voltage,

the adapter output voltage does not

need to be regulated or well ﬁltered,

resulting in lower adapter cost.

A portable system with a high out-

put-voltage adapter, however, requires

that the system’s DC-to-DC converter

functions over a very wide range of

input voltage: from fully discharged

battery voltage to the highest adapter

output voltage.

This problem can be resolved by

using the LT1510 as both the battery

charger and the main step-down con-

verter, as shown in Figure 1. An

important feature of the circuit in

Figure 1 is the glitch-free transfer

from AC operation to battery opera-

tion and back.

The LT1510 battery charger IC is

capable of charge current up to 1.5A

and output (battery) voltage up to

20V. High efﬁciency and small induc-

tor size are achieved by a saturating

switch running at 200kHz. The

LT1510 is capable of charging lithium-

ion and sealed-lead-acid batteries in

the constant-voltage/constant-

current conﬁguration, and nickel-

cadmium and nickel-metal-

hydride batteries in the constant-

current conﬁguration. The LT1510

contains an internal switch and cur-

rent sense resistor. All the designer

needs to do in order to program the

current and voltage is select the cur-

rent-programming resistor and the

voltage-divider resistors.

In the circuit shown in Figure 1,

the system’s DC-to-DC converter is

connected to the SENSE pin. This

way, the internal sense resistor is

bypassed for the system load but is

active in regulating the charge cur-

rent. The sum of the charge current

and system current should not ex-

ceed the maximum output current

allowed (limited by thermal consider-

ations or peak switch current). Since

the DC-to-DC converter circuit has a

large input capacitor, it cannot be

connected directly to the SENSE pin.

This is because the internal sense

resistor between SENSE and BAT pins

will see a large capacitance across it,

which will cause instability. A 2.2mH

inductor, such as the DT1608C-222

by Coilcraft (L2), is used to isolate the

input capacitance of the DC-to-DC

converter. CR5 limits the transient

current through the LT1510’s inter-

nal sense resistor when the system is

operating on battery and turned on.

Q2 (Si9433) is required if the series

resistance of 0.2 between the BA T

pin and SENSE pin is too high. The

Si9433’s on resistance is 0.075 . The

Figure 1. LT1510 battery charger/main step-down converter provides glitch-free transfer between AC and battery operation.

continued on page 36

Linear Technology Magazine • February 1997

DESIGN IDEAS

Combine a Switching Regulator

and an UltraFast Linear Regulator

for a High Performance 3.3V Supply

by Craig Varga

Introduction

It is becoming increasingly necessary

to provide low voltage power to micro-

processor loads at very high current

levels. Many processors also exhibit

high speed load transients. The Pen-

tium

Pro processor from Intel exhibits

both of these requirements. This

processor requires 3.3V ±5% at

approximately 14A peak (9A average)

and is capable of making the transi-

tion from a low power state to full load

in several clock cycles. Generally,

switching regulators are used to sup-

ply such high power devices, because

of the unacceptable power losses

associated with linear regulators.

Unfortunately, switching regulators

exhibit much slower transient re-

sponse than linear regulators. This

greatly increases the output capaci-

tor requirements for switchers.

Circuit Operation

The circuit shown in Figure 1 takes

advantage of a new, ultrahigh speed

linear regulator combined with a

switching regulator to get the best of

both worlds. An LTC1435 synchro-

nous buck regulator is combined with

an LT1575 linear regulator to gener-

ate a 3.3V output from a 12V input

with an overall conversion efﬁciency

of approximately 72%. The output is

capable of current slew rates of

approximately 20A per microsecond.

+ + +

DI1575_01.eps

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150µF

16V

C12

150µF

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C13

150µF

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C3, 0.1µF

C9

1500pF

R5

16.5k

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C8, 68pF

C7, 0.1µF

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CERAMIC

0805 CASE



 L1=COILTRONICS CTX01-13199-X2

 Q2, Q3 =SILICONIX SUD50N03-10

Figure 1. 12V to 3.3V/9A (14A peak) hybrid regulator

continued on page 36

Pentium is a registered trademark of Intel Corp.

Linear Technology Magazine • February 1997

DESIGN IDEAS

What Efﬁciency Curves Don’t Tell

Introduction

In switching regulators’ data sheets,

there are always efﬁciency curves that

show how efﬁcient the regulators are

in transforming one voltage to an-

other. Although these curves are

useful in comparing one regulator to

another, they don’t allow a system

designer to determine accurately how

long batteries will last before they

need to be replaced or recharged when

they are used as the power source.

This complication arises because the

type of batteries used to power the

system and the regulator load char-

acteristic strongly affect the lifetime

of the batteries.

In this article, battery lifetime

curves are obtained for the LTC1174

and the LTC1433.

A Short Introduction to the

LTC1174 and LTC1433

The LTC1174 uses a constant off-

time architecture to switch its internal

P-channel power MOSFET. The in-

put-to-output voltage ratio sets the

on time and requires the inductor

current to reach a preset limit. Even

at low load current, the LTC1174 still

requires the inductor current to reach

the preset limit before it initiates the

off-time cycle. Burst Mode operation

of the LTC1174 enhances efﬁciency

throughout the load-current range

by switching only the required num-

ber of cycles to bring the output into

regulation and then stopping switch-

ing (going into sleep mode). When the

output voltage has dropped slightly,

the switching sequence resumes. By

doing this, switching losses are re-

duced and are minimized when the

load current is low, because the sleep

duration is long.

The LTC1433 is a constant-fre-

quency, current mode, monolithic

switching regulator in which the in-

ductor peak current varies according

to the load current. In place of Burst

Mode operation, the LTC1433 has an

Adaptive Power output stage to en-

hance its efﬁciency at low load current.

Under low load conditions, the

LTC1433 uses only a fraction of its

power MOSFET, effectively reducing

switching losses without introducing

low frequency noise components.

For more information on both parts,

consult the data sheets.

The Setup

The circuits in Figures 1a, 1b and 1c

were used to obtain the lifetime data.

All outputs were set at 3.3V and the

power was supplied by either four AA

alkaline (Eveready No. EN91) or four

AA NiCd (Eveready No. CH15) cells or

a single 9V alkaline (Eveready No.

EN22) battery. A current-sink load

was set up to either draw a constant

400mA or provide a load-step charac-

teristic. The load stepping operated at

0.05Hz, going from 10mA to 410mA

with a duty cycle of 10%, providing an

average load current of 50mA.

In Figure 1b, the LTC1433 was set

up to optimize low load current efﬁ-

ciency by conﬁguring the Adaptive

Power output stage with separate in-

ductors for low and high current

operation.

Efﬁciency curves for each circuit

are shown in Figure 2a and 2b. Fig-

ures 3 through 8 show the battery

voltage and regulator output voltage

versus time for various battery and

load combinations.

4-Cell to 3.3V Conﬁguration

Figures 3 and 4 were obtained with a

load current of 400mA. For Figure 3,

the input power to the regulator was

provided by four AA alkaline batter-

ies, whereas four AA NiCds were used

in Figure 4. The alkaline batteries

lasted longer than the NiCds, due to

by San-Hwa Chee

DI_EFF_01a.eps

IN

PGND



OSC

POR



OSENSE



PROG



16

15

14

13

12

11

10

LTC1433

D1

MBRM5819

C6

100µF

10V

OUT

L1

22µH

C7

0.1µF

1

2

3

4

5

6

7

SSW

NC

BSW

NC

SGND

RUNSS

LB0

LB1

C3

33µF, 20V

C4

0.1µF

R1

5.1k

C5

47pF

C1

6800pF

C2

680pF

L1 = SUMIDA CD54-220

DI_EFF_01b.eps

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PGND



OSC

POR



OSENSE



PROG



16

15

14

13

12

11

10

LTC1433

MBRM520LT1

MBRM5819

C6

100µF

10V

OUT

C7

0.1µF

1

2

3

4

5

6

7

SSW

NC

BSW

NC

SGND

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LB1

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C4

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R1

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C5

47pF

C1

6800pF

C2

680pF

100µH

22µH

L1 = SUMIDA CD54-101

L2 = SUMIDA CD54-220

Figure 1a. LTC1433 single-inductor conﬁguration Figure 1b. LTC1433 dual-inductor conﬁguration

Linear Technology Magazine • February 1997

DESIGN IDEAS

their higher energy capacity. From

Figure 4, it is apparent when the NiCd

gives up, from the cliff-like shape of

the output voltage.

For Figures 5 and 6, a step load

was applied to the regulators instead

of a DC load. Figure 5 and 6 are the

data obtained for alkaline and NiCd

AA cells, respectively. With the aver-

age load one-eighth of the previous

experiment, it would be expected that

the lifetime of the alkaline batteries

would be eight times longer or ap-

DI_EFF_01c.eps

SHDN

OUT



8

1

LTC1174-3.3

GND

D1

MBR0520LT1

C1

100µF

10V

C3

0.1µF

C2

22µF

50V

OUT

3

2

BIN



OUT



PGM

L1



68µH

L1 = SUMIDA CDRH74-680



INPUT VOLTAGE (V)

100

EFFICIENCY (%)

DI_EFF_02a.eps

48765

FIGURE 1A 

FIGURE 1B

FIGURE 1C

INPUT VOLTAGE (V)

100

EFFICIENCY (%)

DI_EFF_02b.eps

48765

FIGURE 1A 

FIGURE 1B

FIGURE 1C

TIME (HOURS)

BATTERY AND OUTPUT VOLTAGE (V)

2.5

DI_EFF_03.eps

0 0.5 1.0 1.5 2.0

LTC1433 WITH DUAL INDUCTORS

LTC1433 WITH SINGLE INDUCTOR

LTC1174HV

OUTPUT

VOLTAGE

BATTERY

VOLTAGE

TIME (HOURS)

BATTERY AND OUTPUT VOLTAGE (V)

2.5

DI_EFF_04.eps

0 0.5 1.0 1.5 2.0

LTC1433 WITH DUAL INDUCTORS

LTC1433 WITH SINGLE INDUCTOR

LTC1174HV

OUTPUT

VOLTAGE

BATTERY

VOLTAGE

proximately 18 hours, but Figure 5

shows a signiﬁcantly better result.

The main reason for this improve-

ment has to do with the internal

resistance of the alkaline cell. At high

constant DC load current, heat is

dissipated by the internal resistance

of the alkaline batteries. The internal

resistance increases as the batteries

voltage decreases, and hence causes

more heat to be dissipated, thus low-

ering the lifetime.

For the NiCd battery, internal re-

sistance is low and remains relatively

constant over its life span. Therefore,

the lifetime of the NiCd batteries for

the load step case comes out to be

approximately the expected eight

times that of a constant DC load

current.

The above result indicates that if

the load is intermittent in nature, the

user can operate the device much

longer if the power is provided by

TIME (HOURS)

BATTERY AND OUTPUT VOLTAGE (V)

DI_EFF_05.eps

0 1020304050

LTC1433 WITH DUAL INDUCTORS

LTC1433 WITH SINGLE INDUCTOR

LTC1174HV

OUTPUT

VOLTAGE

BATTERY

VOLTAGE



TIME (HOURS)

BATTERY AND OUTPUT VOLTAGE (V)

DI_EFF_06.eps

02468 1214161810

LTC1433 WITH DUAL INDUCTORS

LTC1433 WITH SINGLE INDUCTOR

LTC1174HV

OUTPUT

VOLTAGE

BATTERY

VOLTAGE

Figure 1c. LTC1174 test circuit Figure 2a. Efﬁciency curves for Figure 1’s

circuits, I

LOAD

= 400mA Figure 2b. Efﬁciency curves for Figure 1’s

circuits, I

LOAD

= 10mA

Figure 3. Lifetime at I

LOAD

= 400mA—four AA alkaline batteries Figure 4. Lifetime at I

LOAD

= 400mA—four AA NiCd batteries

Figure 5. Lifetime with load step from 10mA to 410mA, 10% duty

cycle, T

PERIOD

= 20s—four AA alkaline batteries Figure 6. Lifetime with load step from 10mA to 410mA, 10% duty

cycle, T

PERIOD

= 20s—four AA NiCd batteries

Linear Technology Magazine • February 1997

DESIGN IDEAS

alkaline batteries. Again, the NiCd

exhibits a sudden “death” at the end

of its life, whereas the alkaline shows

a much gentler decay. The gentle

sloping of the output voltage of Figure

5 towards the end of the battery life

can be attributed to the on-resistance

of the switch when the regulator is in

dropout.

For the above load characteristic,

where the load is light most of the

time, making full use of the Adaptive

Power mode of the LTC1433 by means

of the dual inductor conﬁguration

helps to squeeze an additional 1.5

hours of life compared to the single

inductor LTC1433 conﬁguration.

Another important point to note is

that although the efﬁciency for the

LTC1174 is better than that of the

single inductor conﬁguration of the

LTC1433 at 10mA load current, the

LTC1433 lasted 2.9 hours longer than

TIME (HOURS)

BATTERY AND OUTPUT VOLTAGE (V)

2.5

DI_EFF_07.eps

0 0.5 1.0 1.5 2.0

 LTC1433 WITH DUAL INDUCTORS

 LTC1433 WITH SINGLE INDUCTOR

 LTC1174HV

OUTPUT

VOLTAGE

BATTERY

VOLTAGE

TIME (HOURS)

BATTERY AND OUTPUT VOLTAGE (V)

DI_EFF_08.eps

02468 1214161810

 LTC1433 WITH DUAL INDUCTORS

 LTC1433 WITH SINGLE INDUCTOR

 LTC1174HV

OUTPUT

VOLTAGE

BATTERY

VOLTAGE

the LTC1174 in Figure 5. The reason

for this is that the LTC1174 inductor’s

current always ramps up to the pre-

set value of 600mA whether the load

current is at 10mA or at 410mA. This

high peak inductor current, combined

with the high internal resistance of

the alkaline AA cells, shortens the

lifetime. Figure 6 shows that the use

time is about the same for the

LTC1174 and the LTC1433 because

of the low, constant internal resis-

tance of the NiCd batteries.

9V-to-3.3V

The lifetime graphs are shown in Fig-

ures 7 and 8. Comparing the data

between the 9V and the AA alkaline

cells, the lifetime of the AA cells is

about 2.5 times longer. This is be-

cause the energy capacity of the 9V

alkaline is much smaller than that of

the AA cells. In addition, the internal

resistance of the 9V alkaline is much

higher than the AA cells, causing

more energy to be dissipated as heat.

For the load step case, the battery

lasted 13.8 times longer than a con-

stant 400mA load. The dual inductor

conﬁguration of the LTC1433 lasted

about an hour longer than the single

inductor one.

Conclusion

Although a switching regulator can

be highly efﬁcient, its architecture,

the type of load it is driving and its

power source can have a signiﬁcant

effect on how long batteries will last.

For a load that varies from low load to

heavy load at low duty cycle, it is

worthwhile to consider powering your

system with alkaline batteries, be-

cause they last longer. For a constant

heavy load, NiCd and alkaline come

out to be about the same but NiCds,

of course, can be recharged.

Figure 7. Lifetime at I

LOAD

= 400mA—one 9V alkaline battery Figure 8. Lifetime with load step from 10mA to 410mA—one 9V

alkaline battery (10% duty cycle, T

PERIOD

= 20s)

amount of current ﬂow in the induc-

tor with a sense resistor, the 3.3V

output is regulated by turning on and

off the top and bottom N-channel

MOSFETs to charge and discharge

the output capacitor.

If we replaced the top external N-

channel MOSFET with a P-channel,

the LTC1266-3.3 could generate the

same 3.3V/5A output without the

help of the LTC1263. But, since N-

channel MOSFETs have lower gate

capacitance and lower R

DS(ON)

, their

higher efﬁciency at high currents more

than compensates for the extra com-

plexity in bringing in another higher

input voltage, especially if that sec-

ond input voltage is readily available.

Since both of these devices are very

stingy on quiescent current, their

combination is also very gentle to the

main power supply, especially if that

power supply is a battery. In standby

mode, the LTC1263 and the LTC1266-

3.3 have a total quiescent current of

about 500µA. To conserve even more

current, both of these parts can be

put into shutdown mode by ﬂoating

their shutdown pins or pulling them

high. The total shutdown current is

less than 40µA. When loaded, the

LTC1263 has a 76% efﬁciency,

whereas the LTC1266-3.3 can squeeze

out more than 90%. Together, with a

60mA load at the 12V output and a 5A

load at the 3.3V output, the overall

efﬁciency is 87%.

The LTC1266-3.3 is available in the

16-pin SO package and the LTC1263

is available in the 8-pin SO package.

Together, these two parts provide an

easy and efﬁcient solution for multiple

power supply demands.

Dual-Output, continued from page 26

Linear Technology Magazine • February 1997

DESIGN INFORMATION

Introducing the LT2078/LT2079

and LT2178/LT2179 Single Supply,

Micropower, Precision Ampliﬁers in

Surface Mount Packages by Raj Ramchandani

Introduction

Circuit designers seldom have an

opportunity to revisit a product as

successful as the LT1078/LT1079

and LT1178/LT1179 dual and quad

micropower precision op amps. In the

last decade, these ampliﬁers have

become true industry standards, with

outstanding DC precision and only

55µA and 21µA of supply current per

op amp, respectively. A look at the

data sheet shows that the LT1078

and LT1178 have much better offset

voltage and offset voltage drift in the

dual in-line package (DIP) than in the

small outline surface mount package

(SO) (see Table 1). Since the introduc-

tion of these parts, the demand for

surface mount packages has grown

dramatically; today, the majority of

ICs sold are in surface mount pack-

ages. The new LT2078 and LT2178

deliver the precision performance of

the LT1078 and LT1178 in SO pack-

ages without changing any of the

other characteristics that made these

parts so popular.

Package Limitations

of Precision Specs

An inherent problem with precision

ICs is that as circuit design and

process innovations improve the per-

formance, packaging issues can

become the major limitation in achiev-

ing precision speciﬁcations. Typically,

the best precision is in metal-can

packages; this is because there is

very little mechanical stress on the

die. Plastic packages exert stress on

the top and sides of the die as the

mold compound cools, causing

changes in the offset voltage. A solu-

tion that has been used for many

years is to put a jelly-like coating all

over the die before molding; this keeps

the plastic (and its stress) off the die.

This works well in the DIP package

but in the small SO package there is

not enough room for this coating.

Optimizing Performance

in Surface Mount

A new coating with much lower vis-

cosity before curing was developed so

that a thin (about 50 micron) coat

would cover the top of the die. The

coating runs to the edge of the die,

where it stops due to capillary action.

This coating is cured to a jelly-like

consistency before the plastic is

molded, keeping the stress off the top

of the die. A new dispensing system

was developed to control the quantity

of “thin coat” on each die, since it

must be controlled more accurately

than the old DIP coating. Unfortu-

nately, the sides of the die in the SO

package are still stressed by the mold-

ing compound, although several new

molding compounds have become

available in the last few years that

exert less stress on the die, making

this problem less severe.

We designed a new die in order to

take full advantage of these new sur-

face mount packaging technologies.

The circuit design and process used

for the LT2078 and LT2178 are the

same as their predecessors; only the

die has been changed, resulting in

identical AC performance. The mask

continued on page 36

retemaraP

8S8701TL )8-OScitsalp( 8SC8702TL )8-0Scitsalp( 8SCA8702TL )8-OScitsalp(

VmumixaM

081 µV021µV07µV

VmumixaM

tfirD5.3µC˚/V5.2µC˚/V8.1µC˚/V

S9701TL

)61-OScitsalp( C9702TL

)41-OScitsalp( CA9702TL

)41-OScitsalp(

VmumixaM

003µV051µV001µV

VmumixaM

tfirD0.4µC˚/V5.3µC˚/V0.3µC˚/V

retemaraP

8SA8711TL )8-OScitsalp( 8SC8712TL )8-OScitsalp( 8SCA8712TL )41-OScitsalp(

VmumixaM

081 µV0 21 µV07µV

VmumixaM

tfirD5.3µC˚/V5.2µC˚/V8.1µC˚/V

S9711TL

)61-OScitsalp( SC9712TL

)41-OScitsalp( SCA9712TL

)41-OScitsalp(

VmumixaM

006µV051µV001µV

VmumixaM

tfirD5.4µC˚/V5.3µC˚/V0.3µC˚/V

Table 1. LT1078/LT1079 and LT2078/LT2079 offset voltage performance comparison

Table 2. LT1178/LT1179 and LT2178/LT2179 offset voltage performance comparison

Linear Technology Magazine • February 1997

DESIGN INFORMATION

LTC1387 Single 5V RS232/RS485

Multiprotocol Transceiver by Y.K. Sim

Introduction

The LTC1387 is a new addition to

Linear Technology’s family of multi-

protocol transceivers. It is a single 5V

supply, logic-conﬁgurable, single-port

RS232 or RS485 transceiver. This

part is targeted at handheld comput-

ers or point-of-sale terminals and

features software-controlled multipro-

tocol operation with an emphasis on

ﬂexibility and minimum pin count.

Positioning and Key Features

The LTC1387 complements the dual-

port LTC1334 by providing a single

port in a smaller 20-pin SO or SSOP

package. The LTC1387 offers a ﬂex-

ible combination of two RS232 drivers,

two RS232 receivers, an RS485 driver,

an RS485 receiver and an onboard

charge pump to generate boosted volt-

ages for true RS232 levels from a

single 5V supply. The RS232 trans-

ceivers and RS485 transceiver are

designed to share the same port I/O

pins for both single-ended and differ-

ential signal communication modes.

The RS232 transceiver supports both

RS232 and EIA562 standards,

whereas the RS485 transceiver

supports both RS485 and RS422

standards. Both half-duplex and full-

duplex communication are supported.

A logic input selects between RS485

and RS232 modes. Three additional

control inputs allow the LTC1387 to

be reconﬁgured easily via software to

adapt to various communication

needs, including a one-signal line

1387_01.eps

LTC1387 CONTROLLER

DX1

VCC

120Ω

ARA

RS485

RS232

INTERFACE

RXEN

DXEN

MODE

DZ/SLEW

RXEN

DXEN

485/232

 RS232 RS232 RS485 RS485 SHUTDOWN

TRANSMIT MODE RECEIVE MODE TRANSMIT MODE RECEIVE MODE MODE

 RXEN = 0 RXEN = 1 RXEN = 0 RXEN = 1 RXEN = 0

 DXEN = 1 DXEN = 0 DXEN = 1 DXEN = 0 DXEN = 0

 MODE = 0 MODE = 0 MODE = 1 MODE = 1 MODE =X

1387_02.eps

LTC1387 CONTROLLER

DX1

120Ω

ARA

RS485

RS232

INTERFACE

RXEN

DXEN

MODE

DZ/SLEW

RXEN

DXEN

485/232

 RS232 RS485 RS485 SHUTDOWN

 MODE TRANSMIT MODE RECEIVE MODE MODE

 RXEN = 1 RXEN = 0 RXEN =1 RXEN = 0

 DXEN = 0 DXEN = 1 DXEN =0 DXEN = 0

 MODE = 0 MODE = 1 MODE = 1 MODE = 0

Figure 1. Half-duplex RS232, half-duplex RS485

Figure 2. Full-duplex RS232, half-duplex RS485

Linear Technology Magazine • February 1997

DESIGN INFORMATION

RS232 I/O mode (see function tables

in ﬁgures). Four examples of inter-

face port connections are shown in

Figures 1–4.

A SLEW input pin, active in RS485

mode, changes the driver transition

between normal and slow slew-rate

modes. In normal RS485 slew mode,

the twisted pair cable must be termi-

nated at both ends to minimized signal

reﬂection. In slow-slew mode, the

maximum signal bandwidth is re-

duced, minimizing EMI and signal

reﬂection problems. Slow-slew-rate

systems can often use improperly ter-

minated or even unterminated cables

with acceptable results. If cable ter-

mination is required, external

termination resistors can be con-

nected through switches or relays.

The LTC1387 features micropower

shutdown mode, loopback mode for

self-test, high data rates (120kbaud

for RS232 and 5Mbaud for RS485)

and 7kV ESD protection at the driver

outputs and receiver inputs.

Conclusion

The LTC1387 is ideal for point-of-sale

terminals, computers, multiplexers,

networks or peripherals that need to

adapt on the ﬂy to various I/O con-

ﬁguration requirements without any

hardware adjustments. The LTC1387,

along with the rest of the LTC multi-

protocol interface line, makes

dedicated single-protocol communi-

cation ports obsolete.

1387_03.eps

RX1

LTC1387 CONTROLLER

DX1

120Ω

ARA

RS485

RS232

INTERFACE

RXD

RS232

RXEN

DXEN

MODE

DZ/SLEW

RXEN

DXEN

485/232

RS232 MODE RS485 MODE SHUTDOWN MODE

 RXEN = 1 RXEN = 1 RXEN = 0

 DXEN = 1 DXEN = 1 DXEN = 0

MODE = 0 MODE =1 MODE = X

1387_04.eps

RX1

LTC1387 CONTROLLER

RX2

DX1

120Ω

ARA

RS485

RS232

INTERFACE

RXD

CTS

TXD

RTS

RS232

DX2/SLEW

RXEN

DXEN

MODE

DZ/SLEW

RXEN

DXEN

485/232

TERMINATE

RS232 MODE RS485 MODE SHUTDOWN MODE

 ON = 1 ON = 1 ON = 0

 RXEN = 1 RXEN = 1 RXEN = 0

 DXEN = 1 DXEN = 1 DXEN = 0

MODE = 0 MODE =1 MODE = X

Figure 3. Full-duplex RS232 (1-channel), full-duplex RS422

Figure 4. Full-duplex RS232 (2-channel), full-duplex RS485 with slew and termination control

Linear Technology Magazine • February 1997

design (layout) of the new ICs care-

fully placed the input transistors in

the area of the die where the package-

induced stress is most uniform.

Further, the input transistors are

“cross-coupled” to cancel out any

nonuniform stresses. The wafer level

trimming was tightened so that as-

sembled parts have less offset

variation. Postpackage trimming was

The LT1575 uses an IRLZ44 MOS-

FET as the pass transistor, allowing

the dropout voltage to be less than

550mV. Setting the switching supply’s

output to only 700mV above the out-

put of the linear regulator ensures

output regulation. The switcher is

therefore set up to deliver 4.0V at 14A

from the 12V supply. Conversion

efﬁciency of the switcher is around

90% (depending on load), whereas

the LT1575’s efﬁciency is 82.5% (see

LOAD CURRENT (A)

100

EFFICIENCY (%)

DI1575_02.eps

024681012

SWITCHER EFFICIENCY

TOTAL EFFICIENCY

Figure 2). The 12V input current is

only about 5.5A. At an average cur-

rent of 9A, the power dissipation in

the linear pass transistor is only 6.3W.

A small stamped aluminum heat sink

is adequate.

Figure 3 shows the transient re-

sponse to a 10A load step with a rise

time of approximately 50ns. The only

output capacitance is 40 1µF ceramic

capacitors. No additional bulk ca-

pacitance is required at the

processor. The circuit eliminates ap-

proximately a dozen low ESR tantalum

capacitors at the load, which would

be required without the linear post-

regulator. The switching supply’s

output is decoupled with three alu-

minum electrolytic capacitors.

Because the transient response at

this point is much less critical than at

the load, the long-term degradation

of the aluminum capacitors will not

be as detrimental to the circuit’s per-

formance as it would be if they were

used for load decoupling.

Conclusion

By combining a high efﬁciency switch-

ing regulator and an UltraFast™ linear

regulator, it is possible to achieve

reasonable efﬁciencies with superior

transient dynamics. Power dissipa-

tion can be held down to easily

manageable levels while eliminating

the need for very large amounts of

bulk decoupling capacitance.

Figure 2. Efﬁciency of Figure 1’s circuit Figure 3. Transient response of Figure 1’s

circuit to a 10A load step

optimized for the new SO packages.

(Postpackage trimming is done by

taking an input pin above the supply

pin to forward bias a diode and “zap”

a trim Zener.)

The result is that the LT2078/

LT2079 and LT2178/LT2179 are the

best single supply, precision op amps

available in the standard-pinout

surface mount packages. Table 1 com-

pares the dual LT2078 and quad

LT2079 to their predecessors, the

LT1078/LT1079, whereas Table 2

compares the LT2178/LT2179 to the

LT1178/LT1179 in surface mount

packages. These new parts are avail-

able in the SO-8 and SO-14 packages

for operation over commercial,

industrial and extended temperature

ranges.

3.3V Supply, continued from page 29

LT2078/LT2079, continued from page 33

charge pump comprising C8, C9, CR6,

CR7 and R2 biases the gate of Q2. Q1

and R1 turn Q2 off on AC operation

active). R7 programs the trickle-

charge current (maximum value is

about 100k) and the equivalent value

of R7 and R8 programs the charge

current. The Charge input must be

pulled low at the end of the charge.

The charger in Figure 1 is con-

nected to a 2-cell NiCd battery, BAT1.

The system switching regulator is

LT1300 (U2) based and powers a 5V/

250mA load. The efﬁciency, h, of the

complete system is deﬁned as:

LT1300 Output Power + Battery Charger Power

LT1510 Input Power

h =

The efﬁciency plot is shown in

Figure 2. For the purpose of measure-

ment, the battery voltage is 3.2V and

the charge current is 0.3A.

INPUT VOLTAGE (V)

67.5

68.0

68.5

69.0

69.5

70.0

70.5

71.0

71.5

72.0

72.5

EFFICIENCY (%)

DI1510_02.eps

82313 18

Figure 2. System efﬁciency vs input voltage

Battery Charger, continued from page 28

CONTINUATIONS

200µs/DIV

50mV/DIV

Linear Technology Magazine • February 1997

NEW DEVICE CAMEOS

New Device Cameos

LT1118 Adjustable

Sink/Source Regulator

The LT1118 adjustable regulator joins

the LT1118-2.5, LT1118-2.85 and

LT1118-5 ﬁxed-output versions of the

sink/source regulator. The output

voltage is programmed by customer

selected feedback resistors to provide

any output voltage between 2.0V and

– 1.0V, for V

up to 15V. The

regulator maintains regulation for

load currents between 800mA sourced

into the load to 400mA sunk from the

load.

The adjustable regulator is espe-

cially useful for data bus terminations,

similar to the use of the 2.85V ﬁxed

version in SCSI applications.

A fused-lead SO-8 package pro-

vides a low thermal resistance to PC

board traces. In most applications,

no further heat sinking is required for

the regulator.

A Shutdown pin places the output

in a high impedance state, effectively

disconnecting the load from the regu-

lator. Fault tolerance is provided by

current limiting of both sunk and

sourced output current in addition to

on-chip thermal shutdown.

LT1141A 3-Driver,

5-Receiver RS232 Transceiver

Meets IEC-1000-4-2 ESD

Protection Standards

The LT1141A 3-driver, 5-receiver

RS232 transceiver is the latest Linear

Technology RS232 transceiver to be

upgraded to pass the IEC-1000-4-2

level 4 ESD test. The chip is internally

protected against ±15kV air-gap or

±8kV contact-mode discharges. The

IEC-1000-4-2 test, formerly known

as IEC-801-2, must be passed by all

equipment sold in Europe. The chip

also passes up to ±15kV ESD as tested

by MIL-STD-883, Method 3015. The

on-chip protection of the LT1141A

frees the user from the cost and board

area required by external transient-

suppression devices that are usually

required to successfully meet the IEC

ESD protection requirements.

The enhanced ESD protection has

been achieved without compromising

the electrical performance of the de-

vice. Present LT1141A users will see

no change in electrical performance.

The 3-driver, 5-receiver device retains

all of the electrical performance fea-

tures which make it popular. The

enhanced ESD protection devices do

not degrade operation at rates of up to

120kbaud with full 2500pF loads or

to 250kbaud with 1000pF loads.

The LT1141A is available in 24-pin

DIP, SO and SSOP packages.

LT1180A/LT1181A

2-Driver, 2-Receiver

RS232 Transceivers

Meet IEC-1000-4-2

ESD Protection Standards

The LT1180A and LT1181A, Linear

Technology’s popular 2-driver, 2-re-

ceiver RS232 transceivers, have been

upgraded to meet the IEC-1000-4-2

level 4 ESD standards. The circuits

are internally protected against ±15kV

air-gap or ±8kV contact-mode dis-

charges. The IEC-1000-4-2 test,

formerly known as IEC-801-2, must

be passed by all equipment sold in

Europe. The chip also passes up to

±15kV ESD as tested by MIL-STD-

883 Method 3015. The on-chip

protection of the LT1181A frees the

user from the cost and board area

required by external transient-sup-

pression devices that are usually

required to successfully meet the IEC

ESD protection requirements.

Enhanced ESD protection has been

achieved without compromising the

electrical performance of the devices.

Present LT1180A/LT1181A users will

see no change in electrical perfor-

mance. The devices retain all of the

electrical performance features that

make them popular. The enhanced

ESD protection devices do not de-

grade operation at rates of up to

120kbaud with full 2500pF loads or

to 250kbaud with 1000pF loads.

The LT1181A is available in 16-pin

DIP, SW and SO packages. The

LT1180A, which includes a shutdown

control, is available in 18-pin DIP and

SW packages.

LTC1438-ADJ:

High Efﬁciency, Dual,

Adjustable Output Voltage

Synchronous Switching

Power Supply Controller for

Portable Applications

The LTC1438-ADJ is a dual, adjust-

able output voltage, synchronous

step-down switching regulator con-

troller that drives external N-channel

power MOSFETs in a ﬁxed frequency

architecture.

The LTC1438-ADJ differs from the

LTC1438 in its ability to set both

controller output voltages using ex-

ternal resistive dividers, for output

voltages as low as 1.2V. External-

feedback voltage setting provides

remote sensing of each output volt-

age at the load—often required in

higher current applications.

A 1% voltage reference and load

regulation of ±0.8% are guaranteed

over the full temperature range, elimi-

nating output voltage adjustment in

most applications. The operating cur-

rent levels are user-programmable

via external current sense resistors.

Wide input supply range allows op-

eration from 3.5V to 30V (36V

maximum).

A secondary feedback input can be

used in conjunction with a ﬂyback

winding on the ﬁrst controller to gen-

erate a third output voltage that can

supply power regardless of the load

Linear Technology Magazine • February 1997

NEW DEVICE CAMEOS

on the ﬁrst controller’s primary wind-

ing. This feedback forces continuous

operation on the ﬁrst controller using

a simple voltage mode loop. The input

can also be used as a logic input to

force continuous operation, thereby

suppressing Burst Mode operation

on the ﬁrst controller.

A hysteretic comparator, which has

its inverting input tied to the internal

1.19V reference, is included. The out-

put is an open-drain type and can be

pulled up to any available supply of

up to 10V. A power-on reset timer

(POR) generates a logic-low output

signal during controller start-up,

which persists for 65,536 clock cycles

after the output reaches 7.5% of the

regulated output voltage.

The part is available in a 28-lead

plastic SSOP package.

LTC1605 Single 5V 100ksps

Sampling 16-Bit ADC

The LTC1605 is a complete 100ksps

sampling 16-bit ADC that operates

on a single 5V supply and typically

dissipates 55mW. Its input range is

an industrial standard ±10V. This

gives the user a large input LSB size

of 305µV, which can help ease noise

requirements for the input condition-

ing circuitry. The input signal is

captured by an onboard sample-and-

hold and digitized by a differential,

switched capacitor SAR ADC. The

LTC1605 achieves 16-bit performance

without the autocalibration overhead

required with other types of ADCs.

Maximum DC specifications include

±2.0LSB INL and 16 bits with no

missing codes guaranteed over tem-

perature. The part has an internal

2.5V bandgap reference. An external

reference can also be used.

The ADC has a microprocessor-

compatible 16-bit parallel output port

that can provide data as a 16-bit word

or as two bytes. The LTC1605 is easily

connected to FIFOs, DSPs and micro-

processors using its convert start

input and data ready signal (BUSY).

LT1039A RS232 Transceiver

Meets IEC-1000-4-2 ESD

Protection Standards

The popular LT1039 3-driver 3-

receiver RS232 transceiver has been

upgraded to pass the IEC-1000-4-2

level 4 ESD test. The chip is internally

For further infor mation on any

of the devices mentioned in this

issue of Linear Technology, use

the reader service car d or call

the LTC literatur e service

number:

1-800-4-LINEAR

Ask for the pertinent data sheets

and Application Notes.

protected against ±15kV air-gap or

±8kV contact-mode discharges. The

IEC-1000-4-2 test, formerly known

as IEC-801-2, must be passed by all

equipment sold in Europe. LT1039A’s

on-chip protection eliminates the cost

and board area required by external

transient-suppression devices, which

are usually needed to successfully

meet the IEC ESD protection

requirements.

The enhanced ESD protection is

achieved without compromising the

electrical performance of the device.

Present LT1039 users will see no

change in electrical performance. The

3-driver, 3-receiver device retains all

of the electrical performance features

that make it popular. Operation to

120kbaud with full 2500pF loads,

and up to 250kbaud with 1000pF

loads, is not degraded by the en-

hanced ESD protection devices.

The LT1039A is available in 16-pin

and 18-pin DIP and SO packages.

The 18-pin versions include ON-OFF

control and a BIAS pin to power one

receiver when the device is shut

down.



for

the latest information

on LTC products, 

visit

www.linear-tech.com

Linear Technology Magazine • February 1997

Applications on Disk

Noise Disk — This IBM-PC (or compatible) program

allows the user to calculate circuit noise using LTC op

amps, determine the best LTC op amp for a low noise

application, display the noise data for LTC op amps,

calculate resistor noise and calculate noise using specs

for any op amp. Available at no charge.

SPICE Macromodel Disk — This IBM-PC (or compat-

ible) high density diskette contains the library of LTC

op amp SPICE macromodels. The models can be used

with any version of SPICE for general analog circuit

simulations. The diskette also contains working circuit

examples using the models and a demonstration copy

of PSPICE™ by MicroSim. Available at no charge.

SwitcherCAD™ — The SwitcherCAD program is a pow-

erful PC software tool that aids in the design and

optimization of switching regulators. The program can

cut days off the design cycle by selecting topologies,

calculating operating points and specifying compo-

nent values and manufacturer’s part numbers. 144

page manual included. $20.00

SwitcherCAD supports the following parts: LT1070

series: LT1070, LT1071, LT1072, LT1074 and LT1076.

LT1082. LT1170 series: LT1170, LT1171, LT1172 and

LT1176. It also supports: LT1268, LT1269 and LT1507.

LT1270 series: LT1270 and LT1271. LT1371 series:

LT1371, LT1372, LT1373, LT1375, LT1376 and

LT1377.

Micropower SwitcherCAD™ — The MicropowerSCAD

program is a powerful tool for designing DC/DC con-

verters based on Linear Technology’s micropower

switching regulator ICs. Given basic design param-

eters, MicropowerSCAD selects a circuit topology and

offers you a selection of appropriate Linear Technology

switching regulator ICs. MicropowerSCAD also per-

forms circuit simulations to select the other components

which surround the DC/DC converter. In the case of a

battery supply, MicropowerSCAD can perform a bat-

tery life simulation. 44 page manual included. $20.00

MicropowerSCAD supports the following LTC micro-

power DC/DC converters: LT1073, LT1107, LT1108,

LT1109, LT1109A, LT1110, LT1111, LT1173, LTC1174,

LT1300, LT1301 and LT1303.

Technical Books

1990 Linear Databook, Vol I —This 1440 page collec-

tion of data sheets covers op amps, voltage regulators,

references, comparators, ﬁlters, PWMs, data conver-

sion and interface products (bipolar and CMOS), in

both commercial and military grades. The catalog

features well over 300 devices. $10.00

1992 Linear Databook, Vol II — This 1248 page supple-

ment to the 1990 Linear Databook is a collection of all

products introduced in 1991 and 1992. The catalog

contains full data sheets for over 140 devices. The

1992 Linear Databook, Vol II is a companion to the

1990 Linear Databook, which should not be discarded.

$10.00

1994 Linear Databook, Vol III —This 1826 page

supplement to the 1990 and 1992 Linear Databooks is

a collection of all products introduced since 1992. A

total of 152 product data sheets are included with

updated selection guides. The 1994 Linear Databook

Vol III is a companion to the 1990 and 1992 Linear

Databooks, which should not be discarded. $10.00

1995 Linear Databook, Vol IV —This 1152 page

supplement to the 1990, 1992 and 1994 Linear Da-

tabooks is a collection of all products introduced since

1994. A total of 80 product data sheets are included

with updated selection guides. The 1995 Linear Data-

book Vol IV is a companion to the 1990, 1992 and 1994

Linear Databooks, which should not be discarded.

$10.00

1996 Linear Databook, Vol V —This 1152 page supple-

ment to the 1990, 1992, 1994 and 1995 Linear

Databooks is a collection of all products introduced

since 1995. A total of 65 product data sheets are

included with updated selection guides. The 1996

Linear Databook Vol V is a companion to the 1990,

1992, 1994 and 1995 Linear Databooks, which should

not be discarded. $10.00

1990 Linear Applications Handbook, Volume I —

928 pages full of application ideas covered in depth by

40 Application Notes and 33 Design Notes. This cata-

log covers a broad range of “real world” linear circuitry.

In addition to detailed, systems-oriented circuits, this

handbook contains broad tutorial content together

with liberal use of schematics and scope photography.

A special feature in this edition includes a 22-page

section on SPICE macromodels. $20.00

1993 Linear Applications Handbook, Volume II —

Continues the stream of “real world” linear circuitry

initiated by the 1990 Handbook. Similar in scope to the

1990 edition, the new book covers Application Notes

40 through 54 and Design Notes 33 through 69.

References and articles from non-LTC publications

that we have found useful are also included. $20.00

1997 Linear Applications Handbook, Volume III —

This 976 page handbook maintains the practical outlook

and tutorial nature of previous efforts, while broaden-

ing topic selection. This new book includes Application

Notes 55 through 69 and Design Notes 70 through

144. Subjects include switching regulators, measure-

ment and control circuits, filters, video designs,

interface, data converters, power products, battery

chargers and CCFL inverters. An extensive subject

index references circuits in LTC data sheets, design

notes, application notes and

Linear Technology

maga-

zines. $20.00

Interface Product Handbook — This 424 page hand-

book features LTC’s complete line of line driver and

receiver products for RS232, RS485, RS423, RS422,

V.35 and AppleTalk

applications. Linear’s particular

expertise in this area involves low power consumption,

high numbers of drivers and receivers in one package,

mixed RS232 and RS485 devices, 10kV ESD protec-

tion of RS232 devices and surface mount packages.

Available at no charge

Power Solutions Brochure — This 84 page collection

of circuits contains real-life solutions for common

power supply design problems. There are over 88

circuits, including descriptions, graphs and perfor-

mance speciﬁcations. Topics covered include battery

chargers, PCMCIA power management, microproces-

sor power supplies, portable equipment power supplies,

micropower DC/DC, step-up and step-down switching

regulators, off-line switching regulators, linear regula-

tors and switched capacitor conversion.

Available at no charge.

High Speed Ampliﬁer Solutions Brochure —

This 72 page collection of circuits contains real-life

solutions for problems that require high speed

ampliﬁers. There are 82 circuits including descrip-

tions, graphs and performance speciﬁcations. Topics

covered include basic ampliﬁers, video-related appli-

cations circuits, instrumentation, DAC and photodiode

ampliﬁers, ﬁlters, variable gain, oscillators and current

sources and other unusual application circuits.

Available at no charge

Data Conversion Solutions Brochure — This 52 page

collection of data conversion circuits, products and

selection guides serves as excellent reference for the

data acquisition system designer. Over 60 products

are showcased, solving problems in low power, small

size and high performance data conversion applica-

tions—with performance graphs and speciﬁcations.

Topics covered include ADCs, DACs, voltage refer-

ences and analog multiplexers. A complete glossary

deﬁnes data conversion speciﬁcations; a list of se-

lected application and design notes is also included.

Available at no charge

Telecommunications Solutions Brochure — This 72

page collection of circuits, new products and selection

guides covers a wide variety of products targeted for

the telecommunications industry. Circuits solving real

life problems are shown for central office switching,

cellular phone, base station and other telecom applica-

tions. New products introduced include high speed

amplifiers, A/D converters, power products, interface

transceivers and filters. Reference material includes a

telecommunications glossary, serial interface stan-

dards, protocol information and a complete list of key

application notes and design notes.

Available at no charge

continued on page 40

DESIGN TOOLS

Acrobat is a trademark of Adobe Systems, Inc. AppleTalk

is a registered trademark of Apple Computer, Inc. PSPICE™

is a trademark of MicroSim Corp.

Information furnished by Linear Technology Corporation

is believed to be accurate and reliable. However, Linear

Technology makes no representation that the circuits

described herein will not infringe on existing patent rights.

Linear Technology Magazine • February 1997

LINEAR TECHNOLOGY CORPORATION

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Japan

Phone: 81-3-3267-7891

FAX: 81-3-3267-8510

KOREA

Linear Technology Korea Co., Ltd

Namsong Building, #403

Itaewon-Dong 260-199

Yongsan-Ku, Seoul 140-200

Korea

Phone: 82-2-792-1617

FAX: 82-2-792-1619

SINGAPORE

Linear Technology Pte. Ltd.

507 Yishun Industrial Park A

Singapore 2776

Phone: 65-753-2692

FAX: 65-754-4113

SWEDEN

Linear Technology AB

Sollentunavägen 63

S-191 40 Sollentuna

Sweden

Phone: 46-8-623-1600

FAX: 46-8-623-1650

TAIWAN

Linear Technology Corporation

Rm. 602, No. 46, Sec. 2

Chung Shan N. Rd.

Taipei, Taiwan, R.O.C.

Phone: 886-2-521-7575

FAX: 886-2-562-2285

UNITED KINGDOM

Linear Technology (UK) Ltd.

The Coliseum, Riverside Way

Camberley, Surrey GU15 3YL

United Kingdom

Phone: 44-1276-677676

FAX: 44-1276-64851

World Headquarters

Linear Technology Corporation

1630 McCarthy Boulevard

Milpitas, CA 95035-7417

Phone: (408) 432-1900

FAX: (408) 434-0507

U.S. Area

Sales Ofﬁces

NORTHEAST REGION

Linear Technology Corporation

3220 Tillman Drive, Suite 120

Bensalem, PA 19020

Phone: (215) 638-9667

FAX: (215) 638-9764

Linear Technology Corporation

266 Lowell St., Suite B-8

Wilmington, MA 01887

Phone: (508) 658-3881

FAX: (508) 658-2701

NORTHWEST REGION

Linear Technology Corporation

1900 McCarthy Blvd., Suite 205

Milpitas, CA 95035

Phone: (408) 428-2050

FAX: (408) 432-6331

SOUTHEAST REGION

Linear Technology Corporation

17000 Dallas Parkway

Suite 219

Dallas, TX 75248

Phone: (214) 733-3071

FAX: (214) 380-5138

Linear Technology Corporation

5510 Six Forks Road

Suite 102

Raleigh, NC 27609

Phone: (919) 870-5106

FAX: (919) 870-8831

CENTRAL REGION

Linear Technology Corporation

Chesapeake Square

229 Mitchell Court, Suite A-25

Addison, IL 60101

Phone: (708) 620-6910

FAX: (708) 620-6977

SOUTHWEST REGION

Linear Technology Corporation

21243 Ventura Blvd.

Suite 227

Woodland Hills, CA 91364

Phone: (818) 703-0835

FAX: (818) 703-0517

Linear Technology Corporation

15375 Barranca Parkway

Suite A-211

Irvine, CA 92718

Phone: (714) 453-4650

FAX: (714) 453-4765

CD-ROM

LinearView — LinearView™ CD-ROM is Linear

Technology’s interactive PC-based CD-ROM. It allows

you to instantly access thousands of pages of product

and applications information, covering Linear

Technology’s complete line of high performance ana-

log products, with easy-to-use search tools.

The LinearView CD-ROM includes the complete prod-

uct speciﬁcations from Linear Technology’s Databook

library (Volumes I–IV) and the complete Applications

Handbook collection (Volumes I and II). Our extensive

collection of Design Notes and the complete collection

Linear Technology

magazine are also included.

A powerful search engine built into the LinearView CD-

ROM enables you to select parts by various criteria,

such as device parameters, keywords or part numbers.

All product categories are represented: data conver-

sion, references, ampliﬁers, power products, ﬁlters

and interface circuits. Up-to-date versions of Linear

Technology’s software design tools, SwitcherCAD,

FilterCAD, Noise Disk and Spice Macromodel library,

are also included. Everything you need to know about

Linear Technology’s products and applications is readily

accessible via LinearView. Available at no charge.

World Wide Web Site

Linear Technology Corporation’s customers can now

quickly and conveniently find and retrieve the latest

technical information covering the Company’s prod-

ucts on LTC’s new internet web site. Located at

www.linear-tech.com, this site allows anyone with

internet access and a web browser to search through

all of LTC’s technical publications, including data sheets,

application notes, design notes,

Linear Technology

magazine issues and other LTC publications, to find

information on LTC parts and applications circuits.

Other areas within the site include help, news and

information about Linear Technology and its sales

offices. The site includes a map that eases navigation

through the different information areas.

Other web sites usually require the visitor to download

large document files to see if they contain the desired

information. This is cumbersome and inconvenient. To

save you time and ensure that you receive the correct

information the first time, the first page of each data

sheet, application note and

Linear Technology

maga-

zine is recreated in a fast, download-friendly format.

This allows you to determine whether the document is

what you need, before downloading the entire file.

The site is searchable. Among the possible search

criteria are part numbers, function, topics and applica-

tions. The search is performed on a user-defined

combination of data sheets, application notes, design

notes and

Linear Technology

magazine articles. Any

data sheet, application note, design note or magazine

article can be downloaded or faxed back. (Files are

downloaded in Adobe Acrobat™ PDF format; you will

need a copy of Acrobat Reader to view or print them.

The site includes a link from which you can download

this program.)

DESIGN TOOLS, continued from page 39

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