LM3477MM/NOPB - NATIONAL SEMICONDUCTOR

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

LM3477 High Efficiency High-Side N-Channel Controller for Switching Regulator

Check for Samples: LM3477

1FEATURES DESCRIPTION

The LM3477/A is a high-side N-channel MOSFET

2• 500kHz Switching Frequency switching regulator controller. It can be used in

• Adjustable Current Limit topologies requiring a high side MOSFET such as

• 1.5% Reference buck, inverting (buck-boost) and zeta regulators. The

LM3477/A's internal push pull driver allows

• Thermal Shutdown compatibility with a wide range of MOSFETs. This,

• Frequency Compensation Optimized with a the wide input voltage range, use of discrete power

Single Capacitor and Resistor components and adjustable current limit allows the

• Internal Softstart LM3477/A to be optimized for a wide variety of

applications.

• Current Mode Operation

• Undervoltage Lockout with Hysteresis The LM3477/A uses a high switching frequency of

500kHz to reduce the overall solution size. Current-

• 8-lead (VSSOP-8) Package mode control requires only a single resistor and

capacitor for frequency compensation. The current

APPLICATIONS mode architecture also yields superior line and load

• Local Voltage Regulation regulation and cycle-by-cycle current limiting. A 5µA

shutdown state can be used for power savings and

• Distributed Power for power supply sequencing. Other features include

• Notebook and Palmtop Computers internal soft-start and output over voltage protection.

• Internet Appliances The internal soft-start reduces inrush current. Over

voltage protection is a safety feature to ensure that

• Printers and Office Automation the output voltage stays within regulation.

• Battery operated Devices The LM3477A is similar to the LM3477. The primary

• Cable Modems difference between the two is the point at which the

• Battery Chargers device transitions into hysteretic mode. The hysteretic

threshold of the LM3477A is one-third of the LM3477.

Hysteretic Threshold(1)

LM3477 ≊36% of programmed current limit

LM3477A ≊12% of programmed current limit

(1) See Hysteretic Threshold and PROGRAMMING THE

CURRENT LIMIT/HYSTERETIC THRESHOLD for more

information.

1Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of

Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.

2All trademarks are the property of their respective owners.

Products conform to specifications per the terms of the Texas

Instruments standard warranty. Production processing does not

necessarily include testing of all parameters.

ISEN

GND

VIN

LM3477

COMP/SD

CBYP

0.1PF

CBOOT

0.1PF

CSN

0.1PF

DR FB

GND

VIN

ISEN LM3477

*RFB1 = RFB2* (Vout - 1.26)/1.26

VIN

4.5V - 5.5V

CIN

120PF/20V

RSN

0.02:

FDC653N

30BG100

3.3PHVOUT

2.5V, 3A

RFB1

20.5k

RFB2

20.0k

510 CC1

47nF

COUT

47PF

ceramic

COMP/SD

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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Typical Application Circuit

Figure 1. Typical High Efficiency Step-Down (Buck) Converter

Connection Diagram

Figure 2. 8 Lead (VSSOP-8 Package)

PIN DESCRIPTION

Pin Name Pin Number Description

ISEN 1 Current sense input pin. Voltage generated across an external sense resistor is fed into this

pin.

COMP/SD 2 Compensation pin. A resistor-capacitor combination connected to this pin provides

compensation for the control loop. Pull this pin below 0.65V to shutdown.

FB 3 Feedback pin. The output voltage should be adjusted using a resistor divider to provide

1.270V at this pin.

GND 4 Ground pin.

SW 5 Switch Node. Source of the external MOSFET is connected to this node.

DR 6 Drive pin. The gate of the external MOSFET should be connected to this pin.

CB 7 Boot-strap pin. A capacitor must be connected between this pin and SW pin (pin 5) for proper

operation. The voltage developed across this capacitor provides the gate drive for the external

MOSFET.

VIN 8 Power Supply Input pin.

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

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Absolute Maximum Ratings (1)(2)

Input Voltage 36V

Peak Driver Output Current (<10µs) 1.0A

CB Pin Voltage (3) 43V

ISEN Pin Voltage 500mV

Power Dissipation Internally Limited

Storage Temperature Range −65°C to +150°C

Junction Temperature +150°C

ESD Susceptibilty (4)

Human Body Model 2kV

Machine Model 200V

Lead Temperature for VSSOP Package

Vapor Phase (60 sec.) 215°C

Infared (15 sec.) 220°C

(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which

operation of the device is intended to be functional. For ensured specifications and test conditions, see the Electrical Characteristics.

(2) If Military/Aerospace specified devices are required, please contact the TI Sales Office/Distributors for availability and specifications.

(3) The CB pin must not be higher than 8V above the VSW.

(4) The human body model is a 100 pF capacitor discharged through a 1.5kΩresistor into each pin. The machine model is 200 pF capacitor

discharged directly into each pin.

Operating Ratings (1)

Supply Voltage 2.97V ≤VIN ≤35V

Junction Temperature Range −40°C ≤TJ≤+125°C

(1) Absolute Maximum Ratings are limits beyond which damage to the device may occur. Operating Ratings are conditions under which

operation of the device is intended to be functional. For ensured specifications and test conditions, see the Electrical Characteristics.

Electrical Characteristics(1)

Specifications in Standard type face are for TJ= 25°C, and in bold type face apply over the full Operating Temperature

Range. Unless otherwise specified, VIN = 12V.

Symbol Parameter Conditions Typical Limit Units

VFB Feedback Voltage VCOMP = 1.4V, 1.270 V

2.97V ≤VIN ≤36V 1.260/1.252 V(min)

1.288/1.290 V(max)

ΔVLINE Feedback Voltage Line Regulation 2.97V ≤VIN ≤36V 0.001 %/V

ΔVLOAD Output Voltage Load Regulation ±0.5 %/V (max)

VUVLO Input Undervoltage Lock-out 2.87 V

2.97 V(max)

VUV(HYS) Input Undervoltage Lock-out Hysteresis 180 mV

130 mV (min)

225 mV (max)

FSW Switching Frequency 500 kHz

435 kHz(min)

575 kHz(max)

RDS1 (ON) Driver Switch On Resistance (top) IDR = 0.2A, VIN= 5V 7 Ω

RDS2 (ON) Driver Switch On Resistance (bottom) IDR = 0.2A 4 Ω

(VCB−VSW)max Maximum Boot Voltage VIN < 7.2V VIN V

VIN ≥7.2V 7.2

Dmax Maximum Duty Cycle 93 %

88 %(min)

Tmin (on) Minimum On Time 330 nsec

230 nsec(min)

495 nsec(max)

(1) All limits are ensured at room temperature (standard type face) and at temperature extremes (bold type face). All room temperature

limits are 100% tested. All limits at temperature extremes are ensured via correlation using standard Statistical Quality Control (SQC)

methods. All limits are used to calculate Average Outgoing Quality Level (AOQL).

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Electrical Characteristics(1) (continued)

Specifications in Standard type face are for TJ= 25°C, and in bold type face apply over the full Operating Temperature

Range. Unless otherwise specified, VIN = 12V.

Symbol Parameter Conditions Typical Limit Units

ISUPPLY Supply Current (switching) (2) 2.0 mA

3.0 mA (max)

IQQuiescent Current in Shutdown Mode (3), VIN = 5V 5 µA

8µA (max)

VCL(O) Current Limit Voltage at 0% Duty Cycle LM3477 155 mV

130/125 mV (min)

185/190 mV (max)

LM3477A 165 mV

140/135 mV (min)

195/200 mV (max)

VCL(100) Current Limit Voltage at 100% Duty LM3477 74 mV

Cycle 50/43 mV (min)

98/98 mV (max)

LM3477A 65 mV

41/25 mV (min)

89/98 mV (max)

VSC Short-Circuit Current Limit Sense VIN = 5V, LM3477 350 mV

Voltage 270 mV (min)

420 mV (max)

VIN = 5V, LM3477A 310 mV

260 mV (min)

380 mV (max)

VSL Internal Compensation Ramp Voltage VIN = 5V, LM3477 83 mV

Height VIN = 5V, LM3477A 103

VOVP Output Over-voltage Protection (with VCOMP = 1.4V 50 mV

32/25

respect to feedback voltage) (4) mV(min)

78/85 mV(max)

VOVP(HYS) Output Over-Voltage Protection VCOMP = 1.4V 60 mV

Hysteresis(4) 20 mV(min)

110 mV(max)

Gm Error Amplifier Transconductance VCOMP = 1.4V 750 µmho

IEAO = 100µA (Source/Sink) 600/365 µmho (min)

1000/1265 µmho (max)

AVOL Error Amplifier Voltage Gain VCOMP = 1.4V 38 V/V

IEAO = 100µA (Source/Sink) 30 V/V (min)

42 V/V (max)

IEAO Error Amplifier Output Current (Source/ Source, VCOMP = 1.4V, VFB = 100 µA

Sink) 0V 75/50 µA (min)

130/160 µA (max)

Sink, VCOMP = 1.4V, VFB = 1.4V −140 µA

−110/−95 µA (min)

−170/−180 µA (max)

VEAO Error Amplifier Output Voltage Swing Upper Limit 2.2 V

VFB = 0V 2.0 V(min)

COMP Pin = Floating 2.35 V(max)

Lower Limit 0.75 V

VFB = 1.4V 0.5 V(min)

0.95 V(max)

TSS Internal Soft-Start Delay VFB = 1.2V, VCOMP = Floating 5 msec

TrDrive Pin Rise Time CGS = 3000pF, VDR = 0 to 3V 25 ns

TfDrive Pin Fall Time CGS = 3000pF, VDR = 0 to 3V 25 ns

(2) For this test, the COMP/SD pin must be left floating.

(3) For this test, the COMP/SD pin must be pulled low.

(4) The over-voltage protection is specified with respect to the feedback voltage. This is because the over-voltage protection tracks the

feedback voltage. The overvoltage protection threshold is given by adding the feedback voltage, VFB to the over-voltage protection

specification.

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Electrical Characteristics(1) (continued)

Specifications in Standard type face are for TJ= 25°C, and in bold type face apply over the full Operating Temperature

Range. Unless otherwise specified, VIN = 12V.

Symbol Parameter Conditions Typical Limit Units

VSD Shutdown Threshold (5) Output = High 1.15 V

1.35 V (max)

Output = Low 0.65 V

0.3 V (min)

ISD Shutdown Pin Current VSD = 5V −1µA

VSD = 0V +1

TSD Thermal Shutdown 165 °C

TSH Thermal Shutdown Hysteresis 10 °C

θJA Thermal Resistance MM Package 200 °C/W

(5) The COMP/SD pin should be pulled to ground to turn the regulator off. The voltage on the COMP/SD pin must be below the limit for

Output = Low to keep the regulator off.

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VIN (Volts)

VCB-VSW (Volts)

0 5 10 15 20 25 30 35

1.5

1.75

2.25

2.5

2.75

0510 15 20 25 30 35 40

ISUPPLY (mA)

VIN (Volts)

-40oC

25oC

125oC

200

400

600

-40 040 80 120

TemperatureoC

Frequency (kHz)

0510 15 20 25 30 35 40

VIN (Volts)

IQ (PA)

-40oC

85oC

25oC

1.5

1.75

2.25

2.5

2.75

0510 15 20 25 30 35 40

VIN (Volts)

ISUPPLY (mA)

-40oC

25oC

125oC

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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Typical Performance Characteristics

Unless otherwise specified, VIN = 12V, TJ= 25°C.

IQ(Shutdown) vs Temperature & Supply Voltage ISupply vs Temperature & Supply Voltage (Non-Switching)

Figure 3. Figure 4.

ISupply vs Temperature & Supply Voltage (Switching) Frequency vs Temperature

Figure 5. Figure 6.

VCB−VSW vs Supply Voltage COMP Pin Voltage vs Load Current

Figure 7. Figure 8.

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120

130

140

150

-40 040 80 120

ICOMP (PA)

Temperature (RC)

100

0.01 0.1 1 10

LM3477

LM3477A

LOAD (A)

EFFICIENCY (%)

LM3477

LM3477A

100

0.01 0.1 1 10

LOAD (A)

EFFICIENCY (%)

100

0.01 0.1 1 10

LM3477

LM3477A

LOAD (A)

EFFICIENCY (%)

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

Typical Performance Characteristics (continued)

Unless otherwise specified, VIN = 12V, TJ= 25°C.

Efficiency vs Load Current (VIN = 24V, VOUT = 12V) Efficiency vs Load Current (VIN = 5V, OUT = 3.3V)

Figure 9. Figure 10.

Efficiency vs Load Current (VIN = 12V, VOUT = 3.3V) Error Amplifier Gain

Figure 11. Figure 12.

Error Amplifier Phase Shift COMP Pin Source Current vs Temperature

Figure 13. Figure 14.

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CURRENT LIMIT VOLTAGE (mV)

DUTY CYCLE (%)

0 10 20 30 40 50 60 70 80 90

100

200

300

400

100

LM3477A

LM3477

0.2

0.4

0.6

0.8

1.2

1.4

-40 040 80 120

SHUTDOWN THRESHOLD (V)

TEMPERATURE (°C)

Output = Low

Output = High

RSL (:)

150

250

100

200

02k 3k 4k 5k

'VSL(mV)

100

150

200

250

300

350

400

450

TEMPERATURE (°C)

SHORT CIRCUIT SENSE VOLTAGE (mV)

LM3477A

LM3477

-50 0 50 100 150

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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Typical Performance Characteristics (continued)

Unless otherwise specified, VIN = 12V, TJ= 25°C. Slope Compensation Ramp vs Slope Compensation

Short Circuit vs Temperature Resistor

Figure 15. Figure 16.

Shutdown Threshold Hysteresis vs Temperature Current Sense Voltage vs Duty Cycle

Figure 17. Figure 18.

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Under-

Voltage

Lockout

7.2V

Bias Voltages

Switch

Driver

Cboot Voltage

Detect

Switch

Logic

Thermal

Shutdown

1.26V

Reference Soft-Start

Shutdown

Detect

Slope

Compensation

500kHz

Oscillator

Set / Blankout

One-Shot

Short-Circuit

Detect

PWM

Comparator

Error

Amplifier

Over-voltage

Detect

VIN

GND

COMP

ISEN

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

Functional Block Diagram

Product Folder Links: LM3477

Hysteretic

VCL0

VSC

LM3477A

VSC

LM3477

VHYS

LM3477

(32mV)

VHYS

LM3477A

(11mV)

D(MIN) D(MAX)

Hysteretic

PWM

LM3477A

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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FUNCTIONAL DESCRIPTION

GENERAL DESCRIPTION

The LM3477/A is a switching regulator controller for topologies incorporating a high side switch. The most

common of these topologies is the step-down, or buck, converter. Other topologies such as the inverting (buck-

boost) and inverse SEPIC (zeta) converters can be realized. This datasheet will focus on buck converter

applications.

The LM3477/A employs current mode control architecture. Among the many benefits of this architecture are

superior line and load regulation, cycle-by-cycle current limiting, and simple loop compensation. The LM3477/A

features a patented adjustable slope compensation scheme to enable flexible inductor selection. The LM3477/A

has a combination of features that allow its use in a wide variety of applications. The input voltage can range

from 2.97V to 35V, with the output voltage being positive or negative depending on the topology. The current

limit can be scaled to safely drive a wide range of loads. An internal soft-start is provided to limit initial in-rush

current. Output over voltage and input under voltage protection ensure safe operation of the LM3477/A.

REGIONS OF OPERATION

Pulse width modulation (PWM) is the normal mode of operation. In PWM, the output voltage is well regulated

and has a ripple frequency equal to the switching frequency (500kHz). In low load conditions, the part operates in

hysteretic mode. In this mode, the output voltage is regulated between a high and low value that results in a

higher ripple magnitude and lower ripple frequency than in PWM mode (see OVER VOLTAGE PROTECTION).

Figure 19. Operating Regions of the LM3477/A

The important differences between the LM3477 and the LM3477A are summarized in Figure 19. The voltages in

Figure 19 can be referred to the switch current by dividing through by RSN. The LM3477A has a lower hysteretic

threshold voltage VHYS, and thus will operate in PWM mode for a larger load range than the LM3477. Typically,

VHYS = 32mV for the LM3477, while VHYS = 11mV for the LM3477A. The difference in area between the shaded

regions give a graphical representation of this. The lightly shaded region is the extra PWM operating area gained

by using the LM3477A. Thus the benefits of operating in PWM mode such as a well regulated output voltage with

low noise ripple are extended to a larger load range when the LM3477A is used. While less significant, the other

noteworthy difference between the two parts is in the short circuit current limit VSC.

VSC is a ceiling limit for the peak sense voltage VSNpk (see SHORT CIRCUIT PROTECTION). VSC is lower in the

LM3477A than in the LM3477 (see the Electrical Characteristics for limits).

OVER VOLTAGE PROTECTION

The LM3477/A has over voltage protection (OVP) for the output voltage. OVP is sensed at and is in respect to

the feedback pin (pin 3). If at anytime the voltage at the feedback pin rises to VFB + VOVP, OVP is triggered. See

Electrical Characteristics for limits on VFB and VOVP.

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VOVP (HIGH)

VOVP (LOW)

VFB

Hysteretic Mode:

PWM Mode:

- Light load current

- D < DMIN

- Transient response

overshoot too high

- Normal

operation

Feedback Voltage

VOVP(HYS)

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

OVP will cause the drive pin to go low, forcing the power MOSFET off. With the MOSFET off, the output voltage

will drop. The LM3477/A will begin switching again when the feedback voltage reaches VFB + (VOVP - VOVP(HYS)).

See Electrical Characteristics for limits on VOVP(HYS).

OVP can be triggered by any event that causes the output voltage to rise out of regulation. There are several

common circumstances in which this can happen, and it is beneficial for a designer to be aware of these for

debugging purposes, since the mode of operation changes from the normal Pulse Width Modulation (PWM)

mode to the hysteretic mode. In the hysteretic mode the output voltage is regulated between a high and low

value that results in a higher ripple magnitude and lower ripple frequency than in the PWM mode, see Figure 20.

See different Ripple Components in PWM and Hysteretic Modes.

Figure 20. The Feedback Voltage is related to the Output Voltage

If the load current becomes too low, the LM3477/A will increase the duty cycle, causing the voltage to rise and

trigger the OVP. The reasons for this involve the way the LM3477/A regulates the output voltage, using a control

waveform at the pulse width modulator. This control waveform has upper and lower bounds.

Another way OVP can be tripped is if the input voltage rises higher than the LM3477/A is able to regulate in

pulse width modulation (PWM) mode. The output voltage is related to the input voltage by the duty cycle as:

VOUT = VIN*D. The LM3477/A has a minimum duty cycle of 16.5% (typical), due to the blank-out timing, TMIN. If

the input voltage increases such that the duty cycle wants to be less than DMIN, the duty cycle will hold at DMIN

and the output voltage will increase with the input voltage until it trips OVP.

It is useful to plot the operational boundaries in order to illustrate the point at which the device switches into

hysteretic mode. In Figure 19, the limits shown are with respect to the peak voltage across the sense resistor

RSN, (VSNpk); they can be referred to the peak inductor current by dividing through by RSN. VSNpk is bound to the

shaded regions. In normal circumstances VSNpk is required to be in the shaded region, and the LM3477/A will

operate in the PWM mode. If operating conditions are chosen such that VSNpk would not normally fall in the

shaded regions, then the mode of operation is changed so that VSNpk will be in the shaded region, and the part

will operate in the hysteretic mode. What actually happens is that the LM3477/A will not allow VSNpk to be outside

of the shaded regions, so the duty cycle is adjusted.

The output voltage transient response overshoot can also trigger OVP. As discussed in Output Capacitor

Selection, if the capacitance is too low or ESR too high, the output voltage overshoot will rise high enough to

trigger OVP. However, as long as there is room for the duty cycle to adjust (the converter is not near DMIN or

DMAX), the LM3477/A will return to PWM mode after a few cycles of hysteretic mode operation.

There is one last way that OVP can be triggered. If the unregulated input voltage crosses 7.2V, the output

voltage will react as shown in Figure 21. The internal bias of the LM3477/A switches supplies at 7.2V. When this

happens, a sudden small change in bias voltage is seen by all the internal blocks of the LM3477/A. The control

voltage, VC, shifts because of the bias change, the PWM comparator tries to keep regulation. To the PWM

comparator, the scenario is identical to step change in the load current, so the response at the output voltage is

the same as would be observed in a step load change. Hence, the output voltage overshoot here can also trigger

OVP. The LM3477/A will regulate in hysteretic mode for several cycles, or may not recover and simply stay in

hysteretic mode until the load current drops. Note that the output voltage is still regulated in hysteric mode.

Predicting whether or not the LM3477/A will come out of hysteretic mode in this scenario is a difficult task,

however it is largely a function of the output current and the output capacitance. Triggering hysteretic mode in

this way is only possible at higher load currents. The method to avoid this is to increase the output capacitance.

Product Folder Links: LM3477

ISEN DR

RSL = 0 (nominal) VIN

LM3477 RSN

DCOUT

VOUT

VSEN

Sn Sf

PWM Comparator

Waveforms

Time

Voltage

VIN (V)

VFB (V)

7.2V

OVP

(1.315V)

1.265V

xxxxx

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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The Feedback Voltage Experiences an Oscillation if the Input Voltage Crosses the 7.2V Internal Bias Threshold

Figure 21.

DEFAULT/ADJUSTABLE SLOPE COMPENSATION

The LM3477/A uses a current mode control scheme. There are many advantages in a current mode architecture

including inherent cycle-by-cycle current limiting and simple compensation of the control loop. However, there are

consequences to using current mode control that one must be aware of while selecting circuit components. One

of these consequences is the inherent possibility of subharmonic oscillations in the inductor current. This is a

form of instability and should be avoided.

Figure 22. The Current Sensing Loop and Corresponding Waveforms

As a brief explanation, consider Figure 22. A lot of information is shown here. The top portion shows a schematic

of the current sensing loop. The bottom portion shows the pulse width modulation (PWM) comparator waveforms

for two switching cycles. The two solid waveforms shown are the waveforms compared at the internal pulse width

modulator, used to generate the MOSFET drive signal. The top waveform with the slope Seis the internally

generated control waveform VC. The bottom waveform with slopes Snand Sfis the sensed inductor current

waveform VSEN. These signals are compared at the PWM comparator. There is a feedback loop involved here.

The inductor current is sensed and fed back to the PWM comparator, where it is compared to VC. The output of

the comparator in combination with the R/S latch determine if the MOSFET is on or off, which effectively controls

the amount of current the inductor receives. While VCis higher than VSEN, the PWM comparator outputs a high

signal, driving the external power MOSFET on. When MOSFET is on, the inductor current rises at a constant

slope, generating the sensed voltage VSEN. When VSEN equals VC, the PWM comparator signals to drive the

MOSFET off, and the sensed inductor current decreases with a slope Sf. The process begins again when RS

latch is set by an internal oscillator.

Product Folder Links: LM3477

Se = fS(VSL + 50 x 10-6 x RSL)V

( )

'Se = 50 x 10-6 x fS x RSL V

( )

'In=Sf - Se

Sn + Se'In-1

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

The subharmonic oscillation phenomenon is realized when a load excursion is experienced. The way it is

analyzed is to calculate how the inductor current settles after such an excursion. Take for example the case

when the inductor current experiences a step increase in its average current, shown as the dotted line in

Figure 22. In the switching period that the excursion occurs, the inductor current will change by ΔI0. In the

following switching period, the inductor current will have a difference ΔI1from its original starting value. The

original excursion is being propagated each switching cycle. What is desired is to find out if this propagation is

converging or diverging. It is apparent that the difference in the inductor current from one cycle to the next is a

function of Sn, Sf, and Se, as follows:

(1)

Hence, if the quantity (Sf- Se)/(Sn+ Se)is greater than 1, the inductor current diverges and subharmonic

oscillations result. Notice that as Seincreases, the factor decreases. Also, when the duty cycle is greater than

50%, as the inductance become less, the factor increases.

The LM3477/A internally generates enough slope compensation Seto allow for the use of reasonable

inductances. The height of the compensation slope ramp VSL can be found in the Electrical Characteristics. The

LM3477/A incorporates a patented scheme to increase Seif there is need to use a smaller inductor. With the use

of a single resistor RSL, Se can be increased indefinitely. RSL increases the compensation slope Se by the

amount:

(2)

Therefore,

(3)

When excursions of the inductor current are divergent, the current sensing control loop is unstable and produces

a subharmonic oscillation in the inductor current. This oscillation is viewed as a resonance in the outer voltage

control loop at half the switching frequency. In POWER INDUCTOR SECTION, calculations for minimum

inductance and necessary slope resistance RSL are carried out based on this resonant peaking.

START-UP/SOFT-START

The LM3477/A incorporates an internal soft-start during start-up. The soft-start forces the inductor current to rise

slowly and smoothly as it increases towards the steady-state current. This technique is used to reduce the input

inrush current during soft-start. The soft-start functionality is effective for approximately the first 5ms of start-up.

NOTE

The LM3477/A will not start-up if the output voltage is biased by more than 200mV above

ground.

NOTE

If the slope resistor RSL is used, the hysteretic threshold will be lowered. Therefore, the

LM3477/A may require up to 100mA of pre-load to successfully start up.

SHORT CIRCUIT PROTECTION

When the voltage across the sense resistor (measured as the VIN −ISEN differential voltage) exceeds VSC, short-

circuit current limit gets activated. In the short-circuit protection mode, the external MOSFET is turned off. When

the short is removed, the external MOSFET is turned on after five cycles. The short circuit protection voltage VSC

is specified in the Electrical Characteristics. VSC is lower in the LM3477A than in the LM3477.

SHUTDOWN

The compensation pin (Pin 2) of LM3477/A also functions as a shutdown pin. If a low signal (refer to the

Electrical Characteristics for definition of low signal) appears on the COMP/SD pin, the LM3477/A stops

switching and goes into a low supply current mode. The total supply current of the IC reduces to less than 10µA

under these conditions. Figure 23 shows different implementations of the shutdown function.

Product Folder Links: LM3477

SHUTDOWN

INPUT

OFF

+5V

VCC

COMP/SD

LM3477

COMP/SD

LM3477

SHUTDOWN

INPUT

OFF

+5V

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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Figure 23. Implementing Shutdown in LM3477

Figure 24. Implementing Shutdown in LM3477

Design Section

GENERAL

Power supply design involves making tradeoffs. To achieve performance specifications, limitations will be set on

component selection. The LM3477/A provides many degrees of flexibility in choosing external components to

accommodate various performance/component selection optimizations. For example, the internal slope

compensation can be externally increased to allow smaller inductances to be used. The design procedures that

follow provide instruction on how to select the external components in a typical LM3477/A buck circuit in

continuous conduction mode, as well as aid in the optimization of performance and/or component selection. See

Figure 25 for component reference and typical circuit. The LM3477/A may also be designed to operate in

discontinuous conduction mode.

Product Folder Links: LM3477

D = VOUT + VD

VIN + VD - VQ -VSEN

VOUT

VIN

RSL

LRSN Q1

CIN

COUT

RESR RRFB1

RFB2

VIN VOUT

POWER STAGE

LM3477

ISEN

VIN

GND

FB 3

CB 7

COMP/SD 2

SW 5RDR

CBOOT

RCCC1

COMPENSATION NETWORK

VSEN

CBYP

LOAD

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

Figure 25. LM3477 Buck Converter Reference Schematic

PROGRAMMING THE OUTPUT VOLTAGE

The output voltage can be programmed using a resistor divider between the output and the feedback pins, as

shown in Figure 25. The resistors are selected such that the voltage at the feedback pin is 1.27V. RFB1 and RFB2

can be selected using the equation:

VOUT = 1.27*(1+ RFB1/RFB2) (4)

CALCULATING THE DUTY CYCLE

In buck converter applications, the duty cycle of the LM3477/A may be calculated as:

(5)

Where

VD= forward drop of the power diode ≊0.5V

VQ= VDS of the MOSFET when it is conducting ≊IOUT*RDSON

VSEN = Voltage across the sense resistor = IOUT x RSN

This is the fraction of the switching period that the switch is on. The switch is off for the remainder of the period.

This fraction is expressed as:

D' = 1 −D (6)

The LM3477/A has limits for the maximum and minimum duty cycle (see Electrical Characteristics). The

maximum duty cycle of 93% (typical) will limit how low the input voltage may drop while maintaining a regulated

output voltage (the dropout voltage). In situations where a very low dropout voltage is required, it is necessary to

include VD, VQand VSN losses in the maximum duty cycle calculation. Voltage drops in the inductor will lower the

dropout voltage as well.

The LM3487 provides the FET drive voltage through the voltage developed across Cboot, which is charged when

the SW pin goes low. If Cboot cannot fully recharge, the device will automatically restart when the Cboot voltage

falls below approximately 2V. Therefore, a Cboot value of at least 0.1uF is recommended to ensure normal

operation at high duty cycles.

The minimum duty cycle of the LM3477/A corresponds to the minimum on time, or blank out time (see Electrical

Characteristics).

DMIN = TMIN* fs(7)

This will not limit how high the input voltage can rise, however the LM3477/A will operate in hysteretic mode once

the operating duty cycle decreases to the minimum duty cycle.

Product Folder Links: LM3477

RSN(MAX)=0.135 - 0.6 (0.135 - .025)

1.15 (3) = 0.02:

RSN(MAX) = VCL(0)(MIN) - DMAX (VCL(0)(MIN) - VCL(100)(MIN))

1.15 x IOUT(MAX)

VCL(0)(MIN) - DMAX (VCL(0)(MIN) - VCL(100)(MIN))

VOUT(1-DMAX)

2 x L x fS

IOUT(MAX) +

VSEN(PEAK) = ) (V), RSL = 0

VOUT(1-DMAX)

2 x L x fS

RSN(IOUT(MAX) +

RSN x IOUT(MAX)(1+ 0.15) (V)

CONTROL VOLTAGE (mV)

DUTY CYCLE (%)

0 20 40 60 80

100

200

100

VCL(0)

VCL(D)

Blank-out

Time

VSEN VHYS(LM3477)

VHYS(LM3477A)

VSEN(PEAK) VCL(100)

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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PROGRAMMING THE CURRENT LIMIT/HYSTERETIC THRESHOLD

Definitions

Current Limit: The current limit is the point at which the LM3477/A begins to limit the peak switch current. The

current limit in the LM3477/A varies with duty cycle, which is a function of the VIN −VOUT differential.

Hysteric Threshold: Hysteretic threshold is the current at which the LM3477/A enters the hysteretic mode of

operation (see OVER VOLTAGE PROTECTION). The hysteretic threshold is with respect to the peak switch

current.

SETTING CURRENT LIMIT AND HYSTERETIC THRESHOLD

The adjustable current limit of the LM3477/A is set by the sense resistor RSN. The voltage across RSN is

compared to an internal control voltage VC. The onset of current limiting is when VSEN(peak) equals VC(max), or VCL.

VSEN is defined here as the differential voltage from the VIN pin to the ISEN pin. VCL decreases as the duty cycle

increases, as shown in Figure 26. Therefore, it is important to know both VSEN(peak) and VCL(min) at the maximum

operating duty cycle, or lowest VIN condition.

Figure 26. Current Limit and Hysteretic Threshold vs Duty Cycle

(8)

VCL(MIN) = VCL(0)(MIN) −D(MAX) (VCL(0)(MIN) −VCL(100)(MIN)) (9)

where DMAX is the duty cycle at the lowest VIN condition.

To avoid current limit,

VSEN(peak) < VCL(MIN) (10)

Therefore,

(11)

Example: VIN(MIN) = 4.5V, VOUT = 2.5V, IOUT(MAX) = 3A

(12)

Product Folder Links: LM3477

IHYS =MAX (VHYS - 50 x 10-6 x RSL x DMAX, 0)

RSN (A)

RSN(MAX) = VCL(0)(MIN) - DMAX (VCL(0)(MIN) - (VCL(100)(MIN) - 50 x 10-6 x RSL))

VOUT(1-DMAX)

2 x L x fS

IOUT +

1.15 x IOUT

VCL(0)(MIN) - DMAX (VCL(0)(MIN) - (VCL(100)(MIN) - 50 x 10-6 x RSL))

CONTROL VOLTAGE (mV)

DUTY CYCLE (%)

0 20 40 60 80

100

200

100

VCL(0)

VSEN VSEN(PEAK)

VCL(D)

VSL

Blank-out

Time

VCL(100)

IHYS =0.032

0.02 = 1.6A, LM3477

0.011

0.02 0.55A, LM3477A

0.032

RSN (A), LM3477

0.011

RSN (A), LM3477A

IHYS =VHYS

RSN =

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

The hysteretic threshold is derived in a similar manner, the only difference being that VSEN(peak) is compared

VC(min) (VHYS). Notice that VHYS does not vary with the duty cycle. The hysteretic threshold is predetermined by

the selection of RSN above. The hysteretic threshold is:

(13)

Continuing with the example above,

(14)

If the peak switch current decreases below this threshold, the LM3477/A will operate in hysteretic mode (see

OVER VOLTAGE PROTECTION). In some designs, it will be desired to use RSL so that lower valued inductors

can be used (see DEFAULT/ADJUSTABLE SLOPE COMPENSATION and POWER INDUCTOR SECTION).

Using RSL will lower the current limit and the hysteretic threshold. See Figure 27. RSL effectively adds an

additional slope to the existing slope of the VCwaveform.

Figure 27. Current Limit and Hysteretic Threshold vs Duty Cycle with RSL

When RSL is used, the following equations apply:

(15)

(16)

where MAX(VHYS −50x10-6 x RSL x DMAX, 0) is the smaller of the two values in the parenthesis and VHYS is

0.032V and 0.011V for the LM3477 and LM3477A, respectively. RSL can be used creatively to intentionally lower

the hysteretic threshold, allowing for better performance at lower loads. However, when RSL is used, there may

be a minimum load requirement (see START-UP/SOFT-START).

Product Folder Links: LM3477

fS (VSL + 50 x 10-6 x RSL)

VIN1.8RSN + D-0.5)

SQMAX

(VIN1.8RSN + D-0.5)

(SQMIN

LfS (VSL + 50 x 10-6 x RSL)

mc = Se

1+ =1+ fSL(VSL+ 50 x 10-6 x RSL)

1.8RSNVIN D' - VQ -VSEN

1+ 1.8RSNVIN D'

fSL(VSL+ 50 x 10-6 x RSL)

D = VOUT + VD

VIN + VD - VQ -VSEN

VOUT

VIN

Q = 1

Smc x D' - 0.5)

VOUT (1-D)

L x fS(A)

'iL(Pk-Pk) =

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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POWER INDUCTOR SECTION

The LM3477/A operates at a high switching frequency of 500kHz, which allows the use of small inductors. This is

made apparent in the following set of equations used to calculate the output voltage ripple.

ΔVOUT(Pk-Pk) ≊ΔiL(Pk-Pk) x RESR (V) (17)

(18)

As the switching frequency fs increases, the inductance required for a given output voltage ripple decreases. The

equations above for ΔVOUT and ΔiLprovide criteria for choosing the inductance. The maximum voltage ripple in

steady-state, PWM operation can be controlled by limiting ΔiLwhich in turn is set by the inductance value.

Alternatively, one can simply choose ΔiLas a percentage of the maximum output current. Clearly, the size of the

output capacitor ESR, RESR, will have an affect on which criteria is used to choose the inductance. When the

ESR is relatively low (less than 100mΩ), such as in ceramic, OSCON, and some low ESR tantalum capacitors, it

is convenient to choose the inductance based on setting ΔiLto 30% of Iout(max). If the ESR is high, then it may

be necessary to restrict ΔiLto a lower value so that the output voltage ripple is not too high. Generally speaking,

the former suggestion of setting ΔiLto 30% of IOUT(MAX) is recommended.

The inductance also affects the stability of the converter. The slopes Snand Sfin Figure 22 are functions of the

inductance, while the compensation ramp, Se, is fixed by default. Therefore if the inductance is too small, the

converter may experience sub-harmonic oscillations. The LM3477/A provides sufficient internal slope

compensation to allow for inductances chosen according to the ΔiL= 0.3 x IOUT guideline in most cases. Still, one

should check to make sure the inductance is not too low before continuing the design process. If it is found that

the selected inductance is too low, a patented scheme to increase the compensation ramp, Se, is provided in the

LM3477/A (see DEFAULT/ADJUSTABLE SLOPE COMPENSATION). In the calculations that follow, if it is found

that the chosen inductance is too small, RSL can be used to increase Se so that the inductance can be used.

In a current mode control architecture, there is an inherent resonance at half the switching frequency (see

DEFAULT/ADJUSTABLE SLOPE COMPENSATION ). A convenient indicator of how much resonance exists is

the quality factor Q. If Q is too high, subharmonic oscillations could occur, if Q is too low, the current mode

architecture begins to act like a voltage mode architecture and the necessary compensation becomes more

complex. This is discussed in more detail in Compensation, but here it is important to calculate Q to be sure the

selected inductance will not cause problems to the stability of the converter. The calculations below call for an

inductance that results in Q between 0.15 and 2. See Compensation if the chosen inductance enforces Q to be

out of this range. By default, no extra slope compensation is needed, so RSL = 0. In general, a Q between 0.5

and 1 is optimal.

(19)

Where,

D' = 1−D (20)

(21)

(22)

VQ= VDS of the MOSFET when it is conducting IOUT*RDS(ON).

1.8 = voltage gain of the current sense amp.

VSEN = Voltage across the sense resistor ≊IOUT x RSN

Back solving for L gives a range for acceptable inductances based on a range for Q:

(23)

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It is recommended that:

Q(max) = 2, and

Q(min) = 0.15

Values for VSL can be found in the Electrical Characteristics.

Note: Adding slope compensation with RSL will decrease the current limit. An iterative process may be needed to

meet current limit and stability requirements, see PROGRAMMING THE CURRENT LIMIT/HYSTERETIC

THRESHOLD.

Output Capacitor Selection

A capacitance between 47µF - 100µF is typically used. Skip to CALCULATIONS FOR THE OUTPUT

CAPACITOR for minimum capacitance calculations.

TYPE OF OUTPUT CAPACITORS

Different type of capacitors often have different combinations of capacitance, equivalent series resistance (ESR),

and voltage ratings. High-capacitance multi-layer ceramic capacitors (MLCCs) have a very low ESR, typically

12mΩ, but also relatively low capacitance and low voltage ratings. Tantalum capacitors can have fairly low ESR,

such as 18mΩ, and high capacitance (up to 1mF) at higher voltage ratings than MLCCs. Aluminum capacitors

offer high capacitance and relatively low ESR and are available in high voltage ratings. OSCON capacitors can

achieve ESR values that are even lower than those of MLCCs and with higher capacitance, but the voltage

ratings are low. Other tradeoffs in capacitor technology include temperature stability, surge current capability, and

capacitance density (physical size vs. capacitance).

OUTPUT CAPACITOR CONSIDERATIONS

Skip to CALCULATIONS FOR THE OUTPUT CAPACITOR if a quick design is desired. While it is generally

desired to use as little output capacitance as possible to keep costs down, the output capacitor should be chosen

with care as it directly affects the ripple component of the output voltage as well as other components in the

design. The output voltage ripple is directly proportional to the ESR of the output capacitor (see POWER

INDUCTOR SECTION). Therefore, designs requiring low output voltage ripple should have an output capacitor

with low ESR. Choosing a capacitor with low ESR has the additional benefit of requiring one less component in

the compensation network, as discussed in Compensation.

In addition to the output voltage ripple, the output capacitor directly affects the output voltage overshoot in a load

transient. Two transients are possible: an unloading transient and a loading transient. An unloading transient

occurs when the load current transitions to a higher current, and charge is unloaded from the output capacitor. A

loading transient is when the load transitions to a lower current, and charge is loaded to the output capacitor.

How the output voltage reacts during these transitions is known as the transient response. Both the capacitance

and the ESR of the output capacitor will affect the transient response.

Figure 28. A Loading Transient

The control loop of the LM3477/A can be made fast enough to saturate the duty cycle when the worst case lode

transient occurs. This means the duty cycle jumps to DMIN or DMAX, depending on the type of load transient. In a

loading transient, as shown in Figure 28, the duty cycle drops to DMIN while the inductor current falls to match the

load current. During this time, the regulator is heavily dependent on the output capacitors to handle the load

transient. The initial overshoot is caused by the ESR of the output capacitors. How the output voltage recovers

after that initial excursion depends on how fast the inductor current falls and how large the output capacitance is.

See Figure 29.

Product Folder Links: LM3477

tpeak =

'IOUT(MAX) x L

VOUT-DMINVIN - COUTRESR

'Vq =COUT

'IOUT(MAX) t - 2 X L X COUT t2 (V)

VOUT - DMINVIN

RESR(MAX) =VOS(MAX)

'IOUT(MAX):

'Vr = RESR('IOUT(MAX) - L

VOUT - DMINVIN t)(V)

'Vc2

Time

Voltage

Upper Voltage

Limit

VOS (MAX)

'Vc1

ESR too large

capacitance too small

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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Figure 29. Output Voltage Overshoot Violation

The ESR and the capacitance of the output capacitor must be carefully chosen so that the output voltage

overshoot is within the design's specification VOS(MAX). If the total combined ESR of the output capacitors is not

low enough, the initial output voltage excursion will violate the specification, see ΔVC1. If the ESR is low enough,

but there is not enough output capacitance, the output voltage will travel outside the specification window due to

the extra charge being dumped into the capacitor, see ΔVC2. The LM3477/A has output over voltage protection

(OVP) which could trigger if the transient overshoot is high enough. If this happens, the controller will operate in

hysteretic mode (see OVER VOLTAGE PROTECTION) for a few cycles before the output voltage settles to its

steady state. If this behavior is not desired, substitute VOVP (referred to the output) for VOS(MAX) (VOVP is found in

the Electrical Characteristics) to find the minimum capacitance and maximum ESR of the output capacitor.

CALCULATIONS FOR THE OUTPUT CAPACITOR

During a loading transient, the delta output voltage ΔVchas two changing components. One is the voltage

difference across the ESR (ΔVr), the other is the voltage difference caused by the gained charge (ΔVq). This

gives:

ΔVc=ΔVr+ΔVq(24)

The design objective is to keep ΔVclower than some maximum overshoot (VOS(MAX)). VOS(MAX) is chosen based

on the output load requirements.

Both voltages ΔVrand ΔVqwill change with time. For ΔVrthe equation is:

(25)

where,

RESR = the output capacitor ESR

ΔIOUT = the difference between the load current change IOUT(MAX) −IOUT(MIN)

DMIN = Minimum duty cycle of device (0.165 typical)

Evaluating this equation at t = 0 gives ΔVr(max). Substituting VOS(MAX) for ΔVr(MAX) and solving for RESR gives:

(26)

The expression for ΔVqis:

(27)

From Figure 30 it can be told that ΔVCwill reach its peak value at some point in time and then decrease. The

larger the output capacitance is, the earlier the peak will occur. To find the peak position, let the derivative of ΔVC

go to zero, and the result is:

(28)

Product Folder Links: LM3477

VIN(MIN)

VOUT

Dmax

PCOND(MAX) = I2DMAX 1+ 'i(PK-PK)

> @

12 I

2RDS(ON)

VOS(MAX)-VOS(MAX)2 - ('IOUT(MAX) x RESR)2

COUT(MIN)=(VOUT) RESR2(F)

'Vq

'Vr

'Vc

Time

Voltage

0tpeak

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

Figure 30. Output Voltage Overshoot Peak

The intention is to find the capacitance value that will yield, at tpeak, a ΔVCthat equals VOS(max). Substituting tpeak

for t and equating ΔVCto VOS(max) gives the following solution for COUT(MIN):

(29)

The chosen output capacitance should not be less than 47µF, even if the solution for COUT(MIN) is less than 47µF.

Notice it is already assumed that the total ESR is no greater than RESR(MAX), otherwise the term under the square

root will be a negative number.

Power Mosfet Selection

The drive pin of LM3477/A must be connected to the gate of an external MOSFET. In a buck topology, the drain

of the external N-Channel MOSFET is connected to the input and the source is connected to the inductor. The

CBpin voltage provides the gate drive needed for an external N-Channel MOSFET. The gate drive voltage

depends on the input voltage (see Typical Performance Characteristics). In most applications, a logic level

MOSFET can be used. For very low input voltages, a sub-logic level MOSFET should be used.

The selected MOSFET directly controls the efficiency. The critical parameters for selection of a MOSFET are:

1. Minimum threshold voltage, VTH(MIN)

2. On-resistance, RDS(ON)

3. Total gate charge, Qg

4. Reverse transfer capacitance, CRSS

5. Maximum drain to source voltage, VDS(MAX)

The off-state voltage of the MOSFET is approximately equal to the input voltage. VDS(MAX) of the MOSFET must

be greater than the input voltage. The power losses in the MOSFET can be categorized into conduction losses

and ac switching or transition losses. RDS(ON) is needed to estimate the conduction losses. The conduction loss,

PCOND, is the I2R loss across the MOSFET. The maximum conduction loss is given by:

(30)

where DMAX is the maximum operating duty cycle:

(31)

The turn-on and turn-off transition times of a MOSFET from the MOSFET specifications require tens of nano-

seconds. CRSS and Qgare needed from the MOSFET specifications to estimate the large instantaneous power

loss that occurs during these transitions.

The average amount of gate current required to turn the MOSFET on can be calculated using the formula:

IG= Qg.FS(32)

The required gate drive power to turn the MOSFET on is equal to the switching frequency times the energy

required to deliver the charge to bring the gate charge voltage to VDR (see Electrical Characteristics and Typical

Performance Characteristics for the drive voltage specification).

PDrive = FS.Qg.VDR (33)

Product Folder Links: LM3477

Fc(s)

Power Stage

Compensator

Feedback

VREF VCVO

Loop (T)

Fp(s) x Fh(s)

IRMS_CIN = IOUT VOUT(VIN-VOUT)

VIN

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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It is sometimes helpful or necessary to slow down the turn on transition of the FET so that less switching noise

appears at the ISEN pin. This can be done by inserting a drive resistor RDR in series with the boot-strap capacitor

(see Figure 25). This can help reduce sensing noise that may be preventing designs from operating at or near

the LM3477/A's minimum duty cycle limit. Gate drive resistors from 2.2Ωto 51Ωare recommended.

Power Diode Selection

The output current commutates through the diode when the external MOSFET turns off. The three most

important parameters for the diode are the peak current, peak inverse voltage, and average power dissipation.

Exceeding these ratings can cause damage to the diode. The average current through the diode is given by:

ID(AVG) = IOUT x (1-D) (34)

where D is the duty cycle and IOUT is the output current. The diode must be rated to handle this current.

The off-state voltage across the diode in a buck converter is approximately equal to the input voltage. The peak

inverse voltage rating of the diode must be greater than the off-state voltage of the diode. To improve efficiency,

a low forward drop schottky diode is recommended.

Input Capacitor Selection

In a buck converter, the high side switch draws large ripple currents from the input capacitor. The input capacitor

must be rated to handle this RMS current.

(35)

The power dissipated in the input capacitor is given by:

PD(CIN)=IRMS_CIN2RESR_CIN,

where RESR_CIN is the ESR of the input capacitor. The input capacitor must be selected to handle the rms current

and must be able to dissipate the power. PD(CIN) must be lower than the rated power dissipation of the selected

input capacitor. In many cases, several capacitors have to be paralleled to handle the rms current. In that case,

the power dissipated in each capacitor is given by:

PD(CIN) = (I2RMS_CINRESR_CIN)/n2, where n is the total number of capacitors paralled at the input.

A 0.1µF or 1µF ceramic bypass capacitor is also recommended on the VIN pin (pin 8) of the IC. This capacitor

must be connected very close to pin 8.

Compensation

Figure 31. Control Block Diagram of a Current Mode Controlled Buck Converter

The LM3477/A is a current mode controller, therefore the control block diagram representation involves 2

feedback loops (see Figure 31). The inner feedback loop derives its feedback from the sensed inductor current,

while the outer loop monitors the output voltage. This section will not give a rigorous analysis of current mode

control, but rather a simplified but accurate method to determine the compensation network. The first part reveals

the results of the model, giving expressions for solving for component values in the compensation network.

Product Folder Links: LM3477

H = feedback gain = RFB2

RFB1 RFB2

-60

-40

MAGNITUDE (dB)

102103104105106

FREQUENCY (Hz)

-20

107

fP1

fP2

fZ1

fCfn

-180

-135

-90

-45

-225

fESR

Open Loop (T)

Power Stage

Compensator

(º)

PHASE

RFB1

RFB2

CFF

CC2

CC1

LM3477

RGM

VOUT

LM3477

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SNVS141K –OCTOBER 2000–REVISED MARCH 2013

The compensation network is designed around the power components, or the power stage. An isolated

schematic of the error amplifier and the various compensation components is shown in Figure 32. The error

amplifier in conjunction with the compensation network makes up the compensator block in Figure 31. The

purpose of the compensator block is to stabilize the control loop and achieve high performance in terms of the

transient response, audio susceptibility and output impedance.

Figure 32. LM3477 Compensation Components

Figure 33 shows a bode plot of a typical current mode buck regulator. It is an estimate of the actual plot using the

asymptotic approach. The three plots shown are of the compensator, powerstage, and loop gain, which is the

product of the power stage, compensator, and feedback gain. The loop gain determines both static and dynamic

performance of the converter. The power stage response is fixed by the selection of the power components,

therefore the compensator is designed around the powerstage response to achieve a good loop response.

Specifically, the compensator is added to increase low frequency magnitude, extend the 0dB frequency

(crossover frequency), and improve the phase characteristic.

Poles, Zeros and Important Measurements are Labeled

Figure 33. Typical Open Loop, Compensator, and Power Stage Bode Plots for LM3477 Buck Circuits

There are several different types of compensation that can be used to improve the frequency response of the

control loop. To determine which compensation scheme to use, some information about the power stage is

needed.

Use VIN = VIN(MIN) and R = RMIN (IOUT(MAX)) when calculating compensation components.

(36)

Product Folder Links: LM3477

fESR

3.16

2SfCRC

CC1 1

2Sfp1RC

RC = fC x RGM

ADC x GM x RGM x H x fP1 - fC

Q = 1

Smc x D' - 0.5)

fESR =2SCOUTRESR

1(Hz)

COUTR

fp1 = +(mc x D' - 0.5)

fsLCOUT

2S(Hz)

VIND' x 1.8RSN

Sn =

mc = 1+ Se

1.8RSN R

fSLmc x D' - 0.5

ADC = 1

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

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(37)

(38)

Se= fS(VSL + 50x10−6RSL) (39)

(40)

(41)

(42)

(43)

With the power stage known, a compensator can be designed to achieve improved performance and stability.

The LM3477/A will typically require only a single resistor and capacitor for compensation, but depending on the

power stage it could require three or four external components.

It is a good idea to check that Q is between 0.15 and 2, if it was not already done when selecting the inductor. If

Q is less than 0.15 or greater than 2, skip to SAMPLING POLE QUALITY FACTOR before continuing with the

compensator design.

First, a target crossover frequency (fc) for the loop gain must be selected. The crossover frequency is the

bandwidth of the converter. A higher bandwidth generally corresponds to faster response times and lower

overshoots to load transients. However, the bandwidth should not be much higher than 1/10 the switching

frequency. The LM3477/A operates with a 500kHz switching frequency, so it is recommended to choose a

crossover frequency between 10kHz - 50kHz.

The schematic of the LM3477/A compensator is shown in Figure 32. The default design uses Rcand CC1 to form

a lag (type 2) compensator. The CC2 capacitor can be added to form an additional pole that is typically used to

cancel out the esr zero of the output capacitor. Finally, if extra phase margin is needed, the Cff capacitor can be

added (this does not help at low output voltages, see below).

The strategy taken here for choosing Rcand CC1 is to set the crossover frequency with Rc, and set the

compensator zero with CC1. Using the selected target crossover frequency, fC, set RCto:

(44)

fC= Crossover frequency in Hertz (20kHz - 50kHz is recommended)

RGM = 50x103Ω

GM = 1000x10−6A/V

The compensator zero, fZ1, is set with CC1. When fast transient responses are desired, fZ1 should be placed as

high as possible, however it should not be higher than the selected crossover frequency fC. The guideline

proposed here is to choose CC1 such that fZ1 falls somewhere between the power pole fP1 and ½ decade before

the selected crossover frequency fc:

(45)

In this compensation scheme, the pole created by CC2 is used to cancel out the zero created by the ESR of the

output capacitor. In other schemes such as the methods discussed in SAMPLING POLE QUALITY FACTOR, the

ESR zero is used. For the typical case, use CC2 if:

(46)

Product Folder Links: LM3477

Q = = 0.33

S[(3.36)(0.44) - 0.5]

2S(100 x 10-6)(0.01)

fESR = = 159kHz

fp1 = (500 x 103)(3.3 x 10-6)(100 x 10-6)

(100 x 10-6)(0.83)

(3.36)(0.44) - 0.5

>@= 2.86 kHz

ADC = 0.83

1.8 x 0.02 [(3.36(0.44) - 0.5]

(500 x 103)(3.3 x 10-6)

0.83 = 15.5

H = feedback gain = 1.27

2.5 = 0.508

FC(s) = (sCC1RC + 1)

s2CC1CC2RCRGM + s(CC2RGM + CC1(RGM + RC)) + 1 , CC2 used

FC(s) = (sCC1RC + 1)

sCC1RGM (RGM + RC) + 1 , CC2 not used

Fh(s) = +1

s21

Sfs

2s1

SfsQ+ 1

FP(S) = 1 + s

SfESR

1 + s

Sfp1

RGM + RC

SfESR RGMRC

CC2 = (F)

LM3477

www.ti.com

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

(47)

PLOTTING THE OPEN LOOP RESPONSE

The open loop response is expressed as:

T=ADC x ACM xHxFp(s) x Fc(s)

Where ADC and H are given above and

ACM =GMxRGM

(48)

(49)

(50)

(51)

One can plot the magnitude and phase of the open loop response to analyze the frequency response.

EXAMPLE: COMPENSATION DESIGN

4.5V ≤VIN ≤5.5V

VOUT = 2.5V

IOUT = 3A (R = 0.83Ω)

RSN = 0.02Ω

L = 3.3µH

RSL = 0Ω

COUT = 100µF

RESR = 0.01Ω

First, calculate the power stage parameters using VIN(MIN) and R(MAX):

(52)

(53)

(54)

(55)

(56)

In this example, a crossover frequency of 20kHz is chosen, so: fC= 20000. RCis now calculated using the power

stage information and the target crossover frequency fC:

Product Folder Links: LM3477

-60

-40

101102103104105

Frequency (Hz)

-20

106

-50

-100

-150

-180

-250

-200

PHASE ( )

MAGNITUDE (dB)

BW = 16.7 kHz

PM = 61 degrees

CC2 = 50 x 103 + 900

2S (.159 x 103) (50 x 103) (900)

= 1.1 nF

2SCOUTRESR 250kHz

2S(20 x 103)(900)

3.16 1

CC1

28 nF 62 nF

2S(2.86 x 103)(900)

RC =

= 904:

(15.5)(0.001)(50 x 103)(0.508)(2.86 x 103) - (20 x 103)

(20 x 103)(50 x 103)

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

www.ti.com

(57)

This sets the high frequency gain of the compensator such that a crossover frequency of fCis obtained. The

capacitor CC1 sets the compensator zero, fZ2. Set fZ2 between the power pole fP1 and the ½ decade before the

target crossover frequency fC:

(58)

Choosing CC1 = 62 x 10−9F will set fZ2 = fP1, canceling out the power pole and insuring a −20dB/decade slope in

the low frequency magnitude response. In other words, the phase margin below the crossover frequency will

always be higher than the phase margin at the crossover frequency.

If better transient response times are desired, a second method is to set fZ2 between fP1 and ½ decade before fC,

the target crossover frequency. This trades more low frequency gain for less phase margin, which translates to

faster but more oscillatory step responses. We pick CC1 = 47nF.

If the esr zero of the output capacitor (fESR) is too low or if more phase margin is required, additional components

may be added to increase the flexibility of the compensator.

Use CC2 if fESR <½fS, that is if:

(59)

For this example, fESR = 159 kHz, so use CC2.

(60)

The equations used here for RC, CC1, and CC2 are approximations valid when CC2 << CC1. For exact equations,

see PLOTTING THE OPEN LOOP RESPONSE. In some cases, the desired inductance is several times higher

than the optimal inductance set by the internal slope compensation. This results in a Q lower than 0.15, in which

case additional methods of compensating are presented (see SAMPLING POLE QUALITY FACTOR).

Figure 34. Open Loop Frequency Response for LM3477 Compensation Design Example

Product Folder Links: LM3477

2SRFB1Cff

fpff = RFB2

RFB1 RFB2

+= fzff VOUT

VFB (Hz)

2SRFB1Cff (Hz)

fzff =

MAGNITUDE (dB)

102103104105106

FREQUENCY (Hz)

-120

-80

-40

107

-300

-250

-200

-150

-100

-50

PHASE (°)

Plotted Q values: 0.01, 0.05, 0.1, 0.2, 0.5, 1, 2, 10

Q = 0.01

Q = 10

LM3477

www.ti.com

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

SAMPLING POLE QUALITY FACTOR

In a current mode control architecture, there is an inherent resonace at half the switching frequency. The

LM3477/A internally compensates for this by adding a negative slope to the PWM control waveform (see

DEFAULT/ADJUSTABLE SLOPE COMPENSATION). The factor in the power stage equations above, Q,

describes how much resonance will be observed. Q is a function of duty cycle and mc.Figure 35 shows how the

power stage bode plot is affected as Q is varied from 0.01 to 10. The resonance is caused by two complex poles

at half the switching frequency. If mcis too low, the resonant peaking could become severe coinciding with

subharmonic oscillations in the inductor current. If mcis too high, the two complex poles split and the converter

begins to act like a voltage mode converter and the compensation scheme used above should be changed.

The Quality Factor Q of the Two Complex Poles is used to qualify how much resonant peaking is observed in the

Power Stage Bode Plot

Figure 35.

If Q>2, the sampling poles are imaginary and are approaching the right half of the imaginary plane (the system is

becoming unstable). In this case, Q must be decreased by either increasing the inductance, or more preferably,

adding more slope compensation through the RSL resistor (see DEFAULT/ADJUSTABLE SLOPE

COMPENSATION).

If Q<0.15, it means that one of the sampling poles is decreasing in frequency towards the dominant power pole,

fp1. There are three ways to compensate for this. Decrease the crossover frequency, add a phase lead network,

or use the output capacitor's ESR to cancel out the low frequency sampling pole.

One option is to decrease the crossover frequency so that the phase margin is not as severely decreased by the

sampling pole. Decreasing the crossover frequency to between 1kHz to 10kHz is advisable here. As a result,

there will be a decrease in transient response performance.

Another option is the use of the feed-forward capacitor, Cff. This will provide a positive phase shift (lead) which

can be used to increase phase margin. However, it is important to note that the effectiveness of Cff decreases

with output voltage. This is due to the fact that the frequencies of the zero fzff and pole fpff get closer together as

the output voltage is reduced.

The frequency of the feed-forward zero and pole are:

(61)

(62)

A third option is to strategically place the ESR zero fESR of the output capacitor to cancel out the sampling pole.

In this case, the capacitor CC2 will not be used to cancel out fESR. fESR should be placed around the crossover

frequency fc, but this will depend on how low Q is.

Product Folder Links: LM3477

LM3477

SNVS141K –OCTOBER 2000–REVISED MARCH 2013

www.ti.com

REVISION HISTORY

Changes from Revision J (March 2013) to Revision K Page

• Changed layout of National Data Sheet to TI format .......................................................................................................... 27

Product Folder Links: LM3477

PACKAGE OPTION ADDENDUM

www.ti.com 11-Jan-2021

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead finish/

Ball material

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM3477AMM/NOPB ACTIVE VSSOP DGK 8 1000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 S13A

LM3477AMMX/NOPB ACTIVE VSSOP DGK 8 3500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 S13A

LM3477MM/NOPB ACTIVE VSSOP DGK 8 1000 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 S13B

LM3477MMX NRND VSSOP DGK 8 3500 Non-RoHS

& Green Call TI Call TI -40 to 125 S13B

LM3477MMX/NOPB ACTIVE VSSOP DGK 8 3500 RoHS & Green SN Level-1-260C-UNLIM -40 to 125 S13B

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) RoHS: TI defines "RoHS" to mean semiconductor products that are compliant with the current EU RoHS requirements for all 10 RoHS substances, including the requirement that RoHS substance

do not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, "RoHS" products are suitable for use in specified lead-free processes. TI may

reference these types of products as "Pb-Free".

RoHS Exempt: TI defines "RoHS Exempt" to mean products that contain lead but are compliant with EU RoHS pursuant to a specific EU RoHS exemption.

Green: TI defines "Green" to mean the content of Chlorine (Cl) and Bromine (Br) based flame retardants meet JS709B low halogen requirements of <=1000ppm threshold. Antimony trioxide based

flame retardants must also meet the <=1000ppm threshold requirement.

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

(6) Lead finish/Ball material - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead finish/Ball material values may wrap to two

lines if the finish value exceeds the maximum column width.

PACKAGE OPTION ADDENDUM

www.ti.com 11-Jan-2021

Addendum-Page 2

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

LM3477AMM/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

LM3477AMMX/NOPB VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

LM3477MM/NOPB VSSOP DGK 8 1000 178.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

LM3477MMX VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

LM3477MMX/NOPB VSSOP DGK 8 3500 330.0 12.4 5.3 3.4 1.4 8.0 12.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 29-Sep-2019

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

LM3477AMM/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0

LM3477AMMX/NOPB VSSOP DGK 8 3500 367.0 367.0 35.0

LM3477MM/NOPB VSSOP DGK 8 1000 210.0 185.0 35.0

LM3477MMX VSSOP DGK 8 3500 367.0 367.0 35.0

LM3477MMX/NOPB VSSOP DGK 8 3500 367.0 367.0 35.0

PACKAGE MATERIALS INFORMATION

www.ti.com 29-Sep-2019

Pack Materials-Page 2

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