General Description
The MAX8732A/MAX8733A/MAX8734A dual step-down,
switch-mode power-supply (SMPS) controllers generate
logic-supply voltages in battery-powered systems. The
MAX8732A/MAX8733A/MAX8734A include two pulse-
width modulation (PWM) controllers, adjustable from 2V to
5.5V or fixed at 5V and 3.3V. These devices feature two
linear regulators providing 5V and 3.3V always-on out-
puts. Each linear regulator provides up to 100mA output
current with automatic linear-regulator bootstrapping to
the main SMPS outputs. The MAX8732A/MAX8733A/
MAX8734A include on-board power-up sequencing, a
power-good (PGOOD) output, digital soft-start, and inter-
nal soft-stop output discharge that prevents negative volt-
ages on shutdown. Additionally, the outputs are high
impedance when VCC falls below its UVLO set point while
the outputs are enabled.
Maxim’s proprietary Quick-PWM™ quick-response, con-
stant on-time PWM control scheme operates without
sense resistors and provides 100ns response to load tran-
sients while maintaining a relatively constant switching fre-
quency. The unique ultrasonic pulse-skipping mode
maintains the switching frequency above 25kHz, which
eliminates noise in audio applications. Other features
include pulse skipping, which maximizes efficiency in
light-load applications, and fixed-frequency PWM mode,
which reduces RF interference in sensitive applications.
The MAX8732A features a 200kHz/5V and 300kHz/3.3V
SMPS for highest efficiency, while the MAX8733A fea-
tures a 400kHz/5V and 500kHz/3.3V SMPS for “thin and
light” applications. The MAX8734A provides a pin-
selectable switching frequency, allowing either 200kHz/
300kHz or 400kHz/500kHz operation of the 5V/3.3V
SMPSs, respectively. The MAX8732A/MAX8733A/
MAX8734A are available in 28-pin QSOP packages and
operate over the extended temperature range
(-40°C to +85°C).
The MAX8732A/MAX8733A/MAX8734A are pin-for-pin
upgrades to the MAX1777/MAX1977/MAX1999.
The MAX1999 evaluation kit (EV kit) can be used to
evaluate the MAX8732A/MAX8733A/MAX8734A.
Applications
Notebook and Subnotebook Computers
PDAs and Mobile Communication Devices
3- and 4-Cell Li+ Battery-Powered Devices
Features
♦No Current-Sense Resistor Needed (MAX8734A)
♦Accurate Current Sense with Current-Sense
Resistor (MAX8732A/MAX8733A)
♦1.5% Output Voltage Accuracy
♦3.3V and 5V 100mA Bootstrapped Linear
Regulators
♦Internal Soft-Start and Soft-Stop Output
Discharge
♦Quick-PWM with 100ns Load Step Response
♦3.3V and 5V Fixed or Adjustable Outputs
(Dual Mode™)
♦4.5V to 24V Input Voltage Range
♦Enhanced Ultrasonic Pulse-Skipping Mode
(25kHz min)
♦Power-Good (PGOOD) Signal
♦Overvoltage Protection Enable/Disable
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
________________________________________________________________ Maxim Integrated Products 1
Ordering Information
19-3711; Rev 0; 5/05
For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at
1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.
PART
TEMP RANGE
PIN-
PA CKA GE
5V/3.3V
SWITCHING
FREQUENCY
(kHz)
MAX8732AEEI+ -40°C to +85°C 28 QSOP
200/300
MAX8732AEEI -40°C to +85°C 28 QSOP
200/300
MAX8733AEEI+ -40°C to +85°C 28 QSOP
400/500
MAX8733AEEI -40°C to +85°C 28 QSOP
400/500
Quick-PWM and Dual Mode are trademarks of Maxim
Integrated Products, Inc.
Ordering Information continued at end of data sheet.
+Denotes lead-free package.
28
27
26
25
24
23
22
21
20
19
18
17
16
15
1
2
3
4
5
6
7
8
9
10
11
12
13
14
BST3
LX3
DH3
LDO3
DL3
GND
LX5
OUT3
OUT5
V+
DL5
LDO5
VCC
DH5
BST5
TON
ILIM5
FB5
REF
FB3
ILIM3
ON5
ON3
PGOOD
N.C.
QSOP
TOP VIEW
MAX8734A
SHDN
PRO
SKIP
Pin Configurations
Pin Configurations continued at end of data sheet.
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
2_______________________________________________________________________________________
ABSOLUTE MAXIMUM RATINGS
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)
Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functional
operation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure to
absolute maximum rating conditions for extended periods may affect device reliability.
V+, SHDN to GND ..................................................-0.3V to +25V
BST_ to GND ..........................................................-0.3V to +30V
LX_ to BST_ ..............................................................-6V to +0.3V
CS_ to GND (MAX8732A/MAX8733A only) .................-2V to +6V
VCC, LDO5, LDO3, OUT3, OUT5, ON3, ON5, REF,
FB3, FB5, SKIP, PRO, PGOOD to GND ...............-0.3V to +6V
DH3 to LX3 ..............................................-0.3V to (VBST3 + 0.3V)
DH5 to LX5 ..............................................-0.3V to (VBST5 + 0.3V)
ILIM3, ILIM5 to GND...................................-0.3V to (VCC + 0.3V)
DL3, DL5 to GND....................................-0.3V to (VLDO5 + 0.3V)
TON to GND (MAX8734A only) ................................-0.3V to +6V
LDO3, LDO5, REF Short Circuit to GND ....................Momentary
LDO3 Current (internal regulator) Continuous................+100mA
LDO3 Current (switched over to OUT3) Continuous ......+200mA
LDO5 Current (internal regulator) Continuous................+100mA
LDO5 Current (switched over to OUT5) Continuous ......+200mA
Continuous Power Dissipation (TA= +70°C)
28-Pin QSOP (derate 10.8mW/°C above +70°C).........860mW
Operating Temperature Range ...........................-40°C to +85°C
Junction Temperature......................................................+150°C
Storage Temperature Range .............................-65°C to +150°C
Lead Temperature (soldering, 10s) .................................+300°C
PARAMETER CONDITIONS
MIN
TYP
MAX
MAIN SMPS CONTROLLERS
LDO5 in regulation 6 24
V+ Input Voltage Range V+ = LDO5, VOUT5 < 4.43V 4.5 5.5 V
3.3V Output Voltage in
Fixed Mode V+ = 6V to 24V, FB3 = GND, V SKIP = 5V
3.285 3.330 3.375
V
V+ = 6V to 24V, FB5 = GND, V SKIP = 5V,
MAX8732A/MAX8734A (TON = VCC)
5V Output Voltage in Fixed Mode
V+ = 7V to 24V, FB5 = GND, V SKIP = 5V,
MAX8733A/MAX8734A (TON = GND)
4.975 5.050 5.125
V
Output Voltage in
Adjustable Mode V+ = 6V to 24V, either SMPS
1.975 2.00 2.025
V
Output Voltage Adjust Range Either SMPS 2.0 5.5 V
FB3, FB5 Adjustable-Mode
Threshold Voltage Dual-Mode comparator 0.1 0.2 V
Either SMPS, V SKIP = 5V, 0 to 5A
-0.1
Either SMPS, SKIP = GND, 0 to 5A
-1.5
DC Load Regulation
Either SMPS, V SKIP = 2V, 0 to 5A
-1.7
%
Line Regulation Either SMPS, 6V < V+ < 24V
0.005
%/V
Current-Limit Threshold
(Positive, Default)
ILIM_ = VCC, GND - CS_ (MAX8732A/MAX8733A),
GND - LX_ (MAX8734A) 93
100
107 mV
VILIM_ = 0.5V 40 50 60
VILIM_ = 1V 93
100
107
Current-Limit Threshold
(Positive, Adjustable)
GND - CS_
(MAX8732A/MAX8733A),
GND - LX_ (MAX8734A) VILIM_ = 2V
185 200
215
mV
Zero-Current Threshold SKIP = GND, ILIM_ = VCC, GND - CS_
(MAX8732A/MAX8733A), GND - LX_ (MAX8734A) 3mV
Current-Limit Threshold
(Negative, Default)
SKIP = ILIM_ = VCC, GND - CS_
(MAX8732A/MAX8733A), GND - LX_ (MAX8734A)
-120
mV
Soft-Start Ramp Time Zero to full limit 1.7 ms
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
_______________________________________________________________________________________ 3
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)
PARAMETER CONDITIONS
MIN
TYP
MAX
UNITS
5V SMPS
200
MAX8732A or MAX8734A
(VTON = 5V), SKIP = VCC 3.3V SMPS
300
5V SMPS
400
MAX8733A or MAX8734A
(VTON = 0), SKIP = VCC 3.3V SMPS
500
Operating Frequency
SKIP = REF 25 36
kHz
VOUT5 = 5.05V
1.895 2.105 2.315
MAX8732A or MAX8734A
(VTON = 5V) VOUT3 = 3.33V
0.833 0.925 1.017
VOUT5 = 5.05V
0.895 1.052 1.209
On-Time Pulse Width
MAX8733A or MAX8734A
(VTON = 0) VOUT3 = 3.33V
0.475 0.555 0.635
µs
Minimum Off-Time
250 300
350 ns
VOUT5 = 5.05V 94
MAX8732A or MAX8734A
(VTON = 5V) VOUT3 = 3.33V 91
VOUT5 = 5.05V 88
Maximum Duty Cycle
MAX8733A or MAX8734A
(VTON = 0) VOUT3 = 3.33V 85
%
INTERNAL REGULATOR AND REFERENCE
LDO5 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO5 < 100mA 4.90 5.00 5.10
V
LDO5 Short-Circuit Current LDO5 = GND
190
mA
LDO5 Undervoltage-Lockout
Fault Threshold Falling edge of LDO5, hysteresis = 1% 3.7 4.0 4.3 V
LDO5 Bootstrap Switch Threshold
Falling edge of OUT5, rising edge at OUT5 regulation
point
4.43 4.56 4.69
V
LDO5 Bootstrap
Switch Resistance LDO5 to OUT5, VOUT5 = 5V 1.4 3.2 Ω
LDO3 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO3 < 100mA 3.28 3.35 3.42
V
LDO3 Short-Circuit Current LDO3 = GND
180
mA
LDO3 Bootstrap Switch Threshold
Falling edge of OUT3, rising edge at OUT3 regulation
point
2.80 2.91 3.02
V
LDO3 Bootstrap Switch
Resistance LDO3 to OUT3, VOUT3 = 3.2V 1.5 3.5 Ω
REF Output Voltage No external load
1.980 2.000 2.020
V
REF Load Regulation 0 < ILOAD < 50µA 10 mV
REF Sink Current REF in regulation 10 µA
V+ Operating Supply Current LDO5 switched over to OUT5, 5V SMPS 25 50 µA
V+ Standby Supply Current V+ = 6V to 24V, both SMPSs off, includes ISHDN
150
250 µA
V+ Shutdown Supply Current V+ = 4.5V to 24V 6 15 µA
Quiescent Power Consumption Both SMPSs on, FB3 = FB5 = SKIP = GND, VOUT3 =
3.5V, VOUT5 = 5.3V 34.5 mW
FAULT DETECTION
Overvoltage Trip Threshold FB3 or FB5 with respect to nominal regulation point +8
+11
+14 %
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
4_______________________________________________________________________________________
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)
PARAMETER CONDITIONS
MIN TYP MAX
UNITS
Overvoltage Fault
Propagation Delay FB3 or FB5 delay with 50mV overdrive 10 µs
PGOOD Threshold FB3 or FB5 with respect to nominal output, falling edge,
typical hysteresis = 1% -12
-9.5
-7 %
PGOOD Propagation Delay Falling edge, 50mV overdrive 10 µs
PGOOD Output Low Voltage ISINK = 4mA 0.3 V
PGOOD Leakage Current High state, forced to 5.5V 1 µA
Thermal-Shutdown Threshold
+160
oC
Output Undervoltage
Shutdown Threshold FB3 or FB5 with respect to nominal output voltage 65 70 75 %
Output Undervoltage
Shutdown Blanking Time From ON_ signal 10 22 35 ms
INPUTS AND OUTPUTS
Feedback Input Leakage Current
VFB3 = VFB5 = 2.2V
-200 +40 +200
nA
Low level 0.6
PRO Input Voltage High level 1.5 V
Low level 0.8
Float level 1.7 2.3SKIP Input Voltage
High level 2.4
V
Low level 0.8
TON Input Voltage High level 2.4 V
Clear fault level/SMPS off level 0.8
Delay start level 1.7 2.3ON3, ON5 Input Voltage
SMPS on level 2.4
V
V PRO or VTON = 0 or 5V -1 +1
VON_ = 0 or 5V -2 +2
V SKIP = 0 or 5V -1 +1
V SHDN = 0 or 24V -1 +1
VCS_ = 0 or 5V -2 +2
Input Leakage Current
VILIM3, VILIM5 = 0 or 2V
-0.2 +0.2
µA
Rising edge 1.2 1.6 2.0
SHDN Input Trip Level Falling edge
0.96 1.00 1.04
V
DH_ Gate-Driver
Sink/Source Current DH3, DH5 forced to 2V 2 A
DL_ Gate-Driver Source Current DL3 (source), DL5 (source), forced to 2V 1.7 A
DL_ Gate-Driver Sink Current DL3 (sink), DL5 (sink), forced to 2V 3.3 A
DH_ Gate-Driver On-Resistance BST - LX_ forced to 5V 1.5 4.0 Ω
DL_, high state (pullup) 2.2 5.0
DL_ Gate-Driver On-Resistance DL_, low state (pulldown) 0.6 1.5 Ω
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
_______________________________________________________________________________________ 5
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA= 0°C to +85°C, unless otherwise noted. Typical values are at TA= +25°C.)
PARAMETER CONDITIONS
MIN TYP MAX
UNITS
OUT3, OUT5 Discharge-Mode
On-Resistance 12 40 Ω
OUT3, OUT5 Discharge-Mode
Synchronous Rectifier
Turn-On Level
0.2 0.3 0.4 V
ELECTRICAL CHARACTERISTICS
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, V SHDN = 5V,
TA= -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER CONDITIONS
MIN TYP MAX
UNITS
MAIN SMPS CONTROLLERS
LDO5 in regulation 6 24
V+ Input Voltage Range V+ = LDO5, VOUT5 < 4.41V 4.5 5.5 V
3.3V Output Voltage in
Fixed Mode V+ = 6V to 24V, FB3 = GND, V SKIP = 5V
3.27 3.39
V
V+ = 6V to 24V, FB5 = GND, V SKIP = 5V,
MAX8732A/MAX8734A (TON = VCC)
5V Output Voltage in Fixed Mode
V+ = 7V to 24V, FB5 = GND, V SKIP = 5V,
MAX8733A/MAX8734A (TON = GND)
4.95 5.15
V
Output Voltage in
Adjustable Mode V+ = 6V to 24V, either SMPS
1.97 2.03
V
Output Voltage Adjust Range Either SMPS 2.0 5.5 V
FB3, FB5 Adjustable-Mode
Threshold Voltage Dual-Mode comparator 0.1 0.2 V
Current-Limit Threshold
(Positive, Default)
ILIM_ = VCC, GND - CS_ (MAX8732A/MAX8733A),
GND - LX_ (MAX8734A) 90 110 mV
VILIM_ = 0.5V 40 60
VILIM_ = 1V 90 110
Current-Limit Threshold
(Positive, Adjustable)
GND - CS_
(MAX8732A/MAX8733A),
GND - LX_ (MAX8734A) VILIM_ = 2V
180
220
mV
VOUT5 = 5.05V
1.895 2.315
MAX8732A or MAX8734A
(VTON = 5V) VOUT3 = 3.33V
0.833 1.017
VOUT5 = 5.05V
0.895 1.209
On-Time Pulse Width
MAX8733A or MAX8734A
(VTON = 0) VOUT3 = 3.33V
0.475 0.635
µs
Minimum Off-Time
200
400 ns
INTERNAL REGULATOR AND REFERENCE
LDO5 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO5 < 100mA 4.90 5.10
V
LDO5 Undervoltage-Lockout
Fault Threshold Falling edge of LDO5, hysteresis = 1% 3.7 4.3 V
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
6_______________________________________________________________________________________
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12.0.V, ON3 = ON5 = VCC, V SHDN = 5V,
TA= -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER CONDITIONS
MIN TYP MAX
UNITS
LDO5 Bootstrap Switch Threshold
Falling edge of OUT5, rising edge at OUT5 regulation
point
4.43 4.69
V
LDO5 Bootstrap Switch
Resistance LDO5 to OUT5, VOUT5 = 5V 3.2 Ω
LDO3 Output Voltage
ON3 = ON5 = GND, 6V < V+ < 24V, 0 < ILDO3 < 100mA 3.27 3.43
V
LDO3 Bootstrap Switch Threshold
Falling edge of OUT3, rising edge at OUT3 regulation
point
2.80 3.02
V
LDO3 Bootstrap
Switch Resistance LDO3 to OUT3, VOUT3 = 3.2V 3.5 Ω
REF Output Voltage No external load
1.975 2.025
V
REF Load Regulation 0 < ILOAD < 50µA 10 mV
REF Sink Current REF in regulation 10 µA
V+ Operating Supply Current LDO5 switched over to OUT5, 5V SMPS 50 µA
V+ Standby Supply Current V+ = 6V to 24V, both SMPSs off, includes ISHDN 300 µA
V+ Shutdown Supply Current V+ = 4.5V to 24V 15 µA
Quiescent Power Consumption Both SMPSs on, FB3 = FB5 = SKIP = GND, VOUT3 =
3.5V, VOUT5 = 5.3V 4.5 mW
FAULT DETECTION
Overvoltage Trip Threshold FB3 or FB5 with respect to nominal regulation point +8 +14 %
PGOOD Threshold FB3 or FB5 with respect to nominal output, falling edge,
typical hysteresis = 1% -12 -7 %
PGOOD Output Low Voltage ISINK = 4mA 0.3 V
PGOOD Leakage Current High state, forced to 5.5V 1 µA
Output Undervoltage Shutdown
Threshold FB3 or FB5 with respect to nominal output voltage 65 75 %
Output Undervoltage Shutdown
Blanking Time From ON_ signal 10 40 ms
INPUTS AND OUTPUTS
Feedback Input Leakage Current
VFB3 = VFB5 = 2.2V
-200 +200
nA
Low level 0.6
PRO Input Voltage High level 1.5 V
Low level 0.8
Float level 1.7 2.3
SKIP Input Voltage
High level 2.4
V
Low level 0.8
TON Input Voltage High level 2.4 V
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
_______________________________________________________________________________________ 7
ELECTRICAL CHARACTERISTICS (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12.0.V, ON3 = ON5 = VCC, V SHDN = 5V,
TA= -40°C to +85°C, unless otherwise noted.) (Note 1)
PARAMETER CONDITIONS
MIN
TYP
MAX
UNITS
Clear fault level/SMPS off level 0.8
Delay start level 1.7 2.3ON3, ON5 Input Voltage
SMPS on level 2.4
V
V PRO or VTON = 0 or 5V -1 +1
VON_ = 0 or 5V -1 +1
V SKIP = 0 or 5V -2 +2
V SHDN = 0 or 24V -1 +1
VCS_ = 0 or 5V -2 +2
Input Leakage Current
VILIM3, VILIM5 = 0 or 2V
-0.2 +0.2
µA
Rising edge 1.2 2.0
SHDN Input Trip Level Falling edge
0.96 1.04
V
DH_ Gate-Driver On-Resistance BST - LX_ forced to 5V 4.0 Ω
DL_, high state (pullup) 5.0
DL_ Gate-Driver On-Resistance DL_, low state (pulldown) 1.5 Ω
OUT3, OUT5 Discharge-Mode
On-Resistance 40 Ω
OUT3, OUT5 Discharge-Mode
Synchronous Rectifier
Turn-On Level
0.2 0.4 V
Note 1: Specifications to -40°C are guaranteed by design, not production tested.
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
8_______________________________________________________________________________________
Typical Operating Characteristics
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA= +25°C, unless otherwise noted.)
100
0
0.001 0.01 0.1 1 10
MAX8732A
5V OUTPUT EFFICIENCY
vs. LOAD CURRENT
20
MAX8732A/3A/4A toc01
LOAD CURRENT (A)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
VIN = 7V
ON5 = VCC
ON3 = GND
PFM MODE
PWM MODE
ULTRASONIC MODE
100
0
0.001 0.01 0.1 1 10
MAX8732A
5V OUTPUT EFFICIENCY
vs. LOAD CURRENT
20
MAX8732A/3A/4A toc02
LOAD CURRENT (A)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
VIN = 12V
ON5 = VCC
ON3 = GND
PFM MODE
PWM MODE
ULTRASONIC MODE
100
0
0.001 0.01 0.1 1 10
MAX8732A
5V OUTPUT EFFICIENCY
vs. LOAD CURRENT
20
MAX8732A/3A/4A toc03
LOAD CURRENT (A)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
VIN = 24V
ON5 = VCC
ON3 = GND
PFM MODE
PWM MODE
ULTRASONIC MODE
100
0
0.001 0.01 0.1 1 10
MAX8733A
3.3V OUTPUT EFFICIENCY
vs. LOAD CURRENT
20
MAX8732A/3A/4A toc04
LOAD CURRENT (A)
EFFICIENCY (%)
40
60
80
70
50
30
10
90
ON5 = VCC
ON3 = VCC
VIN = 7V
VIN = 12V
VIN = 24V
PFM MODE
PWM MODE
ULTRASONIC MODE
1
10
100
0.1
710 13 16 19 22 25
MAX8732A
NO-LOAD BATTERY CURRENT
vs. INPUT VOLTAGE
MAX8732A/3A/4A toc05
INPUT VOLTAGE (V)
BATTERY CURRENT (mA)
PWM MODE
ULTRASONIC MODE
PFM MODE
1
10
100
0.1
710 13 16 19 22 25
MAX8733A
NO-LOAD BATTERY CURRENT
vs. INPUT VOLTAGE
MAX8732A/3A/4A toc06
INPUT VOLTAGE (V)
BATTERY CURRENT (mA)
PWM MODE
ULTRASONIC MODE
PFM MODE
170
176
174
172
180
178
188
186
184
182
190
7101316192225
STANDBY INPUT CURRENT
vs. INPUT VOLTAGE
MAX8732A/3A/4A toc07
INPUT VOLTAGE (V)
STANDBY INPUT CURRENT (µA)
MAX8732A
MAX8733A
5.0
6.5
6.0
5.5
7.5
7.0
9.5
9.0
8.5
8.0
10.0
7101316192225
SHUTDOWN INPUT CURRENT
vs. INPUT VOLTAGE
MAX8732A/3A/4A toc08
INPUT VOLTAGE (V)
SHUTDOWN INPUT CURRENT (µA)
MAX8733A
MAX8732A
MAX8732A
5V OUTPUT SWITCHING FREQUENCY
vs. LOAD CURRENT MAX8732A
MAX8732A/3A/4A toc09
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
10.10.01
25
50
75
100
125
150
175
200
225
250
0
0.001 10
VIN = 7V
PWM MODE
PFM MODE
ULTRASONIC MODE
Idle Mode is a trademark of Maxim Integrated Products, Inc.
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
_______________________________________________________________________________________ 9
Typical Operating Characteristics (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA= +25°C, unless otherwise noted.)
3.3V OUTPUT SWITCHING FREQUENCY
vs. LOAD CURRENT (MAX8732A)
MAX8732A/3A/4A toc10
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
10.10.01
40
80
120
160
200
240
280
320
360
0
0.001 10
VIN = 7V
PWM MODE
PFM MODE
ULTRASONIC MODE
250
0
0.001 0.01 0.1 1 10
MAX8732A
5V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT
50
MAX8732A/3A/4A toc11
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
100
150
200
175
125
75
25
225
PWM MODE
PFM MODE
VIN = 24V
ULTRASONIC MODE
360
0
0.001 0.01 0.1 1 10
MAX8732A
3.3V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT
80
MAX8732A/3A/4A toc12
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
160
240
320
280
200
120
40
PWM MODE
PFM MODE
VIN = 24V
ULTRASONIC MODE
450
0
0.001 0.01 0.1 1 10
MAX8733A
5V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT
100
MAX8732A/3A/4A toc13
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
200
300
400
350
250
150
50
VIN = 7V
PWM MODE
PFM MODE
ULTRASONIC MODE
110
0
100
200
400
300
500
550
0.001 0.01 0.1
MAX8733A
3.3V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT
MAX8732A/3A/4A toc14
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
450
350
250
150
50
PWM MODE
PFM MODE
VIN = 7V
ULTRASONIC MODE
450
0
0.001 0.01 0.1 1 10
MAX8733A
5V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT
100
MAX8732A/3A/4A toc15
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
200
300
400
350
250
150
50
PWM MODE
PFM MODE
VIN = 24V
ULTRASONIC MODE
110
0
100
200
400
300
500
550
0.001 0.01 0.1
MAX8733A
3.3V OUTPUT SWITCHING
FREQUENCY vs. LOAD CURRENT
MAX8732A/3A/4A toc16
LOAD CURRENT (A)
SWITCHING FREQUENCY (kHz)
450
350
250
150
50
PWM MODE
PFM MODE
VIN = 24V
ULTRASONIC MODE
MAX8732A
OUT5 VOLTAGE REGULATION
vs. LOAD CURRENT
MAX8732A/3A/4A toc17
LOAD CURRENT (A)
OUTPUT VOLTAGE (V)
10.10.01
5.07
5.09
5.11
5.13
5.15
5.17
5.19
5.05
0.001 10
ULTRASONIC
IDLE MODE
FORCED-PWM
MAX8732A
OUT3 VOLTAGE REGULATION
vs. LOAD CURRENT
MAX8732A/3A/4A toc18
LOAD CURRENT (A)
OUTPUT VOLTAGE (V)
10.10.01
3.34
3.35
3.36
3.37
3.38
3.39
3.41
3.40
3.33
0.001 10
ULTRASONIC
IDLE MODE
FORCED-PWM
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
10 ______________________________________________________________________________________
Typical Operating Characteristics (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA= +25°C, unless otherwise noted.)
4.95
4.96
4.97
4.98
4.99
5.00
020304010 50 60 70 9080 100
LDO5 REGULATOR OUTPUT VOLTAGE
vs. OUTPUT CURRENT
MAX8732A/3A/4A toc19
LDO5 OUTPUT CURRENT (mA)
LDO5 OUTPUT VOLTAGE (V)
3.330
3.334
3.332
3.338
3.336
3.342
3.340
3.344
3.348
3.346
3.350
020304010 50 60 70 9080 100
LDO3 REGULATOR OUTPUT VOLTAGE
vs. OUTPUT CURRENT
MAX8732A/3A/4A toc20
LDO3 OUTPUT CURRENT (mA)
LDO3 OUTPUT VOLTAGE (V)
1.995
1.997
1.996
2.000
1.999
1.998
2.001
2.002
2.004
2.003
2.005
-10 10 20030405060708090100
REFERENCE VOLTAGE
vs. OUTPUT CURRENT
MAX8732A/3A/4A toc21
IREF (µA)
VREF (V)
0
0
0
10V
REF, LDO3, AND LDO5 POWER-UP
MAX8732A/3A/4A toc22
400µs/div
V+
10V/div
LDO5
2V/div
LDO3
2V/div
REF
1V/div
0
0
0
0
5V
DELAYED-START WAVEFORMS
(ON3 = REF)
MAX8732A/3A/4A toc23
100µs/div
ON5
5V/div
OUT5
2V/div
OUT3
2V/div
0
0
0
5V
DELAYED-START WAVEFORMS
(ON5 = REF)
MAX8732A/3A/4A toc24
100µs/div
ON3
5V/div
OUT5
2V/div
OUT3
2V/div
0
5A
0
3.3V
0
5A
SOFT-START WAVEFORMS
MAX8732A/3A/4A toc25
200µs/div
IL5
5A/div
OUT3
5V/div
OUT5
5V/div
IL3
5A/div
0
5V
0
3.3V
0
5V
0
5V
SHUTDOWN WAVEFORMS
MAX8732A/3A/4A toc26
10ms/div
ON3
5V/div
OUT5
5V/div
DL3
5V/div
OUT3
5V/div
0
5V SWITCHING
0
4A
1A
5V
MAX8732A/MAX8734A (TON = VCC)
5V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX8732A/3A/4A toc27
20µs/div
VOUT,
AC-
COUPLED
100mV/div
INDUCTOR
CURRENT
2A/div
DL5
5V/div
5V
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 11
0
4A
1A
5V
MAX8733A/MAX8734A (TON = GND)
5V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX8732A/3A/4A toc28
10µs/div
VOUT,
AC-
COUPLED
100mV/div
INDUCTOR
CURRENT
2A/div
DL5
5V/div
5V
0
4A
1A
3.3V
MAX8732A/MAX8734A (TON = VCC)
3.3V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX8732A/3A/4A toc29
20µs/div
VOUT,
AC-
COUPLED
100mV/div
INDUCTOR
CURRENT
2A/div
DL3
5V/div
5V
0
4A
1A
3.3V
MAX8733A/MAX8734A (TON = GND)
3.3V PWM-MODE
LOAD TRANSIENT RESPONSE
MAX8732A/3A/4A toc30
10µs/div
VOUT,
AC-
COUPLED
100mV/div
INDUCTOR
CURRENT
2A/div
DL3
5V/div
5V
MAX8733A
5V OUTPUT EFFICIENCY
vs. LOAD CURRENT
MAX8732A/3A/4A toc31
LOAD CURRENT (A)
EFFICIENCY (%)
10.10.01
10
20
30
40
50
60
70
80
90
100
0
0.001 10
ON5 = VCC
ON3 = GND
VIN = 7V
VIN = 12V
VIN = 24V
PFM MODE
PWM MODE
ULTRASONIC
MODE
MAX8733A
3.3V OUTPUT EFFICIENCY
vs. LOAD CURRENT
MAX8732A/3A/4A toc32
LOAD CURRENT (A)
EFFICIENCY (%)
10.10.01
10
20
30
40
50
60
70
80
90
100
0
0.001 10
ON5 = VCC
ON3 = VCC
VIN = 7V
VIN = 12V
VIN = 24V
PFM MODE
PWM MODE
ULTRASONIC
MODE
Typical Operating Characteristics (continued)
(Circuit of Figure 1 and Figure 2, no load on LDO5, LDO3, OUT3, OUT5, and REF, V+ = 12V, ON3 = ON5 = VCC, SHDN = V+,
RCS = 7mΩ, VILIM_ = 0.5V, TA= +25°C, unless otherwise noted.)
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
12 ______________________________________________________________________________________
Pin Description
PIN
MAX8732A
MAX8733A MAX8734A NAME
FUNCTION
1—CS3
3.3V SMPS Current-Sense Input. Connect CS3 to a current-sensing resistor from the source
of the synchronous rectifier to GND. The voltage at ILIM3 determines the current-limit
threshold (see the Current-Limit Circuit (ILIM_) section).
—1
N.C.
No Connection. Not internally connected.
22
PGOOD
Power-Good Output. PGOOD is an open-drain output that is pulled low if either output is
disabled or is more than 10% below its nominal value.
33
ON3
3.3V SMPS Enable Input. The 3.3V SMPS is enabled if ON3 is greater than the SMPS on level
and disabled if ON3 is less than the SMPS off level. If ON3 is connected to REF, the 3.3V
SMPS starts after the 5V SMPS reaches regulation (delay start). Drive ON3 below the clear
fault level to reset the fault latches.
44
ON5
5V SMPS Enable Input. The 5V SMPS is enabled if ON5 is greater than the SMPS on level and
disabled if ON5 is less than the SMPS off level. If ON5 is connected to REF, the 5V SMPS
starts after the 3.3V SMPS reaches regulation (delay start). Drive ON5 below the clear fault
level to reset the fault latches.
55
ILIM3
3.3V SMPS Current-Limit Adjustment. The GND-LX current-limit threshold defaults to 100mV if
ILIM3 is connected to VCC. In adjustable mode, the current-limit threshold is 1/10 the voltage
seen at ILIM3 over a 0.5V to 3V range. The logic threshold for switchover to the 100mV
default value is approximately VCC - 1V. Connect ILIM3 to REF for a fixed 200mV threshold.
66
SHDN
Shutdown Control Input. The device enters its 6µA supply current shutdown mode if
V SHDN is less than the SHDN input falling-edge trip level and does not restart until V SHDN is
greater than the SHDN input rising-edge trip level. Connect SHDN to V+ for automatic
startup. SHDN can be connected to V+ through a resistive voltage-divider to implement a
programmable undervoltage lockout.
77FB3 3.3V SMPS Feedback Input. Connect FB3 to GND for fixed 3.3V operation. Connect FB3 to a
resistive voltage-divider from OUT3 to GND to adjust the output from 2V to 5.5V.
88REF
2V Reference Output. Bypass to GND with a 0.22µF (min) capacitor. REF can source up to
100µA for external loads. Loading REF degrades FB_ and output accuracy according to the
REF load-regulation error.
99FB5 5V SMPS Feedback Input. Connect FB5 to GND for fixed 5V operation. Connect FB5 to a
resistive voltage-divider from OUT5 to GND to adjust the output from 2V to 5.5V.
10 10
PRO
Overvoltage and Undervoltage Fault Protection Enable/Disable. Connect PRO to VCC to
disable undervoltage, overvoltage protection, and discharge mode (DL = low in shutdown).
Connect PRO to GND to enable undervoltage and overvoltage protection (see the Fault
Protection section), and output discharge mode.
11 11
ILIM5
5V SMPS Current-Limit Adjustment. The GND-LX current-limit threshold defaults to 100mV if
ILIM5 is connected to VCC. In adjustable mode, the current-limit threshold is 1/10 the voltage
seen at ILIM5 over a 0.5V to 3V range. The logic threshold for switchover to the 100mV
default value is approximately VCC - 1V. Connect ILIM5 to REF for a fixed 200mV threshold.
12 12
SKIP
Low-Noise Mode Control. Connect SKIP to GND for normal Idle-Mode (pulse-skipping)
operation or to VCC for PWM mode (fixed frequency). Connect to REF or leave floating for
ultrasonic mode (pulse skipping, 25kHz min).
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 13
Pin Description (continued)
PIN
MAX8732A
MAX8733A
MAX8734A NAME
FUNCTION
13 — CS5
5V SMPS Current-Sense Input. Connect CS5 to a current-sensing resistor from the source of
the synchronous rectifier to GND. The voltage at ILIM5 determines the current-limit threshold
(see the Current-Limit Circuit (ILIM_) section).
—13
TON
Frequency Select Input. Connect to VCC for 200kHz/300kHz operation and to GND for
400kHz/500kHz operation (5V/3.3V SMPS switching frequencies, respectively).
14 14
BST5
Boost Flying Capacitor Connection for 5V SMPS. Connect to an external capacitor and diode
according to the typical application circuits (Figure 1 and Figure 2). See the MOSFET Gate
Drivers (DH_, DL_) section.
15 15 LX5 Inductor Connection for 5V SMPS. LX5 is the internal lower supply rail for the DH5 high-side
gate driver. LX5 is the current-sense input for the 5V SMPS (MAX8734A only).
16 16
DH5
High-Side MOSFET Floating Gate-Driver Output for 5V SMPS. DH5 swings from LX5 to BST5.
17 17 VCC Analog Supply Voltage Input for PWM Core. Connect VCC to the system supply voltage with a
series 50Ω resistor. Bypass to GND with a 1µF ceramic capacitor.
18 18
LDO5
5V Linear-Regulator Output. LDO5 is the gate-driver supply for the external MOSFETs. LDO5
can provide a total of 100mA, including MOSFET gate-drive requirements and external loads.
The internal load depends on the choice of MOSFET and switching frequency (see the
Reference and Linear Regulators (REF, LDO5, and LDO3) section). If OUT5 is greater than
the LDO5 bootstrap switch threshold, the LDO5 regulator shuts down and the LDO5 pin
connects to OUT5 through a 1.4Ω switch. Bypass LDO5 with a minimum of 4.7µF. Use an
additional 1µF per 5mA of load.
19 19 DL5 5V SMPS Synchronous Rectifier Gate-Drive Output. DL5 swings between GND and LDO5.
20 20 V+
Power-Supply Input. V+ powers the LDO5/LDO3 linear regulators and is also used for the
Quick-PWM on-time, one-shot circuits. Connect V+ to the battery input and bypass with a
0.1µF capacitor.
21 21
OUT5
5V SMPS Output Voltage-Sense Input. Connect to the 5V SMPS output. OUT5 is an input to
the Quick-PWM on-time, one-shot circuit. It also serves as the 5V feedback input in fixed-
voltage mode. If OUT5 is greater than the LDO5 bootstrap-switch threshold, the LDO5 linear
regulator shuts down and LDO5 connects to OUT5 through a 1.4Ω switch.
22 22
OUT3
3.3V SMPS Output Voltage-Sense Input. Connect to the 3.3V SMPS output. OUT3 is an input
to the Quick-PWM on-time, one-shot circuit. It also serves as the 3V feedback input in fixed-
voltage mode. If OUT3 is greater than the LDO3 bootstrap-switch threshold, the LDO3 linear
regulator shuts down and LDO3 connects to OUT3 through a 1.5Ω switch.
23 23
GND
Analog and Power Ground
24 24 DL3 3.3V SMPS Synchronous-Rectifier Gate-Drive Output. DL3 swings between GND and LDO5.
25 25
LDO3
3.3V Linear-Regulator Output. LDO3 powers up after REF is in regulation. LDO3 can provide
a total of 100mA to external loads. If OUT3 is greater than the LDO3 bootstrap-switch
threshold, the LDO3 regulator shuts down and the LDO3 pin connects to OUT3 through a
1.5Ω switch. Bypass LDO3 with a minimum of 4.7µF. Use an additional 1µF per 5mA of load.
26 26
DH3
High-Side MOSFET Floating Gate-Driver Output for 3.3V SMPS. DH3 swings from LX3 to
BST3.
27 27 LX3 Inductor Connection for 3.3V SMPS. LX3 is the current-sense input for the 3.3V SMPS
(MAX8734A only).
28 28
BST3
Boost Flying Capacitor Connection for 3.3V SMPS. Connect to an external capacitor and
diode according to the typical application circuits (Figure 1 and Figure 2). See the MOSFET
Gate Drivers (DH_, DL_) section.
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
14 ______________________________________________________________________________________
Typical Application Circuits
The typical application circuits (Figures 1 and 2) gener-
ate the 5V/5A and 3.3V/5A main supplies in a notebook
computer. The input supply range is 7V to 24V. Table 1
lists component suppliers.
Detailed Description
The MAX8732A/MAX8733A/MAX8734A dual-buck,
BiCMOS, switch-mode power-supply controllers gener-
ate logic supply voltages for notebook computers. The
MAX8732A/MAX8733A/MAX8734A are designed pri-
marily for battery-powered applications where high effi-
ciency and low-quiescent supply current are critical.
The MAX8732A is optimized for highest efficiency with a
5V/200kHz SMPS and a 3.3V/300kHz SMPS, while the
MAX8732A
MAX8733A
VIN 7V TO 24V
10µF1/2
D1
0.1µF
0.1µF
N1
FDS6612A
L5
5V
C5
D3
EP10QY03
N2
IRF7811AV
ON
OFF
VCC
REF
0.22µF4.7µF
3.3V ALWAYS ON
1MΩ
100kΩ
VCC
50Ω
1µF
5V ALWAYS ON
4.7µF
10µF
1µF
10µF
1/2
D1
VCC
CMPSH-3A
0.1µF
N3
FDS6612A
L3 3.3V
C3
D2
EP10QY03
N4
IRF7811AV
LDO5 ILIM3 VCC
V+
BST5
DH5
LX5
CS5
DL5
OUT5
FB5
PGOOD
FB3
OUT3
DL3
LX3
DH3
BST3
ILIM5
ON5
ON3
GND
SHDN
REF LDO3
PRO
SKIP
400kHz/500kHz
MAX8733A
200kHz/300kHz
MAX8732A
5V/3.3V SMPS
SWITCHING FREQUENCY
L3
L5
C3
C5
4.7µH
7.6µH
470µF
330µF
3.0µH
5.6µH
220µF
150µF
FREQUENCY-DEPENDENT COMPONENTS
10Ω
RCS5
20mΩRCS3
20mΩ
10Ω
CS3
Figure 1. MAX8732A/MAX8733A Typical Application Circuit
MANUFACTURER PHONE FAX
Central Semiconductor
516-435-1110
516-435-1824
Dale-Vishay
402-564-3131
402-563-6418
Fairchild
408-721-2181
408-721-1635
International Rectifier
310-322-3331
310-322-3332
NIEC (Nihon)
805-843-7500
847-843-2798
Sanyo
619-661-6835
619-661-1055
Sprague
603-224-1961
603-224-1430
Sumida
847-956-0666
847-956-0702
Taiyo Yuden
408-573-4150
408-573-4159
TDK
847-390-4461
847-390-4405
Table 1. Component Suppliers
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 15
*OPTIONAL CAPACITANCE BETWEEN
LX AND PGND (CLOSE TO THE IC) ONLY
REQUIRED FOR ULTRASONIC MODE
MAX8734A
VIN 7V TO 24V
10µF1/2
D1
0.1µF
0.1µF
N1
FDS6612A
L5
5V
C5
D3
EP10QY03
N2
IRF7811AV
ON
OFF
VCC
REF
0.22µF4.7µF
3.3V ALWAYS ON
1MΩ
100kΩ
VCC
50Ω
1µF
5V ALWAYS ON
4.7µF
10µF
1µF
10µF
1/2
D1
CMPSH-3A
0.1µF
N3
FDS6612A
L3 3.3V
C3
D2
EP10QY03
N4
IRF7811AV
LDO5 ILIM3 VCC
VCC
V+
BST5
DH5
LX5
DL5
OUT5
FB5
PGOOD
FB3
OUT3
DL3
TON
LX3
DH3
BST3
ILIM5
ON5
ON3
GND
SHDN
REF LDO3
PRO
SKIP
400kHz/500kHz
TON = GND
200kHz/300kHz
TON = VCC
5V/3.3V SMPS
SWITCHING FREQUENCY
L3
L5
C3
C5
4.7µH
7.6µH
470µF
330µF
3.0µH
5.6µH
220µF
150µF
FREQUENCY-DEPENDENT COMPONENTS
SEE
TABLE
10Ω
470pF*
10Ω
470pF*
Figure 2. MAX8734A Typical Application Circuit
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
16 ______________________________________________________________________________________
MAX8733A is optimized for “thin and light” applications with
a 5V/400kHz SMPS and a 3.3V/500kHz SMPS. The
MAX8734A provides a pin-selectable switching frequency,
allowing either 200kHz/300kHz or 400kHz/500kHz operation
of the 5V/3.3V SMPSs, respectively.
Light-load efficiency is enhanced by automatic Idle-
Mode operation, a variable-frequency pulse-skipping
mode that reduces transition and gate-charge losses.
Each step-down, the power-switching circuit consists of
two n-channel MOSFETs, a rectifier, and an LC output fil-
ter. The output voltage is the average AC voltage at the
switching node, which is regulated by changing the duty
cycle of the MOSFET switches. The gate-drive signal to
the n-channel, high-side MOSFET must exceed the
battery voltage, and is provided by a flying-capacitor
boost circuit that uses a 100nF capacitor connected
to BST_.
MAX8732A
MAX8733A
MAX8734A
LDO5
DL3
CS3
(MAX8732A/
MAX8733A)
ILIM3
FB3
OUT3
LDO3
ON3
ON5
SHDN
PRO
2.91V
3V
LINEAR
REG
POWER-ON SEQUENCE/
CLEAR FAULT LATCH
EN3
THERMAL
SHUTDOWN
5V
LINEAR
REG
2V
REFERENCE
3.3V
SMPS PWM
CONTROLLER
5V
SMPS PWM
CONTROLLER
4.56V
PGOOD3
PGOOD5
PGOOD
LDO5
BST5
DH5
LX5
EN5
DL5
CS5
(MAX8732A/
MAX8733A)
ILIM5
FB5
OUT5
LDO5
VCC
REF
V+
GND
TON
(MAX8734 ONLY)
BST3
DH3
LX3
Figure 3. Detailed Functional Diagram
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 17
Each PWM controller consists of a Dual-Mode feedback
network and multiplexer, a multi-input PWM comparator,
high-side and low-side gate drivers, and logic. The
MAX8732A/MAX8733A/MAX8734A contain fault-protection
circuits that monitor the main PWM outputs for undervolt-
age and overvoltage conditions. A power-on sequence
block controls the power-up timing of the main PWMs and
monitors the outputs for undervoltage faults. The
MAX8732A/MAX8733A/MAX8734A include 5V and 3.3V
linear regulators. Bias generator blocks include the 5V
(LDO5) linear regulator, 2V precision reference, and auto-
matic bootstrap switchover circuit.
ON-TIME
COMPUTE
tON
tOFF
TRIG
TRIG
ONE SHOT
ONE SHOT
Q
Q
Q
R
S
ERROR
AMPLIFIER
CURRENT
LIMIT
ZERO
CROSSING
Q
R
S
FAULT
LATCH
20ms
BLANKING
OUT
REF
ILIM_
CS_ (MAX8732A/8733A)
LX_ (MAX8734A)
SKIP
OUT_
FB_
0.15V
PRO
0.9 x VREF
1.1 x VREF
0.7 x VREF
OV_FAULT
UV_FAULT
PGOOD
TO DL_ DRIVER
TO DH_ DRIVER
Σ
TON (MAX8734A)
V+
Figure 4. PWM Controller (One Side Only)
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
18 ______________________________________________________________________________________
These internal blocks are not powered directly from the
battery. Instead, the 5V (LDO5) linear regulator steps
down the battery voltage to supply both internal circuit-
ry and the gate drivers. The synchronous-switch gate
drivers are directly powered from LDO5, while the high-
side switch gate drivers are indirectly powered from
LDO5 through an external diode-capacitor boost cir-
cuit. An automatic bootstrap circuit turns off the 5V lin-
ear regulator and powers the device from OUT5 when
OUT5 is above 4.56V.
Free-Running, Constant On-Time PWM
Controller with Input Feed-Forward
The Quick-PWM control architecture is a pseudo-fixed-
frequency, constant on-time, current-mode type with
voltage feed-forward. The Quick-PWM control architec-
ture relies on the output ripple voltage to provide the
PWM ramp signal; thus, the output filter capacitor’s
ESR acts as a current-feedback resistor. The high-side
switch on-time is determined by a one-shot whose peri-
od is inversely proportional to input voltage and directly
proportional to output voltage. Another one-shot sets a
minimum off-time (300ns typ). The on-time, one-shot
triggers when the following conditions are met: the error
comparator is low, the synchronous rectifier current is
below the current-limit threshold, and the minimum off-
time one-shot has timed out.
On-Time, One-Shot (tON)
Each PWM core includes a one-shot that sets the high-
side switch on-time for each controller. Each fast, low-
jitter, adjustable one-shot includes circuitry that varies
the on-time in response to battery and output voltage.
The high-side switch on-time is inversely proportional to
the battery voltage as measured by the V+ input, and
proportional to the output voltage. This algorithm results
in a nearly constant switching frequency despite the
lack of a fixed-frequency clock generator. The benefit
of a constant switching frequency is the frequency can
be selected to avoid noise-sensitive frequency regions:
See Table 2 for approximate K-factors. The constant
0.075V is an approximation to account for the expected
drop across the synchronous-rectifier switch. Switching
frequency increases as a function of load current due
to the increasing drop across the synchronous rectifier,
which causes a faster inductor-current discharge ramp.
On-times translate only roughly to switching frequen-
cies. The on-times guaranteed in the Electrical
Characteristics are influenced by switching delays in
the external high-side power MOSFET. Also, the dead-
time effect increases the effective on-time, reducing the
switching frequency. It occurs only in PWM mode (SKIP
= VCC) and during dynamic output voltage transitions
when the inductor current reverses at light or negative
load currents. With reversed inductor current, the
inductor’s EMF causes LX to go high earlier than nor-
mal, extending the on-time by a period equal to the DH-
rising dead time.
For loads above the critical conduction point, the actual
switching frequency is:
where VDROP1 is the sum of the parasitic voltage drops
in the inductor discharge path, including synchronous
rectifier, inductor, and PC board resistances; VDROP2 is
the sum of the parasitic voltage drops in the charging
path, including high-side switch, inductor, and PC
board resistances, and tON is the on-time calculated by
the MAX8732A/MAX8733A/MAX8734A.
Automatic Pulse-Skipping Switchover
(Idle Mode)
In Idle Mode (SKIP = GND), an inherent automatic
switchover to PFM takes place at light loads. This
switchover is affected by a comparator that truncates
the low-side switch on-time at the inductor current’s
zero crossing. This mechanism causes the threshold
between pulse-skipping PFM and nonskipping PWM
operation to coincide with the boundary between con-
fVV
tV V
OUT DROP
ON DROP
=+
++
()
1
2
tKV V
V
ON OUT
=+
()
+
0 075.
SMPS SWITCHING FREQUENCY
(kHz) K-FACTOR (µs) APPROXIMATE K-
FACTOR ERROR (%)
MAX8732A/MAX8734A (tON = VCC), 5V 200 5.0 ±10
MAX8732A/MAX8734A (tON = VCC), 3.3V 300 3.3 ±10
MAX8733A/MAX8734A (tON = GND), 5V 400 2.5 ±10
MAX8733A/MAX8734A (tON = GND), 3.3V 500 2.0 ±10
Table 2. Approximate K-Factor Errors
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 19
tinuous and discontinuous inductor-current operation
(also known as the critical conduction point):
where K is the on-time scale factor (see the On-Time
One-Shot (tON)section). The load-current level at which
PFM/PWM crossover occurs, ILOAD(SKIP), is equal to 1/2
the peak-to-peak ripple current, which is a function of the
inductor value (Figure 5). For example, in the MAX8732A
Typical Application Circuit with VOUT2 = 5V, V+ = 12V,
L = 7.6µH, and K = 5µs, switchover to pulse-skipping
operation occurs at ILOAD = 0.96A or about 1/5 full load.
The crossover point occurs at an even lower value if a
swinging (soft-saturation) inductor is used.
The switching waveforms may appear noisy and asyn-
chronous when light loading causes pulse-skipping
operation, but this is a normal operating condition that
results in high light-load efficiency. Trade-offs in PFM
noise vs. light-load efficiency are made by varying the
inductor value. Generally, low inductor values produce
a broader efficiency vs. load curve, while higher values
result in higher full-load efficiency (assuming that the
coil resistance remains fixed) and less output voltage
ripple. Penalties for using higher inductor values
include larger physical size and degraded load-tran-
sient response (especially at low input-voltage levels).
DC output accuracy specifications refer to the trip level of
the error comparator. When the inductor is in continuous
conduction, the output voltage has a DC regulation higher
than the trip level by 50% of the ripple. In discontinuous
conduction (SKIP = GND, light load), the output voltage
has a DC regulation higher than the trip level by approxi-
mately 1.5% due to slope compensation.
Forced-PWM Mode
The low-noise, forced-PWM (SKIP = VCC) mode dis-
ables the zero-crossing comparator, which controls the
low-side switch on-time. Disabling the zero-crossing
detector causes the low-side, gate-drive waveform to
become the complement of the high-side, gate-drive
waveform. The inductor current reverses at light loads
as the PWM loop strives to maintain a duty ratio of
VOUT/V+. The benefit of forced-PWM mode is to keep
the switching frequency fairly constant, but it comes at
a cost: the no-load battery current can be 10mA to
50mA, depending on switching frequency and the
external MOSFETs.
Forced-PWM mode is most useful for reducing audio-
frequency noise, improving load-transient response,
providing sink-current capability for dynamic output
voltage adjustment, and improving the cross-regulation
of multiple-output applications that use a flyback trans-
former or coupled inductor.
Enhanced Ultrasonic Mode
(25kHz (min) Pulse Skipping)
Leaving SKIP unconnected or connecting SKIP to REF
activates a unique pulse-skipping mode with a mini-
mum switching frequency of 25kHz. This ultrasonic
pulse-skipping mode eliminates audio-frequency mod-
ulation that would otherwise be present when a lightly
loaded controller automatically skips pulses. In ultra-
sonic mode, the controller automatically transitions to
fixed-frequency PWM operation when the load reaches
the same critical conduction point (ILOAD(SKIP)) that
occurs when normally pulse skipping.
An ultrasonic pulse occurs when the controller detects
that no switching has occurred within the last 28µs.
Once triggered, the ultrasonic controller pulls DL high,
turning on the low-side MOSFET to induce a negative
inductor current. After the inductor current reaches the
negative ultrasonic current threshold, the controller
turns off the low-side MOSFET (DL pulled low) and trig-
gers a constant on-time (DH driven high). When the on-
time has expired, the controller reenables the low-side
MOSFET until the controller detects that the inductor
current dropped below the zero-crossing threshold.
Starting with a DL pulse greatly reduces the peak out-
put voltage when compared to starting with a DH pulse.
The output voltage at the beginning of the ultrasonic
pulse determines the negative ultrasonic current thresh-
old, resulting in the following equation:
where VFB > VREF and RON is the on-resistance of the
synchronous rectifier (MAX8734A) or the current-sense
resistor value (MAX8732A/MAX8733A).
VIRVV
ISONIC L ON REF FB
==−
()
× 058.
IKV
L
VV
V
LOAD SKIP OUT OUT
() __
=×
×
+−
+
2
INDUCTOR CURRENT
ILOAD = IPEAK / 2
ON-TIME0 TIME
-IPEAK
L
V+ - VOUT
∆i
∆t=
Figure 5. Pulse-Skipping/Discontinuous Crossover Point
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
20 ______________________________________________________________________________________
Reference and Linear Regulators
(REF, LDO5, and LDO3)
The 2V reference (REF) is accurate to ±1% over tem-
perature, making REF useful as a precision system
reference. Bypass REF to GND with a 0.22µF (min)
capacitor. REF can supply up to 100µA for external
loads. However, if extremely accurate specifications for
both the main output voltages and REF are essential,
avoid loading REF. Loading REF reduces the LDO5,
LDO3, OUT5, and OUT3 output voltages slightly
because of the reference load-regulation error.
Two internal regulators produce 5V (LDO5) and 3.3V
(LDO3). LDO5 provides gate drive for the external
MOSFETs and powers the PWM controller, logic, refer-
ence, and other blocks within the device. The LDO5
regulator supplies a total of 100mA for internal and
external loads, including MOSFET gate drive, which
typically varies from 10mA to 50mA, depending on
switching frequency and the external MOSFETs. LDO3
powers up when the reference (REF) is in regulation,
and supplies up to 100mA for external loads. Bypass
LDO5 and LDO3 with a minimum 4.7µF load; use an
additional 1µF per 5mA of internal and external load.
When the 5V main output voltage is above the LDO5
bootstrap-switchover threshold, an internal 1.4Ωp-chan-
nel MOSFET switch connects OUT5 to LDO5 while simul-
taneously shutting down the LDO5 linear regulator.
Similarly, when the 3.3V main output voltage is above the
LDO3 bootstrap-switchover threshold, an internal 1.5Ω
p-channel MOSFET switch connects OUT3 to LDO3 while
simultaneously shutting down the LDO3 linear regulator.
These actions bootstrap the device, powering the internal
circuitry and external loads from the output SMPS volt-
ages, rather than through linear regulators from the bat-
tery. Bootstrapping reduces power dissipation due to
gate charge and quiescent losses by providing power
from a 90%-efficient switch-mode source, rather than
from a much-less-efficient linear regulator.
Current-Limit Circuit (ILIM_)
The current-limit circuit employs a “valley” current-sens-
ing algorithm. The MAX8734A uses the on-resistance of
the synchronous rectifier, while the MAX8732A/
MAX8733A use a discrete resistor in series with the
source of the synchronous rectifier as a current-sensing
element. If the magnitude of the current-sense signal at
CS_ (MAX8732A/MAX8733A)/LX_ (MAX8734A) is above
the current-limit threshold, the PWM is not allowed to initi-
ate a new cycle (Figure 7). The actual peak current is
greater than the current-limit threshold by an amount
equal to the inductor ripple current. Therefore, the exact
current-limit characteristic and maximum load capability
are a function of the current-limit threshold, inductor
value, and input and output voltage.
For the MAX8732A/MAX8733A, connect CS_ to the
junction of the synchronous rectifier source and a cur-
rent-sense resistor to GND. With a current-limit threshold
of 100mV, the accuracy is approximately ±7%. Using a
lower current-sense threshold results in less accuracy.
The current-sense resistor only dissipates power when
the synchronous rectifier is on.
For lower power dissipation, the MAX8734A uses the
on-resistance of the synchronous rectifier as the cur-
rent-sense element. Use the worst-case maximum
value for RDS(ON) from the MOSFET data sheet, and
add some margin for the rise in RDS(ON) with tempera-
ture. A good general rule is to allow 0.5% additional
resistance for each °C of temperature rise. The current
limit varies with the on-resistance of the synchronous
rectifier. The reward for this uncertainty is robust, loss-
less overcurrent sensing. When combined with the
ON-TIME (tON)
ZERO-CROSSING
DETECTION
ISONIC
0
40µs (MAX)
INDUCTOR
CURRENT
Figure 6. Ultrasonic Current Waveforms
INDUCTOR CURRENT
ILIMIT
ILOAD
0 TIME
-IPEAK
Figure 7. “Valley” Current-Limit Threshold Point
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 21
undervoltage-protection circuit, this current-limit
method is effective in almost every circumstance.
A negative current limit prevents excessive reverse
inductor currents when VOUT sinks current. The nega-
tive current-limit threshold is set to approximately 120%
of the positive current limit and therefore tracks the
positive current limit when ILIM_ is adjusted.
The current-limit threshold is adjusted with an external
voltage-divider at ILIM_. The current-limit threshold
adjustment range is from 50mV to 300mV. In the
adjustable mode, the current-limit threshold voltage is
precisely 1/10th the voltage at ILIM_. The threshold
defaults to 100mV when ILIM_ is connected to VCC.
The logic threshold for switchover to the 100mV default
value is approximately VCC - 1V.
Carefully observe the PC board layout guidelines to
ensure that noise and DC errors do not corrupt the cur-
rent-sense signals at CS_. Mount or place the device
close to the synchronous rectifier or sense resistor
(whichever is used) with short, direct traces, making a
Kelvin-sense connection to the sense resistor. The cur-
rent-sense accuracy of Figure 8 is degraded if the
Schottky diode conducts during the synchronous recti-
fier on-time. To ensure that all current passes through
the sense resistor, connect the Schottky diode in paral-
lel with only the synchronous rectifier (Figure 9) if the
voltage drop across the synchronous rectifier and
sense resistor exceeds the Schottky diode’s forward
voltage. Note that at high temperatures, the on-resis-
tance of the synchronous rectifier increases and the
forward voltage of the Schottky diode decreases.
MOSFET Gate Drivers (DH_, DL_)
The DH_ and DL_ gate drivers sink 2.0A and 3.3A,
respectively, of gate drive, ensuring robust gate drive for
high-current applications. The DH_ floating high-side
MOSFET drivers are powered by diode-capacitor charge
pumps at BST_. The DL_ synchronous-rectifier drivers are
powered by LDO5.
The internal pulldown transistors that drive DL_ low
have a 0.6Ωtypical on-resistance. These low on-resis-
tance pulldown transistors prevent DL_ from being
pulled up during the fast rise time of the inductor nodes
due to capacitive coupling from the drain to the gate of
the low-side synchronous-rectifier MOSFETs. However,
for high-current applications, some combinations of
high- and low-side MOSFETS may cause excessive
gate-drain coupling, which leads to poor efficiency and
EMI-producing shoot-through currents. Adding a resis-
tor in series with BST_ increases the turn-on time of the
MAX8732A
MAX8733A
V+
DH_
DL_
CS_
LX_ OUT_
Figure 8. Current Sensing Using Sense Resistor
(MAX8732A/MAX8733A)
MAX8732A
MAX8733A
V+
DH_
DL_
CS_
LX_ OUT_
Figure 9. More Accurate Current Sensing with Adjusted
Schottky Connection
MAX8732A
MAX8733A
MAX8734A
5V VIN
10Ω
BST
DH
LX
Figure 10. Reducing the Switching-Node Rise Time
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
22 ______________________________________________________________________________________
high-side MOSFETs at the expense of efficiency, without
degrading the turn-off time (Figure 10).
Adaptive dead-time circuits monitor the DL_ and DH_
drivers and prevent either FET from turning on until the
other is fully off. This algorithm allows operation without
shoot-through with a wide range of MOSFETs, minimiz-
ing delays and maintaining efficiency. There must be
low-resistance, low-inductance paths from the gate dri-
vers to the MOSFET gates for the adaptive dead-time cir-
cuit to work properly. Otherwise, the sense circuitry
interprets the MOSFET gate as “off” when there is actual-
ly charge left on the gate. Use very short, wide traces
measuring 10 to 20 squares (50 mils to 100 mils wide if
the MOSFET is 1in from the device).
POR, UVLO, and Internal Digital
Soft-Start
Power-on reset (POR) occurs when V+ rises above
approximately 2.4V, resetting the undervoltage, over-
voltage, and thermal-shutdown fault latches. LDO5
undervoltage-lockout (UVLO) circuitry inhibits switching
when LDO5 is below 4V (typ). DL_ is low if PRO is dis-
abled; DL_ is high if PRO is enabled. The output volt-
ages begin to ramp up once VCC exceeds its 3.25V
(typ) UVLO threshold and REF is in regulation. The
internal digital soft-start timer begins to ramp up the
maximum-allowed current limit during startup. The
1.7ms ramp occurs in five steps: 20%, 40%, 60%, 80%,
and 100%.
When LD05 falls below its 4V (typ) UVLO threshold,
DH_ and DL_ are immediately forced low, and the out-
puts are high impedance. REF is turned off when VCC
falls below 3.25V (typ). DL_ is forced high again when
VCC falls below its 1V (typ) POR threshold.
Power-Good Output (PGOOD)
The PGOOD comparator continuously monitors both out-
put voltages for undervoltage conditions. PGOOD is
actively held low in shutdown, standby, and soft-start.
PGOOD releases and digital soft-start terminates when
both outputs reach the error-comparator threshold.
PGOOD goes low if EITHER output turns off or is 10%
below its nominal regulation point. PGOOD is a true
open-drain output. Note that PGOOD is independent of
the state of PRO.
Fault Protection
The MAX8732A/MAX8733A/MAX8734A provide
over/undervoltage fault protection. Drive PRO low to
activate fault protection. Drive PRO high to disable fault
protection. Once activated, the devices continuously
monitor for both undervoltage and overvoltage conditions.
Overvoltage Protection
When the output voltage is 11% above the set voltage,
the overvoltage fault protection activates. The synchro-
nous rectifier turns on 100% and the high-side MOSFET
turns off. This rapidly discharges the output capacitors,
decreasing the output voltage. The output voltage may
dip below ground. For loads that cannot tolerate a neg-
ative voltage, place a power Schottky diode across the
output to act as a reverse-polarity clamp. In practical
applications, there is a fuse between the power source
(battery) and the external high-side switches. If the
overvoltage condition is caused by a short in the high-
side switch, turning the synchronous rectifier on 100%
creates an electrical short between the battery and
GND, blowing the fuse and disconnecting the battery
from the output. Once an overvoltage fault condition is
set, it can only be reset by toggling SHDN, ON_, or
cycling V+ (POR).
Undervoltage Protection
When the output voltage is 30% below the set voltage for
over 22ms (undervoltage shutdown blanking time), the
undervoltage fault protection activates. Both SMPSs stop
switching. The two outputs start to discharge (see the
Discharge Mode (Soft-Stop) section). When the output
voltage drops to 0.3V, the synchronous rectifiers turn on,
clamping the outputs to GND. Toggle SHDN or ON_, or
cycle V+ (POR) to clear the undervoltage fault latch.
Thermal Protection
The MAX8732A/MAX8733A/MAX8734A have thermal
shutdown to protect the devices from overheating.
Thermal shutdown occurs when the die temperature
exceeds +160°C. All internal circuitry shuts down during
thermal shutdown. The MAX8732A/MAX8733A/
MAX8734A may trigger thermal shutdown if LDO_ is not
bootstrapped from OUT_ while applying a high input
voltage on V+ and drawing the maximum current
(including short circuit) from LDO_. Even if LDO_ is boot-
strapped from OUT_, overloading the LDO_ causes
large power dissipation on the bootstrap switches, which
may result in thermal shutdown. Cycling SHDN, ON3,
ON5, or a V+ (POR) ends the thermal-shutdown state.
Discharge Mode (Soft-Stop)
When PRO is low and a transition to standby or shut-
down mode occurs, or the output undervoltage fault
latch is set, the outputs discharge to GND through an
internal 12Ωswitch, until the output voltages decrease
to 0.3V. The reference remains active to provide an
accurate threshold and to provide overvoltage protec-
tion. When both SMPS outputs discharge to 0.3V, the
DL_ synchronous rectifier drivers are forced high. The
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 23
synchronous rectifier drivers clamp the SMPS outputs
to GND. When PRO is high, the SMPS outputs do not
discharge and the DL_ synchronous rectifier drivers
remain low.
Shutdown Mode
Drive SHDN below the precise SHDN input falling-edge
trip level to place the MAX8732A/MAX8733A/MAX8734A
in their low-power shutdown state. The MAX8732A/
MAX8733A/MAX8734A consume only 6µA of quiescent
current while in shutdown mode. When shutdown mode
activates, the reference turns off, making the threshold
to exit shutdown inaccurate. To guarantee startup,
drive SHDN above 2V (SHDN input rising-edge trip
level). For automatic shutdown and startup, connect
SHDN to V+. If PRO is low, both SMPS outputs are dis-
charged to 0.3V through a 12Ωswitch before entering
true shutdown. The accurate 1V falling-edge threshold
on SHDN can be used to detect a specific analog volt-
age level and shut down the device. Once in shutdown,
the 1.6V rising-edge threshold activates, providing suffi-
cient hysteresis for most applications. For additional
hysteresis, the undervoltage threshold can be made
dependent on REF or LDO_, which go to 0V in shutdown.
MODE CONDITION COMMENT
Power-Up LDO5 < UVLO threshold Transitions to discharge mode after a V+ POR and after REF becomes valid.
LDO5, LDO3, and REF remain active. DL_ is active if PRO is low.
Run SHDN = high, ON3 or ON5
enabled Normal operation.
Overvoltage
Protection
Either output > 111% of
nominal level, PRO = low
DL_ is forced high. LDO3, LDO5 active. Exited by a V+ POR or by toggling
SHDN, ON3, or ON5.
Undervoltage
Protection
Either output < 70% of
nominal after 22ms time-
out expires and output is
enabled, PRO = low
If PRO is low, DL_ is forced high after discharge mode terminates. LDO3,
LDO5 active. Exited by a V+ POR or by toggling SHDN, ON3, or ON5.
Discharge
PRO is low and either
SMPS output is still high in
either standby mode or
shutdown mode
Discharge switch (12Ω) connects OUT_ to PGND. One output may still run
while the other is in discharge mode. Activates when LDO_ is in UVLO, or
transition to UVLO, standby, or shutdown has begun. LDO3, LDO5 active.
Standby ON5, ON3 < startup
threshold, SHDN = high DL_ stays high if PRO is low. LDO3, LDO5 active.
Shutdown SHDN = low All circuitry off.
Thermal Shutdown
TJ > +160°C All circuitry off. Exited by V+ POR or cycling SHDN, ON3, or ON5.
Table 3. Operating-Mode Truth Table
SHDN
(V)
VON3
(V)
VON5
(V)
LDO5
LDO3 5V SMPS 3V SMPS
Low X X Off Off Off Off
“> 2.4” => High
Low Low
On
On (after REF powers up)
Off Off
“> 2.4” => High
High High
On
On (after REF powers up)
On On
“> 2.4” => High
High Low
On
On (after REF powers up)
Off On
“> 2.4” => High
Low High
On
On (after REF powers up)
On Off
“> 2.4” => High
High
REF On
On (after REF powers up) On (after 3V SMPS is up)
On
“> 2.4” => High
REF
High
On
On (after REF powers up)
On
On (after 5V SMPS is up)
Table 4. Power-Up Sequencing
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
24 ______________________________________________________________________________________
Power-Up Sequencing and
On/Off Controls (ON3, ON5)
ON3 and ON5 control SMPS power-up sequencing.
ON3 or ON5 rising above 2.4V enables the respective
outputs. ON3 or ON5 falling below 1.6V disables the
respective outputs.
Connecting ON3 or ON5 to REF forces the respective
outputs off while the other output is below regulation and
starts after that output regulates. The second SMPS
remains on until the first SMPS turns off, the device shuts
down, a fault occurs, or LDO5 goes into undervoltage
lockout. Both supplies begin their power-down sequence
immediately when the first supply turns off. Driving ON_
below 0.8V clears the overvoltage, undervoltage, and
thermal fault latches.
Adjustable-Output Feedback
(Dual-Mode FB)
Connect FB_ to GND to enable the fixed, preset SMPS
output voltages (3.3V and 5V). Connect a resistive volt-
age-divider at FB_ between OUT_ and GND to adjust
the respective output voltage between 2V and 5.5V
(Figure 11). Choose R2 to be approximately 10kΩ, and
solve for R1 using the equation:
where VFB = 2V nominal.
When using the adjustable-output mode, set the 3.3V
SMPS lower than the 5V SMPS. LDO5 connects to OUT5
through an internal switch only when OUT5 is above the
LDO5 bootstrap-switch threshold (4.56V). LDO3 con-
nects to OUT3 through an internal switch only when
OUT3 is above the LDO3 bootstrap switch threshold
(2.91V). Bootstrapping is most effective when the fixed
output voltages are used. Once LDO_ is bootstrapped
from OUT_, the internal linear regulator turns off. This
reduces internal power dissipation and improves effi-
ciency when LDO_ is powered with a high input voltage.
Design Procedure
Establish the input voltage range and maximum load
current before choosing an inductor and its associated
ripple-current ratio (LIR). The following four factors dic-
tate the rest of the design:
1) Input Voltage Range. The maximum value (V+(MAX))
must accommodate the maximum AC adapter volt-
age. The minimum value (V+(MIN)) must account for
the lowest input voltage after drops due to connec-
tors, fuses, and battery selector switches. Lower input
voltages result in better efficiency.
2) Maximum Load Current. The peak load current
(ILOAD(MAX)) determines the instantaneous compo-
nent stress and filtering requirements, and thus dri-
ves output capacitor selection, inductor saturation
rating, and the design of the current-limit circuit.
The continuous load current (ILOAD) determines the
thermal stress and drives the selection of input
capacitors, MOSFETs, and other critical heat-con-
tributing components.
3) Switching Frequency. This choice determines the
basic trade-off between size and efficiency. The
optimal frequency is largely a function of maximum
input voltage and MOSFET switching losses. The
MAX8732A has a nominal switching frequency of
200kHz for the 5V SMPS and 300kHz for the 3.3V
SMPS. The MAX8733A has a nominal switching fre-
quency of 400kHz for the 5V SMPS and 500kHz for
the 3.3V SMPS. The MAX8734A has a pin-selec-
table switching frequency.
4) Inductor Ripple Current Ratio (LIR). LIR is the
ratio of the peak-to-peak ripple current to the aver-
age inductor current. Size and efficiency trade-offs
must be considered when setting the inductor rip-
ple current ratio. Low inductor values cause large
ripple currents, resulting in the smallest size, but
poor efficiency and high output noise. The minimum
practical inductor value is one that causes the cir-
cuit to operate at critical conduction (where the
inductor current just touches zero with every cycle
RR V
V
OUT
FB
12 1=× −
_
MAX8732A
MAX8733A
MAX8734A
DH_
DL_
GND
OUT_
FB_
V+
R1
R2
VOUT_
Figure 11. Setting VOUT_ with a Resistor-Divider
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 25
at maximum load).Inductor values lower than this
grant no further size-reduction benefit.
The MAX8732A/MAX8733A/MAX8734As’ pulse-skip-
ping algorithm (SKIP = GND) initiates skip mode at the
critical conduction point, so the inductor’s operating
point also determines the load current at which
PWM/PFM switchover occurs. The optimum point is
usually found between 20% and 50% ripple current.
Inductor Selection
The switching frequency (on-time) and operating point
(% ripple or LIR) determine the inductor value as follows:
Example: ILOAD(MAX) = 5A, V+ = 12V, VOUT5 = 5V, f =
200kHz, 35% ripple current or LIR = 0.35:
Find a low-loss inductor with the lowest possible DC
resistance that fits in the allotted dimensions. Ferrite cores
are often the best choice. The core must be large enough
not to saturate at the peak inductor current (IPEAK):
IPEAK = ILOAD(MAX) + [(LIR / 2) x ILOAD(MAX)]
The inductor ripple current also impacts transient-
response performance, especially at low V+ - VOUT_
differences. Low inductor values allow the inductor cur-
rent to slew faster, replenishing charge removed from
the output filter capacitors by a sudden load step. The
peak amplitude of the output transient (VSAG) is also a
function of the maximum duty factor, which can be cal-
culated from the on-time and minimum off-time:
where minimum off-time = 0.350µs (max) and K is from
Table 2.
Determining the Current Limit
The minimum current-limit threshold must be great
enough to support the maximum load current when the
current limit is at the minimum tolerance value. The val-
ley of the inductor current occurs at ILOAD(MAX) minus
half of the ripple current; therefore:
ILIMIT(LOW) > ILOAD(MAX) - [(LIR / 2) x ILOAD(MAX)]
where ILIMIT(LOW) = minimum current-limit threshold
voltage divided by the RDS(ON) of N2/N4 (MAX8734A).
For the MAX8732A/MAX8733A/MAX8734A, the mini-
mum current-limit threshold voltage is 93mV (ILIM_ =
VCC). Use the worst-case maximum value for RDS(ON)
from the MOSFET N2/N4 data sheet and add some
margin for the rise in RDS(ON) with temperature. A good
general rule is to allow 0.5% additional resistance for
each °C of temperature rise.
Examining the 5A circuit example with a maximum
RDS(ON) = 12mΩat high temperature reveals the following:
ILIMIT(LOW) = 93mV / 12mΩ> 5A - (0.35 / 2) 5A
7.75A > 4.125A
7.75A is greater than the valley current of 4.125A, so
the circuit can easily deliver the full-rated 5A using the
fixed 100mV nominal current-limit threshold voltage.
Connect the source of the synchronous rectifier to a
current-sense resistor to GND (MAX8732A/MAX8733A),
and connect CS_ to that junction to set the current limit
for the device. The MAX8732A/MAX8733A/MAX8734A
limit the current with the sense resistor instead of the
RDS(ON) of N2/N4. The maximum value of the sense
resistor can be calculated with the equation:
ILIM_ = 93mV / RSENSE
Output-Capacitor Selection
The output filter capacitor must have low enough equiv-
alent series resistance (ESR) to meet output ripple and
load-transient requirements, yet have high enough ESR
to satisfy stability requirements. The output capaci-
tance must also be high enough to absorb the inductor
energy while transitioning from full-load to no-load con-
ditions without tripping the overvoltage fault latch. In
applications where the output is subject to large load
transients, the output capacitor’s size depends on how
much ESR is needed to prevent the output from dip-
ping too low under a load transient. Ignoring the sag
due to finite capacitance:
where VDIP is the maximum-tolerable transient voltage
drop. In non-CPU applications, the output capacitor’s
size depends on how much ESR is needed to maintain
an acceptable level of output voltage ripple:
RV
I
ESR DIP
LOAD MAX
≤
()
V
ILK
V
V
t
CVK
VV
V
t
SAG
LOAD MAX OUT OFF MIN
OUT OUT OUT OFF MIN
=
∆
()
×++
×× +−
+
−
() _()
__()
2
2
L
VV V
V kHz A
H=−
()
×××
=
512 5
12 200 0 35 5
83
.
.µ
LVVV
VfLIR I
OUT_ OUT_
LOAD(MAX)
=+−
()
+× × ×
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
26 ______________________________________________________________________________________
where VP-P is the peak-to-peak output voltage ripple.
The actual capacitance value required relates to the
physical size needed to achieve low ESR, as well as to
the chemistry of the capacitor technology. Thus, the
capacitor is usually selected by ESR and voltage rating
rather than by capacitance value (this is true of tanta-
lum, OS-CON, and other electrolytic-type capacitors).
When using low-capacity filter capacitors such as
polymer types, capacitor size is usually determined by
the capacity required to prevent VSAG and VSOAR from
tripping the undervoltage and overvoltage fault latches
during load transients in ultrasonic mode.
For low input-to-output voltage differentials (VIN / VOUT
< 2), additional output capacitance is required to main-
tain stability and good efficiency in ultrasonic mode.
The amount of overshoot due to stored inductor energy
can be calculated as:
where IPEAK is the peak inductor current.
Stability Considerations
Stability is determined by the value of the ESR zero
(fESR) relative to the switching frequency (f). The point
of instability is given by the following equation:
where:
For a typical 300kHz application, the ESR zero frequen-
cy must be well below 95kHz, preferably below 50kHz.
Low-ESR capacitors (especially polymer or tantalum),
in widespread use at the time of publication, typically
have ESR zero frequencies lower than 30kHz. In the
design example used for inductor selection, the ESR
needed to support a specified ripple voltage is found
by the equation:
where LIR is the inductor ripple current ratio and ILOAD
is the average DC load. Using LIR = 0.35 and an aver-
age load current of 5A, the ESR needed to support
50mVP-P ripple is 28mΩ.
Do not place high-value ceramic capacitors directly
across the fast-feedback inputs (OUT_ to GND for inter-
nal feedback, FB_ divider point for external feedback)
without taking precautions to ensure stability. Large
ceramic capacitors can have a high-ESR zero frequency
and cause erratic, unstable operation. Adding a discrete
resistor or placing the capacitors a couple of inches
downstream from the junction of the inductor and OUT_
may improve stability.
Unstable operation manifests itself in two related but
distinctly different ways: double pulsing and fast-feed-
back loop instability. Noise on the output or insufficient
ESR may cause double pulsing. Insufficient ESR does
not allow the amplitude of the voltage ramp in the output
signal to be large enough. The error comparator mistak-
enly triggers a new cycle immediately after the 350ns
minimum off-time period has expired. Double pulsing
results in increased output ripple, and can indicate the
presence of loop instability caused by insufficient ESR.
Loop instability results in oscillations or ringing at the
output after line or load perturbations, causing the out-
put voltage to fall below the tolerance limit.
The easiest method for checking stability is to apply a
very fast zero-to-max load transient (refer to the
MAX8734A EV kit data sheet) and observe the output
voltage-ripple envelope for overshoot and ringing.
Monitoring the inductor current with an AC current
probe can also provide some insight. Do not allow
more than one cycle of ringing of under- or overshoot
after the initial step response.
Input-Capacitor Selection
The input capacitors must meet the input-ripple-current
(IRMS) requirement imposed by the switching current.
The MAX8732A/MAX8733A/MAX8734A dual switching
regulators operate at different frequencies. This inter-
leaves the current pulses drawn by the two switches and
reduces the overlap time where they add together. The
input RMS current is much smaller in comparison than
with both SMPSs operating in phase. The input RMS cur-
rent varies with load and the input voltage.
ESR
V
LIR I
RIPPLE P P
LOAD
()
=×
−
fRC
ESR ESR OUT
=1
2π
ff
ESR ≤π
VIL
CV
SOAR PEAK
OUT OUT
=2
2_
RV
LIR I
ESR PP
LOAD MAX
≤×
−
()
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 27
The maximum input capacitor RMS current for a single
SMPS is given by:
When V+ = 2 x VOUT_(D = 50%), IRMS has a maximum
current of ILOAD / 2.
The ESR of the input capacitor is important for deter-
mining capacitor power dissipation. All the power
(IRMS2x ESR) heats up the capacitor and reduces effi-
ciency. Nontantalum chemistries (ceramic or OS-CON)
are preferred due to their low ESR and resilience to
power-up surge currents. Choose input capacitors that
exhibit less than +10°C temperature rise at the RMS
input current for optimal circuit longevity. Place the
drains of the high-side switches close to each other to
share common input bypass capacitors.
Power-MOSFET Selection
Most of the following MOSFET guidelines focus on the
challenge of obtaining high load-current capability
(> 5A) when using high-voltage (> 20V) AC adapters.
Low-current applications usually require less attention.
Choose a high-side MOSFET (N1/N3) that has conduc-
tion losses equal to the switching losses at the typical
battery voltage for maximum efficiency. Ensure that the
conduction losses at the minimum input voltage do not
exceed the package thermal limits or violate the overall
thermal budget. Ensure that conduction losses plus
switching losses at the maximum input voltage do not
exceed the package ratings or violate the overall ther-
mal budget.
Choose a synchronous rectifier (N2/N4) with the lowest
possible RDS(ON). Ensure the gate is not pulled up by the
high-side switch turning on due to parasitic drain-to-gate
capacitance, causing crossconduction problems.
Switching losses are not an issue for the synchronous
rectifier in the buck topology since it is a zero-voltage
switched device when using the buck topology.
MOSFET Power Dissipation
Worst-case conduction losses occur at the duty-factor
extremes. For the high-side MOSFET, the worst-case
power dissipation (PD) due to the MOSFET’s RDS(ON)
occurs at the minimum battery voltage:
Generally, a small high-side MOSFET reduces switch-
ing losses at high input voltage. However, the RDS(ON)
required to stay within package power-dissipation limits
often limits how small the MOSFET can be. The opti-
mum situation occurs when the switching (AC) losses
equal the conduction (RDS(ON)) losses.
Switching losses in the high-side MOSFET can become
an insidious heat problem when maximum battery volt-
age is applied, due to the squared term in the CV2 ✕f
switching-loss equation. Reconsider the high-side
MOSFET chosen for adequate RDS(ON) at low battery
voltages if it becomes extraordinarily hot when subject-
ed to V+(MAX).
Calculating the power dissipation in NH(N1/N3) due to
switching losses is difficult since it must allow for quan-
tifying factors that influence the turn-on and turn-off
times. These factors include the internal gate resis-
tance, gate charge, threshold voltage, source induc-
tance, and PC board layout characteristics. The
following switching-loss calculation provides only a
very rough estimate and is no substitute for bench eval-
uation, preferably including verification using a thermo-
couple mounted on NH(N1/N3):
where COSS is the output capacitance of NH(N1/N3),
QG(SW) is the switch gate charge of NH, and IGATE is
the peak gate-drive source/sink current.
For the synchronous rectifier, the worst-case power dis-
sipation always occurs at maximum battery voltage:
The absolute worst case for MOSFET power dissipation
occurs under heavy overloads that are greater than
ILOAD(MAX) but are not quite high enough to exceed
the current limit and cause the fault latch to trip. To pro-
tect against this possibility, “overdesign” the circuit to
tolerate:
ILOAD = ILIMIT(HIGH) + (LIR / 2 ) x ILOAD(MAX)
where ILIMIT(HIGH) is the maximum valley current
allowed by the current-limit circuit, including threshold
tolerance and resistance variation.
PD N V
VIR
LOUT
IN MAX LOAD DS ON
()
=−
12
_
() ()
PD N Switching
CV f
VIQf
I
H
OSS IN MAX SW
IN MAX LOAD G SW SW
GATE
()
()
()
() ()
=
+
2
2
PD N sis ce V
V
IR
HOUT
IN MIN
LOAD DS ON
(Retan ) _
()
()
=
()
2
II
VVV
V
RMS LOAD
OUT OUT
≈+−
()
+
__
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
28 ______________________________________________________________________________________
Rectifier Selection
Current circulates from ground to the junction of both
MOSFETs and the inductor when the high-side switch is
off. As a consequence, the polarity of the switching
node is negative with respect to ground. This voltage is
approximately -0.7V (a diode drop) at both transition
edges while both switches are off (dead time). The drop
is ILx RDS(ON) when the low-side switch conducts.
The rectifier is a clamp across the synchronous rectifier
that catches the negative inductor swing during the dead
time between turning the high-side MOSFET off and the
synchronous rectifier on. The MOSFETs incorporate a
high-speed silicon body diode as an adequate clamp
diode if efficiency is not of primary importance. Place a
Schottky diode in parallel with the body diode to reduce
the forward-voltage drop and prevent the N2/N4 MOSFET
body diodes from turning on during the dead time.
Typically, the external diode improves the efficiency by
1% to 2%. Use a Schottky diode with a DC current rating
equal to 1/3 of the load current. For example, use an
MBR0530 (500mA-rated) type for loads up to 1.5A, a
1N5819 type for loads up to 3A, or a 1N5822 type for
loads up to 10A. The rectifier’s rated reverse-breakdown
voltage must be at least equal to the maximum input volt-
age, preferably with a 20% derating factor.
Boost Supply Diode
A signal diode, such as a 1N4148, works well in most
applications. Use a small (20mA) Schottky diode for
slightly improved efficiency and dropout characteris-
tics, if the input voltage can go below 6V. Do not use
large power diodes, such as 1N5817 or 1N4001, since
high-junction capacitance can force LDO5 to excessive
voltages.
Applications Information
Dropout Performance
The output voltage-adjust range for continuous-conduc-
tion operation is restricted by the nonadjustable 350ns
(max) minimum off-time, one-shot. Use the slower 5V
SMPS for the higher of the two output voltages for best
dropout performance in adjustable feedback mode. The
duty-factor limit must be calculated using worst-case val-
ues for on- and off-times, when working with low input
voltages. Manufacturing tolerances and internal propaga-
tion delays introduce an error to the tON K-factor. Also,
keep in mind that transient-response performance of
buck regulators operated close to dropout is poor, and
bulk output capacitance must often be added (see the
VSAG equation in the Output-Capacitor Selection section).
The absolute point of dropout occurs when the inductor
current ramps down during the minimum off-time
(∆IDOWN) as much as it ramps up during the on-time
(∆IUP). The ratio h = ∆IUP/∆IDOWN indicates the ability to
slew the inductor current higher in response to
increased load, and must always be greater than 1. As
h approaches 1, the absolute minimum dropout point,
the inductor current is less able to increase during each
switching cycle and VSAG greatly increases unless
additional output capacitance is used.
A reasonable minimum value for h is 1.5, but this can
be adjusted up or down to allow tradeoffs between
VSAG, output capacitance, and minimum operating
voltage. For a given value of h, the minimum operating
voltage can be calculated as:
where VDROP1 and VDROP2 are the parasitic voltage
drops in the discharge and charge paths (see the On-
Time, One-Shot section), tOFF(MIN) is from the EC table,
and K is taken from Table 2. The absolute minimum
input voltage is calculated with h = 1.
Operating frequency must be reduced or h must be
increased and output capacitance added to obtain an
acceptable VSAG if calculated V+(MIN) is greater than
the required minimum input voltage. Calculate VSAG to
be sure of adequate transient response if operation
near dropout is anticipated.
Dropout Design Example
MAX8733A: With VOUT5 = 5V, fsw = 400kHz, K = 2.25µs,
tOFF(MIN) = 350ns, VDROP1 = VDROP2 = 100mV, and h = 1.5,
V
VV
th
K
VV
MIN
OUT DROP
OFF MIN
DROP DROP
+= +
()
−×
+−
()
_
()
1
21
1
MAX8732A
MAX8733A
MAX8734A
V+ 12V
POSITIVE
SECONDARY
OUTPUT
5V
MAIN
OUTPUT
DL_
DH_
T1
10µH
1:2.2
T1 = TRANSPOWER TECHNOLOGIES TTI-5870
MAX1658/
MAX1659
LDO
Figure 12. Transformer-Coupled Secondary Output
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 29
the minimum V+ is:
Calculating with h = 1 yields:
Therefore, V+ must be greater than 6.65V. A practical
input voltage with reasonable output capacitance
would be 7.5V.
Use of Coupled Inductors to Create
Auxiliary Outputs
A coupled inductor or transformer can be substituted for
the inductor in the 5V or 3.3V SMPS to create an auxiliary
output (Figure 12). The MAX8732A/MAX8733A/
MAX8734A are particularly well suited for such applica-
tions because they can be configured in ultrasonic or
forced-PWM mode to ensure good load regulation when
the main supplies are lightly loaded. An additional
postregulation circuit can be used to improve load regula-
tion and limit output current.
The power requirements of the auxiliary supply must be
considered in the design of the main output. The trans-
former must be designed to deliver the required current
in both the primary and the secondary outputs with the
proper turns ratio and inductance. The power ratings of
the synchronous-rectifier MOSFETs and the current limit
in the MAX8732A/MAX8733A/MAX8734A must also be
adjusted accordingly. Extremes of low input-output dif-
ferentials, widely different output loading levels, and high
turns ratios can further complicate the design due to par-
asitic transformer parameters such as interwinding
capacitance, secondary resistance, and leakage induc-
tance. Power from the main and secondary outputs is
combined to get an equivalent current referred to the
main output. Use this total current to determine the cur-
rent limit (see the Determining the Current Limit section):
where ITOTAL is the equivalent output current referred
to the main output and PTOTAL is the sum of the output
power from both the main output and the secondary
output:
where LPRIMARY is the primary inductance, N is the
transformer turns ratio, VSEC is the minimum-required
rectified secondary voltage, VFWD is the forward drop
across the secondary rectifier, VOUT(MIN) is the minimum
value of the main output voltage, and VRECT is the on-
state voltage drop across the synchronous rectifier
MOSFET. The transformer secondary return is often con-
nected to the main output voltage instead of ground to
reduce the necessary turns ratio. In this case, subtract
VOUT from the secondary voltage (VSEC - VOUT) in the
transformer turns-ratio equation above.
The secondary diode in coupled-inductor applications
must withstand flyback voltages greater than 60V, which
usually rules out most Schottky rectifiers. Common sili-
con rectifiers, such as the 1N4001, are also prohibited
because they are too slow. This often makes fast silicon
rectifiers such as the MURS120 the only choice. The fly-
back voltage across the rectifier is related to the VIN -
VOUT difference, according to the transformer turns ratio:
VFLYBACK = VSEC + (VIN - VOUT) ✕N
where N is the transformer turns ratio (secondary wind-
ings/primary windings), VSEC is the maximum secondary
DC output voltage, and VOUT is the primary (main) out-
put voltage. If the secondary winding is returned to VOUT
instead of ground, subtract VOUT from VFLYBACK in the
equation above. The diode’s reverse breakdown voltage
rating must also accommodate any ringing due to leak-
age inductance. The diode’s current rating should be at
least twice the DC load current on the secondary output.
The optional linear postregulator must be selected to
deliver the required load current from the transformer’s
rectified DC output. The linear regulator should be con-
figured to run close to dropout to minimize power dissi-
pation and should have good output accuracy under
those conditions. Input and output capacitors are cho-
sen to meet line regulation, stability, and transient
requirements. There is a wide variety of linear regulators
appropriate for this application; consult the specific lin-
ear-regulator data sheet for details.
Widely different output loads affect load regulation. In
particular, when the secondary output is left unloaded
while the main output is fully loaded, the secondary out-
put capacitor may become overcharged by the leakage
inductance, reaching voltages much higher than intend-
ed. In this case, a minimum load or overvoltage protec-
LVV V
VILIR
NVV
VV
PRIMARY
OUT IN MAX OUT
IN MAX TOTAL
SEC FWD
OUT MIN RECT
=−
×ƒ× ×
=+
+
(()
()
()
IPV
TOTAL TOTAL OUT
=/
V
VV
s
s
VV V
MIN
+= +
()
−µ×
µ
+−=
()
.
.
.
.. .
501
1035 1
225
01 01 604
V
VV
s
s
VV V
MIN
+= +
()
−µ×
µ
+−=
()
.
..
.
.. .
501
1035 15
225
01 01 665
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
30 ______________________________________________________________________________________
tion may be required on the secondary output to protect
any device connected to this output.
PC Board Layout Guidelines
Careful PC board layout is critical to achieve minimal
switching losses and clean, stable operation. This is
especially true when multiple converters are on the
same PC board where one circuit can affect the other.
The switching power stages require particular attention
(Figure 13). Refer to the MAX1999 EV kit I.C. data sheet
for a specific layout example.
Mount all of the power components on the top side of
the board with their ground terminals flush against one
another, if possible. Follow these guidelines for good
PC board layout:
•Isolate the power components on the top side from
the sensitive analog components on the bottom side
with a ground shield. Use a separate PGND plane
under the OUT3 and OUT5 sides (called PGND3 and
PGND5). Avoid the introduction of AC currents into
the PGND3 and PGND5 ground planes. Run the
power plane ground currents on the top side only, if
possible.
•Use a star ground connection on the power plane to
minimize the crosstalk between OUT3 and OUT5.
•Keep the high-current paths short, especially at the
ground terminals. This practice is essential for sta-
ble, jitter-free operation.
•Keep the power traces and load connections short.
This practice is essential for high efficiency. Using
thick copper PC boards (2oz vs. 1oz) can enhance
full-load efficiency by 1% or more. Correctly routing
PC board traces must be approached in terms of
fractions of centimeters, where a single milliohm of
excess trace resistance causes a measurable effi-
ciency penalty.
•CS_ (MAX8732A/MAX8733A)/LX_ (MAX8734A) and
GND connections to the synchronous rectifiers for
current limiting must be made using Kelvin-sense
connections to guarantee the current-limit accuracy.
With 8-pin SO MOSFETs, this is best done by routing
power to the MOSFETs from outside using the top
copper layer, while connecting CS_/LX_ traces inside
(underneath) the MOSFETs.
•When trade-offs in trace lengths must be made, it is
preferable to allow the inductor charging path to be
made longer than the discharge path. For example, it
is better to allow some extra distance between the
input capacitors and the high-side MOSFET than to
allow distance between the inductor and the syn-
chronous rectifier or between the inductor and the
output filter capacitor.
•Ensure that the OUT_ connection to COUT_ is short and
direct. However, in some cases it may be desirable to
deliberately introduce some trace length between the
OUT_ connector node and the output filter capacitor
(see the Stability Considerations section).
•Route high-speed switching nodes (BST_, DH_, LX_,
and DL_) away from sensitive analog areas (REF,
ILIM_, and FB_). Use PGND3 and PGND5 as an EMI
shield to keep radiated switching noise away from the
IC’s feedback divider and analog bypass capacitors.
AGND
PGND
VIA TO OUT5
GROUND OUT3OUT5
VIA TO OUT3
VIA TO PGND
VIA TO LX5 V+ VIA TO LX3
USE AGND PLANE TO:
- BYPASS VCC AND REF
- TERMINATE EXTERNAL FB
DIVIDER (IF USED)
- TERMINATE RILIM
(IF USED)
- PIN-STRAP CONTROL
INPUTS
USE PGND PLANE TO:
- BYPASS LDO_
- CONNECT PGND TO THE TOPSIDE STAR GROUND
VIAS TO GROUND
NOTE: EXAMPLE SHOWN IS FOR DUAL I.C. n-CHANNEL MOSFET.
ANALOG GROUND
PLANE ON INNER LAYER
C4
C3
C1
N4
D1
D2
N2
C2
L1 L2
CONNECT PGND TO AGND
BENEATH THE CONTROLLER AT
ONE POINT ONLY AS SHOWN.
VIA BETWEEN POWER
AND ANALOG GROUND
N3 N1
Figure 13. PC Board Layout Example
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
______________________________________________________________________________________ 31
•Make all pin-strap control input connections (SKIP,
ILIM_, etc.) to GND or VCC of the device.
Layout Procedure
1) Place the power components first with ground ter-
minals adjacent (N2/N4 source, CIN_, COUT_, D1
anode). If possible, make all these connections on
the top layer with wide, copper-filled areas.
2) Mount the controller IC adjacent to the synchronous-
rectifier MOSFETs, preferably on the back side to
keep DH_, GND, and the DL_ gate drive lines short
and wide. The DL_ gate trace must be short and
wide, measuring 50 mils to 100 mils wide if the
MOSFET is 1in from the controller device.
3) Group the gate-drive components (BST_ diode and
capacitor, V+ bypass capacitor) together near the
controller device.
4) Make the DC-DC controller ground connections as
follows: near the device, create a small analog
ground plane. Connect the small analog ground
plane to GND (Figure 13) and use the plane for the
ground connection for the REF and VCC bypass
capacitors, FB dividers, and ILIM resistors (if any).
Create another small ground island for PGND, and
use the plane for the V+ bypass capacitor, placed
very close to the device. Connect the AGND and
PGND planes together at the GND pin of the device.
5) On the board’s top side (power planes), make a
star ground to minimize crosstalk between the two
sides. The top-side star ground is a star connection
of the input capacitors and synchronous rectifiers.
Keep the resistance low between the star ground
and the source of the synchronous rectifiers for
accurate current limit. Connect the top-side star
ground (used for MOSFET, input, and output
capacitors) to the small island with a single short,
wide connection (preferably just a via).
Create PGND islands on the layer just below the
top-side layer (refer to the MAX1999 EV kit for an
example) to act as an EMI shield if multiple layers
are available (highly recommended). Connect each
of these individually to the star ground via, which
connects the top side to the PGND plane. Add one
more solid ground plane under the device to act as
an additional shield, and also connect the solid
ground plane to the star ground via.
6) Connect the output power planes (VCORE and system
ground planes) directly to the output filter capacitor
positive and negative terminals with multiple vias.
Table 5. MAX8732A/MAX8733A/MAX8734A and MAX1777/MAX1977/MAX1999 Differences
MAX8732A/MAX8733A/MAX8734A MAX1777/MAX1977/MAX1999
Line Transient Behavior
Improved line transient behavior requires only
a 0.1µF filter capacitor on V+. Allows fast
rising-edge line transients of 10V/µs and
falling-edge line transients of 5V/µs.
A 4Ω/4.7µF filter capacitor is required on V+ to
limit the dV/dt on the V+ pin.
Ultrasonic Mode
Simplified Z pattern offers better efficiency
and smoother transition into continuous-
conduction mode.
Original “W” pattern conducts through the high-
side MOSFET’s body diode, reducing efficiency.
Transition between ultrasonic mode and
continuous-conduction mode is not as smooth.
LDO3 and LDO5 Sequencing
LDO3 starts only after LDO5 is in regulation,
reducing the inrush current when SHDN goes
high.
LDO3 and LDO5 start up together at the current
limit of each LDO, causing large inrush currents
through the 4Ω series resistor at V+.
Soft-Shutdown Enable Delay
Soft-shutdown (10Ω discharge feature) is
enabled immediately when an output is
enabled, and is not dependent on the 22ms
(typ) startup undervoltage blanking timer.
Soft-shutdown (10Ω discharge feature) is
enabled only after the 22ms (typ) startup
undervoltage blanking time. This causes DL_ to
be driven high if the part is commanded to turn
off before the 22ms timer.
High-Output Impedance in UVLO
When LDO5 falls below its 4V (typ) UVLO
threshold, DH_ and DL_ are immediately
pulled low, and the outputs are high
impedance. The outputs are discharged by
the load.
When LDO5 falls below its 4V (typ) UVLO
threshold, DH_ is immediately pulled low and
DL_ forced high to clamp the output rails. This
causes the outputs to swing below ground.
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
32 ______________________________________________________________________________________
Ordering Information (continued)
PART
TEMP RANGE
PIN-
PA CKA GE
5V/3V
SWITCHING
FREQUENCY
(kHz)
MAX8734AEEI+
-40°C to +85°C 28 QSOP
200/300 or
400/500
MAX8734AEEI
-40°C to +85°C 28 QSOP
400kHz/500
28
27
26
25
24
23
22
21
20
19
18
17
16
15
1
2
3
4
5
6
7
8
9
10
11
12
13
14
BST3
LX3
DH3
LDO3
DL3
GND
LX5
OUT3
OUT5
V+
DL5
LDO5
VCC
DH5
BST5
CS5
ILIM5
FB5
REF
FB3
ILIM3
ON5
ON3
PGOOD
CS3
QSOP
TOP VIEW
MAX8732A
MAX8733A
SHDN
PRO
SKIP
Pin Configurations (continued)
Chip Information
TRANSISTOR COUNT: 8335
PROCESS: BiCMOS
+Denotes lead free package.
MAX8732A/MAX8733A/MAX8734A
High-Efficiency, Quad-Output, Main Power-
Supply Controllers for Notebook Computers
Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses are
implied. Maxim reserves the right to change the circuitry and specifications without notice at any time.
Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600 ____________________ 33
©2005 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products, Inc.
QSOP.EPS
E1
1
21-0055
PACKAGE OUTLINE, QSOP .150", .025" LEAD PITCH
Package Information
(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,
go to www.maxim-ic.com/packages.)
ENGLISH • ???? • ??? • ???
WHAT'S NEW
PRODUCTS
SOLUTIONS
DESIGN
APPNOTES
SUPPORT
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COMPANY
MEMBERS
MAX8734A
Part Number Table
Notes:
See the MAX8734A QuickView Data Sheet for further information on this product family or download the
MAX8734A full data sheet (PDF, 1.1MB).
1.
Other options and links for purchasing parts are listed at: http://www.maxim-ic.com/sales.2.
Didn't Find What You Need? Ask our applications engineers. Expert assistance in finding parts, usually within one
business day.
3.
Part number suffixes: T or T&R = tape and reel; + = RoHS/lead-free; # = RoHS/lead-exempt. More: See full
data sheet or Part Naming Conventions.
4.
* Some packages have variations, listed on the drawing. "PkgCode/Variation" tells which variation the product
uses.
5.
Part Number
Free
Sample
Buy
Direct
Package:
TYPE PINS SIZE
DRAWING CODE/VAR *
Temp
RoHS/Lead-Free?
Materials Analysis
MAX8734AEEI+G104
-40C to +85C
RoHS/Lead-Free: Yes
MAX8734AEEI+T
QSOP;28 pin;.150"
Dwg: 21-0055F (PDF)
Use pkgcode/variation: E28+1*
-40C to +85C
RoHS/Lead-Free: Yes
Materials Analysis
MAX8734AEEI+
QSOP;28 pin;.150"
Dwg: 21-0055F (PDF)
Use pkgcode/variation: E28+1*
-40C to +85C
RoHS/Lead-Free: Yes
Materials Analysis
MAX8734AEEI
QSOP;28 pin;.150"
Dwg: 21-0055F (PDF)
Use pkgcode/variation: E28-1*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
MAX8734AEEI-T
QSOP;28 pin;.150"
Dwg: 21-0055F (PDF)
Use pkgcode/variation: E28-1*
-40C to +85C
RoHS/Lead-Free: No
Materials Analysis
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