LT8304/LT8304-1
1
8304fa
For more information www.linear.com/LT8304
TYPICAL APPLICATION
FEATURES DESCRIPTION
100VIN Micropower
No-Opto Isolated Flyback
Converter with 150V/2A Switch
The LT
®
8304/LT8304-1 are monolithic micropower iso-
lated flyback converters. By sampling the isolated output
voltage directly from the primary-side flyback waveform,
the parts require no third winding or opto-isolator for
regulation. The output voltage is programmed with two
external resistors and a third optional temperature com-
pensation resistor. Boundary mode operation provides a
small magnetic solution with excellent load regulation. Low
ripple Burst Mode operation maintains high efficiency at
light load while minimizing the output voltage ripple. A 2A,
150V DMOS power switch is integrated along with all the
high voltage circuitry and control logic into a thermally
enhanced 8-lead SO package.
The LT8304/LT8304-1 operate from an input voltage range
of 3V to 100V and deliver up to 24W of isolated output
power. The high level of integration and the use of boundary
and low ripple Burst Mode operation result in a simple to
use, low component count, and high efficiency applica-
tion solution for isolated power delivery. The LT8304-1 is
specially optimized for high step-up output applications.
4V to 80VIN/5VOUT Isolated Flyback Converter
APPLICATIONS
n 3V to 100V Input Voltage Range
n 2A, 150V Internal DMOS Power Switch
n Low Quiescent Current:
n 116µA in Sleep Mode
n 390µA in Active Mode
n Quasi-Resonant Boundary Mode Operation at
Heavy Load
n Low Ripple Burst Mode
®
Operation at Light Load
n Minimum Load < 0.5% (Typ) of Full Output
n No Transformer Third Winding or Opto-Isolator
Required for Output Voltage Regulation
n Accurate EN/UVLO Threshold and Hysteresis
n Internal Compensation and Soft-Start
n Temperature Compensation for Output Diode
n Output Short-Circuit Protection
n Thermally Enhanced 8-Lead SO Package
n Isolated Automotive, Industrial, Medical, Telecom
Power Supplies
n Isolated Auxiliary/Housekeeping Power Supplies
L, LT , LT C , LT M , Linear Technology, the Linear logo and Burst Mode are registered trademarks
of Linear Technology Corporation. All other trademarks are the property of their respective
owners. Protected by U.S. Patents, including 5438499, 7463497, 7471522.
Efficiency vs Load Current
VIN
LT8304
SW
40µH
VIN
4V TO 80V 6:1
1.1µH
RFB
RREF
EN/UVLO
220pF
10µF
1µF
100µF
×3
20mA TO 2.4A (VIN = 24V)
20mA TO 3.6A (VIN = 48V)
20mA TO 4.2A (VIN = 72V)
VOUT–
100Ω
309k
•
•
100k 10k
8304 TA01a
GND
INTVCC
TC
VOUT+
5V
LOAD CURRENT (A)
0
0.6
1.2
1.8
2.4
3.0
3.6
4.2
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA01b
VIN = 24V
VIN = 48V
VIN = 72V
LT8304/LT8304-1
2
8304fa
For more information www.linear.com/LT8304
PIN CONFIGURATIONABSOLUTE MAXIMUM RATINGS
SW (Note 2) ............................................................150V
VIN ..........................................................................100V
EN/UVLO ....................................................................VIN
RFB ........................................................VIN – 0.5V to VIN
Current Into RFB ....................................................200µA
INTVCC, RREF, TC .........................................................4V
Operating Junction Temperature TJ Range (Notes 3, 4)
LT8304E/LT8304E-1 .......................... –40°C to 125°C
LT8304I/LT8304I-1 ............................ –40°C to 125°C
LT8304H/LT8304H-1 ......................... –40°C to 150°C
Storage Temperature Range .................. –65°C to 150°C
Lead Temperature (Soldering, 10 sec) ................... 300°C
(Note 1)
1
2
3
4
8
7
6
5
TOP VIEW
TC
RREF
RFB
SW
EN/UVLO
INTVCC
VIN
GND
S8E PACKAGE
8-LEAD PLASTIC SO
9
GND
θJA = 33°C/W
EXPOSED PAD (PIN 9) IS GND, MUST BE SOLDERED TO PCB
ORDER INFORMATION
LEAD FREE FINISH TAPE AND REEL PART MARKING* PACKAGE DESCRIPTION TEMPERATURE RANGE
LT8304ES8E#PBF LT8304ES8E#TRPBF 8304 8-Lead Plastic SO –40°C to 125°C
LT8304IS8E#PBF LT8304IS8E#TRPBF 8304 8-Lead Plastic SO –40°C to 125°C
LT8304HS8E#PBF LT8304HS8E#TRPBF 8304 8-Lead Plastic SO –40°C to 150°C
LT8304ES8E-1#PBF LT8304ES8E-1#TRPBF 83041 8-Lead Plastic SO –40°C to 125°C
LT8304IS8E-1#PBF LT8304IS8I-1#TRPBF 83041 8-Lead Plastic SO –40°C to 125°C
LT8304HS8E-1#PBF LT8304HS8E-1#TRPBF 83041 8-Lead Plastic SO –40°C to 150°C
Consult LTC Marketing for parts specified with wider operating temperature ranges. *The temperature grade is identified by a label on the shipping container
For more information on lead free part marking, go to: http://www.linear.com/leadfree/
For more information on tape and reel specifications, go to: http://www.linear.com/tapeandreel/. Some packages are available in 500 unit reels through
designated sales channels with #TRMPBF suffix.
http://www.linear.com/product/LT8304#orderinfo
LT8304/LT8304-1
3
8304fa
For more information www.linear.com/LT8304
ELECTRICAL CHARACTERISTICS
The l denotes the specifications which apply over the full operating
temperature range, otherwise specifications are at TA = 25°C. VIN = 24V, VEN/UVLO = VIN, CINTVCC = 1µF to GND, unless otherwise
noted.
SYMBOL PARAMETER CONDITIONS MIN TYP MAX UNIT
VIN VIN Voltage Range l3 100 V
IQVIN Quiescent Current VEN/UVLO = 0.2V
VEN/UVLO = 1.1V
Sleep Mode (Switch Off)
Active Mode (Switch On)
1.8
63
116
390
3 µA
µA
µA
µA
EN/UVLO Shutdown Threshold For Lowest Off IQl0.2 0.5 V
EN/UVLO Enable Threshold Falling l1.178 1.214 1.250 V
EN/UVLO Enable Hysteresis 14 mV
IHYS EN/UVLO Hysteresis Current VEN/UVLO = 0.2V
VEN/UVLO = 1.1V
VEN/UVLO = 1.3V
–0.1
2.3
–0.1
0
2.5
0
0.1
2.7
0.1
µA
µA
µA
VINTVCC INTVCC Regulation Voltage IINTVCC = 0mA to 10mA 2.8 3 3.1 V
IINTVCC INTVCC Current Limit VINTVCC = 2.8V 16 mA
INTVCC UVLO Threshold Falling 2.38 2.47 2.56 V
INTVCC UVLO Hysteresis 105 mV
(RFB – VIN) Voltage IRFB = 75µA to 125µA –60 60 mV
RREF Regulation Voltage l0.98 1.00 1.02 V
RREF Regulation Voltage Line Regulation 3V ≤ VIN ≤ 100V 0.02 0.1 %
VTC TC Pin Voltage 1.00 V
ITC TC Pin Current VTC = 1.2V (LT8304)
VTC = 1.2V (LT8304-1)
VTC = 0.8V
12
7
15
10
–200
18
13
µA
µA
µA
fMAX Maximum Switching Frequency l315 350 385 kHz
fMIN Minimum Switching Frequency 8 11 14 kHz
tON(MIN) Minimum Switch-On Time (LT8304)
(LT8304-1)
160
950
ns
ns
ISW(MAX) Maximum Switch Current Limit 2.0 2.4 2.8 A
ISW(MIN) Minimum Switch Current Limit 0.43 0.48 0.53 A
RDS(ON) Switch On-Resistance ISW = 0.8A 0.5 Ω
ILKG Switch Leakage Current VSW = 150V 0.1 0.5 µA
tSS Soft-Start Timer 11 ms
Note 1: Stresses beyond those listed under Absolute Maximum Ratings
may cause permanent damage to the device. Exposure to any Absolute
Maximum Rating condition for extended periods may affect device
reliability and lifetime.
Note 2: The SW pin is rated to 150V for transients. Depending on the
leakage inductance voltage spike, operating waveforms of the SW pin
should be derated to keep the flyback voltage spike below 150V as shown
in Figure 5.
Note 3: The LT8304E/LT8304E-1 are guaranteed to meet performance
specifications from 0°C to 125°C junction temperature. Specifications over
the –40°C to 125°C operating junction temperature range are assured by
design, characterization and correlation with statistical process controls.
The LT8304I/LT8304I-1 are guaranteed over the full –40°C to 125°C
operating junction temperature range. LT8304H/LT304H-1 are guaranteed
over the full –40°C to 150°C operating junction temperature range. High
junction temperatures degrade operating lifetimes. Operating lifetime is
derated at junction temperature greater than 125°C.
Note 4: The LT8304/LT8304-1 includes overtemperature protection that
is intended to protect the device during momentary overload conditions.
Junction temperature will exceed 150°C when overtemperature protection
is active. Continuous operation above the specified maximum operating
junction temperature may impair device reliability.
LT8304/LT8304-1
4
8304fa
For more information www.linear.com/LT8304
TYPICAL PERFORMANCE CHARACTERISTICS
Boundary Mode Waveforms Discontinuous Mode Waveforms Burst Mode Operation Waveforms
VIN Shutdown Current
VIN Quiescent Current,
Sleep Mode
VIN Quiescent Current,
Active Mode
Output Load and Line Regulation Output Temperature Variation
Switching Frequency
vs Load Current
TA = 25°C, unless otherwise noted.
0
0.6
1.2
1.8
2.4
3.0
3.6
4.2
4.80
4.85
4.90
4.95
5.00
5.05
5.10
5.15
5.20
OUTPUT VOLTAGE (V)
8304 G01
FRONT PAGE APPLICATION
VIN = 24V
VIN = 48V
VIN = 72V
TEMPERATURE (°)
–50
–25
0
25
50
75
100
125
150
4.7
4.8
4.9
5.0
5.1
5.2
5.3
OUTPUT VOLTAGE (V)
8304 G02
FRONT PAGE APPLICATION
VIN = 48V
IOUT = 1A
RTC = 100k
RTC = OPEN
0
0.6
1.2
1.8
2.4
3
3.6
4.2
0
100
200
300
400
500
FREQUENCY (kHz)
8304 G03
FRONT PAGE APPLICATION
VIN = 24V
VIN = 48V
VIN = 72V
VSW
50V/DIV
VOUT
50mV/DIV
2µs/DIV
FRONT PAGE APPLICATION
VIN = 48V
IOUT = 3A
8304 G04
VSW
50V/DIV
VOUT
50mV/DIV
2µs/DIV
FRONT PAGE APPLICATION
VIN = 48V
IOUT = 0.5A
8304 G05
VSW
50V/DIV
VOUT
50mV/DIV
20µs/DIV
FRONT PAGE APPLICATION
VIN = 48V
IOUT = 20mA
8304 G06
T
J
= 150°C
T
J
= 25°C
T
J
= –50°C
V
IN
(V)
0
20
40
60
80
100
0
2
4
6
8
10
I
Q
(µA)
IN
8304 G07
T
J
= 150°C
T
J
= 25°C
T
J
= –50°C
V
IN
(V)
0
20
40
60
80
100
90
100
110
120
130
140
150
I
Q
(µA)
Sleep Mode
8304 G08
T
J
= 150°C
T
J
= 25°C
T
J
= –50°C
V
IN
(V)
0
20
40
60
80
100
350
370
390
410
430
450
I
Q
(µA)
Active Mode
8304 G09
LT8304/LT8304-1
5
8304fa
For more information www.linear.com/LT8304
TYPICAL PERFORMANCE CHARACTERISTICS
INTVCC Voltage vs VIN INTVCC UVLO Threshold (RFB – VIN) Voltage
RREF Regulation Voltage RREF Line Regulation TC Pin Voltage
EN/UVLO Enable Threshold EN/UVLO Hysteresis Current INTVCC Voltage vs Temperature
TA = 25°C, unless otherwise noted.
RISING
FALLING
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
1.200
1.205
1.210
1.215
1.220
1.225
1.230
1.235
1.240
V
EN/UVLO
(V)
EN/UVLO Enable Threshold
8304 G10
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
1
2
3
4
5
I
HYST
(µA)
EN/UVLO Hysteresis Current
8304 G11
I
INTVCC
= 0mA
I
INTVCC
= 10mA
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
2.80
2.85
2.90
2.95
3.00
3.05
3.10
V
INTVCC
(V)
CC
8304 G12
I
INTVCC
= 0mA
I
INTVCC
= 10mA
V
IN
(V)
0
20
40
60
80
100
2.80
2.85
2.90
2.95
3.00
3.05
3.10
V
INTVCC
(V)
CC
IN
8304 G13
RISING
FALLING
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
2.2
2.3
2.4
2.5
2.6
2.7
2.8
V
INTVCC
(V)
INTV
CC
UVLO Threshold
8304 G14
I
RFB
= 125µA
I
RFB
= 100µA
I
RFB
= 75µA
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
–40
–30
–20
–10
0
10
20
30
40
VOLTAGE (mV)
(R
FB
- V
IN
) Voltage
8304 G15
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0.990
0.992
0.994
0.996
0.998
1.000
1.002
1.004
1.006
1.008
1.010
V
RREF
(V)
R
REF
Regulation Voltage
8304 G16
V
IN
(V)
0
20
40
60
80
100
0.990
0.992
0.994
0.996
0.998
1.000
1.002
1.004
1.006
1.008
1.010
V
RREF
(V)
R
REF
Line Regulation
8304 G17
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0.7
0.8
0.9
1.0
1.1
1.2
1.3
1.4
1.5
V
TC
(V)
TC Pin Voltage
8304 G18
LT8304/LT8304-1
6
8304fa
For more information www.linear.com/LT8304
Minimum Switching Frequency Minimum Switch-On Time Minimum Switch-Off Time
RDS(ON) Switch Current Limit Maximum Switching Frequency
TYPICAL PERFORMANCE CHARACTERISTICS
TA = 25°C, unless otherwise noted.
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
0.2
0.4
0.6
0.8
1.0
1.2
RESISTANCE (Ω)
DS(ON)
8304 G19
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
0.5
1.0
1.5
2.0
2.5
3.0
ISW (A)
8304 G20
MAXIMUM CURRENT LIMIT
MINIMUM CURRENT LIMIT
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
100
200
300
400
500
FREQUENCY (kHz)
8304 G21
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
4
8
12
16
20
FREQUENCY (kHz)
8304 G22
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
100
200
300
400
TIME (ns)
8304 G23
TEMPERATURE (°C)
–50
–25
0
25
50
75
100
125
150
0
100
200
300
400
500
TIME (ns)
8304 G24
LT8304/LT8304-1
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8304fa
For more information www.linear.com/LT8304
PIN FUNCTIONS
EN/UVLO (Pin 1): Enable/Undervoltage Lockout. The
EN/UVLO pin is used to enable the LT8304. Pull the pin
below 0.2V to shut down the LT8304. This pin has an ac-
curate 1.214V threshold and can be used to program a VIN
undervoltage lockout (UVLO) threshold using a resistor
divider from VIN to ground. A 2.5µA current hysteresis
allows the programming of VIN UVLO hysteresis. If neither
function is used, tie this pin directly to VIN.
INTVCC (Pin 2): Internal 3V Linear Regulator Output. The
INTVCC pin is supplied from VIN and powers the internal
control circuitry and gate driver. Do not overdrive the
INTVCC pin with any external supply, such as a third winding
supply. Locally bypass this pin to ground with a minimum
1µF ceramic capacitor.
VIN (Pin 3): Input Supply. The VIN pin supplies current to
the internal circuitry and serves as a reference voltage for
the feedback circuitry connected to the RFB pin. Locally
bypass this pin to ground with a capacitor.
GND (Pin 4, Exposed Pad Pin 9): Ground. The exposed
pad provides both electrical contact to ground and good
thermal contact to the printed circuit board. Solder the
exposed pad directly to the ground plane.
SW (Pin 5): Drain of the Internal DMOS Power Switch.
Minimize trace area at this pin to reduce EMI and voltage
spikes.
RFB (Pin 6): Input Pin for External Feedback Resistor.
Connect a resistor from this pin to the transformer primary
SW pin. The ratio of the RFB resistor to the RREF resistor,
times the internal voltage reference, determines the output
voltage (plus the effect of any non-unity transformer turns
ratio). Minimize trace area at this pin.
RREF (Pin 7): Input Pin for External Ground Referred Ref-
erence Resistor. The resistor at this pin should be in the
range of 10k, but for convenience in selecting a resistor
divider ratio, the value may range from 9.09k to 11.0k.
TC (Pin 8): Output Voltage Temperature Compensation. The
voltage at this pin is proportional to absolute temperature
(PTAT) with temperature coefficient equal to 3.35mV/°C,
i.e., equal to 1V at room temperature 25°C. The TC pin
voltage can be used to estimate the LT8304 junction tem-
perature. Connect a resistor from this pin to the RREF pin
to compensate the output diode temperature coefficient.
LT8304/LT8304-1
8
8304fa
For more information www.linear.com/LT8304
BLOCK DIAGRAM
OPERATION
–
+
–
+
3
2
8
DRIVER
INTVCC
VIN
T1
N:1
A2 RSENSE
A3
TC
8304 BD
RREF
RREF
REN2
REN1
RTC
RFB
–
+
gm
1.214V
1V
M4
OSCILLATOR
LDO
BOUNDARY
DETECTOR
START-UP,
REFERENCE,
CONTROL
PTAT
VOLTAGE
R M1
GND
4, EXPOSED PAD PIN 9
Q
S
M2
VIN
VIN
CIN
6
RFB
5
SW
L1A L1B COUT
DOUT
VOUT+
VOUT–
M3
25µA
INTVCC
1EN/UVLO
CINTVCC
2.5µA
1:4
7
–
+
A1
•
•
The LT8304 is a current mode switching regulator IC
designed specially for the isolated flyback topology. The
key problem in isolated topologies is how to communicate
the output voltage information from the isolated secondary
side of the transformer to the primary side for regulation.
Historically, opto-isolators or extra transformer windings
communicate this information across the isolation bound-
ary. Opto-isolator circuits waste output power, and the
extra components increase the cost and physical size of
the power supply. Opto-isolators can also cause system
issues due to limited dynamic response, nonlinearity, unit-
to-unit variation and aging over lifetime. Circuits employing
extra transformer windings also exhibit deficiencies, as
using an extra winding adds to the transformer’s physical
size and cost, and dynamic response is often mediocre.
The LT8304 samples the isolated output voltage through
the primary-side flyback pulse waveform. In this manner,
neither opto-isolator nor extra transformer winding is re-
quired for regulation. Since the LT8304 operates in either
boundary conduction mode or discontinuous conduction
mode, the output voltage is always sampled on the SW
pin when the secondary current is zero. This method im-
proves load regulation without the need of external load
compensation components.
LT8304/LT8304-1
9
8304fa
For more information www.linear.com/LT8304
OPERATION
The LT8304 is a simple to use micropower isolated fly-
back converter housed in a thermally enhanced 8-lead
SO package. The output voltage is programmed with two
external resistors. An optional TC resistor provides easy
output diode temperature compensation. By integrating
the loop compensation and soft-start inside, the part
reduces the number of external components. As shown
in the Block Diagram, many of the blocks are similar to
those found in traditional switching regulators including
reference, regulators, oscillator, logic, current amplifier,
current comparator, driver, and power switch. The novel
sections include a flyback pulse sense circuit, a sample-
and-hold error amplifier, and a boundary mode detector,
as well as the additional logic for boundary conduction
mode, discontinuous conduction mode, and low ripple
Burst Mode operation.
Quasi-Resonant Boundary Mode Operation
The LT8304 features quasi-resonant boundary conduction
mode operation at heavy load, where the chip turns on the
primary power switch when the secondary current is zero
and the SW rings to its valley. Boundary conduction mode
is a variable frequency, variable peak-current switching
scheme. The power switch turns on and the transformer
primary current increases until an internally controlled peak
current limit. After the power switch turns off, the voltage
on the SW pin rises to the output voltage multiplied by
the primary-to-secondary transformer turns ratio plus the
input voltage. When the secondary current through the
output diode falls to zero, the SW pin voltage collapses
and rings around VIN. A boundary mode detector senses
this event and turns the power switch back on at its valley.
Boundary conduction mode returns the secondary current
to zero every cycle, so parasitic resistive voltage drops
do not cause load regulation errors. Boundary conduc-
tion mode also allows the use of smaller transformers
compared to continuous conduction mode and does not
exhibit subharmonic oscillation.
Discontinuous Conduction Mode Operation
As the load gets lighter, boundary conduction mode in-
creases the switching frequency and decreases the switch
peak current at the same ratio. Running at a higher switching
frequency up to several MHz increases switching and gate
charge losses. To avoid this scenario, the LT8304 has an
additional internal oscillator, which clamps the maximum
switching frequency to be less than 350kHz (TYP). Once
the switching frequency hits the internal frequency clamp,
the part starts to delay the switch turn-on and operates in
discontinuous conduction mode.
Low Ripple Burst Mode Operation
Unlike traditional flyback converters, the LT8304 has to
turn on and off at least for a minimum amount of time
and with a minimum frequency to allow accurate sampling
of the output voltage. The inherent minimum switch cur-
rent limit and minimum switch-off time are necessary to
guarantee the correct operation of specific applications.
As the load gets very light, the LT8304 starts to fold back
the switching frequency while keeping the minimum switch
current limit. So the load current is able to decrease while
still allowing minimum switch-off time for the sample-and-
hold error amplifier. Meanwhile, the part switches between
sleep mode and active mode, thereby reducing the effec-
tive quiescent current to improve light load efficiency. In
this condition, the LT8304 runs in low ripple Burst Mode
operation. The typical 11kHz minimum switching frequency
determines how often the output voltage is sampled and
also the minimum load requirement.
High Step-Up VOUT Applications
Typically, high step-up output applications have excessive
primary inductor current ringing during primary switch
turn-on due to the huge reflected capacitance on SW node.
Such current ringing can falsely trigger LT8304 current
comparator after 160ns typical blanking time and create
large signal oscillation, especially at high VIN and light
load condition. The LT8304-1, specially optimized for
high step-up output applications, is more immune to the
current ringing without requiring longer blanking time.
For any 1:N step-up transformer turns ratio larger than
or equal to 5, the LT8304-1 is recommended.
LT8304/LT8304-1
10
8304fa
For more information www.linear.com/LT8304
APPLICATIONS INFORMATION
Output Voltage
The RFB and RREF resistors as depicted in the Block Diagram
are external resistors used to program the output voltage.
The LT8304 operates similar to traditional current mode
switchers, except in the use of a unique flyback pulse
sense circuit and a sample-and-hold error amplifier, which
sample and therefore regulate the isolated output voltage
from the flyback pulse.
Operation is as follows: when the power switch M1 turns
off, the SW pin voltage rises above the VIN supply. The
amplitude of the flyback pulse, i.e., the difference between
the SW pin voltage and VIN supply, is given as:
VFLBK = (VOUT + VF + ISEC • ESR) • NPS
VF = Output diode forward voltage
ISEC = Transformer secondary current
ESR = Total impedance of secondary circuit
NPS = Transformer effective primary-to-secondary
turns ratio
The flyback voltage is then converted to a current, IRFB,
by the RFB resistor and the flyback pulse sense circuit
(M2 and M3). This current, IRFB, also flows through the
RREF resistor to generate a ground-referred voltage. The
resulting voltage feeds to the inverting input of the sample-
and-hold error amplifier. Since the sample-and-hold error
amplifier samples the voltage when the secondary current
is zero, the (ISEC • ESR) term in the VFLBK equation can be
assumed to be zero.
The internal reference voltage, VREF, 1.00V, feeds to the
noninverting input of the sample-and-hold error ampli-
fier. The relatively high gain in the overall loop causes the
voltage at the RREF pin to be nearly equal to the internal
reference voltage VREF. The resulting relationship between
VFLBK and VREF can be expressed as:
VFLBK
RFB
•RREF =VREF or
VFLBK =VREF •RFB
RREF
VREF = Internal reference voltage 1.00V
Combination with the previous VFLBK equation yields an
equation for VOUT, in terms of the RFB and RREF resistors,
transformer turns ratio, and diode forward voltage:
VOUT =VREF •RFB
RREF
•1
NPS
– VF
Output Temperature Compensation
The first term in the VOUT equation does not have tempera-
ture dependence, but the output diode forward voltage, VF,
has a significant negative temperature coefficient (–1mV/°C
to –2mV/°C). Such a negative temperature coefficient pro-
duces approximately 200mV to 300mV voltage variation
on the output voltage across temperature.
For higher voltage outputs, such as 12V and 24V, the
output diode temperature coefficient has a negligible ef-
fect on the output voltage regulation. For lower voltage
outputs, such as 3.3V and 5V, however, the output diode
temperature coefficient does count for an extra 2% to 5%
output voltage regulation.
The LT8304 junction temperature usually tracks the output
diode junction temperature to the first order. To compensate
the negative temperature coefficient of the output diode,
a resistor, RTC, connected between the TC and RREF pins
generates a proportional-to-absolute-temperature (PTAT)
current. The PTAT current is zero at 25°C, flows into the
RREF pin at hot temperature, and flows out of the RREF pin
at cold temperature. With the RTC resistor in place, the
output voltage equation is revised as follows:
VOUT =VREF •RFB
RREF
•1
NPS
– VFTO
( )
– VTC / T
( )
•
T –TO
( )
•RFB
RTC
•1
NPS
– VF/ T
( )
•T–TO
( )
TO=Room temperature 25°
°
C
VF/ T
( )
=Output diode forward voltage
temperature coefficient
V
TC
/ T
( )
=3.35mV/ C
LT8304/LT8304-1
11
8304fa
For more information www.linear.com/LT8304
APPLICATIONS INFORMATION
To cancel the output diode temperature coefficient, the
following two equations should be satisfied:
VOUT =VREF •RFB
RREF
•1
NPS
– VFTO
( )
VTC/ T
( )
•RFB
RTC
•1
NPS
=– VF/ T
( )
Selecting Actual RREF, RFB, RTC Resistor Values
The LT8304 uses a unique sampling scheme to regulate
the isolated output voltage. Due to the sampling nature,
the scheme contains repeatable delays and error sources,
which will affect the output voltage and force a re-evaluation
of the RFB and RTC resistor values. Therefore, a simple
2-step sequential process is recommended for selecting
resistor values.
Rearrangement of the expression for VOUT in the previous
sections yields the starting value for RFB:
RFB =RREF •NPS •VOUT +VFTO
( )
( )
V
REF
VOUT = Output voltage
VF (TO) = Output diode forward voltage at 25°C = ~0.3V
NPS = Transformer effective primary-to-secondary
turns ratio
The equation shows that the RFB resistor value is indepen-
dent of the RTC resistor value. Any RTC resistor connected
between the TC and RREF pins has no effect on the output
voltage setting at 25°C because the TC pin voltage is equal
to the RREF regulation voltage at 25°C.
The RREF resistor value should be approximately 10k
because the LT8304 is trimmed and specified using this
value. If the RREF resistor value varies considerably from
10k, additional errors will result. However, a variation in
RREF up to 10% is acceptable. This yields a bit of freedom
in selecting standard 1% resistor values to yield nominal
RFB/RREF ratios.
First, build and power up the application with the starting
RREF, RFB values (no RTC resistor yet) and other compo-
nents connected, and measure the regulated output volt-
age, VOUT(MEAS). The new RFB value can be adjusted to:
RFB(NEW) =
V
OUT
VOUT(MEAS)
•RFB
Second, with a new RFB resistor value selected, the output
diode temperature coefficient in the application can be
tested to determine the RTC value. Still without the RTC
resistor, the VOUT should be measured over temperature
at a desired target output load. It is very important for
this evaluation that uniform temperature be applied to
both the output diode and the LT8304. If freeze spray or
a heat gun is used, there can be a significant mismatch
in temperature between the two devices that causes sig-
nificant error. Attempting to extrapolate the data from a
diode data sheet is another option if there is no method
to apply uniform heating or cooling such as an oven. With
at least two data points spreading across the operating
temperature range, the output diode temperature coef-
ficient can be determined by:
–δVF/δT
( )
=
V
OUT
T1
( )
– V
OUT
T2
( )
T1– T2
Using the measured output diode temperature coefficient,
an exact RTC value can be selected with the following
equation:
RTC =δVTC/δT
( )
–δVF/δT
( )
•RFB
NPS
Once the RREF, RFB, and RTC values are selected, the regula-
tion accuracy from board to board for a given application
will be very consistent, typically under ±5% when includ-
ing device variation of all the components in the system
(assuming resistor tolerances and transformer windings
matching within ±1%). However, if the transformer or
the output diode is changed, or the layout is dramatically
altered, there may be some change in VOUT.
LT8304/LT8304-1
12
8304fa
For more information www.linear.com/LT8304
APPLICATIONS INFORMATION
Output Power
A flyback converter has a complicated relationship between
the input and output currents compared to a buck or a
boost converter. A boost converter has a relatively constant
maximum input current regardless of input voltage and a
buck converter has a relatively constant maximum output
current regardless of input voltage. This is due to the
continuous non-switching behavior of the two currents. A
flyback converter has both discontinuous input and output
currents which make it similar to a nonisolated buck-boost
converter. The duty cycle will affect the input and output
currents, making it hard to predict output power. In ad-
dition, the winding ratio can be changed to multiply the
output current at the expense of a higher switch voltage.
The graphs in Figures 1 to 4 show the typical maximum
output power possible for the output voltages 3.3V, 5V,
12V, and 24V. The maximum output power curve is the
calculated output power if the switch voltage is 110V dur-
ing the switch-off time. 40V of margin is left for leakage
inductance voltage spike. To achieve this power level at
a given input, a winding ratio value must be calculated
to stress the switch to 110V, resulting in some odd ratio
values. The curves below the maximum output power
curve are examples of common winding ratio values and
the amount of output power at given input voltages.
One design example would be a 5V output converter with
a minimum input voltage of 36V and a maximum input
voltage of 75V. A six-to-one winding ratio fits this design
example perfectly and outputs equal to 19.0W at 75V but
lowers to 14.4W at 36V.
Figure 1. Output Power for 3.3V Output
Figure 2. Output Power for 5V Output
Figure 3. Output Power for 12V Output
Figure 4. Output Power for 24V Output
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
100
8304 F02
5
020 40 60 80
25
20
N = 6:1
N = 2:1
N = 8:1
N =4:1
MAXIMUM
OUTPUT
POWER
ASSUME 85% EFFICIENCY
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
100
8304 F03
5
020 40 60 80
MAXIMUM
OUTPUT
POWER
25
20 N = 3:1
N = 1:1
N = 4:1
N = 2:1
ASSUME 90% EFFICIENCY
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
100
8304 F04
5
020 40 60 80
MAXIMUM
OUTPUT
POWER
25
20 N = 3:2
N = 1:2
N = 2:1
N = 1:1
ASSUME 90% EFFICIENCY
INPUT VOLTAGE (V)
0
OUTPUT POWER (W)
10
15
100
8304 F01
5
020 40 60 80
25
20
N = 8:1
N = 4:1
N = 12:1
N = 6:1
MAXIMUM
OUTPUT
POWER
ASSUME 80% EFFICIENCY
LT8304/LT8304-1
13
8304fa
For more information www.linear.com/LT8304
APPLICATIONS INFORMATION
The equations below calculate output power:
POUT = η • VIN • D • ISW(MAX) • 0.5
η = Efficiency = ~85%
D=Duty Cycle =VOUT +VF
( )
•NPS
VOUT +VF
( )
•NPS +VIN
ISW(MAX) = Maximum switch current limit = 2A (MIN)
Primary Inductance Requirement
The LT8304 obtains output voltage information from the
reflected output voltage on the SW pin. The conduction
of secondary current reflects the output voltage on the
primary SW pin. The sample-and-hold error amplifier needs
a minimum 350ns to settle and sample the reflected output
voltage. In order to ensure proper sampling, the second-
ary winding needs to conduct current for a minimum of
350ns. The following equation gives the minimum value
for primary-side magnetizing inductance:
LPRI ≥tOFF(MIN) •NPS • VOUT
+
VF
( )
ISW(MIN)
tOFF(MIN) = Minimum switch-off time = 350ns (TYP)
ISW(MIN) = Minimum switch current limit = 0.48A (TYP)
In addition to the primary inductance requirement for
the minimum switch-off time, the LT8304 has minimum
switch-on time that prevents the chip from turning on
the power switch shorter than approximately 160ns. This
minimum switch-on time is mainly for leading-edge blank-
ing the initial switch turn-on current spike. If the inductor
current exceeds the desired current limit during that time,
oscillation may occur at the output as the current control
loop will lose its ability to regulate. Therefore, the following
equation relating to maximum input voltage must also be
followed in selecting primary-side magnetizing inductance:
LPRI ≥
t
ON(MIN)
• V
IN(MAX)
ISW(MIN)
tON(MIN) = Minimum switch-on time = 160ns (TYP)
In general, choose a transformer with its primary mag-
netizing inductance about 40% to 60% larger than the
minimum values calculated above. A transformer with
much larger inductance will have a bigger physical size
and may cause instability at light load.
Selecting a Transformer
Transformer specification and design is perhaps the most
critical part of successfully applying the LT8304. In addition
to the usual list of guidelines dealing with high frequency
isolated power supply transformer design, the following
information should be carefully considered.
Linear Technology has worked with several leading mag-
netic component manufacturers to produce pre-designed
flyback transformers for use with the LT8304. Table 1
shows the details of these transformers.
Table 1. Predesigned Transformers – Typical Specifications
TRANSFORMER
PART NUMBER
DIMENSION
(W × L × H) (mm)
LPRI (μH)
TYP
LLKG (μH)
TYP (MAX) NP:NSVENDOR
TARGET APPLICATION
VIN (V) VOUT (V) IOUT (A)
750315125 17.75 × 13.46 × 12.70 40 1 (2) 6:1 Wurth Elektronik 36 – 75 5 3
750315126 17.75 × 13.46 × 12.70 40 0.5 (1) 2:1 Wurth Elektronik 36 – 75 12 1.2
750315835 17.75 × 13.46 × 12.70 40 1 (2) 8:1 Wurth Elektronik 36 – 75 3.3 4.2
750315836 17.75 × 13.46 × 12.70 40 0.45 (0.9) 1:1 Wurth Elektronik 36 – 75 24 0.6
750315837 17.75 × 13.46 × 12.70 40 0.5 (1) 1:2 Wurth Elektronik 36 – 75 48 0.3
750315839 17.75 × 13.46 × 12.71 40 0.25 (0.5) 1:10 Wurth Elektronik 4 – 36 200 0.012
13324-T083 18.0 × 13.5 × 12.5 40 (2) 8:1 Sumida 36 – 75 3.3 4.2
13324-T084 18.0 × 13.5 × 12.5 40 (1.2) 1:1 Sumida 36 – 75 24 0.6
13324-T085 18.0 × 13.5 × 12.5 40 (1.2) 1:2 Sumida 36 – 75 48 0.3
13324-T086 18.0 × 13.5 × 12.6 40 (1.2) 1:5 Sumida 4 – 36 200 0.012
13324-T087 18.0 × 13.5 × 12.5 40 (1.2) 1:10 Sumida 4 – 18 400 0.006
LT8304/LT8304-1
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8304fa
For more information www.linear.com/LT8304
APPLICATIONS INFORMATION
Turns Ratio
Note that when choosing an RFB/RREF resistor ratio to set
output voltage, the user has relative freedom in selecting
a transformer turns ratio to suit a given application. In
contrast, the use of simple ratios of small integers, e.g.,
3:1, 2:1, 1:1, etc., provides more freedom in settling total
turns and mutual inductance.
Typically, choose the transformer turns ratio to maximize
available output power. For low output voltages (3.3V
or 5V), a N:1 turns ratio can be used with multiple pri-
mary windings relative to the secondary to maximize the
transformer’s current gain (and output power). However,
remember that the SW pin sees a voltage that is equal
to the maximum input supply voltage plus the output
voltage multiplied by the turns ratio. In addition, leakage
inductance will cause a voltage spike (VLEAKAGE) on top of
this reflected voltage. This total quantity needs to remain
below the 150V absolute maximum rating of the SW pin
to prevent breakdown of the internal power switch. To-
gether these conditions place an upper limit on the turns
ratio, NPS, for a given application. Choose a turns ratio
low enough to ensure
NPS <
150V – V
IN(MAX)
– V
LEAKAGE
V
OUT
+V
F
For larger N:1 step-down turns ratio, choose a transformer
with a larger physical size to deliver additional current. In
addition, choose a large enough inductance value to en-
sure that the switch-off time is long enough to accurately
sample the output voltage. Always choose the LT8304 for
N:1 step-down transformer turns ratio.
For lower output power levels or higher output voltage,
choose a 1:1 or 1:N step-up transformer for the absolute
smallest transformer size. A 1:N step-up transformer will
minimize the magnetizing inductance and size, but will also
limit the available output power. A higher 1:N step-up turns
ratio makes it possible to have very high output voltages
without exceeding the breakdown voltage of the internal
power switch. For any 1:N step-up transformer turns ratio
larger than or equal to 5, the LT8304-1 is recommended.
The turns ratio is an important element in the isolated
feedback scheme, and directly affects the output voltage
accuracy. Make sure the transformer manufacturer speci-
fies turns ratio accuracy within ±1%.
Saturation Current
The current in the transformer windings should not exceed
its rated saturation current. Energy injected once the core is
saturated will not be transferred to the secondary and will
instead be dissipated in the core. When designing custom
transformers to be used with the LT8304, the saturation
current should always be specified by the transformer
manufacturers.
Winding Resistance
Resistance in either the primary or secondary windings
will reduce overall power efficiency. Good output voltage
regulation will be maintained independent of winding re-
sistance due to the boundary/discontinuous conduction
mode operation of the LT8304.
Leakage Inductance and Snubbers
T
ransformer leakage inductance on either the primary or
secondary causes a voltage spike to appear on the primary
after the power switch turns off. This spike is increasingly
prominent at higher load currents where more stored en-
ergy must be dissipated. It is very important to minimize
transformer leakage inductance.
When designing an application, adequate margin should be
kept for the worst-case leakage voltage spikes even under
overload conditions. In most cases shown in Figure5, the
reflected output voltage on the primary plus VIN should
be kept below 110V. This leaves at least 40V margin for
the leakage spike across line and load conditions. A larger
voltage margin will be required for poorly wound trans-
formers or for excessive leakage inductance.
LT8304/LT8304-1
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For more information www.linear.com/LT8304
APPLICATIONS INFORMATION
tOFF > 350ns
VLEAKAGE
VSW
<150V
<110V
TIME 8304 F05
tSP < 250ns
Figure 5. Maximum Voltages for SW Pin Flyback Waveform
In addition to the voltage spikes, the leakage inductance
also causes the SW pin ringing for a while after the power
switch turns off. To prevent the voltage ringing falsely trig-
ger boundary mode detector, the LT8304 internally blanks
the boundary mode detector for approximately 250ns.
Any remaining voltage ringing after 250ns may turn the
power switch back on again before the secondary current
falls to zero. In this case, the LT8304 enters continuous
conduction mode. So the leakage inductance spike ringing
should be limited to less than 250ns.
To clamp and damp the leakage voltage spikes, a
(RC + DZ) snubber circuit in Figure6 is recommended.
The RC (resistor-capacitor) snubber quickly damps the
voltage spike ringing and provides great load regulation
and EMI performance. And the DZ (diode-Zener) ensures
well defined and consistent clamping voltage to protect
SW pin from exceeding its 150V absolute maximum rating.
Figure 6. (RC + DZ) Snubber Circuit
8304 F06
R
CZ
D
Lℓ
•
•
then add capacitance until the period of the ringing is 1.5
to 2 times longer. The change in period determines the
value of the parasitic capacitance, from which the para-
sitic inductance can be also determined from the initial
period. Once the value of the SW node capacitance and
inductance is known, a series resistor can be added to
the snubber capacitance to dissipate power and critically
damp the ringing. The equation for deriving the optimal
series resistance using the observed periods ( tPERIOD and
tPERIOD(SNUBBED)) and snubber capacitance (CSNUBBER) is:
CPAR =
C
SNUBBER
tPERIOD(SNUBBED)
tPERIOD
2
–1
LPAR =tPERIOD2
CPAR • 4π2
RSNUBBER =LPAR
CPAR
Note that energy absorbed by the RC snubber will be
converted to heat and will not be delivered to the load.
In high voltage or high current applications, the snubber
needs to be sized for thermal dissipation. A 220pF capaci-
tor in series with a 100Ω resistor is a good starting point.
For the DZ snubber, proper care should be taken when
choosing both the diode and the Zener diode. Schottky
diodes are typically the best choice, but some PN diodes
can be used if they turn on fast enough to limit the leak-
age inductance spike. Choose a diode that has a reverse-
voltage rating higher than the maximum SW pin voltage.
The Zener diode breakdown voltage should be chosen to
balance power loss and switch voltage protection. The best
compromise is to choose the largest voltage breakdown
with 5V margin. Use the following equation to make the
proper choice:
VZENNER(MAX) ≤ 145V – VIN(MAX)
For an application with a maximum input voltage of 80V,
choose a 62V Zener diode, the VZENER(MAX) of which is
around 65V. The power loss in the DZ snubber determines
the power rating of the Zener diode. A 1.5W Zener diode
is typically recommended.
The recommended approach for designing an RC snub-
ber is to measure the period of the ringing on the SW pin
when the power switch turns off without the snubber and
LT8304/LT8304-1
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APPLICATIONS INFORMATION
Undervoltage Lockout (UVLO)
A resistive divider from VIN to the EN/UVLO pin imple-
ments undervoltage lockout (UVLO). The EN/UVLO enable
falling threshold is set at 1.214V with 14mV hysteresis. In
addition, the EN/UVLO pin sinks 2.5µA when the voltage
on the pin is below 1.214V. This current provides user
programmable hysteresis based on the value of R1. The
programmable UVLO thresholds are:
VIN(UVLO+)=1.228V • R1+R2
( )
R2 +2.5µA •R
1
VIN(UVLO–)=1.214V • R1+R2
( )
R2
Figure 7 shows the implementation of external shutdown
control while still using the UVLO function. The NMOS
grounds the EN/UVLO pin when turned on, and puts the
LT8304 in shutdown with quiescent current less than 3µA.
LT8304
GND
EN/UVLO
R1
RUN/STOP
CONTROL
(OPTIONAL)
R2
VIN
8304 F07
Figure 7. Undervoltage Lockout (UVLO)
Minimum Load Requirement
The LT8304 samples the isolated output voltage from
the primary-side flyback pulse waveform. The flyback
pulse occurs once the primary switch turns off and the
secondary winding conducts current. In order to sample
the output voltage, the LT8304 has to turn on and off for a
minimum amount of time and with a minimum frequency.
The LT8304 delivers a minimum amount of energy even
during light load conditions to ensure accurate output volt-
age information. The minimum energy delivery creates a
minimum load requirement, which can be approximately
estimated as:
ILOAD(MIN) =
L
P
2
RI
•I
SW(MIN)
•f
MIN
2•VOUT
LPRI = Transformer primary inductance
ISW(MIN) = Minimum switch current limit = 0.53A (MAX)
fMIN = Minimum switching frequency = 14kHz (MAX)
The LT8304 typically needs less than 0.5% of its full
output power as minimum load. Alternatively, a Zener
diode with its breakdown of 10% higher than the output
voltage can serve as a minimum load if pre-loading is not
acceptable. For a 5V output, use a 5.6V Zener with cathode
connected to the output. The LT8304-1 requires slightly
higher minimum load, typically 2% of full load.
Output Short Protection
When the output is heavily overloaded or shorted to ground,
the reflected SW pin waveform rings longer than the in-
ternal blanking time. After the 350ns minimum switch-off
time, the excessive ringing falsely triggers the boundary
mode detector and turns the power switch back on again
before the secondary current falls to zero. Under this
condition, the LT8304 runs into continuous conduction
mode at 350kHz (TYP) maximum switching frequency. If
the sampled RREF voltage is still less than 0.6V after 11ms
(typ) soft-start timer, the LT8304 initiates a new soft-start
cycle. If the sampled RREF voltage is larger than 0.6V after
11ms, the switch current may run away and exceed the
2.4A maximum current limit. Once the switch current hits
3.6A over current limit, the LT8304 also initiates a new
soft-start cycle. Under either condition, the new soft-start
cycle throttles back both the switch current limit and switch
frequency. The output short-circuit protection prevents the
switch current from running away and limits the average
output diode current.
LT8304/LT8304-1
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APPLICATIONS INFORMATION
Design Example
Use the following design example as a guide to designing
applications for the LT8304. The design example involves
designing a 5V output with a 2.8A load current and an
input range from 36V to 75V.
VIN(MIN) = 36V, VIN(NOM) = 48V, VIN(MAX) = 75V,
VOUT = 5V, IOUT = 2.8A
Step 1: Select the transformer turns ratio.
NPS <
150V – V
IN(MAX)
– V
LEAKAGE
V
OUT
+V
F
VLEAKAGE = Margin for transformer leakage spike = 40V
VF = Output diode forward voltage = ~0.3V
Example:
NPS <
150V – 75V – 40V
5V +0.3V
=6.6
The choice of transformer turns ratio is critical in determin-
ing output current capability of the converter. Table2 shows
the switch voltage stress and output current capability at
different transformer turns ratio.
Table 2. Switch Voltage Stress and Output Current Capability vs
Turns Ratio
NPS
VSW(MAX) at
VIN(MAX) (V)
IOUT(MAX) at
VIN(MIN) (A) DUTY CYCLE (%)
4:1 96.2 2.27 22 to 37
5:1 101.5 2.59 26 to 42
6:1 106.8 2.87 30 to 47
Clearly, only NPS = 6 can meet the 2.8A output current
requirement, so NPS = 6 is chosen as the turns ratio in
this example.
Step 2: Determine the primary inductance.
Primary inductance for the transformer must be set above
a minimum value to satisfy the minimum switch-off and
switch-on time requirements:
LPRI ≥
t
OFF(MIN)
•N
PS
• V
OUT
+V
F
( )
ISW(MIN)
LPRI ≥tON(MIN) • VIN(MAX)
ISW(MIN)
tOFF(MIN) = 350ns
tON(MIN) = 160ns
ISW(MIN) = 0.48A
Example:
LPRI ≥
350ns •6 • 5V +0.3V
( )
0.48A =23µH
LPRI ≥160ns•75V
0.48A
=25µH
Most transformers specify primary inductance with a toler-
ance of ±20%. With other component tolerance considered,
choose a transformer with its primary inductance 40% to
60% larger than the minimum values calculated above.
LPRI = 40µH is then chosen in this example.
The transformer also needs to be rated for the correct
saturation current level across line and load conditions.
A saturation current rating larger than 2.8A is necessary
to work with the LT8304. The 750315125 from Würth is
chosen as the flyback transformer.
LT8304/LT8304-1
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APPLICATIONS INFORMATION
Step 3: Choose the output diode.
Two main criteria for choosing the output diode include
forward current rating and reverse-voltage rating. The
maximum load requirement is a good first-order guess
at the average current requirement for the output diode.
Under output short-circuit condition, the output diode
needs to conduct much higher current. Therefore, a con-
servative metric is 60% of the maximum switch current
limit multiplied by the turns ratio:
IDIODE(MAX) = 0.6 • ISW(MAX) • NPS
Example:
IDIODE(MAX) = 8.6A
Next calculate reverse voltage requirement using maxi-
mum VIN:
VREVERSE =VOUT +
V
IN(MAX)
N
PS
Example:
VREVERSE =5V +
75V
6
=17.5V
The PDS835L (8A, 35V diode) from Diodes Inc. is chosen.
Step 4: Choose the output capacitor.
The output capacitor should be chosen to minimize the
output voltage ripple while considering the increase in size
and cost of a larger capacitor. Use the following equation
to calculate the output capacitance:
COUT =
L
PRI
•I
SW
2
2• V
OUT
•∆V
OUT
Example:
Design for output voltage ripple less than ±1% of VOUT,
i.e., 100mV.
COUT =40µH• 2.4A
( )
2
2•5V •0.1V
=230µF
Remember ceramic capacitors lose capacitance with
applied voltage. The capacitance can drop up to 40% of
quoted capacitance at the maximum voltage rating. So
three 100µF, 10V rating ceramic capacitors are chosen.
Step 5: Design snubber circuit.
The snubber circuit protects the power switch from leak-
age inductance voltage spike. A (RC + DZ) snubber is
recommended for this application. A 220pF capacitor in
series with a 100Ω resistor is chosen as the RC snubber.
The maximum Zener breakdown voltage is set according
to the maximum VIN:
VZENNER(MAX) ≤ 145V – VIN(MAX)
Example:
VZENNER(MAX) ≤ 145V – 75V = 70V
A 62V Zener with a maximum of 65V will provide optimal
protection and minimize power loss. So a 62V, 1.5W Zener
from Central Semiconductor (CMZ5944B) is chosen.
Choose a diode that is fast and has sufficient reverse
voltage breakdown:
VREVERSE > VSW(MAX)
VSW(MAX) = VIN(MAX) + VZENNER(MAX)
Example:
VREVERSE > 150V
A 150V, 1A diode from Diodes Inc. (DFLS1150) is chosen.
Step 6: Select the RREF and RFB resistors.
Use the following equation to calculate the starting values
for RREF and RFB:
RFB =RREF •NPS •VOUT +V
FTO
( )
( )
V
REF
R
REF
=10k
Example:
RFB =
10k • 6 • 5V +0.3V
( )
1.00V
=318k
For 1% standard values, a 316k resistor is chosen.
LT8304/LT8304-1
19
8304fa
For more information www.linear.com/LT8304
APPLICATIONS INFORMATION
Step 7: Adjust RFB resistor based on output voltage.
Build and power up the application with application com-
ponents and measure the regulated output voltage. Adjust
RFB resistor based on the measured output voltage:
RFB(NEW) =
V
OUT
VOUT(MEASURED)
•RFB
Example:
RFB =
5V
5.11V
•316k =309k
Step 8: Select RTC resistor based on output voltage
temperature variation.
Measure output voltage in a controlled temperature envi-
ronment like an oven to determine the output temperature
coefficient. Measure output voltage at a consistent load
current and input voltage, across the operating tempera-
ture range.
Calculate the temperature coefficient of VF:
–δVF/δT
( )
=VOUT T1
( )
– VOUT T2
( )
T1– T2
RTC =3.35mV/°C
–δVF/δT
( )
•RFB
NPS
Example:
–δVF/δT
( )
=
5.149V – 4.977V
100°C – 0°C
( )
=1.72mV / °C
RTC =3.35mV/°C
1.72mV/°C•309
6
=100k
Step 9: Select the EN/UVLO resistors.
Determine the amount of hysteresis required and calculate
R1 resistor value:
VIN(HYS) = 2.5µA • R1
Example:
Choose 2.5V of hysteresis, R1 = 1M
Determine the UVLO thresholds and calculate R2 resistor
value:
V
IN(UVLO+)=
1.228V •R1+R2
( )
R2
+2.5µA •R
1
Example:
Set VIN UVLO rising threshold to 34.5V:
R2 = 40.2k
VIN(UVLO+) = 34.3V
VIN(UNLO–) = 31.4V
Step 10: Ensure minimum load.
The theoretical minimum load can be approximately
estimated as:
ILOAD(MIN) =40µH• 0.53A
( )
2
•14kHz
2•5V
=15.7mA
Remember to check the minimum load requirement in
real application. The minimum load occurs at the point
where the output voltage begins to climb up as the con-
verter delivers more energy than what is consumed at
the output. The real minimum load for this application is
about 20mA. In this example, a 249Ω resistor is selected
as the minimum load.
LT8304/LT8304-1
20
8304fa
For more information www.linear.com/LT8304
TYPICAL APPLICATIONS
18V to 80VIN/3.3VOUT Isolated Flyback Converter
18V to 80VIN/5VOUT Isolated Flyback Converter
VIN
LT8304
SW
40µH
VIN
18V TO 80V
T1
8:1
0.63µH
RFB
RREF
C3
220pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
330µF
×2
VOUT+
3.3V
25mA TO 3.4A (VIN = 24V)
25mA TO 4.8A (VIN = 48V)
25mA TO 5.6A (VIN = 72V)
VOUT–
R3
100Ω
R4
274k
•
•
R6
100k
R2
88.7k
R1
1M
R5
10k
D1: DIODES DFLS1150
D2: DIODES SBR15U30SP5
T1: SUMIDA 13324-T083
Z1: CENTRAL CMZ5944B
8304 TA02a
TC
EN/UVLO
GND
INTVCC
VIN
LT8304
SW
40µH
VIN
18V TO 80V
T1
6:1
1.1µH
RFB
RREF
C3
220pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
100µF
×3
VOUT+
5V
20mA TO 2.4A (VIN = 24V)
20mA TO 3.6A (VIN = 48V)
20mA TO 4.2A (VIN = 72V)
VOUT–
R3
100Ω
R4
309k
•
•
R6
100k
R2
88.7k
R1
1M
R5
10k
D1: DIODES DFLS1150
D2: DIODES PDS835L
T1: WURTH 750315125
Z1: CENTRAL CMZ5944B
8304 TA03a
TC
EN/UVLO
GND
INTVCC
0
0.8
1.6
2.4
3.2
4.0
4.8
5.6
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA02b
VIN = 24V
VIN = 48V
VIN = 72V
0
0.6
1.2
1.8
2.4
3.0
3.6
4.2
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA03b
VIN = 24V
VIN = 48V
VIN = 72V
Efficiency vs Load Current
Efficiency vs Load Current
LT8304/LT8304-1
21
8304fa
For more information www.linear.com/LT8304
TYPICAL APPLICATIONS
18V to 80VIN/12VOUT Isolated Flyback Converter
18V to 80VIN/24VOUT Isolated Flyback Converter
VIN
LT8304
SW
40µH
VIN
18V TO 80V
T1
2:1
10µH
RFB
RREF
C3
220pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
47µF
VOUT+
12V
10mA TO 1.0A (VIN = 24V)
10mA TO 1.4A (VIN = 48V)
10mA TO 1.6A (VIN = 72V)
VOUT–
R3
100Ω
R4
237k
•
•
R6
OPEN
R2
88.7k
R1
1M
R5
10k
D1: DIODES DFLS1150
D2: DIODES PMEG6030EP
T1: WURTH 750315126
Z1: CENTRAL CMZ5944B
8304 TA04a
TC
EN/UVLO
GND
INTVCC
VIN
LT8304
SW
40µH
VIN
18V TO 80V
T1
1:1
40µH
RFB
RREF
C3
220pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
10µF
VOUT+
24V
5mA TO 0.5A (VIN = 24V)
5mA TO 0.7A (VIN = 48V)
5mA TO 0.8A (VIN = 72V)
VOUT–
R3
100Ω
R4
237k
•
•
R6
OPEN
R2
88.7k
R1
1M
R5
10k
D1: DIODES DFLS1150
D2: DIODES SBR2U150SA
T1: SUMIDA 13324-T084
Z1: CENTRAL CMZ5944B
8304 TA05a
TC
EN/UVLO
GND
INTVCC
0
0.4
0.8
1.2
1.6
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA04b
VIN = 24V
VIN = 48V
VIN = 72V
0
0.1
0.2
0.3
0.4
0.5
0.6
0.8
0.7
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA05b
VIN = 24V
VIN = 48V
VIN = 72V
Efficiency vs Load Current
Efficiency vs Load Current
LT8304/LT8304-1
22
8304fa
For more information www.linear.com/LT8304
18V to 80VIN/48VOUT Isolated Flyback Converter
VIN
LT8304
SW
40µH
VIN
18V TO 80V
T1
1:2
160µH
RFB
RREF
C3
220pF
Z1
D1
D2
C1
10µF
C2
1µF
C4
2.2µF
VOUT+
48V
2mA TO 0.24A (VIN = 24V)
2mA TO 0.34A (VIN = 48V)
2mA TO 0.40A (VIN = 72V)
VOUT–
R3
100Ω
R4
232k
•
•
R6
OPEN
R2
88.7k
R1
1M
R5
10k
D1: DIODES DFLS1150
D2: DIODES SBR1U400P1
T1: SUMIDA 13324-T085
Z1: CENTRAL CMZ5944B
8304 TA06a
TC
EN/UVLO
GND
INTVCC
0
0.1
0.2
0.3
0.4
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA06b
VIN = 24V
VIN = 48V
VIN = 72V
Efficiency vs Load Current
TYPICAL APPLICATIONS
LT8304/LT8304-1
23
8304fa
For more information www.linear.com/LT8304
Efficiency, VOUT = 200V Load Regulation, VOUT = 200V
4V to 36VIN/200VOUT Isolated Flyback Converter
VIN
LT8304-1
SW
40µH
VIN
4V TO 36V
T1
1:5
1mH
RFB
RREF
D1
10µF
50V
1µF
6.3V
0.33µF
250V
VOUT
+
200V
0.3mA TO 12mA (VIN = 4V)
0.5mA TO 35mA (VIN
= 12V)
1.5mA TO 75mA (VIN
= 36V)
VOUT–
392k
•
•
10k 10pF
T1: SUMIDA 13324-T086
D1: CENTRAL CMMR1U-06 TR
8304 TA07a
GND
EN/UVLO
TC
INTVCC
TYPICAL APPLICATIONS
V
IN
= 4V
V
IN
= 12V
V
IN
= 36V
LOAD CURRENT (mA)
0
15
30
45
60
75
60
65
70
75
80
85
90
95
EFFICIENCY (%)
Efficiency, V
OUT
= 200V
8304 TA07b
V
IN
= 4V
V
IN
= 12V
V
IN
= 36V
LOAD CURRENT (mA)
0
15
30
45
60
75
190
195
200
205
210
OUTPUT VOLTAGE (V)
Load Regulation, V
OUT
= 200V
8304 TA07c
LT8304/LT8304-1
24
8304fa
For more information www.linear.com/LT8304
Efficiency, VOUT = 400V Load Regulation, VOUT = 400V
4V to 18VIN/400VOUT Isolated Flyback Converter
VIN
LT8304-1
SW
40µH
VIN
4V TO 18V
T1
1:10
4mH
RFB
RREF
D1
10µF
25V
1µF
6.3V
0.15µF
600V
VOUT+
400V
0.4mA TO 6mA (VIN = 4V)
0.4mA TO 20mA (VIN
= 12V)
0.4mA TO 30mA (VIN
= 18V)
VOUT–
392k
•
•
10k 10pF
T1: SUMIDA 13324-T087
D1: CENTRAL CMMR1U-06 TR
8304 TA08a
GND
EN/UVLO
TC
INTVCC
V
IN
= 4V
V
IN
= 12V
V
IN
= 18V
LOAD CURRENT (mA)
0
5
10
15
20
25
30
40
50
60
70
80
90
EFFICIENCY (%)
OUT
8304 TA08b
V
IN
= 4V
V
IN
= 12V
V
IN
= 18V
LOAD CURRENT (mA)
0
5
10
15
20
25
30
380
390
400
410
420
OUTPUT VOLTAGE (V)
Load Regulation, V
OUT
= 400V
8304 TA08c
LT8304/LT8304-1
25
8304fa
For more information www.linear.com/LT8304
–18V to –80VIN/–12VOUT Negative Buck Converter
Efficiency vs Load Current
Efficiency vs Load Current
–4V to –80VIN/12VOUT Buck-Boost Converter
TYPICAL APPLICATIONS
VIN SW
LT8304
L1
33µH D1
Z1
GND
RFB
RREF
EN/UVLO
C3
47µF
D1: DIODES SBR2U150SA
L1: WURTH 744771133
Z1: CENTRAL CMHZ5243B
C2
1µF
VIN
–4V TO –80V
C1
10µF
R5
10k
8304 TA09a
V
OUT
+12V
5mA TO 0.25A (VIN
= –5V)
5mA TO 0.7A (VIN
= –24V)
5mA TO 0.8A (VIN
= –48V)
5mA TO 0.9A (VIN
= –72V)
R4
121k
INTVCC
0
0.1
0.2
0.3
0.4
0.5
0.6
0.7
0.9
0.8
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA09b
VIN = –5V
VIN = –24V
VIN = –48V
VIN = –72V
VIN
LT8304
L1
33µH
VOUT
–12V
10mA TO 1A
D1 Z1
SW
RREF
EN/UVLO
RFB
EN/UVLO D1: DIODES SBR2U150SA
L1: WURTH 744771133
Z1: CENTRAL CMHZ5243B
C2
1µF
VIN
–18V TO –80V
C1
10µF
C3
47µF
R5
10k
8304 TA10a
R4
121k
R2
71.5k
R1
806k
INTVCC
0
0.2
0.4
0.6
0.8
1.0
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA10b
VIN = –24V
VIN = –48V
VIN = –72V
LT8304/LT8304-1
26
8304fa
For more information www.linear.com/LT8304
PACKAGE DESCRIPTION
.016 – .050
(0.406 – 1.270)
.010 – .020
(0.254 – 0.508)× 45°
0°– 8° TYP
.008 – .010
(0.203 – 0.254)
S8E 1015 REV C
.053 – .069
(1.346 – 1.752)
.014 – .019
(0.355 – 0.483)
TYP
.004 – .010
(0.101 – 0.254)
0.0 – 0.005
(0.0 – 0.130)
.080 – .099
(2.032 – 2.530)
.118 – .139
(2.997 – 3.550)
.050
(1.270)
BSC
1234
.150 – .157
(3.810 – 3.988)
NOTE 3
87
.005 (0.13) MAX
65
.189 – .197
(4.801 – 5.004)
NOTE 3
.228 – .244
(5.791 – 6.197)
.160 ±.005
(4.06 ±0.127)
.118
(2.99)
REF
RECOMMENDED SOLDER PAD LAYOUT
.045 ±.005
(1.143 ±0.127)
.050
(1.27)
BSC
INCHES
(MILLIMETERS)
NOTE:
1. DIMENSIONS IN
2. DRAWING NOT TO SCALE
3. THESE DIMENSIONS DO NOT INCLUDE MOLD FLASH OR PROTRUSIONS.
MOLD FLASH OR PROTRUSIONS SHALL NOT EXCEED .010" (0.254mm)
4. STANDARD LEAD STANDOFF IS 4mils TO 10mils (DATE CODE BEFORE 542)
5. LOWER LEAD STANDOFF IS 0mils TO 5mils (DATE CODE AFTER 542)
S8E Package
8-Lead Plastic SOIC (Narrow .150 Inch) Exposed Pad
(Reference LTC DWG # 05-08-1857 Rev C)
.089
(2.26)
REF
.030 ±.005
(0.76 ±0.127)
TYP
.245
(6.22)
MIN
45
Please refer to http://www.linear.com/product/LT8304#packaging for the most recent package drawings.
LT8304/LT8304-1
27
8304fa
For more information www.linear.com/LT8304
Information furnished by Linear Technology Corporation is believed to be accurate and reliable.
However, no responsibility is assumed for its use. Linear Technology Corporation makes no representa-
tion that the interconnection of its circuits as described herein will not infringe on existing patent rights.
REVISION HISTORY
REV DATE DESCRIPTION PAGE NUMBER
A 02/17 Added LT8304-1 and H-Grade options
Changed TC Pin Current conditions
Changed TC pin description to °C
Added High Step-Up VOUT Applications section
Updated Predesigned Transformers – Typical Specifications table
Revised Turns Ratio section
Added new application circuits and graphs
All
3
7
9
13
14
23, 24
LT8304/LT8304-1
28
8304fa
For more information www.linear.com/LT8304
LINEAR TECHNOLOGY CORPORATION 2016
LT 0217 REV A • PRINTED IN USA
Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 ● FAX: (408) 434-0507 ● www.linear.com/LT8304
RELATED PARTS
TYPICAL APPLICATION
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Efficiency vs Load Current
PART NUMBER DESCRIPTION COMMENTS
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VIN SW
LT8304
L1
150µH D1
Z1
Z2
GND
RFB
RREF
EN/UVLO C3
1µF
D1: DIODES DFLS1200
L1: COILCRAFT DS5022P-154MLB
Z1, Z2: CENTRAL CMHZ5207B
C2
1µF
VIN
4V TO 100V
C1
10µF
R5
3.57k
8304 TA11a
V
OUT
140V
1.5mA TO 25mA (VIN = 5V)
2mA TO 300mA (VIN = 48V)
7mA TO 700mA (VIN = 100V)
R4
499k
R3
1M
INTVCC
LOAD CURRENT (mA)
1
10
100
1000
40
50
60
70
80
90
100
EFFICIENCY (%)
8304 TA11b
VIN = 5V
VIN = 48V
VIN = 100V