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RON LM3404/04HV

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VCC

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An IMPORTANT NOTICE at the end of this data sheet addresses availability, warranty, changes, use in safety-critical applications,

intellectual property matters and other important disclaimers. PRODUCTION DATA.

LM3404

LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

LM3404xx 1-A Constant Current Buck Regulator for Driving High Power LEDs

1 Features

1• Integrated 1-A MOSFET

• VIN Range 6 V to 42 V (LM3404)

• VIN Range 6 V to 75 V (LM3404HV)

• 1.2-A Output Current Overtemperature

• Cycle-by-Cycle Current Limit

• No Control Loop Compensation Required

• Separate PWM Dimming and Low Power

Shutdown

• Supports All-Ceramic Output Capacitors and

Capacitor-less Outputs

• Thermal Shutdown Protection

• SOIC-8 Package, SO PowerPAD™-8 Package

2 Applications

• LED Drivers

• Constant Current Sources

• Automotive Lighting

• General Illumination

• Industrial Lighting

3 Description

The LM3404 and LM3404HV devices are monolithic

switching regulators designed to deliver constant

currents to high power LEDs. Ideal for automotive,

industrial, and general lighting applications, these

devices contain a high-side N-channel MOSFET

switch with a current limit of

1.5-A (typical) for step-down (Buck) regulators.

Hysteretic controlled on-time and an external resistor

allow the converter output voltage to adjust as

needed to deliver a constant current to series and

series-parallel connected LED arrays of varying

number and type. Some features are: LED dimming

using pulse width modulation (PWM), broken or open

LED protection, low-power shutdown, and thermal

shutdown.

Device Information(1)

PART NUMBER PACKAGE BODY SIZE (NOM)

LM3404, LM3404HV SOIC (8) 3.91 mm × 4.90 mm

SO PowerPAD (8) 3.90 mm × 4.89 mm

(1) For all available packages, see the orderable addendum at

the end of the data sheet.

Typical Application Diagram

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Table of Contents

1 Features.................................................................. 1

2 Applications ........................................................... 1

3 Description............................................................. 1

4 Revision History..................................................... 2

5 Pin Configuration and Functions......................... 3

6 Specifications......................................................... 4

6.1 Absolute Maximum Ratings ..................................... 4

6.2 ESD Ratings.............................................................. 4

6.3 Recommended Operating Conditions....................... 4

6.4 Thermal Information.................................................. 5

6.5 Electrical Characteristics........................................... 5

6.6 Switching Characteristics.......................................... 6

6.7 Typical Characteristics.............................................. 7

7 Detailed Description............................................ 10

7.1 Overview................................................................. 10

7.2 Functional Block Diagram....................................... 10

7.3 Feature Description................................................. 11

7.4 Device Functional Modes........................................ 14

8 Application and Implementation ........................ 15

8.1 Application Information............................................ 15

8.2 Typical Applications ................................................ 23

9 Power Supply Recommendations...................... 32

10 Layout................................................................... 32

10.1 Layout Guidelines ................................................. 32

10.2 Layout Example .................................................... 33

11 Device and Documentation Support................. 34

11.1 Device Support .................................................... 34

11.2 Related Links ........................................................ 34

11.3 Community Resources.......................................... 34

11.4 Trademarks........................................................... 34

11.5 Electrostatic Discharge Caution............................ 34

11.6 Glossary................................................................ 34

12 Mechanical, Packaging, and Orderable

Information........................................................... 34

4 Revision History

NOTE: Page numbers for previous revisions may differ from page numbers in the current version.

Changes from Revision F (May 2013) to Revision G Page

• Added ESD Ratings table, Feature Description section, Device Functional Modes,Application and Implementation

section, Power Supply Recommendations section, Layout section, Device and Documentation Support section, and

Mechanical, Packaging, and Orderable Information section.................................................................................................. 1

Changes from Revision E (May 2013) to Revision F Page

• Changed layout of National Data Sheet to TI format ........................................................................................................... 31

RON

2VIN

3BOOT

VCC

GND

DIM 6

CS 5

DAP

RON

2VIN

3BOOT

VCC

GND

DIM 6

CS 5

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5 Pin Configuration and Functions

D Package

8-Pin SOIC

Top View DDA Package

8-Pin SO With PowerPAD

Top View

Pin Functions

PIN I/O DESCRIPTION

NAME NO.

SW 1 O Switch pin. Connect this pin to the output inductor and Schottky diode.

BOOT 2 O MOSFET drive bootstrap pin. Connect a 10-nF ceramic capacitor from this pin to SW.

DIM 3 I Input for PWM dimming. Connect a logic-level PWM signal to this pin to enable and disable

the power MOSFET and reduce the average light output of the LED array.

GND 4 — Ground pin. Connect this pin to system ground.

CS 5 I Current sense feedback pin. Set the current through the LED array by connecting a resistor

from this pin to ground.

RON 6 I On-time control pin. A resistor connected from this pin to VIN sets the regulator controlled

on-time.

VCC 7 O Output of the internal 7-V linear regulator. Bypass this pin to ground with a minimum 0.1-µF

ceramic capacitor with X5R or X7R dielectric.

VIN 8 I Input voltage pin. Nominal operating input range for this pin is 6 V to 42 V (LM3404) or 6 V

to 75 V (LM3404HV).

DAP — PowerPAD. Connect to ground. Place 4-6 vias from DAP to bottom layer ground plane.

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LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

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(1) Stresses beyond those listed under Absolute Maximum Ratings may cause permanent damage to the device. These are stress ratings

only, which do not imply functional operation of the device at these or any other conditions beyond those indicated under Recommended

Operating Conditions. Exposure to absolute-maximum-rated conditions for extended periods may affect device reliability.

(2) If Military or Aerospace specified devices are required, contact the Texas Instruments Semiconductor Sales Office or Distributors for

availability and specifications.

6 Specifications

6.1 Absolute Maximum Ratings

over operating free-air temperature range (unless otherwise noted)(1)(2)

MIN MAX UNIT

VIN to GND LM3404 –0.3 45 V

LM3404HV –0.3 76

BOOT to GND LM3404 –0.3 59 V

LM3404HV –0.3 90

SW to GND LM3404 –1.5 45 V

LM3404HV –1.5 76

BOOT to VCC LM3404 –0.3 45 V

LM3404HV –0.3 76

BOOT to SW –0.3 14 V

VCC to GND –0.3 14 V

DIM to GND –0.3 7 V

CS to GND –0.3 7 V

RON to GND –0.3 7 V

Soldering information Lead temperature (soldering, 10 s) 260 °C

Infrared or convection reflow (15 s) 235

Junction temperature 150 °C

Storage temperature –65 125 °C

(1) JEDEC document JEP155 states that 500-V HBM allows safe manufacturing with a standard ESD control process.

(2) The human body model is a 100-pF capacitor discharged through a 1.5-kΩresistor into each pin.

(3) JEDEC document JEP157 states that 250-V CDM allows safe manufacturing with a standard ESD control process.

6.2 ESD Ratings VALUE UNIT

V(ESD) Electrostatic discharge Human-body model (HBM), per ANSI/ESDA/JEDEC JS-001(1)(2) ±2000 V

Charged-device model (CDM), per JEDEC specification JESD22-C101(3) ±1000

(1) Absolute Maximum Ratings indicate limits beyond which damage to the device may occur. Recommended Operating Conditions indicate

conditions for which the device is intended to be functional, but specific performance is not ensured. For specifications and the test

conditions, see Electrical Characteristics.

6.3 Recommended Operating Conditions

over operating free-air temperature range (unless otherwise noted)(1)

MIN NOM MAX UNIT

VIN LM3404 6 42 V

LM3404HV 6 75

Junction Temperature Range LM3404 –40 125 °C

LM34040HV –40 125

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(1) For more information about traditional and new thermal metrics, see the Semiconductor and IC Package Thermal Metrics application

report, SPRA953.

6.4 Thermal Information

THERMAL METRIC(1) LM3404, LM3404HV

UNITSOIC SO PowerPAD

8 PINS 8 PINS

RθJA Junction-to-ambient thermal resistance 106.8 44.7 °C/W

RθJC(top) Junction-to-case (top) thermal resistance 46.2 51.2 °C/W

RθJB Junction-to-board thermal resistance 48.7 24.5 °C/W

ψJT Junction-to-top characterization parameter 6.7 6.8 °C/W

ψJB Junction-to-board characterization parameter 48 24.4 °C/W

RθJC(bot) Junction-to-case (bottom) thermal resistance N/A 2.6 °C/W

(1) Typical specifications represent the most likely parametric norm at 25°C operation.

(2) VCC provides self bias for the internal gate drive and control circuits. Device thermal limitations limit external loading.

6.5 Electrical Characteristics

VIN = 24 V (unless otherwise noted).

–40°C ≤TJ≤125°C. (1)

PARAMETER CONDITIONS MIN TYP MAX UNIT

REGULATION AND OVERVOLTAGE COMPARATORS

VREF-REG CS Regulation Threshold CS Decreasing, SW turns on 194 200 206 mV

VREF-0V CS Overvoltage Threshold CS Increasing, SW turns off 300 mV

ICS CS Bias Current CS = 0 V 0.1 µA

SHUTDOWN

VSD-TH Shutdown Threshold RON / SD Increasing 0.3 0.7 1.05 V

VSD-HYS Shutdown Hysteresis RON / SD Decreasing 40 mV

INTERNAL REGULATOR

VCC-REG VCC Regulated Output 6.4 7 7.4 V

VIN-DO VIN – VCC ICC = 5 mA, 6 V < VIN < 8 V 300 mV

VCC-BP-TH VCC Bypass Threshold VIN Increasing 8.8 V

VCC-BP-HYS VCC Bypass Hysteresis VIN Decreasing 230 mV

VCC-Z-6 VCC Output Impedance

(0 mA < ICC < 5 mA)

VIN = 6 V 55 Ω

VCC-Z-8 VIN = 8 V 50 Ω

VCC-Z-24 VIN = 24 V 0.4 Ω

VCC-LIM VCC Current Limit (2) VIN = 24 V, VCC = 0 V 16 mA

VCC-UV-TH VCC Undervoltage Lock-out

Threshold VCC Increasing 5.3 V

VCC-UV-HYS VCC Undervoltage Lock-out

Hysteresis VCC Decreasing 150 mV

VCC-UV-DLY VCC Undervoltage Lock-out Filter

Delay 100-mV Overdrive 3 µs

IIN-OP IIN Operating Current Non-switching, CS = 0.5 V 625 900 µA

IIN-SD IIN Shutdown Current RON / SD = 0 V 95 180 µA

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LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

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Electrical Characteristics (continued)

VIN = 24 V (unless otherwise noted).

–40°C ≤TJ≤125°C. (1)

PARAMETER CONDITIONS MIN TYP MAX UNIT

CURRENT LIMIT

ILIM Current Limit Threshold 1.2 1.5 1.8 A

DIM COMPARATOR

VIH Logic High DIM Increasing 2.2 V

VIL Logic Low DIM Decreasing 0.8 V

IDIM-PU DIM Pullup Current DIM = 1.5 V 80 µA

MOSFET AND DRIVER

RDS-ON Buck Switch On Resistance ISW = 200 mA, BST-SW = 6.3 V 0.37 0.75 Ω

VDR-UVLO BST Undervoltage Lock-out

Threshold BST–SW Increasing 1.7 3 4 V

VDR-HYS BST Undervoltage Lock-out

Hysteresis BST–SW Decreasing 400 mV

THERMAL SHUTDOWN

TSD Thermal Shutdown Threshold 165 °C

TSD-HYS Thermal Shutdown Hysteresis 25 °C

6.6 Switching Characteristics

over operating free-air temperature range (unless otherwise noted)

PARAMETER TEST CONDITIONS MIN TYP MAX UNIT

SYSTEM PARAMETERS - LM3404

tON-1 On-time 1 VIN = 10 V, RON = 200 kΩ2.1 2.75 3.4 µs

tON-2 On-time 2 VIN = 40 V, RON = 200 kΩ515 675 835 ns

SYSTEM PARAMETERS - LM3404HV

tON-1 On-time 1 VIN = 10 V, RON = 200 kΩ2.1 2.75 3.4 µs

tON-2 On-time 2 VIN = 70 V, RON = 200 kΩ325 415 505 ns

OFF TIMER

tOFF-MIN Minimum Off-time CS = 0 V 270 ns

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6.7 Typical Characteristics

spacer

VIN = 24 V

Figure 1. VREF vs Temperature

TA= 25°C

Figure 2. VREF vs VIN, LM3404

TA= 25°C

Figure 3. VREF vs VIN, LM3404HV

VIN = 24 V

Figure 4. Current Limit vs Temperature

TA= 25°C

Figure 5. Current Limit vs VIN, LM3404

TA= 25°C

Figure 6. Current Limit vs VIN, LM3404HV

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Typical Characteristics (continued)

spacer

TA= 25°C

Figure 7. TON vs VIN, RON = 100 kΩ

TA= 25°C

Figure 8. TON vs VIN

TA= 25°C

Figure 9. TON vs VIN

TA= 25°C

Figure 10. TON vs RON, LM3404

TA= 25°C

Figure 11. TON vs RON, LM3404HV

TA= 25°C

Figure 12. VCC vs VIN

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Typical Characteristics (continued)

spacer

TA= 25°C

Figure 13. VO-MAX vs fSW, LM3404

TA= 25°C

Figure 14. VO-MIN vs fSW, LM3404

TA= 25°C

Figure 15. VO-MAX vs fSW, LM3404HV

TA= 25°C

Figure 16. VO-MIN vs fSW, LM3404HV

BOOT

VCC

VIN

DIM

RON

GND

VIN

SENSE

7V BIAS

REGULATOR

BYPASS

SWITCH

VCC

UVLO THERMAL

SHUTDOWN

ON TIMER

RON Complete

Start

300 ns MIN

OFF TIMER

Complete

Start

LOGIC

CURRENT

LIMIT OFF

TIMER

BUCK

SWITCH

CURRENT

SENSE

LEVEL

SHIFT

GATE DRIVE

UVLO VIN

-1.5A

0.7V

0.2V

0.3V

1.5V

75 PASD

LM3404

LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

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7 Detailed Description

7.1 Overview

The LM3404 and LM3404HV devices are buck regulators with a wide input voltage range, low voltage reference,

and a fast output enable and disable function. These features combine to make these devices ideal for use as a

constant current source for LEDs with forward currents as high as 1.2 A. The controlled on-time (COT)

architecture is a combination of hysteretic mode control and a one-shot on-timer that varies inversely with input

voltage. Hysteretic operation eliminates the need for small-signal control loop compensation. When the converter

runs in continuous conduction mode (CCM) the controlled on-time maintains a constant switching frequency over

the range of input voltage. Fast transient response, PWM dimming, a low power shutdown mode, and simple

output overvoltage protection round out the functions of the LM3404 LM3404HV devices.

7.2 Functional Block Diagram

IL-MIN =0.2

RSNS L

-VO x tSNS

fSW =VO

1.34 x 10-10 x RON

VO = n x VF + 200 mV

LM3404/04HV

RSNS

One-shot

Comparator

LED 1

LED n

VSNS

VREF

tON = 1.34 x 10-10 xRON

VIN

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7.3 Feature Description

7.3.1 Controlled On-Time Overview

Figure 17 shows the feedback system used to control the current through an array of LEDs. A voltage signal,

VSNS, is created as the LED current flows through the current setting resistor, RSNS, to ground. VSNS is fed back

to the CS pin, where it is compared against a 200-mV reference, VREF. The on-comparator turns on the power

MOSFET when VSNS falls below VREF. The power MOSFET conducts for a controlled on-time, tON, set by an

external resistor, RON, and by the input voltage, VIN. On-time is governed by the Equation 1.

(1)

At the conclusion of tON the power MOSFET turns off for a minimum off-time, tOFF-MIN, of 300 ns. Once tOFF-MIN is

complete, the CS comparator compares VSNS and VREF again, waiting to begin the next cycle.

Figure 17. Comparator and One-Shot

The LM3404 and LM3404HV regulators must be operated in continuous conduction mode (CCM), where inductor

current stays positive throughout the switching cycle. During steady-state CCM operation, the converter

maintains a constant switching frequency that can be selected using Equation 2.

• VF= forward voltage of each LED

• n = number of LEDs in series (2)

7.3.2 Average LED Current Accuracy

The COT architecture regulates the valley of ΔVSNS, the AC portion of VSNS. To determine the average LED

current (which is also the average inductor current), the valley inductor current is calculated using Equation 3.

(3)

In Equation 3, tSNS represents the propagation delay of the CS comparator, and is approximately 220 ns. The

average inductor/LED current is equal to IL-MIN plus one-half of the inductor current ripple, ΔiL:

IF= IL= IL-MIN +ΔiL/ 2 (4)

Detailed information for the calculation of ΔiLis given in Application Information.

TSW

300 ns

VO(MIN) = VIN x

TSW

TSW - 300 ns

VO(MAX) = VIN x

TSW = 1/fSW

nMAX =VF(MAX)

VO(max) - 200 mV

DMAX =tON

tON + tOFF-MIN

VO(max) = DMAX x VIN

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Feature Description (continued)

7.3.3 Maximum Output Voltage

The 300-ns minimum off-time limits the maximum duty cycle of the converter, DMAX, and in turn the maximum

output voltage, VO(MAX), is determined by Equation 5.

(5)

The maximum number of LEDs, nMAX, that can be placed in a single series string is governed by VO(MAX) and the

maximum forward voltage of the LEDs used, VF(MAX), using Equation 6.

(6)

At low switching frequency, the maximum duty cycle and output voltage are higher, allowing the LM3404 and

LM3404HV devices to regulate output voltages that are nearly equal to input voltage. Equation 7 relates

switching frequency to maximum output voltage, and is also shown graphically in Typical Characteristics:

(7)

7.3.4 Minimum Output Voltage

The minimum recommended on-time for the LM3404 and LM3404HV devices is 300 ns. This lower limit for tON

determines the minimum duty cycle and output voltage that can be regulated based on input voltage and

switching frequency. The relationship is determined by Equation 8, shown on the same graphs as maximum

output voltage in Typical Characteristics:

(8)

7.3.5 High Voltage Bias Regulator

The LM3404 and LM3404HV devices contain an internal linear regulator with a 7-V output, connected between

the VIN and the VCC pins. The VCC pin must be bypassed to the GND pin with a 0.1-µF ceramic capacitor

connected as close as possible to the pins of the IC. VCC tracks VIN until VIN reaches 8.8 V (typical) and then

regulates at 7 V as VIN increases. Operation begins when VCC crosses 5.25 V.

7.3.6 Internal MOSFET and Driver

The LM3404 and LM3404HV devices feature an internal power MOSFET as well as a floating driver connected

from the SW pin to the BOOT pin. Both rise time and fall time are 20-ns each (typical) and the approximate gate

charge is 6 nC. The high-side rail for the driver circuitry uses a bootstrap circuit consisting of an internal high-

voltage diode and an external 10-nF capacitor, CB. VCC charges CBthrough the internal diode while the power

MOSFET is off. When the MOSFET turns on, the internal diode reverse biases. This creates a floating supply

equal to the VCC voltage minus the diode drop to drive the MOSFET when its source voltage is equal to VIN.

GND

DIM

BOOT SW

RON LM3404/04HV

VIN

RSNS

RON

CIN

VIN

VCC

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Feature Description (continued)

7.3.7 Fast Shutdown for PWM Dimming

The DIM pin of the LM3404 and LM3404HV devices is a TTL compatible input for low-frequency PWM dimming

of the LED. A logic low (below 0.8 V) at DIM will disable the internal MOSFET and shut off the current flow to the

LED array. While the DIM pin is in a logic low state the support circuitry (driver, bandgap, VCC) remains active to

minimize the time needed to turn the LED array back on when the DIM pin sees a logic high (above 2.2 V). A

75-µA (typical) pullup current ensures that the LM3404 and LM3404HV devices are on when DIM pin is open

circuited, eliminating the need for a pullup resistor. Dimming frequency, fDIM, and duty cycle, DDIM, are limited by

the LED current rise time and fall time and the delay from activation of the DIM pin to the response of the internal

power MOSFET. In general, fDIM must be at least one order of magnitude lower than the steady state switching

frequency to prevent aliasing.

7.3.8 Peak Current Limit

The current limit comparator of the LM3404 and LM3404HV devices will engage whenever the power MOSFET

current (equal to the inductor current while the MOSFET is on) exceeds 1.5-A (typical). The power MOSFET is

disabled for a cool-down time that is approximately 75× the steady-state on-time. At the conclusion of this cool-

down time the system restarts. If the current limit condition persists the cycle of cool-down time and restarting will

continue, creating a low-power hiccup mode, minimizing thermal stress on the LM3404 and LM3404HV devices

and the external circuit components.

7.3.9 Overvoltage and Overcurrent Comparator

The CS pin includes an output overvoltage and overcurrent comparator that will disable the power MOSFET

whenever VSNS exceeds 300 mV. This threshold provides a hard limit for the output current. Output current

overshoot is limited to 300 mV / RSNS by this comparator during transients.

The OVP/OCP comparator can also be used to prevent the output voltage from rising to VO(MAX) in the event of

an output open-circuit. This is the most common failure mode for LEDs, due to breaking of the bond wires. In a

current regulator an output open circuit causes VSNS to fall to zero, commanding maximum duty cycle. Figure 18

shows a method using a Zener diode, Z1, and Zener limiting resistor, RZ, to limit output voltage to the reverse

breakdown voltage of Z1 plus 200 mV. The Zener diode reverse breakdown voltage, VZ, must be greater than

the maximum combined VFof all LEDs in the array. The maximum recommended value for RZis 1 kΩ.

As discussed in Maximum Output Voltage, there is a limit to how high VOcan rise during an output open-circuit

that is always less than VIN. If no output capacitor is used, the output stage of the LM3404 and LM3404HV

devices is capable of withstanding VO(MAX) indefinitely; however, the voltage at the output end of the inductor will

oscillate and can go above VIN or below 0 V. A small (typically 10 nF) capacitor across the LED array dampens

this oscillation. For circuits that use an output capacitor, the system can still withstand VO(MAX) indefinitely as long

as COis rated to handle VIN. The high current paths are blocked in output open-circuit and the risk of thermal

stress is minimal, hence the user may opt to allow the output voltage to rise in the case of an open-circuit LED

failure.

Figure 18. Output Open-Circuit Protection

ON/OFF Q1

2N7000 or

equivalent GND

DIM

BOOT SW

RONLM3404/04HV

VIN

RSNS

RON

CIN

VIN

VCC

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7.4 Device Functional Modes

7.4.1 Low-Power Shutdown

The LM3404 and LM3404HV devices can be placed into a low-power state (IIN-SD = 90 µA) by grounding the

RON pin with a signal-level MOSFET as shown in Figure 19. Low-power MOSFETs like the 2N7000, 2N3904, or

equivalent are recommended devices for putting the LM3404 and LM3404HV devices into low-power shutdown.

Logic gates can also be used to shut down the LM3404 LM3404HV devices as long as the logic low voltage is

below the over temperature minimum threshold of 0.3 V. Noise filter circuitry on the RON pin can cause a few

pulses with longer on-times than normal after RON is grounded or released. In these cases, the OVP/OCP

comparator will ensure that the peak inductor or LED current does not exceed 300 mV / RSNS.

Figure 19. Low-Power Shutdown

7.4.2 Thermal Shutdown

Internal thermal shutdown circuitry is provided to protect the IC in the event that the maximum junction

temperature is exceeded. The threshold for thermal shutdown is 165°C with a 25°C hysteresis (both values

typical). During thermal shutdown the MOSFET and driver are disabled.

VIN - VO

'iL = 'iF = tON

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8 Application and Implementation

NOTE

Information in the following applications sections is not part of the TI component

specification, and TI does not warrant its accuracy or completeness. TI’s customers are

responsible for determining suitability of components for their purposes. Customers should

validate and test their design implementation to confirm system functionality.

8.1 Application Information

8.1.1 Switching Frequency

Switching frequency is selected based on the trade-offs between efficiency (better at low frequency), solution

size and cost (smaller at high frequency), and the range of output voltage that can be regulated (wider at lower

frequency). Many applications place limits on switching frequency due to EMI sensitivity. The on-time of the

LM3404 and LM3404HV devices can be programmed for switching frequencies ranging from the 10’s of kHz to

over 1 MHz. The maximum switching frequency is limited only by the minimum on-time and minimum off-time

requirements.

8.1.2 LED Ripple Current

Selection of the ripple current, ΔiF, through the LED array is analogous to the selection of output ripple voltage in

a standard voltage regulator. Where the output ripple in a voltage regulator is commonly ±1% to ±5% of the DC

output voltage, LED manufacturers generally recommend values for ΔiFranging from ±5% to ±20% of IF. Higher

LED ripple current allows the use of smaller inductors, smaller output capacitors, or no output capacitors at all.

The advantages of higher ripple current are reduction in the solution size and cost. Lower ripple current requires

more output inductance, higher switching frequency, or additional output capacitance. The advantages of lower

ripple current are a reduction in heating in the LED itself and greater tolerance in the average LED current before

the current limit of the LED or the driving circuitry is reached.

8.1.3 Buck Converters Without Output Capacitors

The buck converter is unique among non-isolated topologies because of the direct connection of the inductor to

the load during the entire switching cycle. By definition an inductor will control the rate of change of current that

flows through it, and this control overcurrent ripple forms the basis for component selection in both voltage

regulators and current regulators. A current regulator such as the LED driver for which the LM3404 and

LM3404HV devices was designed focuses on the control of the current through the load, not the voltage across

it. A constant current regulator is free of load current transients, and has no need of output capacitance to supply

the load and maintain output voltage. Referring to Typical Application Diagram on the front page of this data

sheet, the inductor and LED can form a single series chain, sharing the same current. When no output capacitor

is used, the same equations that govern inductor ripple current, ΔiL, also apply to the LED ripple current, ΔiF. For

a controlled on-time converter such as the LM3404 and LM3404HV devices, the ripple current is described by

Equation 9.

(9)

A minimum ripple voltage of 25 mV is recommended at the CS pin to provide good signal to noise ratio (SNR).

The CS pin ripple voltage, ΔvSNS, is described by Equation 10.

ΔvSNS =ΔiF× RSNS (10)

'iF =

ZC = ESR +

'iL

1 + ZC1

2Sx fSW x CO

'iL

ESR

RSNS

'iC'iF

'iL

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Application Information (continued)

8.1.4 Buck Converters With Output Capacitors

A capacitor placed in parallel with the LED or array of LEDs can be used to reduce the LED current ripple while

keeping the same average current through both the inductor and the LED array. This technique is demonstrated

in Design Examples 1 and 2. With this topology the output inductance can be lowered, making the magnetics

smaller and less expensive. Alternatively, the circuit could be run at lower frequency but keep the same inductor

value, improving the efficiency and expanding the range of output voltage that can be regulated. Both the peak

current limit and the OVP/OCP comparator still monitor peak inductor current, placing a limit on how large ΔiL

can be even if ΔiFis made very small. A parallel output capacitor is also useful in applications where the inductor

or input voltage tolerance is poor. Adding a capacitor that reduces ΔiFto well below the target provides

headroom for changes in inductance or VIN that might otherwise push the peak LED ripple current too high.

Figure 20 shows the equivalent impedances presented to the inductor current ripple when an output capacitor,

CO, and its equivalent series resistance (ESR) are placed in parallel with the LED array. The entire inductor

ripple current flows through RSNS to provide the required 25 mV of ripple voltage for proper operation of the CS

comparator.

Figure 20. LED and CORipple Current

To calculate the respective ripple currents the LED array is represented as a dynamic resistance, rD. LED

dynamic resistance is not always specified on the manufacturer’s data sheet, but it can be calculated as the

inverse slope of the LED’s VFvs. IFcurve. Dividing VFby IFwill give an incorrect value that is 5× to 10× too high.

Total dynamic resistance for a string of n LEDs connected in series can be calculated as the rDof one device

multiplied by n. Inductor ripple current is still calculated with the expression from Buck Converters Without Output

Capacitors.Equation 11 can then be used to estimate ΔiFwhen using a parallel capacitor.

(11)

The calculation for ZCassumes that the shape of the inductor ripple current is approximately sinusoidal.

Small values of COthat do not significantly reduce ΔiFcan also be used to control EMI generated by the

switching action of the LM3404 and LM3404HV devices. EMI reduction becomes more important as the length of

the connections between the LED and the rest of the circuit increase.

IIN(rms) = IF x D(1 - D)

CIN (MIN) ='VIN (MAX)

IF x tON

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Application Information (continued)

8.1.5 Input Capacitors

Input capacitors at the VIN pin of the LM3404 and LM3404HV devices are selected using requirements for

minimum capacitance and rms ripple current. The input capacitors supply pulses of current approximately equal

to IFwhile the power MOSFET is on, and are charged up by the input voltage while the power MOSFET is off.

Switching converters such as the LM3404 and LM3404HV devices have a negative input impedance due to the

decrease in input current as input voltage increases. This inverse proportionality of input current to input voltage

can cause oscillations (sometimes called power supply interaction) if the magnitude of the negative input

impedance is greater the the input filter impedance. Minimum capacitance can be selected by comparing the

input impedance to the converter’s negative resistance; however this requires accurate calculation of the input

voltage source inductance and resistance, quantities that can be difficult to determine.

An alternative method to select the minimum input capacitance, CIN(MIN), is to select the maximum input voltage

ripple which can be tolerated. This value, ΔvIN(MAX), is equal to the change in voltage across CIN during the

converter on-time, when CIN supplies the load current. CIN(MIN) can be selected with Equation 12.

(12)

A good starting point for selection of CIN is to use an input voltage ripple of 5% to 10% of VIN. TI recommends a

minimum input capacitance of 2× the CIN(MIN) value for all LM3404 and LM3404HV circuits. To determine the rms

current rating, Equation 13 can be used.

(13)

Ceramic capacitors are the best choice for the input to the LM3404 and LM3404HV devices due to their high

ripple current rating, low ESR, low cost, and small size compared to other types. When selecting a ceramic

capacitor, special attention must be paid to the operating conditions of the application. Ceramic capacitors can

lose one-half or more of their capacitance at their rated DC voltage bias and also lose capacitance with extremes

in temperature. TI recommends a DC voltage rating equal to twice the expected maximum input voltage. In

addition, the minimum quality dielectric which is suitable for switching power supply inputs is X5R, while X7R or

better is preferred.

8.1.6 Recirculating Diode

The LM3404 and LM3404HV devices are non-synchronous buck regulators that require a recirculating diode D1

(see the Typical Application Diagram) to carrying the inductor current during the MOSFET off-time. The most

efficient choice for D1 is a Schottky diode due to low forward drop and near-zero reverse recovery time. D1 must

be rated to handle the maximum input voltage plus any switching node ringing when the MOSFET is on. In

practice all switching converters have some ringing at the switching node due to the diode parasitic capacitance

and the lead inductance. D1 must also be rated to handle the average current, ID, calculated as shown in

Equation 14.

ID= (1 – D) x IF(14)

This calculation must be done at the maximum expected input voltage. The overall converter efficiency becomes

more dependent on the selection of D1 at low duty cycles, where the recirculating diode carries the load current

for an increasing percentage of the time. This power dissipation can be calculating by checking the typical diode

forward voltage, VD, from the I-V curve on the product data sheet and then multiplying it by ID. Diode data sheets

will also provide a typical junction-to-ambient thermal resistance, θJA, which can be used to estimate the

operating die temperature of the device. Multiplying the power dissipation (PD= ID× VD) by θJA gives the

temperature rise. The diode case size can then be selected to maintain the Schottky diode temperature below

the operational maximum.

0 1 2 3 4 5 6

SW VOLTAGE (V)

SW CURRENT (PA)

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Application Information (continued)

8.1.7 LED Current During DIM Mode

The LM3404 contains high speed MOSFET gate drive circuitry that switches the main internal power MOSFET

between on and off states. This circuitry uses current derived from the VCC regulator to charge the MOSFET

during turn-on, then dumps current from the MOSFET gate to the source (the SW pin) during turn-off. As shown

in Figure 19, the MOSFET drive circuitry contains a gate drive undervoltage lockout (UVLO) circuit that ensures

the MOSFET remains off when there is inadequate VCC voltage for proper operation of the driver. This watchdog

circuitry is always running including during DIM and shutdown modes, and supplies a small amount of current

from VCC to SW. Because the SW pin is connected directly to the LEDs through the buck inductor, this current

returns to ground through the LEDs. The amount of current sourced is a function of the SW voltage, as shown in

Figure 21.

Figure 21. LED Current From SW Pin

Though most power LEDs are designed to run at several hundred milliamps, some can be seen to glow with a

faint light at extremely low current levels, as low as a couple microamps in some instances. In lab testing, the

forward voltage was found to be approximately 2 V for LEDs that exhibited visible light at these low current

levels. For LEDs that did not show light emission at very low current levels, the forward voltage was found to be

around 900 mV. It is important to remember that the forward voltage is also temperature dependent, decreasing

at higher temperatures. Consequently, with a maximum Vcc voltage of 7.4 V, current will be observed in the

LEDs if the total stack voltage is less than about 6 V at a forward current of several microamps. No current is

observed if the stack voltage is above 6 V, as shown in Figure 21. The need for absolute darkness during DIM

mode is also application dependent. It will not affect regular PWM dimming operation.

The fix for this issue is extremely simple. Place a resistor from the SW pin to ground according to Table 1.

Table 1. Resistor Value for Number of LEDs

NUMBER OF LEDs RESISTOR VALUE (kΩ)

1 20

2 50

3 90

4 150

5 200

>5 300

The luminaire designer must ensure that the suggested resistor is effective in eliminating the off-state light

output. A combination of calculations based on LED manufacturer data and lab measurements over temperature

will ensure the best design.

LM3404

GND

VTRANSIENT

~ 0.675

Module

Connector

Module

Connector

1 k5

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8.1.8 Transient Protection Considerations

Considerations must be made when external sources, loads or connections are made to the switching converter

circuit due to the possibility of electrostatic discharge (ESD) or electric over stress (EOS) events occurring and

damaging the integrated circuit (IC) device. All IC device pins contain Zener based clamping structures that are

meant to clamp ESD. ESD events are very low energy events, typically less than 5 µJ (microjoules). Any event

that transfers more energy than this may damage the ESD structure. Damage is typically represented as a short

from the pin to ground as the extreme localized heat of the ESD or EOS event causes the aluminum metal on

the chip to melt, causing the short. This situation is common to all integrated circuits and not just unique to the

LM3404x device.

8.1.8.1 CS Pin Protection

When hot swapping in a load (that is, test points, load boards, LED stack), any residual charge on the load will

be immediately transferred through the output capacitor to the CS pin, which is then damaged as shown in

Figure 22. The EOS event due to the residual charge from the load is represented as VTRANSIENT.

Figure 22. CS Pin, Transient Path With Protection

From measurements, we know that the 8-V ESD structure on the CS pin can typically withstand 25 mA of direct

current (DC). Adding a 1-kΩresistor in series with the CS pin, shown in Figure 22, results in the majority of the

transient energy to pass through the discrete sense resistor rather than the device. The series resistor limits the

peak current that can flow during a transient event, thus protecting the CS pin. With the 1-kΩresistor shown, a

33-V, 49-A transient on the LED return connector terminal could be absorbed as calculated by:

V = 25 mA × 1 kΩ+ 8 V = 33 V (15)

I = 33 V / 0.67 Ω= 49 A (16)

This is an extremely high-energy event, so the protection measures previously described should be adequate to

solve this issue.

LM3404

GND

VTRANSIENT

~ 0.675

Module

Connector

Module

Connector

1 k5

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Adding a resistor in series with the CS pin causes the observed output LED current to shift very slightly. The

reason for this is twofold: (1) the CS pin has about 20 pF of inherent capacitance inside it, which causes a slight

delay (20 ns for a 1-kΩseries resistor), and (2) the comparator that is watching the voltage at the CS pin uses a

pnp bipolar transistor at its input. The base current of this pnp transistor is approximately 100 nA which will cause

a 0.1-mV change in the 200-mV threshold. These are both very minor changes and are well understood. The

shift in current can either be neglected or taken into consideration by changing the current sense resistance

slightly.

8.1.8.2 CS Pin Protection With OVP

When designing output overvoltage protection into the switching converter circuit using a Zener diode, transient

protection on the CS pin requires additional consideration. As shown in Figure 23, adding a Zener diode from the

output to the CS pin (with the series resistor) for output overvoltage protection will now again allow the transient

energy to be passed

Adding an additional series resistor to the CS pin as shown in Figure 24 will result in the majority of the transient

energy to pass through the sense resistor thereby protecting the LM3404x device.

Figure 23. CS Pin With OVP, Transient Path

LM3404

GND

VTRANSIENT

~ 0.675

Module

Connector

Module

Connector

50051 k5

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Figure 24. CS Pin With OVP, Transient Path With Protection

8.1.8.3 VIN Pin Protection

The VIN pin also has an ESD structure from the pin to GND with a breakdown voltage of approximately 80 V.

Any transient that exceeds this voltage may damage the device. Although transient absorption is usually present

at the front end of a switching converter circuit, damage to the VIN pin can still occur.

When VIN is hot swapped in, the current that rushes in to charge CIN up to the VIN value also charges (energizes)

the circuit board trace inductance as shown in Figure 25. The excited trace inductance then resonates with the

input capacitance (similar to an under-damped LC tank circuit) and causes voltages at the VIN pin to rise well in

excess of both VIN and the voltage at the module input connector as clamped by the input TVS. If the resonating

voltage at the VIN pin exceeds the 80-V breakdown voltage of the ESD structure, the ESD structure will activate

and then snap-back to a lower voltage due to its inherent design. If this lower snap-back voltage is less than the

applied nominal VIN voltage, then significant current will flow through the ESD structure resulting in the IC being

damaged.

An additional TVS or small Zener diode must be placed as close as possible to the VIN pins of each IC on the

board, in parallel with the input capacitor as shown in Figure 26. A minor amount of series resistance in the input

line would also help, but would lower overall conversion efficiency. For this reason, NTC resistors are often used

as inrush limiters instead.

VIN

LM3404

GND

80V

VIN

Module

Connector

Module

Connector

TVS CIN

Board Trace

Inductance

TVS or

smaller

zener diode

VIN

LM3404

GND

80V

VIN

Module

Connector

Module

Connector

TVS CIN

Board Trace

Inductance

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Figure 25. VIN Pin With Typical Input Protection

Figure 26. VIN Pin With Additional Input Protection

RON =VO

1.34 x 10-10 x fSW

GND

DIM

BOOT SW

RON LM3404

VIN

RSNS

RON

CIN

VIN = 24V IF = 700 mA

VCC

COLED1

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8.1.8.4 General Comments Regarding Other Pins

Any pin that goes off-board through a connector must have series resistance of at least 1 kΩto 10 kΩin series

with it to protect it from ESD or other transients. These series resistors limit the peak current that can flow (or

cause a voltage drop) during a transient event, thus protecting the pin and the device. Pins that are not used

must not be left floating. Instead, the pins must be tied to GND or to an appropriate voltage through resistance.

8.2 Typical Applications

8.2.1 Design Example 1: LM3404

The first example circuit will guide the user through component selection for an architectural accent lighting

application. A regulated DC voltage input of 24 V ±10% will power a 5.4-W warm white LED module that consists

of four LEDs in a 2 × 2 series-parallel configuration. The module will be treated as a two-terminal element and

driven with a forward current of 700 mA ±5%. The typical forward voltage of the LED module in thermal steady

state is 6.9 V, hence the average output voltage will be 7.1 V. The objective of this application is to place the

complete current regulator and LED module in a compact space formerly occupied by a halogen light source.

(The LED will be on a separate metal-core PCB and heatsink.) Switching frequency will be 400 kHz to keep

switching loss low, as the confined space with no air-flow requires a maximum temperature rise of 50°C in each

circuit component. A small solution size is also important, as the regulator must fit on a circular PCB with a 1.5"

diameter. A complete bill of materials can be found in Table 2 at the end of this example.

Figure 27. Schematic for Design Example 1

8.2.1.1 Design Requirements

• Input voltage: 24 V ±10%

• LED forward voltage: 6.9 V

• LED current: 700 mA

• Switching frequency: 400 kHz

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 RON and tON

A moderate switching frequency is needed in this application to balance the requirements of magnetics size and

efficiency. RON is selected from the equation for switching frequency as shown in Equation 17 and Equation 18.

(17)

RON = 7.1 / (1.34 × 10-10 × 4 × 105) = 132.5 kΩ(18)

'iFx rD

'iL - 'iF

ZC =

LMIN =VIN - VO

'iLx tON

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Typical Applications (continued)

The closest 1% tolerance resistor is 133 kΩ. The switching frequency and on-time of the circuit can then be

found using the equations relating RON and tON to fSW, as shown in Equation 19 and Equation 20

fSW = 7.1 / (1.33 × 105× 1.34 × 10-10) = 398 kHz (19)

tON = (1.34 × 10-10 × 1.33 × 105) / 24 = 743 ns (20)

8.2.1.2.2 Output Inductor

Because an output capacitor will be used to filter some of the AC ripple current, the inductor ripple current can be

set higher than the LED ripple current. A value of 40%P-P is typical in many buck converters:

ΔiL= 0.4 × 0.7 = 0.28 A (21)

With the target ripple current determined the inductance can be chosen:

(22)

LMIN = [(24 – 7.1) × 7.43 × 10-7] / (0.28) = 44.8 µH (23)

The closest standard inductor value is 47 µH. The average current rating must be greater than 700 mA to

prevent overheating in the inductor. Separation between the LM3404 drivers and the LED arrays means that heat

from the inductor will not threaten the lifetime of the LEDs, but an overheated inductor could still cause the

LM3404 to enter thermal shutdown.

The inductance of the standard part chosen is ±20%. With this tolerance the typical, minimum, and maximum

inductor current ripples can be calculated:

ΔiL(TYP) = [(24 - 7.1) × 7.43 × 10-7] / 47 × 10-6 = 266 mAP-P (24)

ΔiL(MIN) = [(24 - 7.1) × 7.43 × 10-7] / 56 × 10-6 = 223 mAP-P (25)

ΔiL(MAX) = [(24 - 7.1) × 7.43 × 10-7] / 38 × 10-6 = 330 mAP-P (26)

The peak LED/inductor current is then estimated:

IL(PEAK) = IL+ 0.5 × ΔiL(MAX) (27)

IL(PEAK) = 0.7 + 0.5 × 0.330 = 866 mA (28)

In the case of a short circuit across the LED array, the LM3404 will continue to deliver rated current through the

short but will reduce the output voltage to equal the CS pin voltage of 200 mV. The inductor ripple current and

peak current in this condition would be equal to:

ΔiL(LED-SHORT) = [(24 – 0.2) × 7.43 × 10-7] / 38 × 10-6 = 465 mAP-P (29)

IL(PEAK) = 0.7 + 0.5 × 0.465 = 933 mA (30)

In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit

will engage at a typical peak current of 1.5 A. To prevent inductor saturation during these fault conditions the

inductor’s peak current rating must be above 1.5 A. A 47-µH off-the shelf inductor rated to 1.4 A (peak) and 1.5 A

(average) with a DCR of 0.1 Ωwill be used.

8.2.1.2.3 Using an Output Capacitor

This application does not require high frequency PWM dimming, allowing the use of an output capacitor to

reduce the size and cost of the output inductor. To select the proper output capacitor, the equation from Buck

Converters With Output Capacitors is re-arranged to yield Equation 31.

(31)

The target tolerance for LED ripple current is 100 mAP-P, and a typical value for rDis 1.8 Ωat 700 mA. The

required capacitor impedance to reduce the worst-case inductor ripple current of 333 mAP-P is therefore:

ZC= [0.1 / (0.333 - 0.1] × 1.8 = 0.77Ω(32)

A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 400 kHz:

CO= 1/(2 × π× 0.77 × 4 × 105) = 0.51 µF (33)

RSNS =

VIN - VO

0.2 x L

IF x L + VO x tSNS -x tON

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Typical Applications (continued)

This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series

inductance (ESL) of COis negligible. The closest 10% tolerance capacitor value is 1 µF. The capacitor used must

be rated to 25 V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors with

these specifications in the 0805 case size. A typical value for ESR is 3 mΩ.

8.2.1.2.4 RSNS

A preliminary value for RSNS was determined in selecting ΔiL. This value must be re-evaluated based on the

calculations for ΔiF:

(34)

tSNS = 220 ns, RSNS = 0.33 Ω(35)

Sub-1-Ωresistors are available in both 1% and 5% tolerance. A 1%, 0.33-Ωdevice is the closest value, and a

0.33 W, 1206 size device will handle the power dissipation of 162 mW. With the resistance selected, the average

value of LED current is re-calculated to ensure that current is within the ±5% tolerance requirement. The average

LED current can be found using Equation 36.

IF= 0.2 / 0.33 - (7.1 × 2.2 × 10-7) / 47 × 10-6 + 0.266 / 2 (36)

= 706 mA, 1% above 700 mA (37)

8.2.1.2.5 Input Capacitor

Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 24 V × 2%P-P = 480 mV. The

minimum required capacitance is:

CIN(MIN) = (0.7 × 7.4 × 10-7) / 0.48 = 1.1 µF (38)

To provide additional safety margin the a higher value of 3.3-µF ceramic capacitor rated to 50 V with X7R

dielectric in an 1210 case size will be used. From Application Information, input rms current is:

IIN-RMS = 0.7 × Sqrt(0.28 × 0.72) = 314 mA (39)

Ripple current ratings for 1210 size ceramic capacitors are typically higher than 2 A, more than enough for this

design.

8.2.1.2.6 Recirculating Diode

The input voltage of 24 V ±5% requires Schottky diodes with a reverse voltage rating greater than 30 V. The next

highest standard voltage rating is 40 V. Selecting a 40-V rated diode provides a large safety margin for the

ringing of the switch node and also makes cross-referencing of diodes from different vendors easier.

The next parameters to be determined are the forward current rating and case size. In this example the low duty

cycle (D = 7.1 / 24 = 28%) places a greater thermal stress on D1 than on the internal power MOSFET of the

LM3404. The estimated average diode current is:

ID= 0.706 × 0.72 = 509 mA (40)

A Schottky with a forward current rating of 1 A would be adequate, however reducing the power dissipation is

critical in this example. Higher current diodes have lower forward voltages, hence a 2-A rated diode will be used.

To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in

Application Information. VDfor a case size such as SMB in a 40 V, 2-A Schottky diode at 700 mA is

approximately 0.3 V and the θJA is 75°C/W. Power dissipation and temperature rise can be calculated as:

PD= 0.509 × 0.3 = 153 mW (41)

TRISE = 0.153 × 75 = 11.5°C (42)

8.2.1.2.7 CBand CF

The bootstrap capacitor CBmust always be a 10-nF ceramic capacitor with X7R dielectric. A 25-V rating is

appropriate for all application circuits. The linear regulator filter capacitor CFmust always be a 100-nF ceramic

capacitor, also with X7R dielectric and a 25-V rating.

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Typical Applications (continued)

8.2.1.2.8 Efficiency

To estimate the electrical efficiency of this example the power dissipation in each current carrying element can

be calculated and summed. Electrical efficiency, η, must not be confused with the optical efficacy of the circuit,

which depends upon the LEDs themselves.

Total output power, PO, is calculated as:

PO= IF× VO= 0.706 × 7.1 = 5 W (43)

Conduction loss, PC, in the internal MOSFET:

PC= (IF2× RDSON) × D = (0.7062× 0.8) × 0.28 = 112 mW (44)

Gate charging and VCC loss, PG, in the gate drive and linear regulator:

PG= (IIN-OP + fSW × QG) × VIN PG= (600 × 10-6 + 4 × 105× 6 × 10-9) × 24 = 72 mW (45)

Switching loss, PS, in the internal MOSFET:

PS= 0.5 × VIN × IF× (tR+ tF) × fSW PS= 0.5 × 24 × 0.706 × 40 × 10-9 × 4 × 105= 136 mW (46)

AC rms current loss, PCIN, in the input capacitor:

PCIN = IIN(rms)2× ESR = 0.31720.003 = 0.3 mW (negligible) (47)

DCR loss, PL, in the inductor

PL= IF2× DCR = 0.7062× 0.1 = 50 mW (48)

Recirculating diode loss, PD= 153 mW

Current Sense Resistor Loss, PSNS = 164 mW

Electrical efficiency, η= PO/ (PO+ Sum of all loss terms) = 5 / (5 + 0.687) = 88%

Temperature Rise in the LM3404 IC is calculated as:

TLM3404 = (PC+ PG+ PS) × θJA = (0.112 + 0.072 + 0.136) × 155 = 49.2°C (49)

8.2.1.3 Application Curves

Figure 28. LED Current (green) and SW (blue) Waveforms Figure 29. DIM (blue) and LED Current (green) 250Hz PWM

Waveforms

Table 2. BOM for Design Example 1

ID PART NUMBER TYPE SIZE PARAMETER QTY VENDOR

U1 LM3404 LED Driver SOIC-8 42 V, 1.2 A 1 TI

L1 SLF10145T-470M1R4 Inductor 10 × 10 × 4.5 mm 47 µH, 1.4 A, 120

mΩ1 TDK

RON =VO

1.34 x 10-10 x fSW

GND

BOOT SW

RON LM3404HV

VIN

RSNS

RON

CIN

VIN = 48V ±10% IF = 0.5A

VCC

LED1

LED10

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Typical Applications (continued)

Table 2. BOM for Design Example 1 (continued)

ID PART NUMBER TYPE SIZE PARAMETER QTY VENDOR

D1 CMSH2-40 Schottky Diode SMB 40 V, 2 A 1 Central Semi

Cf VJ0805Y104KXXAT Capacitor 0805 100 nF 10% 1 Vishay

Cb VJ0805Y103KXXAT Capacitor 0805 10 nF 10% 1 Vishay

Cin C3225X7R1H335M Capacitor 1210 3.3 µF, 50V 1 TDK

Co C2012X7R1E105M Capacitor 0805 1 µF, 25V 1 TDK

Rsns ERJ8BQFR33V Resistor 1206 0.33 Ω1% 1 Panasonic

Ron CRCW08051333F Resistor 0805 133 kΩ1% 1 Vishay

8.2.2 Design Example 2: LM3404HV

The second example circuit will guide the user through component selection for an outdoor general lighting

application. A regulated DC voltage input of 48 V ±10% will power ten series-connected LEDs at 500 mA ±10%

with a ripple current of 50 mAP-P or less. The typical forward voltage of the LED module in thermal steady state is

35 V, hence the average output voltage will be 35.2 V. A complete bill of materials can be found in Table 3.

Figure 30. Schematic for Design Example 2

8.2.2.1 Design Requirements

• Input voltage: 48 V ±10%

• LED forward voltage: 35 V

• LED current: 500 mA

• Switching frequency: 225 kHz

8.2.2.2 Detailed Design Procedure

8.2.2.2.1 RON and tON

A low switching frequency, 225 kHz, is needed in this application, as high-efficiency and low-power dissipation

take precedence over the solution size. RON is selected from the equation for switching frequency as shown in

Equation 50 and Equation 51.

(50)

RON = 35.2 / (1.34 × 10-10 × 2.25 × 105) = 1.16 MΩ(51)

'iFx rD

'iL - 'iF

ZC =

LMIN =VIN - VO

'iLx tON

LM3404

LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

www.ti.com

Product Folder Links: LM3404 LM3404HV

The next highest 1% tolerance resistor is 1.18 MΩ. The switching frequency and on-time of the circuit can then

be found using the equations relating RON and tON to fSW, as shown in Equation 52 and Equation 53.

fSW = 35.2 / (1.18 × 106× 1.34 × 10-10) = 223 kHz (52)

tON = (1.34 × 10-10 × 1.18 × 106) / 48 = 3.3 µs (53)

8.2.2.2.2 Output Inductor

Because an output capacitor will be used to filter some of the AC ripple current, the inductor ripple current can be

set higher than the LED ripple current. A value of 30%P-P makes a good trade-off between the current ripple and

the size of the inductor:

ΔiL= 0.3 × 0.5 = 0.15A (54)

With the target ripple current determined the inductance can be chosen:

(55)

LMIN = [(48 – 35.2) × 3.3 × 10-6] / (0.15) = 281 µH (56)

The closest standard inductor value above 281 is 330 µH. The average current rating must be greater than 0.5 A

to prevent overheating in the inductor. In this example the LM3404HV driver and the LED array share the same

metal-core PCB, meaning that heat from the inductor could threaten the lifetime of the LEDs. For this reason the

average current rating of the inductor used must have a derating of about 50%, or 1 A.

The inductance of the standard part chosen is ±20%. With this tolerance the typical, minimum, and maximum

inductor current ripples can be calculated:

ΔiL(TYP) = [(48 - 35.2) × 3.3 × 10-6] / 330 × 10-6 = 128 mAP-P (57)

ΔiL(MIN) = [(48 - 35.2) × 3.3 × 10-6] / 396 × 10-6 = 107 mAP-P (58)

ΔiL(MAX) = [(48 - 35.2) × 3.3 × 10-6] / 264 × 10-6 = 160 mAP-P (59)

The peak inductor current is then estimated:

IL(PEAK) = IL+ 0.5 × ΔiL(MAX) (60)

IL(PEAK) = 0.5 + 0.5 × 0.16 = 0.58A (61)

In the case of a short circuit across the LED array, the LM3404HV will continue to deliver rated current through

the short but will reduce the output voltage to equal the CS pin voltage of 200 mV. The inductor ripple current

and peak current in this condition would be equal to:

ΔiL(LED-SHORT) = [(48 – 0.2) × 3.3 × 10-6] / 264 × 10-6 = 0.598AP-P (62)

IL(PEAK) = 0.5 + 0.5 × 0.598 = 0.8 A (63)

In the case of a short at the switch node, the output, or from the CS pin to ground the short circuit current limit

will engage at a typical peak current of 1.5 A. To prevent inductor saturation during these fault conditions the

inductor’s peak current rating must be above 1.5 A. A 330-µH off-the shelf inductor rated to 1.9 A (peak) and 1 A

(average) with a DCR of 0.56 Ωwill be used.

8.2.2.2.3 Using an Output Capacitor

This application uses sub-1 kHz frequency PWM dimming, allowing the use of a small output capacitor to reduce

the size and cost of the output inductor. To select the proper output capacitor, the equation from buck regulators

with output capacitors is re-arranged to yield Equation 64.

(64)

The target tolerance for LED ripple current is 50 mAP-P, and the typical value for rDis 10 Ωwith ten LEDs in

series. The required capacitor impedance to reduce the worst-case steady-state inductor ripple current of 160

mAP-P is therefore:

ZC= [0.05 / (0.16 - 0.05] × 10 = 4.5 Ω(65)

RSNS =

VIN - VO

0.2 x L

IF x L + VO x tSNS -x tON

LM3404

LM3404HV

www.ti.com

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

Product Folder Links: LM3404 LM3404HV

A ceramic capacitor will be used and the required capacitance is selected based on the impedance at 223 kHz:

CO= 1 / (2 × π× 4.5 × 2.23 × 105) = 0.16 µF (66)

This calculation assumes that impedance due to the equivalent series resistance (ESR) and equivalent series

inductance (ESL) of COis negligible. The closest 10% tolerance capacitor value is 0.15 µF. The capacitor used

must be rated to 50 V or more and have an X7R dielectric. Several manufacturers produce ceramic capacitors

with these specifications in the 0805 case size. ESR values are not typically provided for such low value

capacitors, however is can be assumed to be under 100 mΩ, leaving plenty of margin to meet to LED ripple

current requirement. The low capacitance required allows the use of a 100-V rated, 1206-size capacitor. The

rating of 100 V ensures that the capacitance will not decrease significantly when the DC output voltage is applied

across the capacitor.

8.2.2.2.4 RSNS

A preliminary value for RSNS was determined in selecting ΔiL. This value must be re-evaluated based on the

calculations for ΔiF:

(67)

tSNS = 220 ns, RSNS = 0.43 Ω(68)

Sub-1-Ωresistors are available in both 1% and 5% tolerance. A 1%, 0.43-Ωdevice is the closest value, and a

0.25 W, 0805 size device will handle the power dissipation of 110 mW. With the resistance selected, the average

value of LED current is re-calculated to ensure that current is within the ±10% tolerance requirement. The

average LED current can be found using Equation 69.

IF= 0.2 / 0.33 – (7.1 × 2.2 × 10-7) / 47 × 10-6 + 0.266 / 2 = 505 mA (69)

8.2.2.2.5 Input Capacitor

Following the calculations from the Input Capacitor section, ΔvIN(MAX) will be 48 V × 2%P-P = 960 mV. The

minimum required capacitance is:

CIN(MIN) = (0.5 × 3.3 × 10-6) / 0.96 = 1.7 µF (70)

To provide additional safety margin a 2.2-µF ceramic capacitor rated to 100 V with X7R dielectric in an 1812

case size will be used. From Application Information, input rms current is:

IIN-RMS = 0.5 × Sqrt(0.73 × 0.27) = 222 mA (71)

Ripple current ratings for 1812 size ceramic capacitors are typically higher than 2 A, more than enough for this

design, and the ESR is approximately 3 mΩ.

8.2.2.2.6 Recirculating Diode

The input voltage of 48 V requires Schottky diodes with a reverse voltage rating greater than 50 V. The next

highest standard voltage rating is 60 V. Selecting a 60-V rated diode provides a large safety margin for the

ringing of the switch node and also makes cross-referencing of diodes from different vendors easier.

The next parameters to be determined are the forward current rating and case size. In this example the high duty

cycle (D = 35.2 / 48 = 73%) places a greater thermal stress on the internal power MOSFET than on D1. The

estimated average diode current is:

ID= 0.5 × 0.27 = 135 mA (72)

A Schottky with a forward current rating of 0.5 A would be adequate; however, reducing the power dissipation is

critical in this example. Higher current diodes have lower forward voltages, hence a 1-A rated diode will be used.

To determine the proper case size, the dissipation and temperature rise in D1 can be calculated as shown in

Application Information. VDfor a case size such as SMA in a 60-V, 1-A Schottky diode at 0.5 A is approximately

0.35 V and the θJA is 75°C/W. Power dissipation and temperature rise can be calculated as:

PD= 0.135 × 0.35 = 47 mW TRISE = 0.047 × 75 = 3.5°C (73)

LM3404

LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

www.ti.com

Product Folder Links: LM3404 LM3404HV

8.2.2.2.7 CBand CF

The bootstrap capacitor CBmust always be a 10-nF ceramic capacitor with X7R dielectric. A 25-V rating is

appropriate for all application circuits. The linear regulator filter capacitor CFmust always be a 100-nF ceramic

capacitor, also with X7R dielectric and a 25-V rating.

8.2.2.2.8 Efficiency

To estimate the electrical efficiency of this example the power dissipation in each current carrying element can

be calculated and summed. Electrical efficiency, η, must not be confused with the optical efficacy of the circuit,

which depends upon the LEDs themselves.

Total output power, PO, is calculated as:

PO= IF× VO= 0.5 × 35.2 = 17.6 W (74)

Conduction loss, PC, in the internal MOSFET:

PC= (IF2× RDSON) × D = (0.52× 0.8) × 0.73 = 146 mW (75)

Gate charging and VCC loss, PG, in the gate drive and linear regulator:

PG= (IIN-OP + fSW × QG) × VIN PG= (600 × 10-6 + 2.23 × 105× 6 × 10-9) × 48 = 94 mW (76)

Switching loss, PS, in the internal MOSFET:

PS= 0.5 × VIN × IF× (tR+ tF) × fSW PS= 0.5 × 48 × 0.5 × 40 × 10-9 × 2.23 × 105= 107 mW (77)

AC rms current loss, PCIN, in the input capacitor:

PCIN = IIN(rms)2× ESR = 0.22220.003 = 0.1 mW (negligible) (78)

DCR loss, PL, in the inductor

PL= IF2× DCR = 0.52× 0.56 = 140 mW (79)

Recirculating diode loss, PD= 47 mW

Current Sense Resistor Loss, PSNS = 110 mW

Electrical efficiency, η= PO/ (PO+ Sum of all loss terms) = 17.6 / (17.6 + 0.644) = 96%

Temperature Rise in the LM3404HV IC is calculated as:

TLM3404 = (PC+ PG+ PS) × θJA = (0.146 + 0.094 + 0.107) × 155 = 54°C (80)

8.2.2.3 Application Curves

Figure 31. LED Current (Green) and SW (Blue) Waveforms Figure 32. DIM (Blue) and LED Current (Green) 250Hz PWM

Waveforms

LM3404

LM3404HV

www.ti.com

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

Product Folder Links: LM3404 LM3404HV

Table 3. BOM for Design Example 2

ID PART NUMBER TYPE SIZE PARAMETERS QTY VENDOR

U1 LM3404HV LED Driver SOIC-8 75 V, 1.2 A 1 TI

L1 DO5022P-334 Inductor 18.5 × 15.4 × 7.1 mm 330 µH, 1.9 A, 0.56 Ω1 Coilcraft

D1 CMSH1-60M Schottky Diode SMA 60 V, 1 A 1 Central Semi

Cf VJ0805Y104KXXAT Capacitor 0805 100 nF 10% 1 Vishay

Cb VJ0805Y103KXXAT Capacitor 0805 10 nF 10% 1 Vishay

Cin C4532X7R2A225M Capacitor 1812 2.2 µF, 100 V 1 TDK

Co C3216X7R2A154M Capacitor 1206 0.15 µF, 100 V 1 TDK

Rsns ERJ6BQFR43V Resistor 0805 0.43 Ω1% 1 Panasonic

Ron CRCW08051184F Resistor 0805 1.18 MΩ1% 1 Vishay

LM3404

LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

www.ti.com

Product Folder Links: LM3404 LM3404HV

9 Power Supply Recommendations

Use any DC output power supply with a maximum voltage high enough for the application. The power supply

must have a minimum current limit of at least 1 A.

10 Layout

10.1 Layout Guidelines

The performance of any switching converter depends as much upon the layout of the PCB as the component

selection. The following guidelines will help the user design a circuit with maximum rejection of outside EMI and

minimum generation of unwanted EMI.

10.1.1 Compact Layout

Parasitic inductance can be reduced by keeping the power path components close together and keeping the

area of the loops that high currents travel small. Short, thick traces or copper pours (shapes) are best. In

particular, the switch node (where L1, D1, and the SW pin connect) must be just large enough to connect all

three components without excessive heating from the current it carries. The LM3404 and LM3404HV devices

operate in two distinct cycles whose high current paths are shown in Figure 33:

Figure 33. Buck Converter Current Loops

The dark grey, inner loop represents the high current path during the MOSFET on-time. The light grey, outer loop

represents the high current path during the off-time.

10.1.2 Ground Plane and Shape Routing

The diagram of Figure 33 is also useful for analyzing the flow of continuous current versus the flow of pulsating

currents. The circuit paths with current flow during both the on-time and off-time are considered to be continuous

current, while those that carry current during the on-time or off-time only are pulsating currents. Preference in

routing must be given to the pulsating current paths, as these are the portions of the circuit most likely to emit

EMI. The ground plane of a PCB is a conductor and return path, and it is susceptible to noise injection just as

any other circuit path. The continuous current paths on the ground net can be routed on the system ground plane

with less risk of injecting noise into other circuits. The path between the input source and the input capacitor and

the path between the recirculating diode and the LEDs and current sense resistor are examples of continuous

current paths. In contrast, the path between the recirculating diode and the input capacitor carries a large

pulsating current. This path must be routed with a short, thick shape, preferably on the component side of the

PCB. Multiple vias in parallel must be used right at the pad of the input capacitor to connect the component side

shapes to the ground plane. A second pulsating current loop that is often ignored is the gate drive loop formed

by the SW and BOOT pins and capacitor CB. To minimize this loop at the EMI it generates, keep CBclose to the

SW and BOOT pins.

BOOT

DIM

GND

VIN

VCC

RON

THERMAL/POWER VIA

GND

VIN

L1 D1

RON

RSNS

CIN

LED+

LED-

LM3404

LM3404HV

www.ti.com

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

Product Folder Links: LM3404 LM3404HV

Layout Guidelines (continued)

10.1.3 Current Sensing

The CS pin is a high-impedance input, and the loop created by RSNS, RZ(if used), the CS pin and ground must

be made as small as possible to maximize noise rejection. Therefore, RSNS must be placed as close as possible

to the CS and GND pins of the IC.

10.1.4 Remote LED Arrays

In some applications, the LED or LED array can be far away (several inches or more) from the LM3404 and

LM3404HV devices, or on a separate PCB connected by a wiring harness. When an output capacitor is used and

the LED array is large or separated from the rest of the converter, the output capacitor must be placed close to

the LEDs to reduce the effects of parasitic inductance on the AC impedance of the capacitor. The current sense

resistor must remain on the same PCB, close to the LM3404 and LM3404HV devices.

10.2 Layout Example

Figure 34. Layout Example

LM3404

LM3404HV

SNVS465G –OCTOBER 2006–REVISED SEPTEMBER 2015

www.ti.com

Product Folder Links: LM3404 LM3404HV

11 Device and Documentation Support

11.1 Device Support

11.1.1 Third-Party Products Disclaimer

TI'S PUBLICATION OF INFORMATION REGARDING THIRD-PARTY PRODUCTS OR SERVICES DOES NOT

CONSTITUTE AN ENDORSEMENT REGARDING THE SUITABILITY OF SUCH PRODUCTS OR SERVICES

OR A WARRANTY, REPRESENTATION OR ENDORSEMENT OF SUCH PRODUCTS OR SERVICES, EITHER

ALONE OR IN COMBINATION WITH ANY TI PRODUCT OR SERVICE.

11.2 Related Links

The table below lists quick access links. Categories include technical documents, support and community

resources, tools and software, and quick access to sample or buy.

Table 4. Related Links

PARTS PRODUCT FOLDER SAMPLE & BUY TECHNICAL

DOCUMENTS TOOLS &

SOFTWARE SUPPORT &

COMMUNITY

LM3404 Click here Click here Click here Click here Click here

LM3404HV Click here Click here Click here Click here Click here

11.3 Community Resources

The following links connect to TI community resources. Linked contents are provided "AS IS" by the respective

contributors. They do not constitute TI specifications and do not necessarily reflect TI's views; see TI's Terms of

Use.

TI E2E™ Online Community TI's Engineer-to-Engineer (E2E) Community. Created to foster collaboration

among engineers. At e2e.ti.com, you can ask questions, share knowledge, explore ideas and help

solve problems with fellow engineers.

Design Support TI's Design Support Quickly find helpful E2E forums along with design support tools and

contact information for technical support.

11.4 Trademarks

PowerPAD, E2E are trademarks of Texas Instruments.

All other trademarks are the property of their respective owners.

11.5 Electrostatic Discharge Caution

These devices have limited built-in ESD protection. The leads should be shorted together or the device placed in conductive foam

during storage or handling to prevent electrostatic damage to the MOS gates.

11.6 Glossary

SLYZ022 —TI Glossary.

This glossary lists and explains terms, acronyms, and definitions.

12 Mechanical, Packaging, and Orderable Information

The following pages include mechanical, packaging, and orderable information. This information is the most

current data available for the designated devices. This data is subject to change without notice and revision of

this document. For browser-based versions of this data sheet, refer to the left-hand navigation.

PACKAGE OPTION ADDENDUM

www.ti.com 25-Jun-2015

Addendum-Page 1

PACKAGING INFORMATION

Orderable Device Status

(1)

Package Type Package

Drawing Pins Package

Qty Eco Plan

(2)

Lead/Ball Finish

(6)

MSL Peak Temp

(3)

Op Temp (°C) Device Marking

(4/5)

Samples

LM3404HVMA/NOPB ACTIVE SOIC D 8 95 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L3404

HVMA

LM3404HVMAX/NOPB ACTIVE SOIC D 8 2500 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L3404

HVMA

LM3404HVMR/NOPB ACTIVE SO PowerPAD DDA 8 95 Green (RoHS

& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L3404

HVMR

LM3404HVMRX/NOPB ACTIVE SO PowerPAD DDA 8 2500 Green (RoHS

& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L3404

HVMR

LM3404MA/NOPB ACTIVE SOIC D 8 95 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L3404

LM3404MAX/NOPB ACTIVE SOIC D 8 2500 Green (RoHS

& no Sb/Br) CU SN Level-1-260C-UNLIM -40 to 125 L3404

LM3404MR/NOPB ACTIVE SO PowerPAD DDA 8 95 Green (RoHS

& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L3404

LM3404MRX/NOPB ACTIVE SO PowerPAD DDA 8 2500 Green (RoHS

& no Sb/Br) CU SN Level-3-260C-168 HR -40 to 125 L3404

(1) The marketing status values are defined as follows:

ACTIVE: Product device recommended for new designs.

LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.

NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in a new design.

PREVIEW: Device has been announced but is not in production. Samples may or may not be available.

OBSOLETE: TI has discontinued the production of the device.

(2) Eco Plan - The planned eco-friendly classification: Pb-Free (RoHS), Pb-Free (RoHS Exempt), or Green (RoHS & no Sb/Br) - please check http://www.ti.com/productcontent for the latest availability

information and additional product content details.

TBD: The Pb-Free/Green conversion plan has not been defined.

Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements for all 6 substances, including the requirement that

lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.

Pb-Free (RoHS Exempt): This component has a RoHS exemption for either 1) lead-based flip-chip solder bumps used between the die and package, or 2) lead-based die adhesive used between

the die and leadframe. The component is otherwise considered Pb-Free (RoHS compatible) as defined above.

Green (RoHS & no Sb/Br): TI defines "Green" to mean Pb-Free (RoHS compatible), and free of Bromine (Br) and Antimony (Sb) based flame retardants (Br or Sb do not exceed 0.1% by weight

in homogeneous material)

(3) MSL, Peak Temp. - The Moisture Sensitivity Level rating according to the JEDEC industry standard classifications, and peak solder temperature.

PACKAGE OPTION ADDENDUM

www.ti.com 25-Jun-2015

Addendum-Page 2

(4) There may be additional marking, which relates to the logo, the lot trace code information, or the environmental category on the device.

(5) Multiple Device Markings will be inside parentheses. Only one Device Marking contained in parentheses and separated by a "~" will appear on a device. If a line is indented then it is a continuation

of the previous line and the two combined represent the entire Device Marking for that device.

(6) Lead/Ball Finish - Orderable Devices may have multiple material finish options. Finish options are separated by a vertical ruled line. Lead/Ball Finish values may wrap to two lines if the finish

value exceeds the maximum column width.

Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is provided. TI bases its knowledge and belief on information

provided by third parties, and makes no representation or warranty as to the accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and

continues to take reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on incoming materials and chemicals.

TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited information may not be available for release.

In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI to Customer on an annual basis.

TAPE AND REEL INFORMATION

*All dimensions are nominal

Device Package

Type Package

Drawing Pins SPQ Reel

Diameter

(mm)

Reel

Width

W1 (mm)

(mm) B0

(mm) K0

(mm) P1

(mm) W

(mm) Pin1

Quadrant

LM3404HVMAX/NOPB SOIC D 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1

LM3404HVMRX/NOPB SO

Power

PAD

DDA 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1

LM3404MAX/NOPB SOIC D 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1

LM3404MRX/NOPB SO

Power

PAD

DDA 8 2500 330.0 12.4 6.5 5.4 2.0 8.0 12.0 Q1

PACKAGE MATERIALS INFORMATION

www.ti.com 25-Jun-2015

Pack Materials-Page 1

*All dimensions are nominal

Device Package Type Package Drawing Pins SPQ Length (mm) Width (mm) Height (mm)

LM3404HVMAX/NOPB SOIC D 8 2500 367.0 367.0 35.0

LM3404HVMRX/NOPB SO PowerPAD DDA 8 2500 367.0 367.0 35.0

LM3404MAX/NOPB SOIC D 8 2500 367.0 367.0 35.0

LM3404MRX/NOPB SO PowerPAD DDA 8 2500 367.0 367.0 35.0

PACKAGE MATERIALS INFORMATION

www.ti.com 25-Jun-2015

Pack Materials-Page 2

www.ti.com

PACKAGE OUTLINE

6X 1.27

8X 0.51

0.31

3.81

TYP

0.25

0.10

0 - 8 0.15

0.00

2.71

2.11

3.4

2.8 0.25

GAGE PLANE

1.27

0.40

4214849/A 08/2016

PowerPAD SOIC - 1.7 mm max heightDDA0008B

PLASTIC SMALL OUTLINE

NOTES:

1. All linear dimensions are in millimeters. Any dimensions in parenthesis are for reference only. Dimensioning and tolerancing

per ASME Y14.5M.

2. This drawing is subject to change without notice.

3. This dimension does not include mold flash, protrusions, or gate burrs. Mold flash, protrusions, or gate burrs shall not

exceed 0.15 mm per side.

4. This dimension does not include interlead flash. Interlead flash shall not exceed 0.25 mm per side.

5. Reference JEDEC registration MS-012.

PowerPAD is a trademark of Texas Instruments.

0.25 C A B

PIN 1 ID

AREA

NOTE 4

SEATING PLANE

0.1 C

SEE DETAIL A

DETAIL A

TYPICAL

SCALE 2.400

EXPOSED

THERMAL PAD

TYP

6.2

5.8

1.7 MAX

NOTE 3

5.0

4.8

B4.0

3.8

www.ti.com

EXAMPLE BOARD LAYOUT

(5.4)

(1.3) TYP

( ) TYP

VIA

0.2

(R ) TYP0.05

0.07 MAX

ALL AROUND 0.07 MIN

ALL AROUND

8X (1.55)

8X (0.6)

6X (1.27)

(2.95)

NOTE 9

(4.9)

NOTE 9

(2.71)

(3.4)

SOLDER MASK

OPENING

(1.3)

TYP

4214849/A 08/2016

PowerPAD SOIC - 1.7 mm max heightDDA0008B

PLASTIC SMALL OUTLINE

SYMM

SEE DETAILS

LAND PATTERN EXAMPLE

SCALE:10X

SOLDER MASK

OPENING

METAL COVERED

BY SOLDER MASK

SOLDER MASK

DEFINED PAD

NOTES: (continued)

6. Publication IPC-7351 may have alternate designs.

7. Solder mask tolerances between and around signal pads can vary based on board fabrication site.

8. This package is designed to be soldered to a thermal pad on the board. For more information, see Texas Instruments literature

numbers SLMA002 (www.ti.com/lit/slma002) and SLMA004 (www.ti.com/lit/slma004).

9. Size of metal pad may vary due to creepage requirement.

10. Vias are optional depending on application, refer to device data sheet. If any vias are implemented, refer to their locations shown

on this view. It is recommended that vias under paste be filled, plugged or tented.

METAL

SOLDER MASK

OPENING

NON SOLDER MASK

DEFINED

SOLDER MASK DETAILS

PADS 1-8

OPENING

SOLDER MASK METAL UNDER

SOLDER MASK

DEFINED

www.ti.com

EXAMPLE STENCIL DESIGN

(R ) TYP0.05

8X (1.55)

8X (0.6)

6X (1.27)

(5.4)

(2.71)

(3.4)

BASED ON

0.125 THICK

STENCIL

4214849/A 08/2016

PowerPAD SOIC - 1.7 mm max heightDDA0008B

PLASTIC SMALL OUTLINE

2.29 X 2.870.175 2.47 X 3.100.150 2.71 X 3.40 (SHOWN)0.125 3.03 X 3.800.1

SOLDER STENCIL

OPENING

STENCIL

THICKNESS

NOTES: (continued)

11. Laser cutting apertures with trapezoidal walls and rounded corners may offer better paste release. IPC-7525 may have alternate

design recommendations.

12. Board assembly site may have different recommendations for stencil design.

SOLDER PASTE EXAMPLE

EXPOSED PAD

100% PRINTED SOLDER COVERAGE BY AREA

SCALE:10X

SYMM

BASED ON

0.125 THICK

STENCIL

BY SOLDER MASK

METAL COVERED SEE TABLE FOR

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