12-Bit, 25MHz Sampling
ANALOG-TO-DIGITAL CONVERTER
FEATURES
●NO MISSING CODES
●LOW POWER: 270mW
●INTERNAL REFERENCE
●WIDEBAND TRACK-AND-HOLD: 65MHz
●SINGLE +5V SUPPLY
DESCRIPTION
The ADS801 is a low-power, monolithic 12-bit, 25MHz Ana-
log-to-Digital (A/D) converter utilizing a small geometry CMOS
process. This complete converter includes a 12-bit quantizer,
wideband track-and-hold, reference, and three-state outputs.
It operates from a single +5V power supply and can be
configured to accept either single-ended or differential input
signals.
The ADS801 employs digital error correction to provide
excellent Nyquist differential linearity performance for de-
manding imaging applications. Its low distortion, high SNR,
and high oversampling capability give it the extra margin
needed for telecommunications, instrumentation, and video
applications.
This high-performance A/D converter is specified over tem-
perature for AC and DC performance at a 25MHz sampling
rate. The ADS820 is available in an SO-28 package.
APPLICATIONS
●IF AND BASEBAND DIGITIZATION
●DIGITAL COMMUNICATIONS
●TEST INSTRUMENTATION
●CCD IMAGING
Copiers
Scanners
Cameras
●VIDEO DIGITIZING
●GAMMA CAMERAS
Pipeline
A/D
Converter
Timing
Circuitry
Error
Correction
Logic
3-State
Outputs
T/H 12-Bit
Digital
Data
CLK
+1.25V
+3.25V
MSBI OE
IN
IN
REFT
CM
REFB
ADS801
SBAS036B – MAY 1995 – REVISED FEBRUARY 2005
www.ti.com
Copyright © 1995-2005, Texas Instruments Incorporated
Please be aware that an important notice concerning availability, standard warranty, and use in critical applications of
Texas Instruments semiconductor products and disclaimers thereto appears at the end of this data sheet.
PRODUCTION DATA information is current as of publication date.
Products conform to specifications per the terms of Texas Instruments
standard warranty. Production processing does not necessarily include
testing of all parameters.
ADS801U
All trademarks are the property of their respective owners.
ADS801
2SBAS036B
www.ti.com
ELECTRICAL CHARACTERISTICS
At TA = +25°C, VS = +5V, Sampling Rate = 25MHz, and with a 50% duty cycle clock having a 2ns rise-and-fall time, unless otherwise noted.
ABSOLUTE MAXIMUM RATINGS(1)
+VS.......................................................................................................+6V
Analog Input ............................................................0V to (+VS + 300mV)
Logic Input ............................................................... 0V to (+VS + 300mV)
Case Temperature......................................................................... +100°C
Junction Temperature.................................................................... +150°C
Storage Temperature .................................................................... +125°C
External Top Reference Voltage (REFT).................................+3.4V Max
External Bottom Reference Voltage (REFB) ............................ +1.1V Min
ELECTROSTATIC
DISCHARGE SENSITIVITY
This integrated circuit can be damaged by ESD. Texas
Instruments recommends that all integrated circuits be handled
with appropriate precautions. Failure to observe proper han-
dling and installation procedures can cause damage.
ESD damage can range from subtle performance degrada-
tion to complete device failure. Precision integrated circuits
may be more susceptible to damage because very small
parametric changes could cause the device not to meet its
published specifications.
NOTE: (1) Stresses above these ratings may permanently damage the device.
SPECIFIED
PACKAGE TEMPERATURE PACKAGE ORDERING TRANSPORT
PRODUCT PACKAGE-LEAD DESIGNATOR RANGE MARKING NUMBER MEDIA, QUANTITY
ADS801U SO-28 DW –40°C to +85°C ADS801U ADS801U Rails, 28
ADS801U """"ADS801U/1K Tape and Reel, 1000
NOTE: (1) For the most current package and ordering information, see the Package Option Addendum at the end of this document, or see the TI website at
www.ti.com.
PACKAGE/ORDERING INFORMATION(1)
ADS800U
PARAMETER CONDITIONS TEMP MIN TYP MAX UNITS
Resolution 12 Bits
Specified Temperature Range TAMBIENT –40 +85 °C
ANALOG INPUT
Differential Full-Scale Input Range Both Inputs, +1.25 +3.25 V
180° Out-of-Phase
Common-Mode Voltage +2.25 V
Analog Input Bandwidth (–3dB)
Small-Signal –20dBFS(1) Input +25°C 400 MHz
Full-Power 0dBFS Input +25°C65MHz
Input Impedance 1.25 || 4 MΩ || pF
DIGITAL INPUT
Logic Family TTL/HCT Compatible CMOS
Convert Command Start Conversion Falling Edge
ACCURACY(2) fS = 2.5MHz
Gain Error +25°C±0.6 ±1.5 %
Full ±1.0 ±2.5 %
Gain Tempco ±85 ppm/°C
Power-Supply Rejection of Gain ∆ +VS = ±5% +25°C 0.03 0.15 %FSR/%
Input Offset Error Full ±2.1 ±3.0 %
Power-Supply Rejection of Offset ∆ +VS = ±5% +25°C 0.05 0.15 %FSR/%
CONVERSION CHARACTERISTICS
Sample Rate 10k 25M Sample/s
Data Latency 6.5 Convert Cycle
DYNAMIC CHARACTERISTICS
Differential Linearity Error
f = 500kHz +25°C±0.3 ±1.0 LSB
0°C to +85°C±0.4 ±1.0 LSB
f = 10MHz +25°C±0.3 ±1.0 LSB
0°C to +85°C±0.4 ±1.0 LSB
No Missing Codes 0°C to +85°C Tested LSB
Integral Linearity Error at f = 500kHz Full ±1.7 LSB
Spurious-Free Dynamic Range (SFDR)
f = 500kHz (–1dBFS input) +25°C 63 77 dBFS
Full 62 73 dBFS
f = 10MHz (–1dBFS input) +25°C 57 61 dBFS
Full 55 61 dBFS
NOTES: (1) dBFS refers to dB below Full-Scale. (2) Percentage accuracies are referred to the internal A/D converter Full-Scale Range of 4Vp-p. (3) IMD is referred
to the larger of the two input signals. If referred to the peak envelope signal (≈0dB), the intermodulation products will be 7dB lower. (4) No “rollover” of bits.
ADS801 3
SBAS036B www.ti.com
ELECTRICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VS = +5V, Sampling Rate = 25MHz, and with a 50% duty cycle clock having a 2ns rise-and-fall time, unless otherwise noted.
NOTES: (1) dBFS refers to dB below Full-Scale. (2) Percentage accuracies are referred to the internal A/D converter Full-Scale Range of 4Vp-p. (3) IMD is referred
to the larger of the two input signals. If referred to the peak envelope signal (≈0dB), the intermodulation products will be 7dB lower. (4) No “rollover” of bits.
ADS800U
PARAMETER CONDITIONS TEMP MIN TYP MAX UNITS
DYNAMIC CHARACTERISTICS (Cont.)
2-Tone Intermodulation Distortion (IMD)(3)
f = 4.4MHz and 4.5MHz (–7dBFS each tone) +25°C –64 dBc
Full –63 dBc
Signal-to-Noise Ratio (SNR)
f = 500kHz (–1dBFS input) +25°C6466 dB
Full 61 64 dB
f = 10MHz (–1dBFS input) +25°C6265 dB
Full 58 64 dB
Signal-to-(Noise + Distortion) (SINAD)
f = 500kHz (–1dBFS input) +25°C6366 dB
Full 60 63 dB
f = 10MHz (–1dBFS input) +25°C5659 dB
Full 54 58 dB
Differential Gain Error NTSC or PAL +25°C 0.5 %
Differential Phase Error NTSC or PAL +25°C 0.1 degrees
Aperture Delay Time +25°C2ns
Aperture Jitter +25°C 7 ps rms
Over-Voltage Recovery Time(4) 1.5x Full-Scale Input +25°C2ns
OUTPUTS
Logic Family TTL/HCT Compatible CMOS
Logic Coding Logic Selectable Falling Edge
Logic Levels Logic LOW, Full 0 0.4 V
CL = 15pF max
Logic HIGH, Full +2.5 +VSV
CL = 15pF max
3-State Enable Time 20 40 ns
3-State Disable Time Full 2 10 ns
POWER-SUPPLY REQUIREMENTS
Supply Voltage: +VSOperating Full +4.75 +5.0 +5.25 V
Supply Current: +ISOperating +25°C5465mA
Operating Full 54 68 mA
Power Consumption Operating +25°C 270 325 mW
Operating Full 270 340 mW
Thermal Resistance,
θ
JA
SO-28 75 °C/W
ADS801
4SBAS036B
www.ti.com
PIN CONFIGURATION
Top View SO
TIMING DIAGRAM
SYMBOL DESCRIPTION MIN TYP MAX UNITS
tCONV Convert Clock Period 40 100µsns
tLClock Pulse LOW 19 20 ns
tHClock Pulse HIGH 19 20 ns
tDAperture Delay 2 ns
t1Data Hold Time, CL = 0pF 3.9 ns
t2New Data Delay Time, CL = 15pF max 12.5 ns
PIN DESIGNATOR DESCRIPTION
1 GND Ground
2 B1 Bit 1, Most Significant Bit (MSB)
3 B2 Bit 2
4 B3 Bit 3
5 B4 Bit 4
6 B5 Bit 5
7 B6 Bit 6
8 B7 Bit 7
9 B8 Bit 8
10 B9 Bit 9
11 B10 Bit 10
12 B11 Bit 11
13 B12 Bit 12, Least Significant Bit (LSB)
14 GND Ground
15 +VS+5V Power Supply
16 CLK Convert Clock Input, 50% Duty Cycle
17 +VS+5V Power Supply
18 OE HIGH: High-Impedance State. LOW or Floating:
Normal Operation. Internal pull-down resistors.
19 MSBI Most Significant Bit Inversion, HIGH: MSB in-
verted for complementary output. LOW or Float-
ing: Straight output. Internal pull-down resistors.
20 +VS+5V Power Supply
21 REFB Bottom Reference Bypass. For external bypass-
ing of internal +1.25V reference.
22 CM Common-Mode Voltage. It is derived by (REFT +
REFB)/2.
23 REFT Top Reference Bypass. For external bypassing
of internal +3.25V reference.
24 +VS+5V Power Supply
25 GND Ground
26 IN Input
27 IN Complementary Input
28 GND Ground
PIN DESCRIPTIONS
NOTE: (1) “ ” indicates the portion of the waveform that will stretch out at slower sample rates.
GND
B1
B2
B3
B4
B5
B6
B7
B8
B9
B10
B11
B12
GND
GND
IN
IN
GND
+V
S
REFT
CM
REFB
+V
S
MSBI
OE
+V
S
CLK
+V
S
1
2
3
4
5
6
7
8
9
10
11
12
13
14
28
27
26
25
24
23
22
21
20
19
18
17
16
15
ADS801
Track Hold
“N”Hold
“N + 1”Hold
“N + 2”Hold
“N + 3”
Hold
“
N + 4
”Hold
“
N + 5
”
Hold
“
N + 6
”
Track
Data Valid
N – 7 Data Valid
N – 6
Internal
Track-and-Hold
Convert
Clock
Output
Data
t
D
t
2
t
1
DATA LATENCY
(6.5 Clock Cycles)
t
CONV
t
L
t
H
Track Track Track Track
N – 3N – 5N – 4N – 2N – 1 N
Track Track
Data Valid
N – 8
(1)
Data Invalid
ADS801 5
SBAS036B www.ti.com
TYPICAL CHARACTERISTICS
At TA = +25°C, VS = +5V, Sampling Rate = 25MHz, and with a 50% duty cycle clock having a 2ns rise-and-fall time, unless otherwise noted.
SPECTRAL PERFORMANCE
0
Amplitude (dB)
Frequency (MHz)
2.5 5.0 7.5 10.0 12.5
0
–20
–40
–60
–80
–100
–120
f
IN
= 500kHz
SPECTRAL PERFORMANCE
0
Amplitude (dB)
Frequency (MHz)
2.5 5.0 7.5 10.0 12.5
0
–20
–40
–60
–80
–100
–120
f
IN
= 10MHz
2.0
1.0
0
–1.0
–2.0
DIFFERENTIAL LINEARITY ERROR
Code
DLE (LSB)
0 1024 2048 3072 4096
f
IN
= 500kHz
0 1024 2048 3072 4096
2.0
1.0
0
–1.0
–2.0
DIFFERENTIAL LINEARITY ERROR
Code
DLE (LSB)
f
IN
= 10MHz
0
–20
–40
–60
–80
–100
–120
2-TONE INTERMODULATION
Amplitude (dB)
0.0 3.13 6.25 9.38 12.50
Frequency (MHz)
f
1
= 4.5MHz
f
2
= 4.4MHz
INPUT FREQUENCY vs DYNAMIC PERFORMANCE
Frequency (MHz)
SFDR, SNR (dB)
80
75
70
65
60
55
500.1 1 10 100
SFDR
SNR
ADS801
6SBAS036B
www.ti.com
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VS = +5V, Sampling Rate = 25MHz, and with a 50% duty cycle clock having a 2ns rise-and-fall time, unless otherwise noted.
Counts
800k
600k
400k
200k
0.0
Code
N – 2N – 1 N N + 1 N + 2
OUTPUT NOISE HISTOGRAM (NO SIGNAL)
100
80
60
40
20
0
Input Amplitude (dBm)
SFDR (dBFS)
SWEPT POWER SFDR
–50 –40 –30 –20 –10 0 10
fIN = 10MHz
80
60
40
20
0
Input Amplitude (dBm)
SWEPT POWER SNR
–50 –40 –30 –20 –10 0 10
SNR (dB)
f
IN
= 10MHz
4.0
2.0
0
–2.0
–4.0
INTEGRAL LINEARITY ERROR
ILE (LSB)
0 1024 2048 3072 4096
Code
fIN = 500kHz
65
60
55
50
DYNAMIC PERFORMANCE vs
SINGLE-ENDED FULL-SCALE INPUT RANGE
Dynamic Range (dB)
12345
Single-Ended Full-Scale Input Range (Vp-p)
SFDR (f
IN
= 10MHz)
SNR (f
IN
= 10MHz)
70
65
60
55
DYNAMIC PERFORMANCE vs
DIFFERENTIAL FULL-SCALE INPUT RANGE
Dynamic Range (dB)
12345
Differential Full-Scale Input Range (Vp-p)
SFDR (f
IN
= 10MHz)
SNR (f
IN
= 10MHz)
ADS801 7
SBAS036B www.ti.com
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VS = +5V, Sampling Rate = 25MHz, and with a 50% duty cycle clock having a 2ns rise-and-fall time, unless otherwise noted.
DIFFERENTIAL LINEARITY ERROR vs TEMPERATURE
Temperature (°C)
DLE (LSBs)
0.5
0.4
0.3
0.2
0.1–50 –25 0 25 50 75 100
fIN = 10MHz
fIN = 500kHz
SPURIOUS-FREE DYNAMIC RANGE (SFDR) vs
TEMPERATURE
Temperature (°C)
SFDR (dBFS)
80
75
70
65
60
55–50 –25 0 25 50 75 100
f
IN
= 500kHz
f
IN
= 10MHz
SIGNAL-TO-NOISE RATIO vs TEMPERATURE
Temperature (°C)
SNR (dB)
70
68
66
64
62–50 –25 0 25 50 75 100
fIN = 500kHz
fIN = 10MHz
SIGNAL-TO-(NOISE + DISTORTION) vs
TEMPERATURE
Temperature (°C)
SINAD (dB)
70
65
60
55
50
45–50 –25 0 25 50 75 100
f
IN
= 500kHz
f
IN
= 10MHz
SUPPLY CURRENT vs TEMPERATURE
Temperature (°C)
IQ (mA)
52
51
50
49–50 –25 0 25 50 75 100
POWER DISSIPATION vs TEMPERATURE
Temperature (°C)
Power (mW)
265
260
255
250
245–50 –25 0 25 50 75 100
ADS801
8SBAS036B
www.ti.com
TYPICAL CHARACTERISTICS (Cont.)
At TA = +25°C, VS = +5V, Sampling Rate = 25MHz, and with a 50% duty cycle clock having a 2ns rise-and-fall time, unless otherwise noted.
GAIN ERROR vs TEMPERATURE
Temperature (°C)
Gain (% FSR)
–0.05
–0.55
–1.05
–1.55–50 –25 0 25 50 75 100
OFFSET ERROR vs TEMPERATURE
Temperature (°C)
Offset (% FSR)
–1.50
–1.75
–2.0
–2.25–50 –25 0 25 50 75 100
TRACK-MODE SMALL-SIGNAL INPUT BANDWIDTH
Frequency (Hz)
Track-Mode Input Response (dB)
10k
1
0
–1
–2
–3
–4
–5100k 1M 10M 100M 1G
ADS801 9
SBAS036B www.ti.com
FIGURE 2. Pipeline A/D Converter Architecture.
FIGURE 1. Input Track-and-Hold Configuration with Timing
Signals.
THEORY OF OPERATION
The ADS801 is a high-speed, sampling A/D converter with
pipelining. It uses a fully differential architecture and digital
error correction to ensure 12-bit resolution. The differential
track-and-hold circuit is shown in Figure 1. The switches are
controlled by an internal clock that is a non-overlapping 2-
phase signal, φ1 and φ2. At the sampling time, the input
signal is sampled on the bottom plates of the input capaci-
tors. In the next clock phase, φ2, the bottom plates of the
input capacitors are connected together and the feedback
capacitors are switched to the op amp output. At this time,
the charge redistributes between CI and CH, completing one
track-and-hold cycle. The differential output is a held DC
representation of the analog input at the sample time. The
track-and-hold circuit can also convert a single-ended input
signal into a fully differential signal for the quantizer.
The pipelined quantizer architecture has 11 stages with each
stage containing a 2-bit quantizer and a 2-bit Digital-to-
Analog Converter (DAC), as shown in Figure 2. Each 2-bit
quantizer stage converts on the edge of the sub-clock, which
is twice the frequency of the externally applied clock. The
output of each quantizer is fed into its own delay line to time-
φ1
φ1φ2φ1
φ1φ1
φ1
φ1
φ2
φ1φ2φ1
φ2
IN
IN
OUT
OUT
Op Amp
Bias V
CM
Op Amp
Bias V
CM
C
H
C
I
C
I
C
H
Input Clock (50%)
Internal Non-Overlapping Clock
B1 (MSB)
B2
B3
B4
B5
B6
B7
B8
B9
B10
B11
B12 (LSB)
2-Bit
DAC
2-Bit
Flash
Input
T/H Digital Delay
x2
x2
2-Bit
DAC
2-Bit
Flash
Digital Delay
2-Bit
Flash Digital Delay
2-Bit
DAC
2-Bit
Flash
Digital Delay
x2
Digital Error Correction
IN
IN
STAGE 1
STAGE 2
STAGE 10
STAGE 11
Σ
+–
Σ
+–
Σ
+–
ADS801
10 SBAS036B
www.ti.com
align it with the data created from the following quantizer
stages. This aligned data is fed into a digital error correction
circuit that can adjust the output data based on the informa-
tion found on the redundant bits. This technique gives the
ADS801 excellent differential linearity and ensures no miss-
ing codes at the 12-bit level.
Since there are two pipeline stages per external clock cycle,
there is a 6.5 clock cycle data latency from the start convert
signal to the valid output data. The output data is available in
Straight Offset Binary (SOB) or Binary Two’s Complement
(BTC) format.
THE ANALOG INPUT AND INTERNAL REFERENCE
The analog input of the ADS801 can be configured in various
ways and driven with different circuits, depending on the
nature of the signal and the level of performance desired. The
ADS801 has an internal reference that sets the full-scale input
range of the A/D converter. The differential input range has
each input centered around the common-mode of +2.25V,
with each of the two inputs having a full-scale range of +1.25V
to +3.25V. Since each input is 2Vp-p and 180° out-of-phase
with the other, a 4V differential input signal to the quantizer
results. As shown in Figure 3, the positive full-scale reference
(REFT) and the negative full-scale (REFB) are brought out for
external bypassing. In addition, the common-mode voltage
(CM) may be used as a reference to provide the appropriate
offset for the driving circuitry. However, care must be taken not
to appreciably load this reference node. For more information
regarding external references, single-ended input, and ADS801
drive circuits, refer to the applications section.
APPLICATIONS
DRIVING THE ADS801
The ADS801 has a differential input with a common-mode of
+2.25V. For AC-coupled applications, the simplest way to
create this differential input is to drive the primary winding of
a transformer with a single-ended input. A differential output
is created on the secondary if the center tap is tied to the
common-mode voltage of +2.25V, as per Figure 4. This
transformer-coupled input arrangement provides good high-
+FS (IN = +3.25V, IN = +1.25V) 111111111111 011111111111
+FS – 1LSB 111111111111 011111111111
+FS – 2LSB 111111111110 011111111110
+3/4 Full-Scale 111000000000 011000000000
+1/2 Full-Scale 110000000000 010000000000
+1/4 Full-Scale 101000000000 001000000000
+1LSB 100000000001 000000000001
Bipolar Zero (IN = IN = +2.25V) 100000000000 000000000000
–1LSB 011111111111 111111111111
–1/4 Full-Scale 011000000000 111000000000
–1/2 Full-Scale 010000000000 110000000000
–3/4 Full-Scale 001000000000 101000000000
–FS + 1LSB 000000000001 100000000001
–FS (IN = +1.25V, IN = +3.25V) 000000000000 100000000000
NOTE: (1) In the single-ended input mode, +FS = +4.25V and –FS = +0.25V.
DIGITAL OUTPUT DATA
The 12-bit output data is provided at CMOS logic levels. The
standard output coding is Straight Offset Binary (SOB) where
a full-scale input signal corresponds to all “1s” at the output,
as shown in Table I. This condition is met with pin 19 “LO”
or Floating, due to an internal pull-down resistor. By applying
a logic “HI” voltage to this pin, a Binary Two’s Complement
(BTC) output will be provided where the most significant bit
is inverted. The digital outputs of the ADS801 can be set to
a high-impedance state by driving
OE
(pin 18) with a logic
HIGH. Normal operation is achieved with pin 18 LOW or
Floating, due to internal pull-down resistors. This function is
provided for testability purposes and is not meant to drive
digital buses directly or be dynamically changed during the
conversion process.
TABLE I. Coding Table for the ADS801.
CLOCK REQUIREMENTS
The CLK pin accepts a CMOS level clock input. The rising
and falling edges of the externally applied convert command
clock controls the various interstage conversions in the
pipeline. Therefore, the duty cycle of the clock should be held
at 50% with low jitter and fast rise-and-fall times of 2ns or
less. This is particularly important when digitizing a high-
frequency input and operating at the maximum sample rate.
Deviation from a 50% duty cycle will effectively shorten some
of the interstage settling times, thus degrading the SNR and
DNL performance.
OUTPUT CODE
SOB BTC
PIN 19 PIN 19
DIFFERENTIAL INPUT(1)
FLOATING or LOW
HIGH
FIGURE 4. AC-Coupled Single-Ended to Differential Drive
Circuit Using a Transformer.
+1.25V
+3.25V
2kΩ
2kΩ
0.1µF
0.1µF
+2.25V
REFT
REFB
CM
ADS801
To
Internal
Comparators
21
22
23
FIGURE 3. Internal Reference Structure.
Mini-Circuits
TT1-6-KK81
or equivalent
22
26
27
CM
IN
IN
ADS801
AC Input
Signal 22pF
22pF
0.1µF
ADS801 11
SBAS036B www.ti.com
frequency AC performance. It is important to select a trans-
former that gives low distortion and does not exhibit core
saturation at full-scale voltage levels. Since the transformer
does not appreciably load the ladder, there is no need to
buffer the Common-Mode (CM) output in this instance. In
general, it is advisable to keep the current draw from the CM
output pin below 0.5µA to avoid nonlinearity in the internal
reference ladder. A FET input operational amplifier, such as
the OPA130, can provide a buffered reference for driving
external circuitry. The analog IN and
IN
inputs should be
bypassed with 22pF capacitors to minimize track-and-hold
glitches and to improve high input frequency performance.
Figure 5 illustrates another possible low-cost interface circuit
that utilizes resistors and capacitors in place of a transformer.
Depending on the signal bandwidth, the component values
should be carefully selected in order to maintain the product
performance outlined. The input capacitors, CIN, and the input
resistors, RIN, create a high-pass filter with the lower corner
frequency at fC = 1/(2pRINCIN). The corner frequency can be
reduced by either increasing the value of RIN or CIN. If the
circuit operates with a 50Ω or 75Ω impedance level, the
resistors are fixed and only the value of the capacitor can be
increased. Usually AC-coupling capacitors are electrolytic or
tantalum capacitors with values of 1µF or higher. It should be
noted that these large capacitors become inductive with
increased input frequency, which could lead to signal ampli-
tude errors or oscillation. To maintain a low AC-coupling
impedance throughout the signal band, a small value (e.g.
1µF) ceramic capacitor could be added in parallel with the
polarized capacitor.
Capacitors CSH1 and CSH2 are used to minimize current
glitches resulting from the switching in the input track-and-
hold stage and to improve signal-to-noise performance. These
capacitors can also be used to establish a low-pass filter and
effectively reduce the noise bandwidth. In order to create a
real pole, resistors RSER1 and RSER2 were added in series with
each input. The cutoff frequency of the filter is determined by
fC = 1/(2pRSER • (CSH + CADC)), where RSER is the resistor in
series with the input, CSH is the external capacitor from the
input to ground, and CADC is the internal input capacitance of
the A/D converter (typically 4pF).
Resistors R1 and R2 are used to derive the necessary
common-mode voltage from the buffered top and bottom
references. The total load of the resistor string should be
selected so that the current does not exceed 1mA. Although
the circuit in Figure 5 uses two resistors of equal value so
that the common-mode voltage is centered between the top
and bottom reference (+2.25V), it is not necessary to do so.
In all cases the center point, VCM, should be bypassed to
ground in order to provide a low-impedance AC ground.
If the signal needs to be DC-coupled to the input of the
ADS801, an operational amplifier input circuit is required. In
the differential input mode, any single-ended signal must be
modified to create a differential signal. This can be accom-
plished by using two operational amplifiers; one in the
noninverting mode for the input and the other amplifier in the
inverting mode for the complementary input. The low distor-
tion circuit in Figure 6 will provide the necessary input
shifting required for signals centered around ground. It also
employs a diode for output level shifting to ensure a low
distortion +3.25V output swing. Other amplifiers can be used
in place of the OPA842s if the lowest distortion is not
necessary. If output level shifting circuits are not used, care
must be taken to select operational amplifiers that give the
necessary performance when swinging to +3.25V with a ±5V
supply operational amplifier.
The ADS801 can also be configured with a single-ended
input full-scale range of +0.25V to +4.25V by tying the
complementary input to the common-mode reference volt-
age (see Figure 7). This configuration will result in increased
even-order harmonics, especially at higher input frequen-
cies. However, this tradeoff may be quite acceptable for
time-domain applications. The driving amplifier must give
adequate performance with a +0.25V to +4.25V output
swing in this case.
FIGURE 5. AC-Coupled Differential Input Circuit.
ADS8xx
RSER1(1)
49.9Ω
R3
1kΩ
R2
(6kΩ)
R1
(6kΩ)
C2
0.1µF
CSH1
22pF
CSH2
22pF
C3
0.1µF
C1
0.1µF
CIN
0.1µF
VCM
CIN
0.1µF
RIN1
25Ω
RIN2
25ΩRSER2(1)
49.9Ω
+3.25V
Top Reference
+1.25V
Bottom Reference
IN
NOTE: (1) Indicates optional component.
IN
ADS801
12 SBAS036B
www.ti.com
FIGURE 6. A Low-Distortion, DC-Coupled, Single-Ended to Differential Input Driver Circuit.
FIGURE 7. Single-Ended Input Connection.
EXTERNAL REFERENCES AND
ADJUSTMENT OF FULL-SCALE RANGE
The internal reference buffers are limited to approximately
1mA of output current. As a result, these internal +1.25V and
+3.25V references may be overridden by external references
that have at least 18mA (at room temperature) of output drive
capability. In this instance, the common-mode voltage will be
set halfway between the two references. This feature can be
used to adjust the gain error, improve gain drift, or to change
the full-scale input range of the ADS801. Changing the full-
scale range to a lower value has the benefit of easing the
swing requirements of external input amplifiers. The external
references can vary as long as the value of the external top
reference (REFTEXT) is less than or equal to +3.4V, the value
of the external bottom reference (REFBEXT) is greater than or
equal to +1.1V, and the difference between the external
references are greater than or equal to 1.5V.
For the differential configuration, the full-scale input
range will be set to the external reference values that are
selected. For the single-ended mode, the input range is
2 • (REFTEXT – REFBEXT), with the common-mode being
centered at (REFTEXT + REFBEXT)/2. Refer to the typical
characteristics for expected performance versus full-scale
input range.
The circuit in Figure 8 works completely on a single +5V
supply. As a reference element, it uses micro-power refer-
ence REF1004-2.5 that is set to a quiescent current of
0.1mA. Amplifier A2 is configured as a follower to buffer the
+1.25V generated from the resistor divider. To provide the
necessary current drive, a pull-down resistor (RP) is added.
Amplifier A1 is configured as an adjustable-gain stage, with
a range of approximately 1 to 1.32. The pull-up resistor again
relieves the op amp from providing the full current drive. The
value of the pull-up, pull-down resistors is not critical and can
be varied to optimize power consumption. The need for pull-
up, pull-down resistors depends only on the drive capability
of the selected drive amplifier, and thus can be omitted.
PC-BOARD LAYOUT AND BYPASSING
A well-designed, clean pc-board layout will assure proper
operation and clean spectral response. Proper grounding and
bypassing, short lead lengths, and the use of ground planes
are particularly important for high-frequency circuits. Multilayer
pc-boards are recommended for best performance, but if
carefully designed, a two-sided pc-board with large, heavy
ground planes can give excellent results. It is recommended
that the analog and digital ground pins of the ADS801 be
connected directly to the analog ground plane. In our experi-
ence, this gives the most consistent results. The A/D converter
power-supply commons should be tied together at the analog
ground plane. Power supplies should be bypassed with 0.1µF
ceramic capacitors as close to the pin as possible.
604Ω
301Ω
301Ω
301Ω
604Ω
49.9Ω
301Ω
604Ω
2.49kΩ
2.49kΩ+2.25V
OPA842
OPA130
301Ω
0.1µF
OPA842
OPA842
+5V
–5V
+5V
(2)
+5V
–5V
+5V
+5V
+5V
–5V
BAS16
(1)
BAS16
(1)
301Ω
24.9ΩInput Level
Shift Buffer
Optional
High Impedance
Input Amplifier
DC-Coupled
Input Signal
26 IN
22 CM
27 IN
ADS801
NOTES: (1) A Philips BAS16 diode or equivalent
may be used. (2) Supply bypassing not shown.
22pF
22pF
604Ω
0.1µF
0.1µF
22
26
27
CM
IN
IN
ADS801
0.1µF
Single-Ended
Input Signal
Full-Scale = +0.25V to +4.25V with internal references.
22pF
ADS801 13
SBAS036B www.ti.com
DYNAMIC PERFORMANCE TESTING
The ADS801 is a high-performance converter and careful
attention to test techniques is necessary to achieve accurate
results. Highly accurate phase-locked signal sources allow
high-resolution FFT measurements to be made without using
data windowing functions. A low-jitter signal generator, such
as the HP8644A for the test signal, phase-locked with a low-
jitter HP8022A pulse generator for the A/D converter clock,
gives excellent results. Low-pass filtering (or bandpass filter-
ing) of test signals is absolutely necessary to test the low
distortion of the ADS801. Using a signal amplitude slightly
lower than full-scale will allow a small amount of “headroom”
so that noise or DC-offset voltage will not overrange the A/D
converter and cause clipping on signal peaks.
DYNAMIC PERFORMANCE DEFINITIONS
1. Signal-to-Noise-and-Distortion Ratio (SINAD):
10 15
log SinewaveSignalPower
Noise HarmonicPower first harmonics+
(
)
2. Signal-to-Noise Ratio (SNR):
10log SinewaveSignalPower
NoisePower
3. Intermodulation Distortion (IMD):
10 5
log PrHighestIMD oductPower to th order
SinewaveSignalPower −
(
)
IMD is referenced to the larger of the test signals f1 or f2. Five
“bins” either side of peak are used for calculation of funda-
mental and harmonic power. The “0” frequency bin (DC) is
not included in these calculations, as it is of little importance
in dynamic signal processing applications.
FIGURE 8. Optional External Reference to Set the Full-Scale Range Utilizing a Dual, Single-Supply Op Amp.
2kΩ
+2.5V to +3.25V
+5V
+5V
RP
220Ω
RP
220Ω
10kΩ
6.2kΩ
0.1µF+2.5V
10kΩ
1/2
OPA2234
A
1
A
2
1/2
OPA2234
Bottom
Reference
Top
Reference
REF1004
+1.25V
10kΩ
10kΩ(1)
10kΩ(1)
NOTE: (1) Use parts alternatively for adjustment capability.
ADS801
14 SBAS036B
www.ti.com
FIGURE 9. ADS801 Interface Schematic with AC-Coupling and External Buffers.
GND
LSB
MSB
GND
11
12
13
14
15
16
17
18
9
8
7
6
5
4
3
2
1
19
+V
S
CLK
+V
S
OE
MSBI
+V
S
REFB
CM
REFT
+V
S
GND
IN
IN
GND
15
16
17
18
19
20
21
22
23
24
25
26
27
28
14
13
12
11
10
9
8
7
6
5
4
3
2
1
ADS801
0.1µF
0.1µF0.1µF
0.1µF
0.1µF
0.1µF0.1µF
R
1
50Ω
R
2
50Ω
Ext
Clk
AC Input
Signal
11
12
13
14
15
16
17
18
9
8
7
6
5
4
3
2
Dir
G+
1
19 Dir
G+
–541
–541
Mini-Circuits
TT1-6-KK81
or equivalent
22pF
22pF
(1)
NOTE: (1) All capacitors should be located as close to the pins as the manufacturing
process will allow. Ceramic X7R surface-mount capacitors or equivalent are recommended.
+5V
PACKAGING INFORMATION
Orderable Device Status (1) Package
Type Package
Drawing Pins Package
Qty Eco Plan (2) Lead/Ball Finish MSL Peak Temp (3)
ADS801E OBSOLETE SSOP DB 28 None Call TI Call TI
ADS801E/1K OBSOLETE SSOP DB 28 None Call TI Call TI
ADS801U ACTIVE SOIC DW 28 28 None CU SNPB Level-3-220C-168 HR
(1) The marketing status values are defined as follows:
ACTIVE: Product device recommended for new designs.
LIFEBUY: TI has announced that the device will be discontinued, and a lifetime-buy period is in effect.
NRND: Not recommended for new designs. Device is in production to support existing customers, but TI does not recommend using this part in
a new design.
PREVIEW: Device has been announced but is not in production. Samples may or may not be available.
OBSOLETE: TI has discontinued the production of the device.
(2) Eco Plan - May not be currently available - please check http://www.ti.com/productcontent for the latest availability information and additional
product content details.
None: Not yet available Lead (Pb-Free).
Pb-Free (RoHS): TI's terms "Lead-Free" or "Pb-Free" mean semiconductor products that are compatible with the current RoHS requirements
for all 6 substances, including the requirement that lead not exceed 0.1% by weight in homogeneous materials. Where designed to be soldered
at high temperatures, TI Pb-Free products are suitable for use in specified lead-free processes.
Green (RoHS & no Sb/Br): TI defines "Green" to mean "Pb-Free" and in addition, uses package materials that do not contain halogens,
including bromine (Br) or antimony (Sb) above 0.1% of total product weight.
(3) MSL, Peak Temp. -- The Moisture Sensitivity Level rating according to the JEDECindustry standard classifications, and peak solder
temperature.
Important Information and Disclaimer:The information provided on this page represents TI's knowledge and belief as of the date that it is
provided. TI bases its knowledge and belief on information provided by third parties, and makes no representation or warranty as to the
accuracy of such information. Efforts are underway to better integrate information from third parties. TI has taken and continues to take
reasonable steps to provide representative and accurate information but may not have conducted destructive testing or chemical analysis on
incoming materials and chemicals. TI and TI suppliers consider certain information to be proprietary, and thus CAS numbers and other limited
information may not be available for release.
In no event shall TI's liability arising out of such information exceed the total purchase price of the TI part(s) at issue in this document sold by TI
to Customer on an annual basis.
PACKAGE OPTION ADDENDUM
www.ti.com 14-Feb-2005
Addendum-Page 1
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