March 2009 1 M9999-032409
MIC2169 Micrel
MIC2169
500kHz PWM Synchronous Buck Control IC
General Description
The MIC2169 is a high-effi ciency, simple to use 500kHz PWM
synchronous buck control IC housed in a small MSOP-10
package. The MIC2169 allows compact DC/DC solutions
with a minimal external component count and cost.
The MIC2169 operates using a 3V to 14.5V input, without
the need for any additional bias voltage. The output voltage
can be precisely regulated down to 0.8V. The adaptive all
N-Channel MOSFET drive scheme allows effi ciencies, over
95%, across a wide load range.
The MIC2169 senses current across the high-side N-Channel
MOSFET, eliminating the need for an expensive and lossy
current-sense resistor. Current limit accuracy is maintained
via a positive temperature coeffi cient that tracks the increas-
ing RDS(ON) of the external MOSFET. Additional cost and
space are saved by the internal in-rush-current limiting and
digital soft-start.
The MIC2169 is available in a 10-pin MSOP package, with a
wide junction operating range of –40°C to +125°C.
All support documentation can be found on Micrel’s web site
at: www.micrel.com.
Typical Application
2.5μH3.3V
VIN = 5V
VDD
COMP/EN
VIN
CS
FB
GND
LSD
BST 1kΩ
10kΩ
4kΩ3.24kΩ
4.7μF
100μF
0.1μF
100nF IRF7821
SD103BWS
IRF7821
150pF
HSD
VSW
MIC2169
150μF x 2
MIC2169 Adjustable Output 500kHz Converter
Features
• 3V to 14.5V input voltage range
• Adjustable output voltage down to 0.8V
• Up to 95% effi ciency
• 500kHz PWM operation
• Adjustable current limit senses high-side N-Channel
MOSFET current
• No external current-sense resistor
• Adaptive gate drive increases effi ciency
• Fast transient response
– Externally compensated
• Overvoltage protection protects the load in fault
conditions
• Dual mode current limit speeds up recovery time
• Hiccup mode short-circuit protection
• Small size MSOP 10-lead package
Applications
• Point-of-load DC/DC conversion
• Set-top boxes
• Graphic cards
• LCD power supplies
• Telecom power supplies
• Networking power supplies
• Cable modems and routers
Micrel, Inc. • 2180 Fortune Drive • San Jose, CA 95131 • USA • tel + 1 (408) 944-0800 • fax + 1 (408) 474-1000 • http://www.micrel.com
50
55
60
65
70
75
80
85
90
95
100
0246810121416
EFFICIENCY (%)
I
LOAD
(A)
MIC2169 Efficienc
y
VIN = 5V
VOUT = 3.3V
MIC2169 Micrel
M9999-032409 2 March 2009
Pin Confi guration
FB GND65
1VIN
VDD
CS
COMP
10 BST
HSD
VSW
LSD
9
8
7
2
3
4
10-Pin MSOP (MM)
Pin Description
Pin Number Pin Name Pin Function
1 VIN Supply Voltage (Input): 3V to 14.5V.
2 VDD 5V Internal Linear Regulator (Output): VDD is the external MOSFET gate
drive supply voltage and an internal supply bus for the IC. When VIN is <5V,
this regulator operates in dropout mode.
3 CS Current Sense (Input): Current-limit comparator noninverting input. The cur-
rent limit is sensed across the MOSFET during the ON time. The current can
be set by the resistor in series with the CS pin.
4 COMP Compensation (Input): Pin for external compensation. .
5 FB Feedback (Input): Input to error amplifi er. Regulates error amplifi er to 0.8V.
6 GND Ground (Return).
7 LSD Low-Side Drive (Output): High-current driver output for external synchro-
nous MOSFET.
8 VSW Switch (Return): High-side MOSFET driver return.
9 HSD High-Side Drive (Output): High-current output-driver for the high-side MOS-
FET. When VIN is between 3.0V to 5V, 2.5V threshold MOSFETs should be
used. At VIN > 5V, 5V threshold MOSFETs should be used.
10 BST Boost (Input): Provides the drive voltage for the high-side MOSFET driver.
The gate-drive voltage is higher than the source voltage by VIN minus a
diode drop.
Ordering Information
Part Number Pb-Free Part Number Frequency Junction Temp. Range Package
MIC2169BMM MIC2169YMM 500kHz –40°C to +125°C 10-lead MSOP
March 2009 3 M9999-032409
MIC2169 Micrel
Absolute Maximum Ratings(1)
Supply Voltage (VIN) ...................................................15.5V
Booststrapped Voltage (VBST) .................................VIN +5V
Junction Temperature (TJ) ..................–40°C ≤ TJ ≤ +125°C
Storage Temperature (TS) ........................ –65°C to +150 °C
Operating Ratings(2)
Supply Voltage (VIN) ..................................... +3V to +14.5V
Output Voltage Range ..........................0.8V to VIN × DMAX
Package Thermal Resistance
θJA 10-lead MSOP .............................................180°C/W
Electrical Characteristics(3)
TJ = 25°C, VIN = 5V; bold values indicate –40°C < TJ < +125°C; unless otherwise specifi ed.
Parameter Condition Min Typ Max Units
Feedback Voltage Reference (± 1%) 0.792 0.8 0.808 V
Feedback Voltage Reference (± 2% over temp) 0.784 0.8 0.816 V
Feedback Bias Current 30 100 nA
Output Voltage Line Regulation 0.03 % / V
Output Voltage Load Regulation 0.5 %
Output Voltage Total Regulation 3V ≤ VIN ≤ 14.5V; 1A ≤ IOUT ≤ 10A; (VOUT = 2.5V)(4) 0.6 %
Oscillator Section
Oscillator Frequency 450 500 550 kHz
Maximum Duty Cycle 92 %
Minimum On-Time(4) 30 60 ns
Input and VDD Supply
PWM Mode Supply Current VCS = VIN –0.25V; VFB = 0.7V (output switching but excluding 1.5 3 mA
external MOSFET gate current.)
Digital Supply Voltage (VDD) VIN ≥ 6V 4.7 5 5.3 V
Error Amplifi er
DC Gain 70 dB
Transconductance 1 ms
Soft-Start
Soft-Start Current After timeout of internal timer. See “Soft-Start” section. 8.5 μA
Current Sense
CS Over Current Trip Point VCS = VIN –0.25V 160 200 240 μA
Temperature Coeffi cient +1800
ppm/°C
Output Fault Correction Thresholds
Upper Threshold, VFB_OVT (relative to VFB) +3 %
Lower Threshold, VFB_UVT (relative to VFB) –3 %
Notes:
1. Absolute maximum ratings indicate limits beyond which damage to the component may occur. Electrical specifi cations do not apply when operating
the device outside of its operating ratings. The maximum allowable power dissipation is a function of the maximum junction temperature, TJ(max),
the junction-to-ambient thermal resistance, θJA, and the ambient temperature, TA. The maximum allowable power dissipation will result in excessive
die temperature, and the regulator will go into thermal shutdown.
2. Devices are ESD sensitive, handling precautions required.
3. Specifi cation for packaged product only.
4. Guaranteed by design.
MIC2169 Micrel
M9999-032409 4 March 2009
Electrical Characteristics(5)
Parameter Condition Min Typ Max Units
Gate Drivers
Rise/Fall Time Into 3000pF at VIN > 5V 30 ns
Output Driver Impedance Source, VIN = 5V 6 Ω
Sink, VIN = 5V 6 Ω
Source, VIN = 3V 10 Ω
Sink, VIN = 3V 10 Ω
Driver Non-Overlap Time Note 6 10 20 ns
Notes:
5. Specifi cation for packaged product only.
6. Guaranteed by design.
March 2009 5 M9999-032409
MIC2169 Micrel
Typical Characteristics
VIN = 5V
0.5
0.7
0.9
1.1
1.3
1.5
1.7
1.9
2.1
2.3
2.5
2.7
2.9
-40 -20 0 20 40 60 80 100120140
IDD (mA)
TEMPERATURE (°C)
PWM Mode Supply Current
vs. Temperature
0.5
1.0
1.5
2.0
0 5 10 15
QUIESCENT CURRENT (mA)
SUPPLY VOLTAGE (V)
PWM Mode Supply Current
vs. Suppl
y
Voltage
0.7980
0.7985
0.7990
0.7995
0.8000
0.8005
0.8010
051015
V
FB
(V)
V
IN
(V)
V
FB
Line Regulation
0.792
0.794
0.796
0.798
0.800
0.802
0.804
0.806
-60 -30 0 30 60 90 120 150
VFB (V)
TEMPERATURE (°C)
VFB vs. Temperature
0
1
2
3
4
5
6
051015
V
DD
(V)
V
IN
(V)
VDD Line Regulation
4.90
4.92
4.94
4.96
4.98
5.00
5.02
0 5 10 15 20 25 30
VDD REGULATOR VOLTAGE (V)
LOAD CURRENT (mA)
VDD Load Regulation
0.0
0.5
1.0
1.5
2.0
2.5
3.0
3.5
4.0
4.5
5.0
-60 -30 0 30 60 90 120 150
V
DD
LINE REGULATION (%)
TEMPERATURE ( °C)
V
DD
Line Regulation
vs. Temperature
450
460
470
480
490
500
510
520
530
540
550
-60 -30 0 30 60 90 120 150
FREQUENCY (kHz)
TEMPERATURE ( °C)
Oscillator Frequency
vs. Temperature
-1.5
-1.0
-0.5
0
0.5
1.0
1.5
051015
FREQUENCY VARIATION (%)
V
IN
(V)
Oscillator Frequency
vs. Suppl
y
Voltage
0
1
2
3
4
0246810
V
OUT
(V)
I
LOAD
(A)
Current Limit Foldback
R
CS
= 1kΩ
Top MOSFET = Si4800
100
120
140
160
180
200
220
240
260
-60 -30 0 30 60 90 120 150
ICS ( μA)
TEMPERATURE ( °C)
Overcurrent Trip Point
vs. Temperature
MIC2169 Micrel
M9999-032409 6 March 2009
Functional Description
The MIC2169 is a voltage mode, synchronous step-down
switching regulator controller designed for high power without
the use of an external sense resistor. It includes an internal
soft-start function (which reduces the power supply input
surge current at start-up by controlling the output voltage rise
time), a PWM generator, a reference voltage, two MOSFET
drivers, and short-circuit current limiting circuitry to form a
complete 500kHz switching regulator.
Theory of Operation
The MIC2169 is a voltage mode step-down regulator. The
block diagram, above, illustrates the voltage control loop. The
output voltage variation due, to load or line changes, will be
sensed by the inverting input of the transconductance error
amplifi er via the feedback resistors R3, and R2 and compared
to a reference voltage at the non-inverting input. This will cause
a small change in the DC voltage level at the output of the
error amplifi er which is the input to the PWM comparator. The
other input to the comparator is a 5V triangular waveform. The
comparator generates a rectangular waveform whose width
tON is equal to the time from the start of the clock cycle t0 until
t1, the time the triangle crosses the output waveform of the
error amplifi er. To illustrate the control loop, assume the output
voltage drops due to sudden load turn-on, this would cause
the inverting input of the error amplifi er which is a divided
down version of VOUT to be slightly less than the reference
Functional Diagram
Current Limit
Reference
Current Limit
Comparator
Error
Amp
Low-Side
Driver
High-Side
Driver
PWM
Comparator
FB
COMP GND
LSD
VREF +3%
VREF 3%
HSD
VDD
CBST
CS
VDD
5V
5V
5V
C2 C1
R1
5V
0.8V
VIN
SW
Q2
Q1
L1
Driver
Logic
0.8V
BG Valid
Clamp &
Startup
Current
Enable
Error
Loop
Hys
Comparator
5V LDO
Bandgap
Reference
Soft-Start &
Digital Delay
Counter
MIC2169
Ramp
Clock
BOOST
R2
R3
2Ω
RSW
RCS
COUT
VOUT
1000pF
1.4Ω
CIN
D1
MIC2169 Block Diagram
voltage. This causes the output voltage of the error amplifi er
to go high. This will also increase the PWM comparator tON
time of the top side MOSFET, causing the output voltage to
go up and bringing VOUT back in regulation.
Soft-Start
The COMP pin on the MIC2169 is used for the following two
functions:
1. External compensation to stabilize the voltage
control loop.
2. Soft-start.
For better understanding of the soft-start feature, assume VIN
= 12V. The COMP pin has an internal 6.5μA current source
that charges the external compensation capacitor. As soon as
this voltage rises to 180mV (t = Cap_COMP × 0.18V/8.5μA),
the MIC2169 allows the internal VDD linear regulator to power
up and as soon as it crosses the undervoltage lockout of 2.6V,
the chip’s internal oscillator starts switching. At this point, the
COMP pin current source increases to 40μA and an internal
1 1-bit counter starts counting. This takes approximately 2ms
to complete. During counting, the COMP voltage is clamped
at 0.65V. After this counting cycle, the COMP current source
is reduced to 8.5μA and the COMP pin voltage rises from
0.65V to 0.95V, the bottom edge of the saw-tooth oscillator.
This is the beginning of 0% duty cycle which increases slowly
causing the output voltage to rise slowly. The MIC2169 has
March 2009 7 M9999-032409
MIC2169 Micrel
two hysteretic comparators that are enabled when VOUT is
within ±3% of steady state. When the output voltage reaches
97% of programmed output voltage, then the gm error amplifi er
is enabled along with the hysteretic comparator. From this
point onwards, the voltage control loop (gm error amplifi er) is
fully in control and will regulate the output voltage.
Soft-start time can be calculated approximately by adding
the following four time frames:
t1 = Cap_COMP × 0.18V/8.5μA
t2 = 12 bit counter, approx 2ms
t3 = Cap_COMP × 0.3V/8.5μA
t4 VV0.5 Cap_COMP
8.5 A
OUT
IN
=⎛
⎝
⎜⎞
⎠
⎟×× μ
Soft-Start Time(Cap_COMP=100nF) = t1 + t2 + t3 +
t4 = 2.1ms + 2ms + 3.5ms + 1.8ms = 10ms
Current Limit
The MIC2169 uses the RDS(ON) of the top power MOSFET
to measure output current. Since it uses the drain to source
resistance of the power MOSFET, it is not very accurate.
However, this scheme is adequate to protect the power supply
and external components during a fault condition by cutting
back the time the top MOSFET is on if the feedback voltage
is greater than 0.67V. In case of a hard short when feedback
voltage is less than 0.67V, the MIC2169 discharges the COMP
capacitor to 0.65V, resets the digital counter and automatically
shuts off the top gate drive, and the gm error amplifi er and the
–3% hysteretic comparators are completely disabled and the
soft-start cycles restarts. This mode of operation is called the
“hiccup mode” and its purpose is to protect the down stream
load in case of a hard short. The circuit in Figure 1 illustrates
the MIC2169 current limiting circuit.
L1 Inductor
V
IN
V
OUT
HSD
LSD
RCS
CS
200μA
C2
C
IN
C1
C
OUT
Q1
MOSFET N
Q2
MOSFET N
1.4Ω
1000pF
2Ω
0.1μF
Figure 1. The MIC2169 Current Limiting Circuit
The current limiting resistor RCS is calculated by the follow-
ing equation:
RRI
200 A
CS DS(ON) Q1 L
=×
μ Equation (1)
II 1
2 Inductor Ripple Current
LLOAD
=+
()
where:
Inductor Ripple Current =
VV–V
VF L
OUT IN OUT
IN SWITCHING
×
()
××
F
SWITCHING = 500kHz
200μA is the internal sink current to program the MIC2169
current limit.
The MOSFET RDS(ON) varies 30% to 40% with temperature;
therefore, it is recommended that a 50% margin be added
to the load current (ILOAD) in the above equation to avoid
false current limiting due to increased MOSFET junction
temperature rise. It is also recommended to connect the
RCS resistor directly to the drain of the top MOSFET Q1,
and the RSW resistor to the source of Q1 to accurately sense
the MOSFETs RDS(ON). To make the MIC2169 insensitive to
board layout and noise generated by the switch node. For
this a 1.4Ω resistor and a 1000pF capacitor is recommended
between the switch node and ground. A 0.1μF capacitor, in
parallel with RCS, should be connected to fi lter some of the
switching noise.
Internal VDD Supply
The MIC2169 controller internally generates VDD for self bias-
ing and to provide power to the gate drives. This VDD supply
is generated through a low-dropout regulator and generates
5V from VIN supply greater than 5V. For supply voltage less
than 5V, the VDD linear regulator is approximately 200mV in
dropout. Therefore, it is recommended to short the VDD supply
to the input supply through a 5Ω resistor for input supplies
between 2.9V to 5V.
MOSFET Gate Drive
The MIC2169 high-side drive circuit is designed to switch an
N-Channel MOSFET. The block diagram on page 6 shows a
bootstrap circuit, consisting of D1 and CBST. It supplies energy
to the high-side drive circuit. Capacitor CBST is charged while
the low-side MOSFET is on and the voltage on the VSW pin
is approximately 0V. When the high-side MOSFET driver is
turned on, energy from CBST is used to turn the MOSFET
on. As the MOSFET turns on, the voltage on the VSW pin
increases to approximately VIN. Diode D1 is reversed biased
and CBST fl oats high while continuing to keep the high-side
MOSFET on. When the low-side switch is turned back on,
CBST is recharged through D1. The drive voltage is derived
from the internal 5V VDD bias supply. The nominal low-side
gate drive voltage is 5V and the nominal high-side gate drive
voltage is approximately 4.5V due the voltage drop across D1.
An approximate 20ns delay between the high- and low-side
driver transition is used to prevent current from simultane-
ously fl owing unimpeded through both MOSFETs.
MOSFET Selection
The MIC2169 controller works from input voltages of 3V to
13.2V and has an internal 5V regulator to provide power to
turn the external N-Channel power MOSFETs for high- and
low-side switches. For applications where VIN < 5V, the internal
VDD regulator operates in dropout mode, and it is necessary
that the power MOSFETs used are sub-logic level and are in
full conduction mode for VGS of 2.5V. For applications when
VIN > 5V; logic-level MOSFETs, whose operation is specifi ed
MIC2169 Micrel
M9999-032409 8 March 2009
at VGS = 4.5V must be used.
It is important to note the on-resistance of a MOSFET increases
with rising temperature. A 75°C rise in junction temperature
will increase the channel resistance of the MOSFET by 50%
to 75% of the resistance specifi ed at 25°C. This change in
resistance must be accounted for when calculating MOSFET
power dissipation and in calculating the value of current-sense
(CS) resistor . Total gate charge is the charge required to turn
the MOSFET on and off under specifi ed operating conditions
(VDS and VGS). The gate charge is supplied by the MIC2169
gate-drive circuit. At 500kHz switching frequency and above,
the gate charge can be a signifi cant source of power dissipa-
tion in the MIC2169. At low output load, this power dissipation
is noticeable as a reduction in effi ciency. The average current
required to drive the high-side MOSFET is:
IQf
G[high-side](avg) G S
=×
where:
IG[high-side](avg) = average high-side MOSFET gate
current.
QG = total gate charge for the high-side MOSFET taken from
manufacturer’s data sheet for VGS = 5V.
The low-side MOSFET is turned on and off at VDS = 0 because
the freewheeling diode is conducting during this time. The
switching loss for the low-side MOSFET is usually negligible.
Also, the gate-drive current for the low-side MOSFET is
more accurately calculated using CISS at VDS = 0 instead
of gate charge.
For the low-side MOSFET:
ICVf
G[low-side](avg) ISS GS S
=××
Since the current from the gate drive comes from the input
voltage, the power dissipated in the MIC2169, due to gate
drive, is:
PVI I
GATEDRIVE IN G[high-side](avg) G[low-side](avg)
=+
()
A convenient fi gure of merit for switching MOSFETs is the on
resistance times the total gate charge RDS(ON) × QG. Lower
numbers translate into higher effi ciency. Low gate-charge
logic-level MOSFETs are a good choice for use with the
MIC2169.
Parameters that are important to MOSFET switch selection
are:• Voltage rating
• On-resistance
• Total gate charge
The voltage ratings for the top and bottom MOSFET are
essentially equal to the input voltage. A safety factor of 20%
should be added to the VDS(max) of the MOSFET s to account
for voltage spikes due to circuit parasitics.
The power dissipated in the switching transistor is the sum
of the conduction losses during the on-time (PCONDUCTION)
and the switching losses that occur during the period of time
when the MOSFETs turn on and off (PAC).
PP P
SW CONDUCTION AC
=+
where:
PIR
CONDUCTION SW(rms) SW
2
=×
PP P
AC AC(off) AC(on)
=+
R
SW = on-resistance of the MOSFET switch
D duty cycle V
VO
IN
=⎛
⎝
⎜⎞
⎠
⎟
Making the assumption the turn-on and turn-off transition times
are equal; the transition times can be approximated by:
tCVC V
I
TISS GS OSS IN
G
=×+ ×
where:
C
ISS and COSS are measured at VDS = 0
I
G = gate-drive current (1A for the MIC2169)
The total high-side MOSFET switching loss is:
P(VV)Itf
AC IN D PK T S
=+×××
where:
t
T = switching transition time (typically 20ns to 50ns)
V
D = freewheeling diode drop, typically 0.5V
f
S it the switching frequency, nominally 500kHz
The low-side MOSFET switching losses are negligible and
can be ignored for these calculations.
Inductor Selection
V alues for inductance, peak, and RMS currents are required
to select the output inductor. The input and output voltages
and the inductance value determine the peak-to-peak induc-
tor ripple current. Generally, higher inductance values are
used with higher input voltages. Larger peak-to-peak ripple
currents will increase the power dissipation in the inductor
and MOSFET s. Larger output ripple currents will also require
more output capacitance to smooth out the larger ripple cur-
rent. Smaller peak-to-peak ripple currents require a larger
inductance value and therefore, a larger and more expensive
inductor. A good compromise between size, loss and cost is
to set the inductor ripple current to be equal to 20% of the
maximum output current. The inductance value is calculated
by the equation below.
LV(VmaxV)
V max f 0.2 I max
OUT IN OUT
IN S OUT
=×−
×× ×
()
() ()
where:
f
S = switching frequency, 500kHz
0.2 = ratio of AC ripple current to DC output current
V
IN(max) = maximum input voltage
The peak-to-peak inductor current (AC ripple current) is:
IV(VmaxV)
Vmax f L
PP OUT IN OUT
IN S
=×−
××
()
()
The peak inductor current is equal to the average output current
plus one half of the peak-to-peak inductor ripple current.
March 2009 9 M9999-032409
MIC2169 Micrel
I I max 0.5 I
PK OUT PP
=+×()
The RMS inductor current is used to calculate the I2 × R
losses in the inductor.
IImax1
1
3I
Imax
INDUCTOR(rms) OUT P
OUT
2
=×+
⎛
⎝
⎜⎞
⎠
⎟
() ()
Maximizing effi ciency requires the proper selection of core
material and minimizing the winding resistance. The high
frequency operation of the MIC2169 requires the use of fer-
rite materials for all but the most cost sensitive applications.
Lower cost iron powder cores may be used but the increase
in core loss will reduce the effi ciency of the power supply.
This is especially noticeable at low output power. The winding
resistance decreases effi ciency at the higher output current
levels. The winding resistance must be minimized although
this usually comes at the expense of a larger inductor. The
power dissipated in the inductor is equal to the sum of the
core and copper losses. At higher output loads, the core
losses are usually insignifi cant and can be ignored. At lower
output currents, the core losses can be a signifi cant con-
tributor. Core loss information is usually available from the
magnetics vendor. Copper loss in the inductor is calculated
by the equation below:
PI R
INDUCTORCu INDUCTOR(rms) WINDING
2
=×
The resistance of the copper wire, RWINDING, increases with
temperature. The value of the winding resistance used should
be at the operating temperature:
R R 1 0.0042 (T T )
WINDING(hot) WINDING(20 C) HOT 20 C
=×+×−
()
°°
where:
T
HOT = temperature of the wire under operating load
T
20°C = ambient temperature
RWINDING(20°C) is room temperature winding resistance (usu-
ally specifi ed by the manufacturer)
Output Capacitor Selection
The output capacitor values are usually determined capacitors
ESR (equivalent series resistance). Voltage and RMS current
capability are two other important factors to consider when
selecting the output capacitor. Recommended capacitors are
tantalum, low-ESR aluminum electrolytics, and POSCAPS.
The output capacitor’s ESR is usually the main cause of
output ripple. The output capacitor ESR also affects the
overall voltage feedback loop from stability point of view. See:
“Feedback Loop Compensation” section for more information.
The maximum value of ESR is calculated:
RV
I
ESR OUT
PP
≤Δ
where:
V
OUT = peak-to-peak output voltage ripple
I
PP = peak-to-peak inductor ripple current
The total output ripple is a combination of the ESR output
capacitance. The total ripple is calculated below:
ΔVI(1D)
Cf IR
OUT PP
OUT S
2
PP ESR 2
=×−
×
⎛
⎝
⎜⎞
⎠
⎟+×
()
where:
D = duty cycle
C
OUT = output capacitance value
f
S = switching frequency
The voltage rating of capacitor should be twice the voltage for
a tantalum and 20% greater for an aluminum electrolytic.
The output capacitor RMS current is calculated below:
II12
CPP
OUT(rms) =
The power dissipated in the output capacitor is:
PIR
DISS(C C ESR(C )
OUT OUT(rms)2OUT
)=×
Input Capacitor Selection
The input capacitor should be selected for ripple current rating
and voltage rating. Tantalum input capacitors may fail when
subjected to high inrush currents, caused by turning the input
supply on. To maximize reliability, tantalum input capacitor
voltage rating should be at least two times the maximum in-
put voltage. Aluminum electrolytic, OS-CON, and multilayer
polymer fi lm capacitors can handle the higher inrush currents
without voltage derating. The input voltage ripple will primar-
ily depends upon the input capacitor’s ESR. The peak input
current is equal to the peak inductor current, so:
ΔVI R
IN INDUCTOR(peak) ESR(C )
IN
=×
The input capacitor must be rated for the input current ripple.
The RMS value of input capacitor current is determined at
the maximum output current. Assuming the peak-to-peak
inductor ripple current is low:
I I max D (1 D)
C (rms) OUT
IN ≈××−()
The power dissipated in the input capacitor is:
PIR
DISS(C ) C (rms) ESR(C )
IN IN 2IN
=×
Voltage Setting Components
The MIC2169 requires two resistors to set the output voltage
as shown in Figure 2.
MIC2169 Micrel
M9999-032409 10 March 2009
Error
Amp 7
MIC2169 [adj.]
FB
VREF
0.8V
R2
R1
Figure 2. Voltage-Divider Confi guration
Where:
V
REF for the MIC2169 is typically 0.8V
The output voltage is determined by the equation:
VV 1
R1
R2
OREF
=×+
⎛
⎝
⎜⎞
⎠
⎟
A typical value of R1 can be between 3kΩ and 10kΩ. If R1 is
too large, it may allow noise to be introduced into the voltage
feedback loop. If R1 is too small, in value, it will decrease the
effi ciency of the power supply , especially at light loads. Once
R1 is selected, R2 can be calculated using:
R2 VR1
VV
REF
OREF
=×
−
External Schottky Diode
An external freewheeling diode is used to keep the inductor
current fl ow continuous while both MOSFETs are turned off.
This dead time prevents current from fl owing unimpeded
through both MOSFETs and is typically 15ns. The diode
conducts twice during each switching cycle. Although the
average current through this diode is small, the diode must
be able to handle the peak current.
I I 2 15ns f
D(avg) OUT S
=×××
The reverse voltage requirement of the diode is:
VV
DIODE(rrm) IN
=
The power dissipated by the Schottky diode is:
PIV
DIODE D(avg) F
=×
where:
V
F = forward voltage at the peak diode current
The external Schottky diode, D1, is not necessary for circuit
operation since the low-side MOSFET contains a parasitic
body diode. The external diode will improve effi ciency and
decrease high frequency noise. If the MOSFET body diode
is used, it must be rated to handle the peak and average cur-
rent. The body diode has a relatively slow reverse recovery
time and a relatively high forward voltage drop. The power
lost in the diode is proportional to the forward voltage drop
of the diode. As the high-side MOSFET starts to turn on, the
body diode becomes a short circuit for the reverse recovery
period, dissipating additional power . The diode recovery and
the circuit inductance will cause ringing during the high-side
MOSFET turn-on. An external Schottky diode conducts at
a lower forward voltage preventing the body diode in the
MOSFET from turning on. The lower forward voltage drop
dissipates less power than the body diode. The lack of a
reverse recovery mechanism in a Schottky diode causes
less ringing and less power loss. Depending upon the circuit
components and operating conditions, an external Schottky
diode will give a 1/2% to 1% improvement in effi ciency.
Feedback Loop Compensation
The MIC2169 controller comes with an internal transcon-
ductance error amplifi er used for compensating the voltage
feedback loop by placing a capacitor (C1) in series with a
resistor (R1) and another capacitor C2 in parallel from the
COMP pin-to-ground. See “Functional Block Diagram.”
Power Stage
The power stage of a voltage mode controller has an induc-
tor, L1, with its winding resistance (DCR) connected to the
output capacitor, COUT, with its electrical series resistance
(ESR) as shown in Figure 3. The transfer function G(s), for
such a system is:
ESR
C
OUT
V
O
DCRL
Figure 3. The Output LC Filter in a Voltage Mode
Buck Converter
G(s) 1 ESR s C
DCRsCs LC1ESRsC
2
=+××
()
×× + ×× ++ ××
⎛
⎝
⎜⎞
⎠
⎟
Plotting this transfer function with the following assumed values
(L=2 μH, DCR=0.009Ω, COUT=1000μF, ESR=0.025Ω) gives
lot of insight as to why one needs to compensate the loop by
adding resistor and capacitors on the COMP pin. Figures 4
and 5 show the gain curve and phase curve for the above
transfer function.
100 1.10
3
1.10
4
1.10
5
1.10
6
-37.5
-15
7.5
30
30
-80
-80
GAIN
1000000100 f
Figure 4. The Gain Curve for G(s)
March 2009 11 M9999-032409
MIC2169 Micrel
Figure 5. Phase Curve for G(s)
It can be seen from the transfer function G(s) and the gain
curve that the output inductor and capacitor create a two pole
system with a break frequency at:
f1
2LC
LC OUT
=××π
Therefore, fLC = 3.6kHz
By looking at the phase curve, it can be seen that the output
capacitor ESR (0.025Ω) cancels one of the two poles (LCOUT)
system by introducing a zero at:
f1
2 ESR C
ZERO OUT
=×× ×π
Therefore, FZERO = 6.36kHz.
From the point of view of compensating the voltage loop, it
is recommended to use higher ESR output capacitors since
they provide a 90° phase gain in the power path. For com-
parison purposes, Figure 6 shows the same phase curve
with an ESR value of 0.002Ω.
Figure 6. The Phase Curve with ESR = 0.002Ω
It can be seen from Figure 5 that at 50kHz, the phase is
approximately –90° versus Figure 6 where the number is
–150°. This means that the transconductance error ampli-
fi er has to provide a phase boost of about 45° to achieve a
closed-loop phase margin of 45° at a crossover frequency
of 50kHz for Figure 4, versus 105° for Figure 6. The simple
RC and C2 compensation scheme allows a maximum error
amplifi er phase boost of about 90°. Therefore, it is easier to
stabilize the MIC2169 voltage control loop by using high ESR
value output capacitors.
gm Error Amplifi er
It is undesirable to have high error amplifi er gain at high
frequencies because high frequency noise spikes would be
picked up and transmitted at large amplitude to the output,
thus, gain should be permitted to fall off at high frequencies. At
low frequency , it is desireable to have high open-loop gain to
attenuate the power line ripple. Thus, the error amplifi er gain
should be allowed to increase rapidly at low frequencies.
The transfer function with R1, C1, and C2 for the internal
gm error amplifi er can be approximated by the following
equation:
Error Amplifier(z) g 1R1 SC1
sC1C21R1
C1 C2 S
C1 C2
m
=× +××
×+
()
+× ××
+
⎛
⎝
⎜⎞
⎠
⎟
⎡
⎣
⎢
⎢
⎢
⎢
⎤
⎦
⎥
⎥
⎥
⎥
The above equation can be simplified by assuming
C2<<C1,
Error Amplifier(z) g 1R1 SC1
s C1 1 R1 C2 S
m
=× +××
×
()
+× ×
()
⎡
⎣
⎢⎤
⎦
⎥
From the above transfer function, one can see that R1 and
C1 introduce a zero and R1 and C2 a pole at the following
frequencies:
Fzero=
1/2 π × R1 × C1
Fpole =
1/2 π × C2 × R1
Fpole@origin =
1/2 π × C1
Figures 7 and 8 show the gain and phase curves for the above
transfer function with R1 = 9.3k, C1 = 1000pF, C2 = 100pF,
and gm = .005Ω–1. It can be seen that at 50kHz, the error
amplifi er exhibits approximately 45° of phase margin.
Figure 7. Error Amplifi er Gain Curve
MIC2169 Micrel
M9999-032409 12 March 2009
Figure 8. Error Amplifi er Phase Curve
Total Open-Loop Response
The open-loop response for the MIC2169 controller is easily
obtained by adding the power path and the error amplifi er
gains together, since they already are in Log scale. It is
desirable to have the gain curve intersect zero dB at tens of
kilohertz, this is commonly called crossover frequency; the
phase margin at crossover frequency should be at least 45°.
Phase margins of 30° or less cause the power supply to have
substantial ringing when subjected to transients, and have
little tolerance for component or environmental variations.
Figures 9 and 10 show the open-loop gain and phase margin.
It can be seen from Figure 9 that the gain curve intersects
the 0dB at approximately 50kHz, and from Figure 10, that at
50kHz, the phase shows approximately 50° of margin.
100 1.10
3
1.10
4
1.10
5
1.10
6
50
0
50
100
71.607
42.933
OPEN LOOP GAIN MARGIN
1000000100 f
Figure 9. Open-Loop Gain Margin
10 100 1.10
3
1.10
4
1.10
5
1.10
6
350
300
250
269.097
360
100000010 f
OPEN LOOP PHASE MARGIN
Figure 10. Open-Loop Phase Margin
March 2009 13 M9999-032409
MIC2169 Micrel
Design Example
Layout and Checklist:
1. Connect the current limiting (R2) resistor directly
to the drain of top MOSFET Q3.
2. Use a 5Ω resistor from the input supply to the VIN
pin on the MIC2169. Also, place a 1μF ceramic
capacitor from this pin to GND, preferably not thru
a via.
3. The feedback resistors R3 and R4/R5/R6 should
be placed close to the FB pin. The top side of R3
should connect directly to the output node. Run
this trace away from the switch node (junction of
Q3, Q2, and L1). The bottom side of R3 should
connect to the GND pin on the MIC2169.
4. The compensation resistor and capacitors should
be placed right next to the COMP pin and the other
side should connect directly to the GND pin on the
MIC2169 rather than going to the plane.
5. Add a 1.4Ω resistor and a 1000pF capacitor from
the switch node to ground pin. See page 7, Current
Limiting section for more detail.
6. Add place holders for gate resistors on the top and
bottom MOSFET gate drives. If necessary, gate
resistors of 10Ω or less should be used.
7. Low gate charge MOSFETs should be used to
maximize effi ciency , such as Si4800, Si4804BDY,
IRF7821, IRF8910, FDS6680A and FDS6912A,
etc.
8. Compensation component GND, feedback resistor
ground, chip ground should all run together and
connect to the output capacitor ground. See demo
board layout, top layer.
9. The 10μF ceramic capacitor should be placed
between the drain of the top MOSFET and the
source of the bottom MOSFET.
10. The 10μF ceramic capacitor should be placed right
on the VDD pin without any vias.
11. The source of the bottom MOSFET should connect
directly to the input capacitor GND with a thick
trace. The output capacitor and the input capacitor
should connect directly to the GND plane.
12. Place a 0.01μF to 0.1μF ceramic capacitor in parallel
with the CS resistor to fi lter any switching noise.
12
+
C6
330uF/6.3V
12
+
C7
330uF
12
+C2
68uF/20V
4
81
2
37
6
5
Q2
IRF7821
4
81
2
37
6
5
Q3
IRF7821
C5
0.1uF/25V
C12
0.1uF/25V
C9
Open
COMP/EN
4
Vdd 2
BST 10
CS 3
FB 5
GND
6
LSD 7
HSD 9
VSW 8
Vin 1
U1
MIC2169
1 2
34
5 6
JP2
HEA DER 3X2
R2
470 ohm
+VIN
C11
Open
C4
10uF/6V
21
D1
SD103BWS
C13
1uF/16V
R9
10
2 1
D2
1N5819HW
C15
100pF
C16
0.1uF
1.5V2.5V3.3V
Cout=AVX TPSD337M006R0045
Cin=AVX TPSD686M020R0070
C10
0.1uF
12
+C3
68uF
1.0uH
R3
10K
R5
4.64K R6
11.3K
R8
4.02K
R7
100K
R10
4R02 Ohm
R12
47
R4
3.16k
1
J2
SHDN
12
+
C8
Open R14
Open
2 1
L1
CDRH127 / LD-1R0-MC
CBA
20V C1
10uF/16V
C14
DIN
1
32
Q1
2N7002E
R1
0 Ohm
R11
RES
R13
RES
1
J1
+Vin 5-12V
1
J3
GND
1
J5
GND
1
J4
Vout
MIC2169BMM Evaluation Board Schematic
March 2009 14 M9999-032409
MIC2169 Micrel
MIC2169BMM Bill of Materials
Item Part Number Manufacturer Description Qty.
U1 MIC2169-YMM Micrel, Inc. Buck controller 1
Q2, Q3 IRF7821-TR IR 30V, N channel HEXFET , Power MOSFET 2
SI4390DY Vishay OR 0
D1 SD103BWS Vishay 30V , Schottky Diode 1
D2 1N5819HW Diodes Inc. 40V , Schottky Diode 1
SL04 Vishay OR 0
CMMSH1-40 Central Semi OR 0
L1 CDRH127LDNP-1R0NC Sumida 1.0uH, 10A inductor 1
HC5-1R0 Cooper Electronic OR 0
SER1360-1R0 Coilcraft OR 0
C1 C3225X7R1C106M TDK 10uF/16V, X7R Ceramic cap. 1
C2 , C3. TPSD686M020R0070 AVX 68uF, 20V Tantalum 2
594D686X0020D2T Vishay/Sprague OR 0
C4 C2012X5R0J106M TDK 10uF/6.3V, 0805 Ceramic cap. 1
CM21X5R106M06AT AVX OR 0
C5, C10 , C12 VJ1206Y104KXXAT Vishay Victramon 0.1uF/25V Ceramic cap. 3
C6, C7 TPSD337M006R0045 AVX 330uF, 6.3V, Tantalum 2
C8 594D337X06R3D2T Vishay/Sprague Open 0
C9 ,C11. Vishay Dale open 0
C13 C2012X7R1C105K TDK 1uF/16V, 0805 Ceramic cap. 1
GRM21BR71C105KA01B. muRata OR 0
VJ1206S105KXJAT Vishay Victramon OR 0
C14 DIN 0
C15 VJ0603A102KXXAT Vishay Victramon 1000pF /25V, 0603 , NPO 1
C16 VJ0603Y104KXXAT Vishay Victramon 0.1uF/25V Ceramic cap. 1
R2 CRCW06034700JRT1 Vishay 470 Ohm , 0603, 1/16W, 5%. 1
R3 CRCW08051002FRT1 Vishay 10K / 0805 1/10W, 1% 1
R4 CRCW08053161FRT1 Vishay 3.16K /0805, 1/10W , 1% 1
R5 CRCW08054641FRT1 Vishay 4.64K /0805, 1/10W , 1% 1
R6 CRCW08051132FRT1 Vishay 11.3K / 0805, 1/10W, 1% 1
R8 CRCW06034021FRT1 Vishay 4.02K ,0603,1/16W, 1% 1
R9, CRCW12065R00FRT1 Vishay 5 ohm , 1/8W , 1206 , 1% 2
R10 CRCW12062R00FRT1 Vishay 2 Ohm , 1/8 W , 1206 , 1% 1
R12 CRCW12061R40FRT1 Vishay 1.4 Ohm , 1/8 W , 1206 , 1% 1
R14 Open 0
J1, J3, J4, J5 2551-2-00-01-00-00-07-0 MilMax Turret Pins 4
Notes:
Micrel.Inc 408-944-08001. Vishay corp 206-452-56642. Diodes. Inc 805-446-48003. Sumida 408-321-96604. TDK 847-803-61005. muRata 800-831-91726. AVX 843-448-94117. International Rectifi er 847-803-61008. Fairchild Semiconductor 207-775-81009. Cooper Electronic 561-752-500010. Coilcraft 1-800-322-264511. Central Semi 631-435-111012.
March 2009 15 M9999-032409
MIC2169 Micrel
Package Information
10-Pin MSOP (MM)
MICREL, INC. 2180 FORTUNE DRIVE SAN JOSE, CA 95131 USA
TEL + 1 (408) 944-0800 FAX + 1 (408) 944-0970 WEB http://www.micrel.com
The information furnished by Micrel in this datasheet is believed to be accurate and reliable. However, no responsibility is assumed by Micrel for its use.
Micrel reserves the right to change circuitry and specifi cations at any time without notifi cation to the customer.
Micrel Products are not designed or authorized for use as components in life support appliances, devices or systems where malfunction of a product can
reasonably be expected to result in personal injury. Life support devices or systems are devices or systems that (a) are intended for surgical implant into
the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a signifi cant injury to the user. A Purchaser’s
use or sale of Micrel Products for use in life support appliances, devices or systems is at Purchaser’s own risk and Purchaser agrees to fully indemnify
Micrel for any damages resulting from such use or sale.
© 2005 Micrel, Incorporated.
